WO2020196580A1 - Radar device and signal processing method for radar signal - Google Patents

Abstract

This radar device is provided with: a transmission circuit for transmitting, from a plurality of transmission antenna elements, each of a plurality of radar signals in which a first phase rotation is cyclically applied to each of the plurality of transmission antenna elements and in which said plurality of radar signals have a phase difference corresponding to the transmission antenna elements; a reception circuit that receives, by means of a plurality of reception antenna elements, reflected wave signals resulting from the plurality of radar signals being reflected at an object to be detected; and a removal circuit for removing a Doppler frequency component in which the power in an estimated direction of arrival of Doppler frequency components has decreased relatively with respect to the phase differences from among Doppler frequency components remaining in signals in which second phase rotations that are the inverse of the first phase rotations of each of the transmission antenna elements and that are of a rotation amount corresponding to the first phase rotations and the phase differences are applied in each of the reflected wave signals received by the plurality of reception antenna elements.

Description

Radar device and signal processing method of radar signal

The present disclosure relates to a radar device and a signal processing method for radar signals.

In recent years, the utilization of radar devices has been in full swing, mainly in advanced driver assistance systems for automobiles and autonomous vehicles. Radar devices are required not only to detect moving object information such as vehicles, motorcycles, bicycles, and pedestrians, but also to detect environmental information such as roads, bollards, and curbs, and the need for radar devices that can achieve high resolution is increasing. ing. As a configuration of a radar device capable of realizing high resolution, there is a MIMO (Multiple-Input and Multiple-Output) radar.

In MIMO radar, by devising the arrangement of antenna elements in the transmission / reception array antenna, a virtual reception array antenna (hereinafter referred to as a virtual array) that is equal to the product of the number of transmission antenna elements and the number of reception antenna elements is configured. it can. As a result, the effective aperture length of the array antenna can be increased with a small number of elements.

Japanese Unexamined Patent Publication No. 2013-7756 Japanese Unexamined Patent Publication No. 2014-119344

The non-limiting examples of the present disclosure contribute to the provision of an improved radar device and a signal processing method for radar signals with improved detection accuracy of a detection target.

The radar device according to one aspect of the present disclosure periodically performs the first phase rotation on each of the plurality of transmitting antenna elements, and produces a plurality of radar signals having a phase difference according to the transmitting antenna element. The transmission circuit that transmits from the plurality of transmitting antenna elements, the receiving circuit that receives the reflected wave signal reflected by the plurality of radar signals in the detection target by the plurality of receiving antenna elements, and the receiving circuit that receives the reflected wave signals by the plurality of receiving antenna elements. Each of the reflected wave signals has a phase rotation opposite to that of the first phase rotation of each transmitting antenna element, and a second phase rotation amount corresponding to the first phase rotation and the phase difference. Among the Doppler frequency components remaining in the signal to which the phase rotation is applied, the removal circuit for removing the Doppler frequency component whose power in the estimated arrival direction of the Doppler frequency component is relatively reduced according to the phase difference is provided. Take the composition.

In the signal processing method of the radar signal according to one aspect of the present disclosure, a plurality of radars are periodically subjected to the first phase rotation for each of the plurality of transmitting antenna elements and have a phase difference according to the transmitting antenna element. The process of transmitting signals from the plurality of transmitting antenna elements, the process of receiving the reflected wave signal reflected by the plurality of radar signals in the detection target by the plurality of receiving antenna elements, and the processing of receiving the plurality of receiving antenna elements. Each of the reflected wave signals received by the transmission antenna element has a phase rotation opposite to that of the first phase rotation of each transmitting antenna element, and a number of rotation amounts corresponding to the first phase rotation and the phase difference. Of the Doppler frequency components remaining in the signal given the phase rotation of 2, the process of removing the Doppler frequency component in which the power in the estimated arrival direction of the Doppler frequency component is relatively reduced according to the phase difference is performed. Take a configuration that includes.

It should be noted that these comprehensive or specific embodiments may be realized in a system, device, method, integrated circuit, computer program, or recording medium, and the system, device, method, integrated circuit, computer program, and recording medium. It may be realized by any combination of.

According to the non-limiting embodiment of the present disclosure, it is possible to provide an improved radar device and a signal processing method of a radar signal with improved detection accuracy of a detection target.

Further advantages and effects in the non-limiting examples of the present disclosure will be apparent from the specification and drawings. Such advantages and / or effects are provided by some embodiments and features described in the specification and drawings, respectively, but not all need to be provided in order to obtain one or more identical features. There is no.

Block diagram showing an example of the configuration of the radar device according to the first embodiment The figure explaining an example of the virtual image which concerns on Embodiment 1. The figure explaining an example of the phase difference of the signal arriving at a virtual reception array antenna The figure explaining an example of the relationship between the position of a virtual receiving antenna element and the phase difference of an incident signal. The figure explaining the influence of the residual error in the arrival direction estimation which concerns on Embodiment 1. A flowchart illustrating an example of the operation of the virtual image removal processing unit according to the first embodiment. A flowchart illustrating an example of the operation of the virtual image removal processing unit in steps S106 and S110 of FIG. A flowchart illustrating an example of the operation of the virtual image peak mask processing unit in step S112 of FIG.

In MIMO radar, in order to distinguish the transmitting antenna that transmitted the received radio wave, for example, a method of distinguishing the transmitting antenna by switching the transmitting antenna in time division is used.

In addition, MIMO radar may detect the distance to an object by using a chirp signal that linearly raises or lowers the frequency according to the passage of time. For example, the chirp signal transmitted by the MIMO radar is reflected by the object to be detected. Next, the distance to the object to be detected is detected based on the signal obtained by synthesizing the received signal including the reflected wave by the object to be detected and the transmitted signal received by the MIMO radar with a mixer. In addition, the MIMO radar detects the relative velocity of the object with respect to the MIMO radar by executing a transmission sequence in which the chirp signal is transmitted in a time-division manner for each transmission antenna a plurality of times.

When increasing the resolution of the MIMO radar, or when realizing a 3D radar that can detect not only the horizontal direction but also the vertical direction, the number of transmitting antennas may be increased. However, in the time-division switching method, as the number of transmitting antennas increases, the transmission sequence period may increase and the relative speed that can be detected may decrease. Also, shortening the chirp signal period can reduce the distance resolution.

For example, the following two techniques are considered as alternative technologies for switching the transmitting antenna in a time-division manner. The first technique enables simultaneous transmission by diffusing each transmitting antenna with a different orthogonal code. In the second technique, simultaneous transmission is enabled by Doppler shifting each transmitting antenna by a frequency larger than the Doppler bandwidth.

However, in the first technique, when backdiffusion is performed using the orthogonal code assigned to each transmitting antenna in the receiving circuit of the radar device, interference caused by the periodicity of the code or the like is caused by multiplying different codes. Ingredients are generated, leading to false detection. Further, in the second technique, since the transmitting antenna is separated on the Doppler frequency axis, the relative speed that can be detected is limited.

In the embodiment described below, in a radar device capable of simultaneous transmission using an orthogonal code, deterioration of distance detection performance and speed detection performance is prevented or suppressed when the number of transmitting antennas increases. This makes it possible to provide a MIMO radar that improves the azimuth resolution and suppresses false detection due to the orthogonal code.

Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. The embodiments described below are examples, and the present disclosure is not limited to the following embodiments.

(Embodiment 1)
FIG. 1 is a block diagram showing an example of the configuration of the radar device 10 according to the first embodiment.

The radar device 10 includes, for example, a transmission control unit 100, a transmission signal generation unit 101, a phase rotation unit 102, a transmission high frequency unit 103, and a transmission array antenna 104. The transmission control unit 100, the transmission signal generation unit 101, the phase rotation unit 102, the transmission high frequency unit 103, and the transmission array antenna 104 form a transmission circuit.

The radar device 10 transmits a radar signal a plurality of times at predetermined intervals of several microseconds to several tens of microseconds, detects the Doppler frequency of the received signal, and thereby reflects the radar signal (reflecting object). , The detection target) detects the relative velocity with respect to the radar device 10. Here, the transmission interval of the radar signal (chirp signal) is Td, and the number of transmissions is Nd. In this case, the maximum value fdmax of the Doppler frequency that can be detected is equal to 1 / (2 × Td).

The transmission control unit 100 controls the radar transmission timing for transmitting at a predetermined transmission interval and the radar signal phase to be transmitted at the transmission timing for each transmission system. The transmission control unit 100 notifies the transmission signal generation unit 101 of the radar transmission timing. In one example, the radar transmission timing is the transmission interval Td. Further, in one example, the radar signal phase is determined based on Ntx codes (code length Nd bits) orthogonal to each other. Here, Ntx is an arbitrary integer of 2 or more. The Ntx codes that are orthogonal to each other are, for example, Walsh codes. In one example, the sign bit value 0 is assigned a phase of 0 radians (0 degrees) and the bit value 1 is assigned a phase of π radians (180 degrees). The transmission control unit 100 notifies the phase rotation unit 102 of a phase control signal indicating the phase assigned to one bit in the code for each transmission interval Td.

The transmission signal generation unit 101 generates a radar transmission signal at the radar transmission timing notified from the transmission control unit 100. The radar transmission signal is, for example, a chirp signal or a coded pulse signal used in the FCM (Fast Chip Modulation) method.

The phase rotation unit 102 phase-rotates the radar transmission signal generated by the transmission signal generation unit 101 for each transmission system based on the phase control signal generated by the transmission control unit 100. Hereinafter, the phase rotation amount of the ncth transmission in the transmission antenna Ntj is defined as θj (nc).

The transmission high frequency unit 103 adjusts the radar signal phase-rotated for each transmission system by the phase rotation unit 102 to a preset transmission power, and outputs the radar signal to each of the plurality of transmission antenna elements of the transmission array antenna 104.

The transmission array antenna 104 transmits the radar signal generated by the transmission high frequency unit 103 as radio waves in the air. The transmission array antenna 104 includes Ntx transmission antenna elements. In the example shown in FIG. 1, Ntx = 2. Hereinafter, for the sake of simplicity, the present disclosure will be described with respect to the case where the transmitting array antenna 104 includes two transmitting antenna elements At1 and At2.

Reference numeral 115 indicates a location where a phase error can occur in the transmission radar signal (a location where a phase error occurs). The phase error can occur for each transmission system.

Further, the radar device 10 includes, for example, a reception array antenna 105, a reception high frequency unit 106, and an A / D converter (ADC) 107. The receiving array antenna 105, the receiving high frequency unit 106, and the ADC 107 form a receiving circuit.

The receiving array antenna 105 receives the reflected wave transmitted from the transmitting array antenna 104, reflected by the reflecting object, and returned to the radar device 10. The receiving array antenna 105 includes Nrx transmitting antenna elements. Here, Nrx is an arbitrary integer of 2 or more. In the example shown in FIG. 1, Nrx = 2. Hereinafter, for the sake of simplicity, the present disclosure will be described with respect to the case where the receiving array antenna 105 includes two receiving antenna elements Ar1 and Ar2.

The receiving antenna elements Ar1 and Ar2 receive the reflected waves of the radio waves transmitted from the transmitting antenna elements At1 and At2, respectively. In other words, the receiving antenna elements Ar1 and Ar2 passed through the path of At1 → reflective object → Ar1, the path of At2 → reflective object → Ar1, the path of At1 → reflective object → Ar2, and the path of At2 → reflective object → Ar2. , Ntx × Nrx system radar signal is received. In the radar device 10, the orientation of the reflecting object is detected based on these plurality of path differences (phase differences caused by the arrival time difference) and the relative positions between the transmitting antenna element and the receiving antenna element. For example, the beamformer method or the MUSIC (MUSIC Signal Classification) method is used to detect the orientation of the reflecting object. In the MIMO radar, by appropriately arranging the transmission / reception antenna spacing, the effective aperture length of the array antenna can be virtually expanded with a small number of elements, and the angular resolution can be improved.

The reception high frequency unit 106 performs demodulation processing by multiplying the signal received by the reception antenna elements Ar1 and Ar2 of the reception array antenna 105 with the transmission signal, and generates a reception signal (beat signal) for each reception system.

The ADC 107 converts the received signal received by the receiving high frequency unit 106 into analog / digital for each receiving system.

Further, the radar device 10 further includes, for example, a distance detection processing unit 108, a phase reverse rotation unit 109, a speed detection processing unit 110, a peak detection processing unit 111, a virtual image removal processing unit (removal circuit) 112, and an azimuth detection. A processing unit (detection circuit) 113 and an object detection processing unit 114 are provided.

The distance detection processing unit 108 detects the distance (arrival distance) from the beat signal to the reflecting object for each receiving system. For detecting the distance to the reflecting object, for example, an FFT (Fast Fourier Transform) calculation using a beat signal within one transmission interval Td as an input is used. The frequency difference between the transmitted chirp signal and the received chirp signal is obtained based on the output of the FFT calculation, and the arrival time is calculated based on the frequency difference and converted into the arrival distance. The input beat signal of the relative time t within the ncth transmission interval Td in the receiving antenna element Ari is represented by r (t, nc; i). Further, the distance detection FFT calculation result for the input beat signal r (t, nc; i) is represented by Fr (nr, nc; i). Here, the distance index nr corresponds to the arrival distance.

The phase reverse rotation unit 109 performs phase reverse rotation on the reception signal of each reception system based on the phase control signal generated for each transmission system by the transmission control unit 100. The number of systems of this output is Ntx × Nrx, and the signal of each output system is a signal Frot (nr) obtained by performing a phase rotation of the FFT calculation result Fr (nr, nc; i) for distance detection by −θj (nc). , Nc; i, j).

The velocity detection processing unit 110 estimates the Doppler frequency for each distance based on the plurality of beat signals generated by receiving the reflected waves of the chirp signals transmitted a plurality of times, and estimates the relative velocity of the reflecting object. For example, the Doppler frequency can be obtained by performing an FFT calculation on the Nd signal Frot (nr, nc; i, j) over the chirp index nc for the signal of each distance index. Doppler frequency is converted to relative velocity. The speed detection FFT calculation result at the distance index nr between the receiving antenna element Ari and the transmitting antenna element Atj is represented by Fv (nr, nv; i, j). Here, the velocity index nv corresponds to the relative velocity of the reflecting object.

The peak detection processing unit 111 extracts the peak signal of the speed detection FFT calculation result Fv (nr, nv; i, j), removes the noise component, and generates information (peak information) for specifying the peak signal. The peak information is, for example, a cell (peak cell) corresponding to the peak (maximum value) of the speed detection FFT calculation result Fv (nr, nv; i, j). Here, the cell is a pair (nr, nv) of the distance index nr and the velocity index nv. The peak detection processing unit 111 extracts a peak signal by, for example, extracting a peak cell of a speed detection FFT calculation result Fv (nr, nv; i, j) by a CFAR (Constant False Allarm Rate) algorithm. The output signal of the peak detection processing unit 111 includes, for example, the speed detection FFT calculation result Fv (nr, nv; i, j) and the generated peak information.

The virtual image removal processing unit 112 removes the interference component between the codes generated when the received signal is separated for each signal component of the transmission system by the phase reverse rotation unit 109, that is, the virtual image component.

The virtual image removal processing unit 112 includes, for example, a phase correction unit 1121, a simple arrival direction estimation unit 1122, a frequency axis power comparison unit 1123, and a virtual image peak mask processing unit 1124.

The phase correction unit 1121 corrects the phase error of the received signal by performing a phase rotation preset for each transmission system. The phase error to be corrected is, for example, a phase error generated at the phase error occurrence location 115. The processing content of the phase correction unit 1121 will be described later with reference to FIGS. 3A, 3B, and 4.

The simple arrival direction estimation unit 1122 simply estimates the arrival direction of the reflected wave and outputs the power (maximum power) in the direction having the maximum likelihood. For the simple arrival direction estimation, for example, an FFT calculation for azimuth detection with coarse azimuth accuracy or a DFT (Discrete Fourier Transform) calculation for azimuth detection is used. The processing content of the simple arrival direction estimation unit 1122 will be described later with reference to FIG.

The frequency axis power comparison unit 1123 compares the maximum powers of the corresponding reflected wave components in the estimated arrival direction of the peak cell and the peak cell deviated by the velocity index offset value at which the virtual image peak appears, and determines the virtual image peak. To detect. The velocity index offset value is a preset value. The processing content of the frequency axis power comparison unit 1123 will be described later with reference to FIG.

The virtual image peak mask processing unit 1124 removes (masks) the component corresponding to the virtual image peak in the speed detection FFT calculation result Fv. As a result, erroneous detection of the radar device 10 due to the virtual image is suppressed or reduced.

The azimuth detection processing unit 113 estimates the azimuth of the reflecting object based on the speed detection FFT calculation result Fv masked by the virtual image peak mask processing unit 1124. For example, the beamformer method or the MUSIC method is used to estimate the azimuth.

The object detection processing unit 114 estimates at least one of the position, size, orientation, and speed of the reflecting object based on the speed detection FFT calculation result Fv after the mask processing and the estimated azimuth. For example, the object detection processing unit 114 determines the distance to the reflective object and the relative velocity of the reflective object shown in the masked speed detection FFT calculation result Fv, and the orientation of the reflective object estimated by the orientation detection processing unit 113. Clustering processing and tracking processing are performed on the point group information of the object indicating.

FIG. 2 is a diagram illustrating an example of a virtual image according to the first embodiment.

Example 201 of a radar signal (chirp signal) transmission pattern shows an example of a chirp transmission having a transmission interval of Td and a number of repeated transmissions of Nd. For example, the transmission signal for the transmission antenna element At1 is phase-rotated using a code having Nd bit values of 0. In the transmission signal for the transmission antenna element At2, bit value 0 and bit value 1 are alternately repeated, and phase rotation is performed using a code that continues Nd in total. The transmitting antenna element At1 and the transmitting antenna element At2 transmit these transmission signals.

For example, it is assumed that there is a reflecting object having a velocity of the Doppler frequency fd, and the reflected wave by the reflecting object is received by the receiving antenna element Ar1. Let r1 (t, nc) be the beat signal of the reflected wave at the time t and the chirp index nc of the chirp signal transmitted independently from the transmitting antenna element At1 without phase rotation. Let r2 (t, nc) be the beat signal of the reflected wave at the time t and the chirp index nc of the chirp signal transmitted independently from the transmitting antenna element At2 without phase rotation.

In this case, the received signal r (t, nc) of the receiving antenna element Ar1 is the reflected wave of the chirp signal transmitted from the transmitting antenna element At1:

And the reflected wave of the chirp signal transmitted from the transmitting antenna element At2:

And, including,

Will be. Here, j represents an imaginary unit. In the example shown in FIG. 2, θ1 (nc) = 0, and

Is.

By performing the distance detection FFT process on the received signal r (t, nc) over a time t, the received signal:

Is obtained. Here, the distance index nr is an index corresponding to the distance r.

Next, in order to extract the received signal component from the transmitting antenna element At1, the received signal Fr (nr, nc ⁾ is multiplied by ^{ej × (−θ1 (nc))} and rotated in the opposite phase.

Is obtained. As shown in this equation, the signal Frot (nr, nc; 1) includes the received signal component Fr2 (nr, nc) from the transmitting antenna element At2 and ej ^{× (θ2 (n) -θ1 (n)).} The component multiplied by is left.

Similarly, in order to extract the received signal component Fr2 (nr, nc) from the transmitting antenna element At2, the received signal Fr (nr, nc ⁾ is multiplied by ^{ej × (−θ2 (nc))} and rotated in opposite phase. By the signal:

Is obtained. As shown in this equation, the signal Frot (nr, nc; 2) includes the received signal component Fr1 (nr, nc) from the transmitting antenna element At1 and ej ^{× (θ1 (nc) −θ2 (nc)).} The component multiplied by is left.

Here, pay attention to the value of the received signal at the distance index nr corresponding to the distance to the reflecting object. In graphs 202 and 203 of FIG. 2 showing an example of the change in the phase of the received signal with time, the horizontal axis represents the time represented by the chirp index nc, and the vertical axis represents the phase of the received signal.

Graph 202 shows the result of anti-phase rotation by multiplying the received signal component Fr1 (nr, nc) from the transmitting antenna element At1 by ej ^{× (−θ1 (nc))} (signal Frot (nr, nc; 1)). The phase change of the first term) of) is shown. In the graph 202, the received signal component Fr1 (nr, nc) is correctly subjected to anti-phase rotation based on the phase rotation amount θ1 (nc) for the same transmitting antenna element At1, and has a Doppler frequency fd of the reflecting object. The resulting phase change can be confirmed. Here, the correct phase reverse rotation is the phase rotation of the phase rotation amount that cancels the phase rotation amount of the transmission signal.

On the other hand, the graph 203 shows the result of anti-phase rotation by multiplying the received signal component Fr2 (nr, nc) from the transmitting antenna element At2 by ej ^{× (−θ1 (nc))} (signal Frot (nr, nc;; The phase change of the second term) of 1) is shown. In graph 203, the received signal component Fr2 (nr, nc) is erroneously subjected to anti-phase rotation based on the phase rotation amount θ1 (nc) for different transmitting antenna elements At1. The erroneous phase reverse rotation is a phase rotation of a phase rotation amount that does not cancel the phase rotation amount of the transmitted signal.

In the graph 203, it can be confirmed that the phase change caused by the Doppler frequency fd of the reflecting object indicated by the white circle has a phase change shifted by a phase π radian, which is a difference in the amount of phase rotation between the transmitting antenna element At1 and the transmitting antenna element At2. As a result of not removing the phase rotation in the transmitting antenna element At2, it can be confirmed in the graph 203 that the phase change caused by the Doppler frequency fd + fs / 2 obtained by adding fs / 2 (fs = 1 / Td) to the Doppler frequency fd.

Graph 204 shows the result of FFT processing for speed detection of the signal Frot (nr, nc; 1). In the graph 204, a peak indicating the existence of a reflecting object is formed at the Doppler frequency fd, and a peak of an object that does not actually exist is also formed at the Doppler frequency fd + fs / 2. This peak of an object that does not actually exist is called a virtual image peak.

FIG. 3A is a diagram illustrating an example of the phase difference of the signal arriving at the virtual reception array antenna.

The virtual receiving array antenna is obtained by superimposing all the transmitting antenna elements At1 and At2 in which the receiving antenna elements Ar1 and Ar2 are shifted according to the deviation from the reference position of the transmitting antenna elements At1 and At2. It is a receiving array antenna. The reference position is, for example, the position of the transmitting antenna element At1. For example, when the distance between the transmitting antenna elements At1 and At2 is a length of 2 × d and the distance between the receiving antenna elements Ar1 and Ar2 is a length d, the transmitting antenna elements At1 and At2 and the receiving antenna elements Ar1 and Ar2 are configured. A 2x2 virtual receive array antenna is configured. The antenna elements (virtual receiving antenna elements) of the virtual receiving array antenna are arranged in a horizontal row with an antenna interval d.

As shown in Example 301 of the incident of the reflected wave on the virtual receiving antenna element shown in FIG. 3A, when the reflected wave from the reflecting object in the orientation θ is incident on the virtual receiving antenna element, the incident of the virtual receiving antenna element A phase difference that is an integral multiple of the phase difference α occurs between the signals.

The orientation of the reflecting object that generates the reflected wave can be detected by using the phase difference of the incident signal. The azimuth detection is, for example, a beamformer method using a Fourier transform. The beamformer method is a method of calculating the phase difference of the ideal incident signal of each virtual receiving antenna element in each direction θ, calculating the correlation with the actual received signal, and estimating the direction based on the correlation.

FIG. 3B is a diagram illustrating an example of the relationship between the position of the virtual receiving antenna element and the phase difference of the incident signal.

In FIG. 3B, the virtual receiving antenna element is arranged on a straight line, the horizontal axis represents the relative position of the virtual receiving antenna element with respect to a certain position on the straight line, and the vertical axis represents the phase difference of the incident signal. .. The ideal phase difference of the incident signal is proportional to the length of the spacing between the virtual receiving antenna elements. Therefore, as shown in the graph 302 of FIG. 3B, the positions of the white circles indicating the phase difference with respect to the relative position of the virtual receiving antenna element are aligned on a straight line.

However, in reality, for example, a phase error may occur in the power supply line between the radar transmission / reception integrated circuit and the transmission / reception antenna element in terms of design, manufacturing, or mounting. For example, a phase error β1 occurs in the combination of the transmitting antenna element At1 and the receiving antenna element Ar1. Further, a phase error β2 occurs in the combination of the transmitting antenna element At2 and the receiving antenna element Ar1. Further, a phase error β3 occurs in the combination of the transmitting antenna element At1 and the receiving antenna element Ar2. Further, a phase error β4 occurs in the combination of the transmitting antenna element At2 and the receiving antenna element Ar2. As a result, the phase difference of the incident signal does not line up on a straight line as represented by the position of the black circle, and the correlation becomes low. The phase errors β1 to β4 can be measured in advance by, for example, calibration.

By adding the phase errors β1 to β4, the received signal (beat signal) r'(t, nc; i) in the receiving antenna element Ari input by the distance detection processing unit 108 is expressed by the following equation:

When,

Can be expressed by.

Here, t represents time and nc represents the chirp index. Further, r (t, nc; i, j) represents a component of the ideal incident signal r (t, nc; i) of the receiving antenna element Ari that corresponds to the transmitting signal of the transmitting antenna element Atj.

The distance detection processing unit 108 performs the distance detection FFT process on the received signal r'for a time t, and the distance detection FFT signal:

When,

And get. Here, the distance index nr is an index corresponding to the distance r.

The phase reverse rotation unit 109 rotates the distance detection FFT signal Fr'in a reverse phase (phase rotation corresponding to the first phase rotation of the second phase rotation) to obtain a signal:

When,

When,

When,

And get.

The speed detection processing unit 110 performs speed detection FFT processing over the chirp index nc on the signal Frot', and the speed detection FFT calculation result:

When,

When,

When,

And get. Here, nv represents a speed index. The phase rotation based on the phase rotation before transmission and the anti-phase rotation after reception of the radar signal is included in the FFT calculation result Fv1 and is not included in the FFT calculation result Fv2.

The peak detection processing unit 111 extracts a peak signal from the speed detection FFT calculation result Fv'. Explaining in light of an example shown in FIG. 2, the first term of the signal Fv'(nr, nv; 1,1) includes a peak (real image peak) signal at the velocity index nv corresponding to the Doppler frequency fd. Exists. Further, in the second term of the signal Fv'(nr, nv; 1,1), there is a peak (virtual image peak) signal at the velocity index nv + Nv / 2 corresponding to the Doppler frequency fd + fs / 2. Here, Nv is the number of points of the speed detection FFT.

FIG. 4 is a diagram for explaining the influence of the residual error in the arrival direction estimation according to the first embodiment.

An example of the processing result 405 is a speed detection FFT processing result Fv'(nr, nv; 1,1) in which the received signal received by the receiving antenna element Ar1 is rotated in the opposite phase assuming the phase rotation of the transmitting antenna element At1. This is an example. An example of the processing result 406 is a speed detection FFT processing result Fv'(nr, nv; 1, 2) in which the received signal received by the receiving antenna element Ar1 is rotated in the opposite phase assuming the phase rotation of the transmitting antenna element At2. This is an example. An example of the processing result 407 is a speed detection FFT processing result Fv'(nr, nv; 2, 1) in which the received signal received by the receiving antenna element Ar2 is rotated in the reverse phase assuming the phase rotation of the transmitting antenna element At1. This is an example. An example of the processing result 408 is a speed detection FFT processing result Fv'(nr, nv; 2, 2) in which the received signal received by the receiving antenna element Ar2 is rotated in the opposite phase assuming the phase rotation of the transmitting antenna element At2. This is an example.

For example, in the speed detection FFT processing result Fv'for each virtual receiving antenna element, peaks 401 and 402 appear in the distance index nr corresponding to the distance to the reflecting object, respectively. The peak 401 is a peak (real image peak) generated by performing a phase reverse rotation in which the phase reverse rotation in the phase reverse rotation unit 109 is correct, that is, a phase reverse rotation that cancels the phase rotation in the phase rotation unit 102, and is reflected. It corresponds to the Doppler frequency of the object, that is, the velocity of the reflecting object. The peak 402 is a peak generated by performing a phase reverse rotation in which the phase reverse rotation in the phase reverse rotation unit 109 is erroneous, that is, a phase reverse rotation that does not cancel the phase rotation in the phase rotation unit 102, and is a virtual image component (virtual image). Peak).

The phase correction unit 1121 corrects the phase error (phase difference according to the transmitting antenna element) β1, β2, β3, β4 for the signals shown in Examples 405, 406, 407, and 408 of the processing result (No. 1). Of the two phase rotations, the phase rotation corresponds to the phase difference according to the transmitting antenna element). For example, the phase-corrected signals Fv1'' to Fv4'' of the processing results example 405, 406, 407, and 408 have the following equations, respectively:

When,

When,

When,

Is represented by.

From this equation, it can be seen that, for example, the phase error (β2-β1) remains in the virtual image peak represented by the second term of Fv1 ″. Further, for example, it can be seen that a phase error (β1-β2) remains in the virtual image peak indicated by the first term of Fv2 ″. Further, for example, it can be seen that a phase error (β4-β3) remains in the virtual image peak indicated by the second term of Fv3 ″. Further, for example, it can be seen that a phase error (β3-β4) remains in the virtual image peak indicated by the first term of Fv4 ″.

Example 409 of the phase difference between the virtual receiving antennas is an example of the phase difference of the signals Fv1 ″ to Fv4 ″ after the phase correction in the speed index corresponding to the peak 401. At peak 401, the absolute value of the component for which the phase error remains becomes small, so the phase difference indicated by the white circle in Example 409 of the phase difference is a straight line corresponding to the arrival direction of the component corresponding to peak 401 in the reflected wave. Asymptote to.

On the other hand, the example 410 of the phase difference between the virtual receiving antennas is an example of the phase difference of the signals Fv1 ″ to Fv4 ″ after the phase correction in the speed index corresponding to the peak 402. At peak 402, the absolute value of the component with residual phase error becomes large, so the phase difference indicated by the white circle in Example 410 of the phase difference is a straight line corresponding to the arrival direction of the component corresponding to peak 402 in the reflected wave. Deviation from.

In one example, the simple arrival direction estimation unit 1122 uses the phase-corrected signals Fv1'' to Fv4'' to reach the peak cell (nr, nv) via the FFT calculation for azimuth detection represented by the following equation. Power for each direction θ of the corresponding reflected wave component (Doppler frequency component):

And maximum power:

And ask. Here, λ (nr) represents the wavelength of the beat signal corresponding to the distance index nr, and d (i-1) represents the distance from the reference antenna position to the position of the antenna i. The simple arrival direction estimation unit 1122, for example, sets the direction θ in which the power P (θ; nr, nv) takes the maximum power Pmax (nr, nv) to the reflected wave component (Doppler frequency component) corresponding to the peak cell (nr, nv). ) Is the arrival direction (estimated arrival direction) (simple arrival direction estimation).

Example 403 of the simple arrival direction estimation result is an example of the result of performing simple arrival direction estimation in the velocity index corresponding to the peak 401 which is the real image peak. Example 404 of the simple arrival direction estimation result is an example of the result of performing simple arrival direction estimation in the velocity index corresponding to the peak 402 which is the virtual image peak. In the example 404 of the simple arrival direction estimation result corresponding to the peak 402, the maximum in the simple arrival direction estimation result 403 corresponding to the peak 401 by the residual phase error components ± (β2-β1) and ± (β4-β3). The maximum power P2 is lower than the power P1.

The difference between the maximum power P1 and the maximum power P2 depends on the magnitude (error amount) of the residual phase error components ± (β2-β1) and ± (β4-β3). In the example shown in FIG. 4, the power difference becomes maximum in the case where the error amount is 45 degrees. The design of the feeding line on the substrate may be adjusted so that the amount of error is controlled to an optimum value (for example, 45 degrees), or the phase rotating unit 102 may adjust the phase of the radar signal.

The frequency axis power comparison unit 1123 detects a real image peak corresponding to the maximum power P1 and a virtual image peak corresponding to the maximum power P2 based on the difference appearing between the maximum powers P1 and P2. Of the two maximum powers P1 and P2, the frequency axis power comparison unit 1123 detects the peak corresponding to the lower maximum power P2 as a virtual image peak. In one example, the frequency axis power comparison unit 1123 detects a peak corresponding to a lower maximum power as a virtual image peak when the difference in maximum power is larger than a predetermined threshold value. This makes it possible to prevent one peak from being mistakenly detected as a virtual image peak when a reflecting object actually exists at both peak positions.

The virtual image peak mask processing unit 1124 masks the detected virtual image peak from the output signal of the peak detection processing unit 111.

FIG. 5 is a flowchart illustrating an example of the operation of the virtual image removal processing unit 112 according to the first embodiment.

In step S102, the virtual image removal processing unit 112 initializes the candidate peak list. The candidate peak list is a list containing peak information that identifies a real or virtual peak and the corresponding maximum power. The peak information is information for specifying the peak of the signal after phase correction, and is, for example, information indicating a peak cell corresponding to the peak. In one example, the virtual image removal processing unit 112 sets a flag indicating that the peak is an unprocessed peak for all the peak cells included in the peak information generated by the peak detection processing unit 111.

In step S104, the virtual image removal processing unit 112 determines whether or not an unprocessed peak exists. In one example, the virtual image removal processing unit 112 searches for a peak cell in which a flag indicating that it is an unprocessed peak is set.

When an unprocessed peak exists (step S104: Yes), in step S106, the virtual image removal processing unit 112 sets the peak information and the corresponding maximum power for the unprocessed peak in the candidate peak list. Details of the process in step S106 will be described later with reference to FIG.

In step S108, the virtual image removal processing unit 112 determines whether or not there is an unprocessed peak at the pseudo peak generation position. Here, the pseudo peak generation position corresponds to the frequency fs / 2 (fs = 1 / Td) generated in response to the erroneous reverse phase rotation by the phase reverse rotation unit 109 with respect to the cell at the peak generation position. This is the position specified by the cell whose velocity index is shifted.

In one example, the virtual image removal processing unit 112 searches for a peak cell that is at a pseudo-peak generation position with respect to the peak cell searched in step S106 and has a flag indicating that it is an unprocessed peak. For example, the cell searched in step S108 is a cell at a pseudo peak generation position corresponding to the distance index nr and the speed index nv + Nv / 2, with respect to the distance index nr and the speed index nv of the peak cell searched in step S106. .. Here, Nv is the number of points of the speed detection FFT.

When there is an unprocessed peak at the pseudo peak generation position (step S108: Yes), in step S110, the virtual image removal processing unit 112 adds peak information and the corresponding maximum power for the pseudo peak to the candidate peak list. The details of the process in step S110 will be described later with reference to FIG. The virtual image removal processing unit 112 repeatedly executes steps S108 to S110 while the unprocessed peak at the pseudo peak generation position exists.

When there is no unprocessed peak at the pseudo peak generation position (step S108: No), in step S112, the frequency axis power comparison unit 1123 sets this peak list based on the candidate peak list. This peak list is a list used by the virtual image peak mask processing unit 1124 when masking a virtual image peak from the output signal of the peak detection processing unit 111. Details of the process in step S112 will be described later with reference to FIG. After that, the virtual image removal processing unit 112 returns the processing to step S104.

On the other hand, when there is no unprocessed peak (step S104: Yes), in step S114, the virtual image peak mask processing unit 1124 masks the virtual image peak from the output signal of the peak detection processing unit 111 based on this peak list. For example, the virtual image peak mask processing unit 1124 sets the value of the speed detection FFT calculation result Fv'in the peak cell indicated by the peak information not included in the peak list to 0. After that, the virtual image removal processing unit 112 ends the processing.

FIG. 6 is a flowchart illustrating an example of the operation of the virtual image removal processing unit 112 in steps S106 and S110 of FIG.

In step S202, the virtual image removal processing unit 112 acquires a peak. The peak is, for example, an unprocessed peak in step S106 and a pseudo-peak candidate in step S110. In one example, the virtual image removal processing unit 112 resets the unprocessed flag of the peak cell corresponding to the acquired peak.

In step S204, the phase correction unit 1121 performs phase correction processing on the peak cell corresponding to the acquired peak.

In step S206, the simple arrival direction estimation unit 1122 estimates the simple arrival direction for the data after phase correction.

In step S208, the simple arrival direction estimation unit 1122 calculates the maximum power corresponding to the direction estimated in the simple arrival direction estimation result.

In step S210, the virtual image removal processing unit 112 stores the peak information acquired in step S202 and the maximum power calculated in step S208 as a pair in the candidate peak list. After that, the virtual image removal processing unit 112 ends the processing of step S106 or step S110.

FIG. 7 is a flowchart illustrating an example of the operation of the frequency axis power comparison unit 1123 in step S112 of FIG.

In step S302, the frequency axis power comparison unit 1123 searches for the top two maximum powers from the candidate peak list.

In step S304, the frequency axis power comparison unit 1123 determines whether or not the difference between the two maximum powers searched in step S302 is larger than a predetermined threshold value.

When the difference between the two maximum powers is larger than a predetermined threshold value (step S304: Yes), in step S306, the frequency axis power comparison unit 1123 has peak information corresponding to the larger maximum power of the two maximum powers. Is added to this peak list. After that, the virtual image peak mask processing unit 1124 ends the processing in step S112.

On the other hand, when the difference between the two maximum powers is not larger than a predetermined threshold value (step S304: No), in step S308, the frequency axis power comparison unit 1123 sets all the peak information included in the candidate peak list as the current peak. Add to list. After that, the virtual image removal processing unit 112 ends the process of step S112.

According to the first embodiment, by detecting and suppressing the virtual image component caused by the interference between the orthogonal codes, it is possible to prevent the deterioration of the speed detection performance and reduce the false detection of the reflecting object. Further, by removing the virtual image component before the orientation detection processing having a relatively large amount of processing, the candidate points of the reflecting object can be reduced, and the calculation amount and power consumption of the orientation detection processing can be reduced.

(Other embodiments)
In the first embodiment described above, the anti-phase rotating unit 109 rotates the distance detection FFT signal in anti-phase rotation. Instead of this, an embodiment in which the phase reverse rotation unit 109 performs the reverse phase rotation on the signal before the distance detection FFT process is also conceivable. In other words, an embodiment in which the order of the distance detection processing unit 108 and the phase reverse rotation unit 109 is reversed can be considered.

In the above-described embodiment, the notation "... part" used for each component is "... circuitry", "... device", "... unit", or "... unit". It may be replaced with another notation such as "... module".

Although various embodiments have been described above with reference to the drawings, it goes without saying that the present disclosure is not limited to such examples. It is clear that a person skilled in the art can come up with various modifications or modifications within the scope of the claims, which naturally belong to the technical scope of the present disclosure. Understood. Moreover, each component in the said embodiment may be arbitrarily combined within the range which does not deviate from the purpose of disclosure.

This disclosure can be realized by software, hardware, or software linked with hardware. Each functional block used in the description of the above embodiment is partially or wholly realized as an LSI which is an integrated circuit, and each process described in the above embodiment is partially or wholly. It may be controlled by one LSI or a combination of LSIs. The LSI may be composed of individual chips, or may be composed of one chip so as to include a part or all of functional blocks. The LSI may include data input and output. LSIs may be referred to as ICs, system LSIs, super LSIs, and ultra LSIs depending on the degree of integration. The method of making an integrated circuit is not limited to LSI, and may be realized by a dedicated circuit, a general-purpose processor, or a dedicated processor. Further, an FPGA (Field Programmable Gate Array) that can be programmed after the LSI is manufactured, or a reconfigurable processor that can reconfigure the connection and settings of the circuit cells inside the LSI may be used. The present disclosure may be realized as digital processing or analog processing. Furthermore, if an integrated circuit technology that replaces an LSI appears due to advances in semiconductor technology or another technology derived from it, it is natural that the functional blocks may be integrated using that technology. There is a possibility of applying biotechnology.

This disclosure can be implemented in all types of devices, devices, and systems (collectively referred to as communication devices) having communication functions. Non-limiting examples of communication devices include telephones (mobile phones, smartphones, etc.), tablets, personal computers (PCs) (laptops, desktops, notebooks, etc.), cameras (digital stills / video cameras, etc.). ), Digital players (digital audio / video players, etc.), wearable devices (wearable cameras, smart watches, tracking devices, etc.), game consoles, digital book readers, telehealth telemedicines (remote health) Care / medicine prescription) devices, vehicles with communication functions or mobile transportation (automobiles, airplanes, ships, etc.), and combinations of the above-mentioned various devices can be mentioned.

Communication devices are not limited to those that are portable or mobile, but are not portable or fixed, any type of device, device, system, such as a smart home device (home appliances, lighting equipment, smart meters or It also includes measuring instruments, control panels, etc.), vending machines, and any other "Things" that can exist on the IoT (Internet of Things) network.

Communication includes data communication using a combination of these, in addition to data communication using a cellular system, wireless LAN system, communication satellite system, etc. Communication devices also include devices such as controllers and sensors that are connected or connected to communication devices that perform the communication functions described in the present disclosure. For example, it includes controllers and sensors that generate control and data signals used by communication devices that perform the communication functions of the communication device.

Communication devices also include infrastructure equipment that communicates with or controls these non-limiting devices, such as base stations, access points, and any other device, device, or system. ..

In the radar device according to the present disclosure, a plurality of radar signals, each of which is periodically subjected to the first phase rotation for each of the plurality of transmitting antenna elements and has a phase difference corresponding to the transmitting antenna element, are produced. The transmission circuit transmitted from the transmission antenna element of the above, the reception circuit for receiving the reflected wave signal reflected by the plurality of radar signals in the detection target by the plurality of reception antenna elements, and the reception circuit received by the plurality of reception antenna elements. For each of the reflected wave signals, a second phase rotation of a rotation amount corresponding to the first phase rotation and the phase difference, which is a phase rotation opposite to the first phase rotation of each transmitting antenna element, is performed. Among the Doppler frequency components remaining in the given signal, the removal circuit for removing the Doppler frequency component whose power in the estimated arrival direction of the Doppler frequency component is relatively reduced according to the phase difference is provided.

The radar device according to the present disclosure includes a detection circuit that detects the direction of the detection target based on the signal output by the removal circuit.

In the radar device according to the present disclosure, the difference between the Doppler frequency corresponding to the Doppler frequency component removed by the removal circuit and the Doppler frequency corresponding to any of the remaining Doppler frequency components is a predetermined frequency. is there.

In the radar device according to the present disclosure, the removal circuit removes the Doppler frequency component whose power is relatively small when the difference in power is larger than a predetermined threshold value.

In the radar device according to the present disclosure, the plurality of transmitting antenna elements and the plurality of receiving antenna elements constitute a virtual receiving array antenna.

In the radar device according to the present disclosure, the first phase rotation is performed based on patterns different from each other among the plurality of transmitting antenna elements.

In the radar device according to the present disclosure, the different patterns constitute an orthogonal code.

In the radar device according to the present disclosure, when the bit value included in the orthogonal code is 0, the rotational phase amount of the first phase rotation is 0 radians, and the bit value included in the orthogonal code is 1. The rotational phase amount of the first phase rotation is π radians.

In the radar device according to the present disclosure, the radar signal is a periodic chirp signal.

In the signal processing method of the radar signal according to the present disclosure, a plurality of radar signals which are periodically subjected to the first phase rotation for each of the plurality of transmitting antenna elements and have a phase difference according to the transmitting antenna element are processed. A process of transmitting from the plurality of transmitting antenna elements, a process of receiving the reflected wave signal reflected by the plurality of radar signals in the detection target by the plurality of receiving antenna elements, and a process of receiving the reflected wave signals by the plurality of receiving antenna elements, respectively. Each of the reflected wave signals has a phase rotation opposite to that of the first phase rotation of each transmitting antenna element, and a second phase of a rotation amount corresponding to the first phase rotation and the phase difference. Among the Doppler frequency components remaining in the rotated signal, the process of removing the Doppler frequency component in which the power in the estimated arrival direction of the Doppler frequency component is relatively reduced according to the phase difference is included.

The disclosures of the specifications, drawings and abstracts contained in the Japanese application of Japanese Patent Application No. 2019-064928 filed on March 28, 2019 are all incorporated herein by reference.

This disclosure is useful for radar systems.

100 Transmission control unit 101 Transmission signal generation unit 102 Phase rotation unit 103 Transmission high frequency unit 104 Transmission array antenna 105 Reception array antenna 106 Reception high frequency unit 107 ADC
108 Distance detection processing unit 109 Phase reverse rotation unit 110 Speed detection processing unit 111 Peak detection processing unit 112 Phantom image removal processing unit 1121 Phase correction unit 1122 Simple arrival direction estimation unit 1123 Frequency axis power comparison unit 1124 Phantom image peak mask processing unit 113 Direction detection Processing unit 114 Object detection processing unit

Claims (10)

1. Transmission in which a first phase rotation is periodically performed for each of the plurality of transmitting antenna elements, and a plurality of radar signals having a phase difference corresponding to the transmitting antenna element are transmitted from the plurality of transmitting antenna elements, respectively. Circuit and
   A receiving circuit that receives the reflected wave signal reflected by the plurality of radar signals in the detection target by the plurality of receiving antenna elements, and
   Each of the reflected wave signals received by the plurality of receiving antenna elements has a phase rotation opposite to that of the first phase rotation of each transmitting antenna element, and the first phase rotation and the phase difference are obtained. Of the Doppler frequency components remaining in the signal given the second phase rotation of the corresponding rotation amount, the Doppler frequency component in which the power in the estimated arrival direction of the Doppler frequency component is relatively reduced according to the phase difference is used. The removal circuit to remove and
   A radar device equipped with.
2. A detection circuit for detecting the direction of the detection target based on the signal output by the removal circuit is provided.
   The radar device according to claim 1.
3. The difference between the Doppler frequency corresponding to the Doppler frequency component removed by the removal circuit and the Doppler frequency corresponding to any of the remaining Doppler frequency components is a predetermined frequency.
   The radar device according to claim 1 or 2.
4. The removal circuit removes the Doppler frequency component, which has a relatively small power, when the difference in power is greater than a predetermined threshold.
   The radar device according to claim 1.
5. The plurality of transmitting antenna elements and the plurality of receiving antenna elements constitute a virtual receiving antenna array.
   The radar device according to any one of claims 1 to 4.
6. The first phase rotation is performed between the plurality of transmitting antenna elements based on patterns different from each other.
   The radar device according to any one of claims 1 to 5.
7. The different patterns constitute an orthogonal sign,
   The radar device according to claim 6.
8. When the bit value included in the orthogonal code is 0, the rotation phase amount of the first phase rotation is 0 radians, and when the bit value included in the orthogonal code is 1, the rotation phase amount of the first phase rotation The amount of rotational phase is π radians,
   The radar device according to claim 7.
9. The radar signal is a periodic chirp signal.
   The radar device according to any one of claims 1 to 8.
10. A process in which a plurality of radar signals that are periodically subjected to the first phase rotation for each of the plurality of transmitting antenna elements and have a phase difference corresponding to the transmitting antenna elements are transmitted from the plurality of transmitting antenna elements. When,
    The process of receiving the reflected wave signal reflected by the plurality of radar signals in the detection target by the plurality of receiving antenna elements, and
    Each of the reflected wave signals received by the plurality of receiving antenna elements has a phase rotation opposite to that of the first phase rotation of each transmitting antenna element, and the first phase rotation and the phase difference are obtained. Of the Doppler frequency components remaining in the signal given the second phase rotation of the corresponding rotation amount, the Doppler frequency component in which the power in the estimated arrival direction of the Doppler frequency component is relatively reduced according to the phase difference is used. The process to remove and
    Signal processing methods for radar signals, including.

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