WO2007136747A2

WO2007136747A2 - Closely coupled antennas for supergain and diversity

Info

Publication number: WO2007136747A2
Application number: PCT/US2007/011902
Authority: WO
Inventors: Yuanxun Ethan Wang
Original assignee: The Regents Of The University Of California
Priority date: 2006-05-18
Filing date: 2007-05-17
Publication date: 2007-11-29
Also published as: WO2007136747A3

Abstract

Parallel information channels are discovered on dipole antennas closely coupled to each other. There channels are supported by different radiation modes in the form of multipoles or combination of multipoles. The spacing between antennas only impacts the radiation impedances of the higher order modes. Through a multiport impedance matching network, these modes can be separated into different driving ports and matched individually. This finding can lead to practical superdirective arrays for beamforming and superdiverse antennas for MIMO applications. A pair of closely coupled dipole antennas is used as an example to illustrate this concept. Analysis has shown, with a pair of half-wave dipoles at 0.1 λ spacing, double of the channel capacity resulted from the diversity increase and more than 3 times of the gain and directivity of a single half-wave dipole can be achieved. The theory can be extended for a large number of closely coupled antenna elements.

Description

CLOSELY COUPLED ANTENNAS FOR SUPERGAIN AND DIVERSITY

CROSS-REFERENCE TO RELATED APPLICATIONS This application claims the benefit under 35 U.S.C. Section 119(e) of co- pending U.S. provisional patent application, serial number 60/801,635, filed May 18, 2006, entitled "CLOSELY COUPLED ANTENNAS FOR SUPER DIRECTIVITY AND DIVERSITY," by Yuanxun E. Wang, which application is incorporated by reference herein.

BACKGROUND OF THE INVENTION

1. Field of the Invention.

This invention is related to antennas, and, in particular, to closely coupled antennas for supergain and diversity.

2. Description of the Related Art.

(Note: This application references a number of different publications as indicated throughout the specification by one or more reference numbers within brackets, e.g., [x]. A list of these different publications ordered according to these reference numbers can be found below in the section entitled "References. " Each of these publications is incorporated by reference herein.)

Wireless communications based on multiple antennas such as beamformiπg and Multiple Input Multiple Output (MIMO) systems have received great attention recently as they promise significant improvements in information capacity without occupying extra frequency spectrums [1-3]. This is achieved by taking advantage of the parallel information channels existing in multiple antennas. One limitation in the physical layer of these systems is the antenna spacing is normally required to be greater than 0.5λ in order to avoid the mutual coupling and the spatial correlation among the parallel information channels. During the past, a number of researches have been carried out to study the mutual coupling and spatial correlation effects and their impacts on the system performance [4-10]. It is commonly believed these effects will degenerate the number of information channels when the antennas are pushed to be really close to each other. Possible ways to combat the mutual coupling and effects in closely spaced antennas have been studied [11-19]. Signal processing approaches [11, 12] can correct the signal errors due to the coupling. However, they do not improve the system capacity performance since no improvements are made to the system's signal to noise ratio. Others have proposed hardware approaches to reduce mutual coupling [13-17]. However, mutual coupling and correlation between two antennas occur through the free space and are related to the non-orthogonality of the radiation patterns of the antennas. It is part of the intrinsic property of the radiation physics and can not be physically removed. Based on the network theory, Wallace and Jensen proposed an intriguing approach [18, 19] that can optimize a multi-antenna system with the mutual coupling and correlation effects with an optimum, lossless multiport matching network. The so-called optimum Hermitian matching network can maximize the signal power received by the coupled antennas and decorrelate the received signals at the outputs of the network. However, the analysis in [18, 19] did not lead to a viable realization scheme of such an optimum matching network. Nor did it provide physical insights on the full potential of the system performance in terms of diversity and directivity with closely coupled antennas. On the other hand, Anderson and Rasmussen proposed a practical transmission line network to decorrelate antennas with purely imaginary mutual impedances in his pioneering work [22]. Several recent papers further advance this approach to decouple and decorrelate coupled antenna pairs with either transmission line or capacitive decouplers [23-26]. In [26], it is pointed out that a 180 degree hybrid can be used to decouple a pair of dipole antennas since it separates the common mode and difference mode. However, the mode dependent impedance behavior that can seriously affects the performance of the system is not considered in their approach. Furthermore, a systematic way of using the decoupled antennas for supergain applications is not yet known until the disclosure of the present invention.

SUMMARY OF THE INVENTION To overcome the limitations in the prior art described above, and to overcome other limitations that will become apparent upon reading and understanding the present specification, the present invention discloses an antenna system. A system in accordance with the present invention comprises a plurality of antenna elements, each element in the plurality of antenna elements is excited jointly with other elements to form a common mode of excitation and a difference mode of excitation, or combination of common and difference modes in different orientations if the total number of antennas is more than two. A passive network coupled to the plurality of antenna elements, the other end of the passive network having a plurality of input/output ports and each of the ports corresponding to a mode of excitation in the plurality of antenna elements; the ports being the inputs and outputs of multiple information channels that are independent from each other. An impedance matching circuit being coupled to each port of the passive network maximizing the power of that channel.

The passive network being one or more 180 degree hybrid couplers, which can be a ring coupler, a coupled line coupler, a transformer coupler or a tapered line coupler, the information channels carry different information and are used by the antenna system to increase diversity, the information channels are combined coherently to increase the gain and directivity of the antenna system, the information channels can selectively be combined or separated to selectively increase gain and increase diversity, the antenna system being used in a wireless communications network, and the antenna system being used in a radio or TV receiver, the antenna system being used in a radar, the antenna system being used in a sensor or sensor network. BRIEF DESCRIPTION OF THE DRAWINGS

Referring now to the drawings in which like reference numbers represent corresponding parts throughout:

FIG. 1 illustrates a closely coupled antenna and the common mode and difference mode excitations of the closely coupled antenna;

FIG. 2 illustrates the common mode and difference mode radiation patterns of directed dipole antennas;

FIG. 3 illustrates the eigenvalues of the channel matrix for a MIMO link comprisin ccoupled dipole pairs in accordance with the present invention; FIG. 4 illustrates a capacity plot for the MIMO link of a closely coupled antenna ir accordance with the present invention;

FIG. 5 illustrates equivalent T networks in different modes for a two-port antenna i accordance with the present invention;

FIG. 6 illustrates a block diagram for the antenna excitation through a hybrid in accordance with the present invention;

FIG. 7 illustrates radiation impedance frequency responses of antennas in accordar with the present invention;

FIG. 8 illustrates simulated S-parameter frequency response for an antenna in accordance with the present invention; FIG. 9 illustrates a graph of directivity versus power ratio for a coherently combined radiation mode of an antenna in accordance with the present invention; and FIG. 10 illustrates a radiation pattern of a coupled half- wave dipole antenna in accordance with the present invention.

DETAILED DESCRIPTION OF THE INVENTION

In the following description of the preferred embodiment, reference is made to the accompanying drawings which form a part hereof, and in which is shown by way of illustration a specific embodiment in which the invention may be practiced. It is to be understood that other embodiments may be utilized and structural changes may be made without departing from the scope of the present invention.

Overview The present invention allows antenna systems to achieve much stronger transmissions and receptions of electromagnetic signals by using multiple closely coupled antennas deployed in a limited space. This can result in significant increases in power of transmission and reception in a line of sight environment or in data capacity in environments with rich scattering and multipaths. The present invention uses the parallel information channels of multiple radiation modes formed in specially arranged closely coupled antennas. The excitation to each antenna can be controlled individually. With certain arrangement of excitations, the antennas can be combined to support one radiation mode and suppress the others. A simple example is a pair of coupled dipoles aligned in parallel to each other. By dividing the common mode and difference mode in the current excitations to these two antennas, two parallel information channels can be utilized.

These two channels can be used to either double the data capacity if the information channels are used to carry different information, or triple the directivity of the antenna system if the information channels are combined coherently to carry the same information. In order to separate these two independent information channels from the outputs of the two antennas, a multiport passive network is used to decouple the signals from the two antennas.

One way of implementing this multiport network is to use a 180 degree hybrid, which can be realized in a number of ways such as ring couplers, coupled line couplers, transformer couplers or tapered line couplers. The outputs of such a network correspond to the independent radiation modes existing in the coupled antennas. The capacity of each information channel needs to be maximized by adding an impedance matching circuit dedicated to that particular channel. A preferred way to implement the invention is to develop densely packed antenna arrays for handheld or any other wireless applications which are space conscious. These arrays serve two purposes depending on the application scenarios. The first purpose is to maximize the diversity performance communication in rich scattering environments. This can help to increase the data rate of Multiple-Input Multiple-Output (MIMO) systems with space-time coding techniques. One example of commercial applications is the Wireless LAN at home.

The second purpose is to maximize the antenna gain in relatively clean communication environments with less scattering and obstructions between the users and the base stations. The antenna arrays resulted from the invention can be made very compact but with enhanced transmission efficiency and reception sensitivity through beamforming techniques. This can find good use in mobile phone networks. Both purposes can be realized with the same hardware platform and they can bring significant advantages in performance, robustness and power usages of the wireless system.

The present invention exploits the full potential of parallel information channels on multiple antennas. The present invention allows for better directivity and diversity performance over current antenna systems. As such, with a number of specially arranged antennas and a decoupling network, the present invention increases information capacity and antenna gain by an order of magnitude compared to what can be achieved with existing practices.

Approach of the Present Invention

The approach of the present invention relates the impedance behavior to the radiation physics of closely coupled antennas for a clear physical explanation of the coupling mechanism. The present invention demonstrates that the mutual coupling can be utilized to realize multi-antenna systems with supergain and diversity by providing parallel non-degenerating information channels. In order to fully exploit those channels in closely coupled antennas, one can no longer separate them onto different antennas. Instead, the antennas have to be considered as a whole and its radiation behavior can be represented by the well known multipole expansions in classical electromagnetic theory [27]. It is found that the current excitations in closely coupled dipoles can be arranged and combined to excite the radiations of different multipoles alone [27].

As the radiation patterns of those multipoles are orthogonal to each other, each of them can support an independent information channel. In rich scattering scenarios, one can use these channels to transmit or receive different channels of information, which is the concept of pattern diversity. On the other hand, a supergain beamforming array can also be obtained if these multipoles are excited in a coherent way.

Here the property of a closely coupled dipole pair is studied as the simplest example of multipolar antennas. It is well known that such an antenna pair can support two radiation modes, e.g. the common mode and the difference mode. From the perspective of multipolar radiations, the common mode corresponds to the radiation of an electric dipole while the radiation of difference mode can be considered as from the combination of a magnetic dipole and an electric quadruple [27]. As the higher order multipoles usually have much smaller radiation resistances than the fundamental dipole mode, the radiation resistance of the difference mode can be much smaller than that of the common mode. This raises two issues in practice. First, these multipolar modes have to be separated from the antenna ports and matched independently, which requires a multiport impedance matching network. The conventional one to one conjugate match to the antennas will not work as the energy of the multipoles will be lost because of mismatch. In fact, this leads to the same conclusion as in [18, 19]. Particular to the coupled dipole pair, a 180 degree hybrid based on either rat-race coupler or magic T is proposed to serve the purpose to separate the common and difference modes, which can also be considered as a simple realization scheme of the optimum Hermitian matching networks predicted in [18, 19]. Second, since the radiation resistances of the higher order multipoles become very small, low-loss designs of antennas and impedance matching circuits are necessary in order to push the antennas to be really close to each other or to use very high order of multipoles. The integration of antennas with transmitters may be preferable in order to minimize the loss. Other issues include the elevated Q value for higher order multipolar antennas [29], which may limit them for narrow band operations only.

//. Multipoles and Pattern Orthogonality:

FIG. 1 illustrates a closely coupled antenna and the common mode and difference mode excitations of the closely coupled antenna of the present invention.

FIG. 1 illustrates two Hertzian dipoles, dipole A 102 and dipole B 104, parallel to each other and pointing to the z-axis, excited by two independent sources Pl 106 and P2 108. The length of the dipoles 102 and 104 is length / 110. The spacing between the two antennas 102 and 104 is distance d 1 12, which is small comparing to the wavelength to allow tight coupling between dipoles 102 and 104.

The currents flowing on these two dipoles 102 and 104 can thus be decomposed into a common mode 114 and a difference mode 116 as shown in FIG. 1, although these modes are typically combined in normal transmission and are decomposed upon reception of the common mode 114 and difference mode 116. The radiation in the common mode 114 is dominated by the first order Taylor expansion of the integral in (1), which is the electric dipole mode according to the multipole theory [27]. In the difference mode 116, the first order radiation is cancelled as the currents flow in opposite directions and the second order expansion of the radiated field becomes dominant which consists of a magnetic dipole and an electric quadruple as shown in [27],

FIG. 2 illustrates the common mode and difference mode radiation patterns of directed dipole antennas of the present invention.

Therefore, the common mode 114 and the difference mode 116 should have orthogonal radiation patterns according to the property of the multipoles. The common mode radiation pattern 200 and difference mode radiation pattern 202 of z- axis oriented dipoles 102 and 104 are drawn in FIG.2.

The orthogonality can be analytically verified between any two modes or polarizations, which is in the form of

<$P^~_k . PΪT^~ _k ^'}tΩ = 0 (1)

where the ij = cm, dm; k=x,y,z; and Ω is the solid angle. The property of pattern orthogonality is very important, since it assures that the channels will not degenerate for co-located multipolar antennas. The radiation resistances for both modes can thus be calculated by integrating the radiation patterns over all the angles, which yields

The above formulas show that when the antenna spacing distance d 112 reduces, the radiation resistance also reduces according to a square relationship for the difference mode 116. For even higher order multipoles, the radiation resistance is expected to drop even faster. However, for legitimately small antenna spacings 112 in the order of 0.1 λ, the radiation resistance of the difference mode 116 is not outrageously small comparing to that of the common mode 114 and can be used if desired. For example, in the case /=£/=0.1 λ, the radiation resistance is 7.9Ω for the common mode 114 and 1.0Ω for the difference mode 116. Other spacings, which approximate the O.lλ spacing, can also be used in accordance with the present invention.

FIG. 3 illustrates the eigenvalues of the channel matrix for a MIMO link comprising coupled dipole pairs in accordance with the present invention, and FIG. 4 illustrates a capacity plot for the MIMO link of a closely coupled antenna in accordance with the present invention. The orthogonal patterns of the multipoles can be used for pattern diversity in a straightforward way if they are driven independently. Considering a rich scattering environment with uniform distribution in full angular spread, it is easy to prove that the common mode 114 and the difference mode 116 offer equally good information channels which can maximize the diversity gain of the MIMO system. To examine the performance under different angular spread, simulations are carried out for a MEMO link.

The receiver is assumed to be surrounded by scatterers uniformly distributed on a spherical surface. The transmitter illuminates the sphere with a certain angular spread. Both the transmitter and the receiver are coupled O.lλ long dipole pairs aligned in the end-fire direction. The spacing between the coupled dipoles is O.lλ. With the assumption of equal power excitation between the two modes, FIG.3 shows two comparable eigenvalues 300 and 302 in the normalized channel matrix which correspond to contributions from both the common mode 114 and difference mode 116. These two values 300 and 302 approach each other as expected when the angular spread increases.

The system capacity is shown in FIG.4. A 1x1 Rayleigh channel 400, 2x2 Rayleigh channel 402, common mode and difference mode 404, and 3x3 Rayleigh channel are shown. The common mode and difference mode 404 of the present invention almost exactly coincides with that of a 2x2 Rayleigh fading channel, which is the optimal case in rich scattering environments. It should be noted that though the simulations are carried out for coupled dipoles at 0.1 λ spacing and length, the results should be independent of the spacing 112 and the length 110 of the antennas 102 and 104, once the antennas 102 and 104 are small and close enough to be tightly coupled. This independence of the results versus the spacing 112 and length 110 is based on the constant patterns of multipolar antennas. ///. Passive Mode Decomposition Network:

FIG. 5 illustrates equivalent T networks in different modes for a two-port antenna in accordance with the present invention.

The crucial step of utilizing the parallel channels provided by multipoles in the coupled antennas is a multiport impedance matching network that can drive the different radiation modes independently. In fact, multiple radiation modes always exist in almost any N-element arrays, regardless coupled or not. However, there may not be a simple and easy way to separate them like those in multipoles. Particularly for the coupled dipole pair, a 3dB 180 degree hybrid can separate the common and difference mode and match them respectively [30]. In order to verify this statement, an equivalent T network can be derived from method of moments to represent the general coupling behavior of the antennas, which is shown in FIG.5.

T network 500 is a model of the two-port antenna, and as discussed herein, is operated in two different modes; an even mode and an odd mode. The even mode equivalent network 502 results in the common mode 114 excitation, and the odd mode 504 equivalent network results in the difference mode 116 excitation of antennas 102 and 104.

By performing the even and odd mode analysis, the impedance matrix of the antenna [Z_onl] can be defined as follows, 1 ,-.

J where R_CM. R D_M are the common-mode and difference mode radiation resistance. If R_CM=R_DM, the non-diagonal terms in the matrix will become zero. From (9) it is evident that the difference in radiation resistances between the two modes is the cause of the mutual coupling. The two-port impedance matrix can be transformed to two- port S parameters [-?„„,]. The matching network realized by a 3dB 180 degree hybrid has the following scattering parameters:

FIG. 6 illustrates a block diagram for the antenna excitation through a hybrid in accordance with the present invention.

Common mode signal 600 and difference mode signal 602 are generated by hybrid module 604. As discussed above, the preferred hybrid module 604 is a 180 degree hybrid module 604, but other hybrid modules can be used without departing from the scope of the present invention. Typically, the 180 degree hybrid is a ring coupler, but coupled line couplers, transformer couplers, tapered line couplers, or other couplers can be used without departing from the scope of the present invention. The inputs of hybrid module 604 are power splitting ports 606 and 608, which are coupled to the coupled antennas 610.

So, in practice, coupled antennas 610 receive both common mode 114 and difference mode 116 signals, and passes these signals 606 and 608 to hybrid module 604, which decomposes signals 606 and 608 into signals 600 and 602. Similarly, the operation can be run in reverse to combine signals 600 and 602 into a single signal that is transmitted by antenna 612 if signals 600 and 602 are generated by a transmitter.

If the two power splitting ports are connected to the antenna ports as shown in FIG.6, the common mode and difference mode are obtained separately from the common port or the difference port of the hybrid module 604. This is shown by decomposing the above four-port hybrid matrix into two 2x2 matrix equations,

At the same time, the S-parameters [S_anι] of the antenna is defined as,

v; "11 "12

O C₂, O C₂₂ (6)

The two-port S-parameters for Σ port and Δ port can now be derived from the following matrix equation.

From (14), one can see that the original coupled [S_ant] matrix is actually diagonalized through the linear transformations defined by the two hybrid coupler matrices. Therefore, the impedance matrix of two driving ports are finally obtained in a diagonal form,

where Z_ΣΣ = and Z Δ_AΔ_A =

2/2, CM 2R DM From the above derivation, one can see the 180 degree hybrid indeed decouples the common mode and difference mode from the antenna ports through a diagonalization process. This is also consistent with the orthogonality of the radiated power of these two modes. Now if the two driving ports are conjugately matched to different impedances through matching circuits 601 and 603, there will be no mismatch in the system. Therefore, the received power is maximized in a way similar to what is described in [18, 19]. IV. Closely Coupled Half-Wave Dipole Pair:

As the short dipoles are difficult to drive in general, a more practical example is a closely coupled half-wave dipole pair. Similar to the short dipoles, the coupled half-wave dipoles can also be decomposed into the common mode and the difference mode.

The sum and difference patterns of coupled half- wave dipoles are similar to the common mode and difference mode patterns of Hertzian dipoles in FIG.2 except their main lobes in θ plane are narrower. One can also derive the orthogonal property between these two modes. To validate the theory, the radiation resistances for common and difference modes are obtained through both analytical approaches and numerical simulations based on Ansoft HFSS. In simulations, the antennas are chosen to be 150mm long, which is half-wavelength at IGHz. The radius of each dipole is lmm. The Z impedance matrix is first generated and the parameters are then converted to the radiation impedances. The impedance result varies according to different values of antenna spacing, which is listed in Table 1.

Table 1. Radiation Resistances from analyses and simulations

From Table 1, one can see that the reactive components of the impedances that were missing in the analytical formulas can also be found through numerical simulations. The resistive parts of the impedances agree reasonably well between the analytical approaches and the simulations especially for small antenna spacing. The radiation impedances are then converted to the port impedances looking from the 180 degree hybrid inputs with 50Ω characteristic impedance, as shown in Table 2.

The difference port impedances obtained from the analytical approach are surprisingly high because of the reactance are not included in the analysis. The numerical simulations have shown more reasonable results. It is noticed that the resistive parts in the port impedances are very similar between the common mode and the difference mode. However, there are significant amount of reactive components in

Table 2. Driving port impedances from analyses and simulations the difference mode port impedances which have to be compensated when designing the matching circuit.

FIG. 7 illustrates radiation impedance frequency responses of antennas in accordance with the present invention, and FIG. 8 illustrates simulated S-parameter frequency response for an antenna in accordance with the present invention.

To further study the stability of the impedance match when the frequency changes, the antenna radiation impedances versus frequency for £7=0. lλ are plotted in FIG.7 for both modes. Graphs 700 and 702 are shown; graph 700 illustrates the common mode 114 resistance 704 and reactance 706 components of impedance versus frequency, while graph 702 shows the difference mode 1 16 resistance 708 and reactance 710 components of impedance versus frequency.

While the common mode 114 behaves more like an ordinary half- wave dipole as shown in graph 700, the reactive component 710 has a more dramatic change versus frequency in the difference mode 116, shown in graph 702. This implies a much higher Q value for this mode. It has been well understood that higher-order multipoles are associated with higher Q values [29]. To better estimate the Q value and the bandwidth of impedance match for this particular case, the driving ports of the hybrid are matched to the resonance point of the difference mode 116, which is slightly lower than 1 GHz.

The return loss of the matching for both common 114 and difference 116 modes are plotted in FIG. 8. Graph 800 shows return loss 802 for the difference mode 116 and return loss 804 for the common mode 114. The difference mode return loss 802 shows a much narrower bandwidth. The figure 800 shows about 2.1% of 1OdB return loss bandwidth and approximately a Q value of 15.8 for the difference mode 116.

Another practical consideration determines whether this scheme can be useful is the radiation efficiency when the ohmic loss of the antennas is counted. The ohmic

loss of a thin wire is given by R_ohmic = where a is the radius of the wire

antenna, σ_c and μ_* are the conductivity and permeability of the conductor and/is the operating frequency. Since the current distribution on the half-wavelength dipoles are approximately sinusoidal and two wires are placed in parallel for the coupled dipoles, the Roh_mi_c is further reduced by a factor of 4. Hence, the radiation efficiency is given by e_r the efficiencies for

half-wave dipoles made of copper are e_/-_icw⁼99.93% and e_ri£>Λ_/=98.36%. V. Supergain Array:

Supergain arrays have been studied for the past half century. Although people have concluded that any directivity can be realized with a densely spaced linear array [31 ], only minor improvements in antenna gain seems to be realizable mainly because the mutual coupling effects in those closely spaced antennas eventually corrupt the impedance matching and degrade the efficiency [32]. There are also disadvantages of extremely high current requirements and high sensitivities in conventional superdirective antennas [31]. It should be noted the traditional superdirective array synthesis techniques assume a constant impedance termination for every antenna [33] the mutual coupling effect is not included in their models. This, in fact, diminishes the contributions of higher order radiation modes and exaggerates the difficulty of realizing a high gain superdirective antenna.

From the analysis of the above coupled half-wave dipoles at d=0Λ\ the directivity of the sum pattern and the difference pattern are calculated to be

respectively. It is obvious that the difference mode alone can achieve more directivity than two uncoupled dipole antennas but with a much smaller antenna separation. In fact, it is not surprising that superdirective antennas can be realized with a tightly coupled antenna as the Yagi-Uda antenna is a perfect example of that [34]. However, with the understanding of the mutual coupling effects in the form of multipoles, the potential of achieving superdirectivity from a given boundary can be fully exploited. One may notice from the radiation patterns that the two radiation modes can be added coherently in the endfire direction. As these two modes can be independently driven with the 180 degree hybrid, the optimal directivity can be achieved through a certain ratio of power and phase matching between these two modes, like that in a beamforming antenna. Based on the power orthogonality, the end-fire directivity from the combination of the common and difference modes can be characterized by the following equation, _D

where P C_M, P D_M are the radiated power of the common mode and difference mode, and η is the ratio of radiated power between two modes, e.g.

The maximum directivity Z)_mαc=Z)c_Λ/ ⁺-θ_£>Λr⁼5.6(7.48dBi) is achieved when

This corresponds to a current amplitude ratio of approximately 1 :8 between the common mode and the difference mode.

FIG. 9 illustrates a graph of directivity versus power ratio for a coherently combined radiation mode of an antenna in accordance with the present invention, and FIG. 10 illustrates a radiation pattern of a coupled half- wave dipole antenna in accordance with the present invention.

The relationship between different power ratios and the combined directivity is shown in graph 900 of FIG.9, and the radiation pattern for the maximum directivity is plotted in graph 1000 of FIG.10. It should be noted that unlike conventional Hanson- Woodyard antennas [25], the proposed coupled dipole pair is indeed a "supergain" antenna as there is no impedance mismatch in the system. The only factor affects the efficiency is the ohmic loss of the antenna. Plugging in the efficiency numbers calculated in the last section, one can find the maximum achievable gain for the coupled half-wave dipole made of copper at 0.1 λ spacing to be 5.53 (7.42dBi).

The antenna system described herein has several uses, such as a wireless communications network, a radio or television receiver, a radar system, or sensor networks. Further, the antenna system described herein has other applications, e.g., wherever the information channels of the common and difference modes are combined or separated to increase the gain and directivity or diversity of the system.

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Conclusion

The parallel channels in a closely coupled antenna pair are investigated and it is found that the radiation behaviors can be best characterized by different radiation modes in the form of multipoles. Instead of trying to reduce the mutual coupling and separate the signals on each antenna like in conventional approaches, it is proposed to consider the information channels carried by the different radiation modes, particularly common and difference modes for the dipole pair. These modes are orthogonal to each other in radiation patterns. A 3dB 180 degree hybrid is used to act as an optimum multiport impedance matching network to separate these two modes and to match them individually. A practical example made of a coupled half- wave dipole pair demonstrates that it not only can increase the data capacity in a MIMO system but also can form a supergain antenna with 5.3 dB higher gain than an ordinary dipole with a moderate current ratio and decent bandwidth.

This concludes the description of the preferred embodiment of the present invention. The foregoing description of one or more embodiments of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not by this detailed description, but rather by the claims appended hereto.

Claims

WHAT IS CLAIMED IS:

1. An antenna system, comprising: a plurality of antenna elements, each element in the plurality of antenna elements is excited jointly with other elements to form a common mode of excitation and a difference mode of excitation; a passive network, coupled to the plurality of antenna elements through a plurality of ports of the passive network, each port of the plurality of ports corresponding to a mode of excitation in the plurality of antenna elements; wherein each port in the plurality of ports is an information channel independent of each other port in the plurality of ports; and a plurality of impedance matching circuits coupled to the passive network, each circuit providing a conjugate match to an input impedance of each port of the passive network coupled to the plurality of antenna elements.

2. The antenna system of claim 1, wherein the passive network is a 180 degree hybrid coupler.

3. The antenna system of claim 2, wherein the 180 degree hybrid is a ring coupler.

4. The antenna system of claim 2, wherein the 180 degree hybrid is a coupled line coupler.

5. The antenna system of claim 2, wherein the 180 degree hybrid is a transformer coupler.

6. The antenna system of claim 2, wherein the 180 degree hybrid is a tapered line coupler.

7. The antenna system of claim 1, wherein each information channel carries different information and is used by the antenna system to increase diversity.

8. The antenna system of claim 1, wherein each information channel is combined coherently by the antenna system to increase gain and directivity of the antenna system.

9. The antenna system of claim 1, wherein each information channel is selectively combined by the passive network to selectively increase gain and diversity of the antenna system.

10. The antenna system of claim 1, wherein the antenna system is used in a wireless communications network.

11. The antenna system of claim 1, wherein the antenna system is used in a radio or TV receiver.

12. The antenna system of claim 1, wherein the antenna system is used in a radar system.

13. The antenna system of claim 1, wherein the antenna system is used in a sensor network.

14. The antenna system of claim 1, wherein the plurality of antenna elements are closely coupled.

15. The antenna system of claim 14, wherein the closely coupled antenna elements have a spacing of approximately O.lλ.