WO2007031691A1

WO2007031691A1 - Device and method for configuring a receiver in a communication device

Info

Publication number: WO2007031691A1
Application number: PCT/FR2006/050884
Authority: WO
Inventors: Eric Hardouin; Jean-Marie Chaufray; Thierry Clessienne
Original assignee: France Telecom
Priority date: 2005-09-14
Filing date: 2006-09-13
Publication date: 2007-03-22

Abstract

The invention concerns a communication device comprising a receiver configurable into a plurality of configurations, adapted to receive a signal derived from a transmission channel and delivered via at least one antenna (A<SUB>q</SUB>). It comprises means (20) for estimating absolute performances of said receiver (10) in different configurations and means for configuring the receiver in accordance with a configuration based on the results of said evaluations.

Description

Method and device for configuring a receiver in a communication device

BACKGROUND OF THE INVENTION The field of the invention is that of digital telecommunications.

The invention finds particular application in the field of radio frequency digital communications between a base station and a mobile terminal. But it can also apply to all transmissions (copper line, underwater acoustics, etc.).

More particularly, the invention relates to a method and a device for selecting, as a function of the transmission conditions, the configuration parameters of a receiver in a communication device.

In this document, "receiver" is understood to mean the detector as well as the reception chains associated with the antennas necessary for the reception of the signals to be processed, the "detector" designating the part of the modem that performs the baseband processing of the signals received. in order to estimate the transmitted symbols, while a "reception chain" designates all the processing carried out between an antenna and the detector, and notably comprises radiofrequency processing and Nyquist root filtering.

In known manner, several types of receivers can be implemented in a mobile terminal. These receivers may be single or multi-antenna, the associated detectors may be linear or not.

The different types of receivers are distinguished from each other by their performance (or efficiency) in a particular environment defined in particular by the signal transmission conditions. However, in a mobile communication network, the transmission environment is likely to vary significantly over time, particularly because of the movement of the mobile and objects in its environment.

This results in particular in a variation of the received useful power, the type of predominant interference (multiple paths for a mobile not far from its base station, intercellular interference for a mobile at the edge of the cell, etc.), and the variation of the transmission channel (selective or not in frequency).

A solution for maintaining the terminal's good performance in any situation consists in dynamically adapting the receiver used as a function of the transmission environment, by selecting the most appropriate receiver for the environment at a given moment, among a plurality of implanted receivers. in the mobile terminal.

Y. Karasawa, Y. Kamïya, Y. Loue and S. Denno "Algorithm diversity in software antenna" IEICE Trans. Common. June 2000 proposes to select a multi-antenna receiver according to two criteria namely a criterion relating to the frequency selectivity of the transmission channel, and a criterion relating to the spatial distribution of the signal and interference sources. Based on these two criteria, the most efficient detector is selected from four types of detectors, differentiated according to their ability to fight against or against the frequency selectivity of the channel and against spatially localized interference.

The method described in the aforementioned document does not allow to evaluate the performance gain of a given receiver relative to another receiver. In particular, it does not make it possible to evaluate the gain provided by a multi-antenna receiver with respect to a single-antenna receiver.

This method therefore leads to selecting the most efficient receiver, even if this receiver is much more complex to implement or consumer of energy than the others and that the performance gain obtained by this receiver is minimal.

It is also known, in particular by the document PJ. Black and MA Howard "Communication Receiver with Hybrid Equalizer", US Patent Application No. 200040240531, December 2004, a method in which a RAKE-type detector and linear linear error minimizer (LMMSE, for linear minimum mean square error) are paralleled, linear equalizer, more powerful but more complex, being activated only when it is able to bring a significant performance gain over the RAKE type detector. According to this method, the most appropriate receiver is selected not from an environmental criterion, but function of a performance measurement performed at the output of the detectors implanted in the terminal. However, this approach is expensive in terms of power consumption since both detectors, and a fortiori LMMSE linear equalizer, must operate in parallel during the selection periods.

Object and summary of the invention

The present invention solves the aforementioned problems by providing a device and a method of selecting the most appropriate receiver according to the absolute performance of the various receivers in the current environment, this selection being made with reduced consumption.

For this purpose and more specifically, it relates to a communication device adapted to receive a signal from a transmission channel, this signal being delivered by at least one antenna, this device comprising a configurable receiver according to a first configuration and at least a second configuration, this receiver being adapted to obtain samples of the signal and to process samples.

This device further comprises: means for estimating, using at least one parameter of the channel, and without implementing the receiver in the first configuration, a first variable representing an absolute performance at the output of the receiver, for processing samples, in the first configuration;

means for obtaining, for each of the at least one second configuration, a second variable representative of an absolute performance at the output of the receiver for processing samples in the at least one second configuration;

means for selecting one of the configurations according to the first and at least one second variables; and - means for configuring the receiver according to the selected configuration to process samples of the signal.

Within the meaning of the invention, a performance of the receiver in a configuration is called "absolute", as long as this performance is characterized independently of the performance of the receiver in another configuration. By way of example, a "signal-to-noise ratio" and "a mean square error" are absolute performance indicators within the meaning of the invention.

Unlike the various solutions of the state of the art, the communication device according to the invention is adapted to configure the receiver according to the most efficient configuration, without the need to activate all the detectors to make this selection, and with absolute knowledge of the performance that can be obtained at the output of the receiver for the different configurations available.

In an alternative embodiment, the second variables representative of the performance of the receiver are estimated, for each second configuration of the receiver, from the measurement of parameters of the transmission environment, and without implementing the receiver in any of the configurations. .

In another variant embodiment, a second variable representing the performance of the receiver is estimated, for each receiver configuration to the exclusion of the current configuration, from the measurement of parameters of the transmission environment and without implement the receiver in any configuration other than the current configuration, the variable representative of the performance of the receiver in the current configuration being directly measured at the output of the receiver.

Indeed, since the receiver is already active to process samples, measuring its performance requires little additional complexity.

In another variant embodiment, second variables representing the performance of the receiver are estimated, for second configurations of the receiver, from the measurement of parameters of the transmission environment, and other second variables representative of the performance. receiver in other second configurations are directly measured at the output of the receiver.

For Rake receivers and when the transmission channel estimate is made from pilot symbols, it is indeed possible to perform, jointly and without increasing complexity, the estimation of the channel and the estimation of the pilot symbols on each path of the channel, so that the performance of the receiver configured according to a Rake configuration can be measured from the pilot symbols estimated at the output of the Rake, the latter being obtained from the pilot symbols estimated on each path and weightings of the Rake at the price of a low complexity. It will be understood that since this method involves pilot symbols estimated at the output of the Rake receiver, it amounts to activating the receiver and measuring its performance at its output.

It should be noted, however, that the performance of the Rake receivers can also be calculated from their weightings and parameters of the transmission environment without estimating the pilot symbols.

Preferably, the selection means select the configuration taking into account the absolute performance that can be obtained but also the energy consumption of the receiver in the first and second configurations.

As mentioned above, this characteristic is particularly advantageous in the case of a mobile terminal since it makes it possible to manage a compromise between the performance and the consumption of the terminal.

Thus, in an alternative embodiment, the selection means select the configuration for which the absolute performance is maximum and the performance gain obtained with respect to another configuration of the lower consumption receiver is greater than a threshold.

This threshold can be predetermined or calculated.

In a second variant embodiment, the selection means select the configuration for which the absolute performance is greater than a target performance and the consumption is minimal.

This target performance can also be predetermined or calculated.

This variant advantageously makes it possible to configure the receiver so as to achieve a given quality of service, determined by the aforementioned target performance, this quality of service being able for example to be associated with a service subscribed by the user of the terminal. It will be readily understood that a strategy that would be limited to selecting the most efficient receiver as in the method proposed in the prior art could lead to the selection of a receiver that is too efficient in relation to the required quality of service, would result in wasted energy if a less complex receiver could achieve the required performance.

The first estimated variable (respectively the second variable obtained) representative of an absolute performance at the output of the receiver in each of the first and second configurations for processing the samples can be chosen in particular from:

a signal to interference and noise ratio (SINR);

- an average squared error; or

a bit error rate. In particular, when one of the possible configurations is that of a linear detector associated with one or more antennas in a terminal used in a downlink established with a base station of a DS-CDMA system, the signal-to-interference ratio and noise can be calculated based on: - parameters of the transmission channel;

- coefficients of the configuration;

a spatio-temporal correlation matrix of the received signal;

the spreading factor associated with the spreading code of interest;

the power associated with the spreading code of interest; and - the total power transmitted by the base station.

In a preferred embodiment, this interference to noise signal ratio is calculated using the ratio p between the transmit power of a pilot channel transmitted by the base station and the total power transmitted by that base station. This preferred embodiment has the advantage of facilitating the calculation of the SINR, because the powers emitted by the base station (total power and power of the pilot channel) can be problematic to estimate at the terminal.

In a particular embodiment, in which the signal is received by at least two antennas, the receiver is at least configurable according to the following four configurations: a mono-antenna RAKE receiver, which designates a receiver comprising a single active reception channel and a single-antenna RAKE detector;

multi-antenna RAKE receiver, which designates a receiver comprising a plurality of active reception channels and a detector of the type

RAKE multi-antennas;

a mono-antenna linear equalizer type receiver, which designates a receiver comprising a single active reception channel and a mono-antenna linear equalizer type detector; and a multi-antenna linear equalizer type receiver, which designates a receiver comprising a plurality of active reception channels and a multi-antenna linear equalizer type detector.

Correlatively, the invention relates to a method of configuring a configurable receiver according to a first configuration and at least a second configuration in a communication device adapted to receive a signal coming from a transmission channel and delivered by at least one antenna, the receiver being adapted to obtain and process samples of the signal. This method comprises: an estimation step, carried out using at least one parameter of the channel, and without implementing the receiver in the first configuration, a first variable representative of an absolute performance at the output of the receiver, for process the samples in the first configuration; a step of obtaining, for each of the at least one second configuration, a variable representative of an absolute performance at the output of the receiver, for processing samples in the at least one second configuration;

a step of selecting one of the configurations according to the first and at least one second variables; and

a step of configuring the receiver according to the configuration selected to process samples of the signal.

In a preferred implementation, the various steps of the configuration method are determined by computer program instructions. Consequently, the invention also relates to a computer program on an information medium, this program being capable of being implemented in a computer / device, this program comprising instructions adapted to the implementation of a configuration method as briefly described above.

This program can use any programming language, and be in the form of source code, object code, or intermediate code between source code and object code, such as in a partially compiled form, or in any other form desirable shape.

The invention also relates to a computer-readable information medium, comprising instructions of a computer program as mentioned above.

The information carrier may be any entity or device capable of storing the program. For example, the medium may comprise storage means, such as a ROM, for example a CD ROM or a microelectronic circuit ROM, or a magnetic recording medium, for example a floppy disk or a disk. hard. On the other hand, the information medium may be a transmissible medium such as an electrical or optical signal, which may be conveyed via an electrical or optical cable, by radio or by other means. The program according to the invention can be downloaded in particular on an Internet type network. Alternatively, the information carrier may be an integrated circuit in which the program is incorporated, the circuit being adapted to execute or to be used in the execution of the method in question.

Brief description of the drawings

Other features and advantages of the present invention will emerge from the description given below, with reference to the accompanying drawings which illustrate an embodiment having no limiting character. In the figures: - Figure 1 shows a communication device according to the invention in a preferred embodiment, and the antennas with which it is associated; and

- Figure 2 shows, in flowchart form, the main steps of a configuration method according to the invention in a preferred embodiment.

Detailed description of an embodiment

Throughout the description: - the notation x * denotes the complex conjugate of the scalar x;

- the notation | x | denotes the modulus of the scalar x;

- the notation E {x} denotes the mathematical expectation of the random variable x;

M ^τ denotes the transpose of the matrix M; and - M ^H denotes the conjugated transpose of the matrix M.

FIG. 1 represents a communication device according to the invention as well as the antennas with which it is associated. In the embodiment described here, this device comprises a multi-antenna receiver (10) comprising a linear detector. This receiver (10) is associated with a plurality of antennas, Q in the mode shown in Figure 1, each of these antennas being referenced Ao, ... A _Q -I. Behind each of these antennas Ai, this linear receiver comprises:

a radiofrequency channel CR adapted in particular to bring the signal back to carrier frequency in baseband;

an analog / digital converter CAN adapted to deliver digital signal samples;

a NY filter Nyquist root, which allows in particular to reduce the contribution of noise in the band of the signal carrying the information; and

sampling means adapted to sample samples delivered by the Nyquist root filter at a chip rate.

In the remainder of the description, y _n ^{(C |)} will be noted the sample of index n delivered by the chip sampling means at the output of the filter NY associated with the antenna Aq. The chip rate signals originating from the antenna q are filtered by a linear filter at the chip rate p ^(q) specific to the considered receiver, whose coefficients of the impulse response are contained in the length vector Lp

_D (? ⁾ L (? ⁾ _D ⁽ 9) _D ⁽ 1 ⁾ ] (I)

F WA ^{• • •} ro ^{• • •} FLp-AI i \ ^X J

where Δ is the filter restoration delay.

The outputs of the different filters are then summed, then the resulting signal is descrambled (multiplier 44) and filtered by a filter 46 adapted to the desired spreading code, the output of this filter being taken at the symbol rate by sampling means 48 to deliver estimated symbols.

According to the invention, the receiver 10 is configurable according to a plurality of configurations. The number of active reception channels is in particular configurable.

Those skilled in the art will understand that when switching from a receiver using two antennas to a receiver using only one antenna, the unused antenna and all the elements that follow it (RF chain, Nyquist root filter) , sampler) are disabled to save energy.

In the exemplary embodiment described here, this device is implemented in the context of the WCDMA standard. For more information on this standard, the skilled person may refer the document H. Holma and A. Toskala, WCDMA for UMTS, ^2nd ed. Wiley, 2002. In this example, we will consider that this receiver can be configured according to the following four configurations, numbered in the following configurations 1, 2, 3 and 4 respectively:

- RAKE mono-antenna type receiver, which designates a receiver comprising a single active reception channel and a detector of the type

RAKE mono-antenna (configuration 1);

multi-antenna RAKE receiver, which designates a receiver comprising a plurality of active reception channels and a multi-antenna RAKE detector (configuration 2); - mono-antenna linear equalizer type receiver, which designates a receiver comprising a single active reception channel and a mono-antenna linear equalizer type detector (configuration 3); and

multi-antenna linear equalizer type receiver, which designates a receiver comprising a plurality of active reception channels and a multi-antenna linear equalizer type detector (configuration 4).

It will be understood that a multi-antenna detector requires that the reception channels of each antenna are activated, whereas in the case of a single-antenna detector, the reception channels which are not necessary are deactivated in order to save money. energy. Thus, in this exemplary embodiment, the configuration of the receiver is entirely determined by the choice of detector. Therefore, we will call the receiver the name of the detector with which it is associated.

In this embodiment, it will be assumed that the variables representing the performance of the receiver configured according to each of the configurations are estimated from the measurement of parameters of the transmission environment as well as the parameters of this configuration, without the receiver being set implemented according to any of the configurations to process samples.

In known manner, each of these four receivers can indeed be implemented by a receiver comprising a linear detector made according to the detector structure described in FIG.

Indeed, the RAKE type detectors can also be made in the form of a linear filter followed by a correlator, as shown in the document M. Lenardi and D.T.M. Slock "A SINR maximizing 2D RAKE receiver for multi-sensor WCDMA mobile terminais", Vehicular Technology Conference Spring, Vol. 2, May 2001, pp. 1474-1478.

Thus, we will consider that regardless of the configuration of the receiver 10 the signals from the necessary antennas are brought back into baseband and then sampled at the chip rate after the usual reception processes (which notably include Nyquist root filtering) in order to give samples y _n ^(q) .

The coefficients of the impulse responses of the filters associated with the different antennas can be gathered in the configuration vectors p of length QLp. '(°)

P = (a)

. ((H)

Each receiver is characterized by a specific filter p.

For example, the linear equalizer minimizing the single-antenna mean squared error (LMMSE), applied behind the antenna O, will have for impulse response:

p IMMSb (b)

with

where H ^(q) is a Toeplitz matrix (L _h + L _p -l) ^χ L _p gathering the elements of the impulse response of the transmission channel in the form

n foW ₀ ) h ^(q \ h "Lh ^{q - ⁾ I 0 0

_≡ "I) T ₌ 0 hn ^(q ₀ ⁾ h ^(q \ ... h" Lh ^{q - ⁾ I (3)

0

0 • • • 0 hn ^(q ₀ ⁾ h ^iq ] h "Lh ^(q - ⁾ I

In these formulas, Z ^ is the length of the impulse response of the transmission channel expressed in number of chip durations, [h ^{9 ₀ ⁾ , h ⁽⁹ \, -; h J ₁ ^ ₁ ] are the coefficients of the response of the transmission channel on the sensor q, r ^(q) is the correlation matrix of the noise on the sensor q and e _Δ is a column vector of length L _{p of} which all the elements are zero except the Δ + 1 th.

It is recalled that in the particular case of the downlink of the WCDMA systems, the noise includes both thermal noise and intercellular interference. In the general case, σ] is the transmission power of the desired signal.

In the particular case of the downlink of the WCDMA systems, σ] is the total power of the signal transmitted by the base station.

It will be noted that the matrix σ j H H H ^ r ^ ^ n'est n'est is nothing other than the temporal correlation matrix of the signal received on the antenna q, which will be denoted R 1 in the following.

Similarly, the multi-antenna linear equalizer that we consider will have for impulse response the vector of the following filter coefficients:

_D (0)

PlMMSE

PΛ / 7V IAiMSE (C)

VlMMSE

which amounts to independently implementing an LMMSE equalizer 15 on each of the Q antennas, and wherein p ^ _Mo is given by equation (2).

The corresponding receiver will be called multi-antenna temporal LMMSE receiver (MT-LMMSE) in the following, while the receiver implementing a LMMSE mono-antenna equalizer will be called T-LMMSE receiver.

For more information on the coefficients of the LMMSE type filters, a person skilled in the art can refer to the document T.P.

Krauss, W.J. Hillery and M.D. Zoltowski "Downlink Specifies Linear Equalization for Selective Frequency CDMA Cellular Systems" J. VLSI Signal Processing, Vol. 30, pp. 143-161, March 2002.

In the case of multi-antenna RAKE detectors, L _p = L _h and the P ^(q) filters are strictly anticausal, which means that A = L _p -i and leads to the expression of the p ^(q) filters.

^• 1 ^J 0 ^U P n ^(q) - ~ \ IPn- ^(q L ⁾ h + l ^{• • •} 'P

n- ^iq \ ⁾ 'P no ^{q) ï J

where the coefficients

depend on the weighting type chosen for the RAKE detector.

For example, in the case of a mono-antenna Rake detector, we have 35 (E)

In the case of multi-antenna Rake detectors, the signals from the different antennas are combined before the finger treatment of the detector. For further information, those skilled in the art may refer to C. Liberti, Jr and T. S. Rappaport, Smart Antennas for Wireless Communications: IS-95 and Third Generation CDMA Applications, Prentice Hall, 1999.

Several types of weightings can be applied to the multi-antenna Rake detector depending on the chosen treatment, for example the optimal combination as described in the document

R. G. Vaughan, "On Optimum Combination at the Mobile", IEEE Trans. Veh.

Technol. Flight. 37, No. 4, Nov. 1988).

Thus, in the case of optimal combination, we have:

(0

Those skilled in the art will note that, in general, the Rake detectors only take into account the most powerful paths of the channel. This can easily be taken into account in the above model by canceling the weights corresponding to the paths not taken into account.

According to the invention, the communication device comprises means 20 for calculating, from the samples y _n ^(q) and for each of the four configurations, a variable J representative of the absolute performance obtainable by the receiver 10 configured according to each of the four configurations.

The variables representative of the absolute performance of the receiver 10 configured according to the configurations 1, 2, 3 and 4 are respectively denoted by J1, J2, J3 and J4. According to this variant of the invention, these variables J1,

J2, J3 and J4 are calculated without activating p ^(q) filters or detectors Mono-or multi-antenna rakes and linear mono- or multi-antenna equalizers for processing samples.

In the embodiment described here, the communication device according to the invention comprises a clock 15 which pauses the phases of reconfiguration of the receiver 10.

In the embodiment described here, we will assume that the reconfiguration of the receiver is done every ten WCDMA frames, that is every 100 milliseconds.

In a first variant embodiment, the calculation of the variables J1, 32, 33, 34 intended to configure the receiver for a period of ten frames is practiced from the signal received at the first slot of this period.

In a second variant, it will be possible to use the signal received at the last slot of a frame to configure the receiver to be active during the following ten frames.

FIG. 2 represents the main steps of a method of configuring the receiver 10 according to the invention. This method may in particular be implemented by the calculation means 20. During a first step ElO, the calculation means 20 estimate the parameters h of the transmission channel and the relevant parameters of the environment. In the preferred embodiment described here, these estimated parameters are:

the impulse response of the transmission channel on each of the Q sensors, the elements of which are noted

;

the power of the signal received on each sensor (q), denoted af ⁽² ;

the spatial intercorrelation between each pair of different sensors (q, v), denoted r ₀ ^{(9> v)} .

These parameters and the manner of obtaining them will be described later with reference to step E30, respectively by formulas (8), (12.1) and (12.2).

This estimation step ElO is followed by a step E20 during which the calculation means 20 determine the coefficients of the filters involved in the various configurations of the receiver 10. This step E20 consists of calculating, in the usual way, the responses pulses p corresponding to formulas (b) (c) (e) and (f). This step E20 for determining the coefficients of the different configurations is followed by a step E30 during which the calculation means 20 determine, for each of the four aforementioned configurations, a variable J representative of an absolute performance of the receiver 10 configured in this configuration. to process the samples y _n ^(q) .

In the preferred embodiment described here, this variable J representative of the absolute performance of the receiver in a given configuration is the signal to interference and noise ratio (SINR) at the output of the receiver 10.

For this purpose, the formula

SINR (4)

this formula allowing, as described in the document M. Lenardi and D.T.M. Slock "A SINR maximizing 2D RAKE receiver for multi-sensor WCDMA mobile terminais", Vehicular Technology Conference Spring, Vol. 2, May 2001, pp. 1474-1478, to calculate the SINR on the symbols obtained at the output of any linear multi-antenna detector in the downlink of a DS-CDMA system in the presence of long scrambling sequences.

In this formula P _CO and SF _CO are respectively the transmitted power and the spreading factor associated with the spreading code of the symbols for which the SINR is to be predicted, P _BS is the total power emitted by the desired base station, p is the vector gathering the coefficients of the impulse responses of the filters put into play by the receiver whose SINR is to be predicted and g ^ is a bias term affecting the estimated multi-user chip on the sensor q, given by

not

RST is the spatio-temporal correlation matrix of the received signal, defined by R (0.0) R (O, 1) ^rt 67 K ₅₇ .

R (0.1) K n Rd (l (W) ^tf ST _57,. I)

R ST (5)

R1 (W0, 0J - 1>)> D (i.o-i) R (H, CH) ST ^K S7 '

with

(9. ^V ) ^ (v, 9) * "(", </) *

'1' lp- \

(9- ^V ) _Λ (9.v)

(9, v) _ (6)

- (9, v), (9, V),. (9, ^V ) O

It will be noted that the matrix R £ ^(q) is none other than the temporal correlation matrix of the signal received on the antenna q. The temporal correlation matrices and their elements will be noted respectively in the following

R (9) - J) UT - ^κ s S: T (7)

and

"(9) _ _r (? //)

'm' m

This method of calculation is particularly precise because it takes into account all the parameters of the environment influencing the SINR.

However, those skilled in the art will understand that the calculation of the SINR by the formula (4) can not be used directly in a mobile terminal, where the transmission powers (P _B s and P _∞d e) as well as the real coefficients of the channel are unknown to the receiving device.

Indeed, the estimated channel coefficients generally differ from the exact coefficients by an unknown scale factor of the receiving device. In the preferred embodiment described here, we assume that the channel coefficients are estimated by correlation with the pilot channel of the WCDMA by:

In this formula, _p is the value of the pilot symbols carried by the pilot channel, S _n is the n th sample of the scrambling sequence such that | s _n | = 1 and N is the length of the signal sequence received on which is the estimation. N is typically of the order of the length of the W-CDMA slot, ie 2560 chips.

The relationship between the estimated coefficients and the exact coefficients is then given by

n, - y] * _p ιi ₀ , _e rlι -T t ₁ \ V)

where Ppϋote is the power of the pilot channel sent by the base station, whose value is unknown to the terminal, and ε, is an estimation error.

Using the relation (9), it is shown that in the absence of estimation error on the channel and on the spatio-temporal correlation matrix, that is to say if ε, = 0 V / and R ₆₇ , = R ₆₇ , where R ₅₇ denotes the estimated spatio-temporal correlation matrix, the SINR of formula (4) expressed on the pilot symbols is exactly found by the formula:

SINR p _n i, l ,, o, te (10)

and p is defined as the ratio between the transmit power of the pilot channel and the total power transmitted by the base station

where P _B s is the total power emitted by the base station.

This method therefore requires the ratio /? is known to the receiving device, for example by being transmitted by the base station.

Note that in the case of single-antenna receivers, the spatio-temporal correlation matrix is reduced to a temporal correlation matrix (defined by (7)).

Moreover, it will be noted that the formula (10) can sometimes be simplified so as to only involve the terms p and g _o ^(q) . This is particularly the case when the detector is the LMMSE monoantenne equalizer. Formula (10) can then be simplified as

where q is the number of the antenna used.

In the general case, the formula (10) requires to have an estimate of the spatio-temporal correlation matrix, denoted R _sτ . We will first notice that the Hermitian symmetry of the spatio-temporal correlation matrix Rsτ, the Toeplitz structure of the matrices denoted by R ^ ^{v) as} well as the Hermitian symmetry and the Toeplitz structure of the correlation matrices denoted by R ^ contained in the matrix R _ST makes it possible to reduce the terms to be estimated on the first rows of the matrices R ^ and on the first row and the first column of the matrices R ^: ^v) located in the upper triangle of R _sτ .

It will also be noted that in the case of detectors of the Rake type that do not demodulate all the paths of the channel, certain terms of the spatio-temporal correlation matrix do not intervene in the calculation of the SINR (10) because of the presence of zeros. in the vector p. This makes it possible to restrict the number of terms of the matrix to be estimated only to the terms that then intervene in the calculation of the SINR.

It is known that the structure of the matrix defined by the formula (6) is given by

R ^v) = σ _d ² n ^{(v) H} V L ^(q) + r ^ι ( ^v) (12) where r ^(q - ^v) is the spatio-temporal correlation matrix of the noise between the sensors q and v defined according to the model of (6). We will also note

F ^(?) = R ^ - ^

In the method described here, the spatio-temporal correlation matrix is calculated from the knowledge of the structure of the matrices

R ^ and R ^: ^v) , given by the formulas (2) and (12) respectively, and estimates of the impulse responses of the transmission channels associated with the different antennas obtained in the estimating step ElO.

In order to simplify the calculation, it is assumed that the noise correlation matrices are diagonal, i.e.

and

where σ _w ^{(q) 2} is the noise power on the sensor q, and σj ⁽ - ^{v) 2} is the cross correlation between the noise on the sensor q and the noise on the sensor v.

This assumption is common in the absence of knowledge about the noise correlation matrix, or when the implementation complexity is preferred. The matrices R ^ and R <f / ^} are then rewritten as

and

The elements of these matrices can then be calculated to a real positive coefficient by, for example by the method that we will now describe, inspired by that described in the document E. Hardouin "Leveling at the chip level for the downlink of the communications systems mobile DS-CDMA ", Doctoral Thesis, University of Rennes 1, May 2004, pp. 110-114, in the mono-antenna case. This method also requires knowledge of the ratio p as defined above (see formula (H)). Let R _sr , R <* ^{> v)} and Rf be the respective estimates of the matrices R _sr , R ⁽ _s * - ^V) and R ^ obtained by this method.

The elements of

are calculated by

where the coefficients h, are the coefficients of the transmission channel estimated by correlation with the pilot channel, by the formula (8). The term σj ^{() 2} is calculated by

σf = σ ^ω -α: i'iΫ (12.1)

where σ ^{ f ² is the power of the signal received on the sensor (q), estimated by

NOT-\ .

\ - (i) ^Δ (</

where N is typically of the order of the length of a W-CDMA slot, ie N = 2560 chips, and σf ^{() 2} is given by

The relationship between the temporal correlation matrix thus calculated and the exact temporal correlation matrix, under the assumption of temporally white noise and in the absence of estimation errors, is given by

The matrices R ^{ ^v) are calculated in a slightly different manner, in order to integrate a value as faithful as possible of the terms σj ⁽ - ^{v) 2} , which represent the intercorrelations between the contributions of the noise on two different sensors. Indeed, these terms can not be calculated since we generally ignore the statistical properties of the noise. The method therefore consists of directly estimating the term r ₀ ^{(?> V)} by

p ( _? .v) _! v. (< _/ ) (V) * (12.2)

^ "= 0

but to calculate the other terms by

, ..., 4 -l

In the application considered, N will typically be of the order of the length of the W-CDMA slot, or 2560 chips. Alternatively, this length may be shorter as 256 chips at the cost of a possible reduction in the quality of the estimate. In the absence of estimation errors and under the assumption of a spatio-temporal correlation matrix of the noise between the sensors q and v diagonal, the relation between the calculated terms r ^ ^{/> v)} and the exact terms r _m ^{q ' ^v) is given by

? ™ = pr ™

In order to take into account the measured value f ^ ^v) in the matrix R% ' ^v) , we therefore set:

7 ₀ ^(q ' ^v) = p ξ ^q ' ^v) In the absence of estimation errors and under the assumption of a spatio-temporal correlation matrix of the noise between the sensors q and v diagonal, the relation between the calculated matrix R ^ ^'10 and the exact matrix R ^ ^v) is thus given by:

R ^{(<?> V)} = oR \ ⁽ S ^q 7- ^v)

Once the matrices R ^ and R ^ ^{* 0 have been} estimated, the matrix R _{sr is formed} according to the model of equation (5), and finally the matrix R ₅₇ intervening in equation (10) is obtained by the relation

R \ _C S7 ₇ - - R Sl

P

Once the spatio-temporal correlation matrix is estimated, the SINR is calculated on the pilot symbols by the formula (10).

As mentioned above, in an alternative embodiment, the selection means 20 selects the configuration for which the absolute performance is greater than a target performance and the receiver consumption is minimal. In this case, the signal to interference and noise ratio SINR target that the receiver must achieve to ensure the quality of service is usually relative to the information symbols. In order to be compared to the target SINR, the predicted SINR must then be the SINR on the information symbols and not the SINR on the pilot symbols. We describe below how to predict the SINR on the information symbols from the predicted SINR on the pilot symbols.

In the case of linear detectors, the SINR on the information symbols, noted SINR _m f ₀ is deduced from the SINR on the pilots, noted SINRpdote by the relation

_ ^ Mζ _πf o_

SINRpiote

where SFmfo and SF _f xiots are the spreading factors associated with the information symbols and the pilot symbols respectively, whereas P _mf0 and Pp, iote are the transmitted powers of the signals respectively carrying the information symbols and the pilot symbols. . Since the spreading factors of the pilot symbols and the information symbols are known, the prediction of the SINR on the information symbols according to the SINR on the pilots therefore requires the knowledge of the P / n / v IPpiiotes ratio, which can be estimated by the communication device from the previous slot when the pilot symbols considered belong to the pilot channel, for example using the formula

In this formula, M and M ¹ are respectively the number of pilot symbols and information symbols in a slot, â ^ ^f ° and â, ^drivers respectively are the information symbols and the pilot symbols of a slot, estimated at the output of one of the fingers of the Rake, or at the output of a linear equalizer type detector, by

where ^χ _n is the nth chip of the signal on which the estimation is made (for example, x _n may be the signal received on an antenna y _n ^(q) in the case of a finger of the Rake, or the output signal of the equalizer at the chip level in the case of a detector of the linear equalizer type), c _n = ± 1 is the nth element of the spreading code of the considered symbols (pilots or information), s _n is the n th sample of the scrambling sequence as

= 1 and SF is the spreading factor of the considered symbols.

The step E30 for calculating the variables J1 to J4 representative of the absolute performance of the receiver 10 configured according to each of the four configurations (namely here the value of the SINR) is followed by a step E40 during which the calculation means 20 select receiver configuration 10.

In a first embodiment, the receiver is configured with the configuration to obtain the best performance while respecting a predefined performance / consumption ratio. In other words, we select the best performing receiver provided it offers a higher performance gain than predefined gains over less complex receivers. These predefined gains are a function of the consumption of the compared receivers, and express the loss of performance that one is ready to accept in exchange for a reduction in consumption. With these predefined gains, you can control the performance / consumption of the terminal.

We now detail an example of a selection process according to this first mode.

The calculation means 20 first determine the receiver associated with the maximum predicted SINR.

Assume that this receiver is the receiver Ro and note SINRR ₀ the value of its predicted SINR.

The calculation means 20 then calculate the performance gain provided by the receiver R ₀ with respect to each other receiver whose consumption is lower, that is to say each pair ΔSINRRO / R _Π = SINRRO-SINRRΠ such that the consumption the receiver R _n is less than the consumption of the receiver R ₀ , where SINR _Rn is the predicted SINR for the receiver R _n .

These gains are then compared with predefined gains ΔSINR ^ref _Rm / _R n, which define the minimum gain that the receiver R _m must bring relative to a receiver R _n , more economical in terms of consumption, to be selected.

If the above values ΔSINRRO / R _Π are all greater than or equal to the predefined gains ΔSINR ^ref Ro / Rn, then the receiver Ro is selected.

Otherwise, the selected receiver is the receiver R _n of lower consumption among those such as ΔSINRRO / RΠ <ΔSINR ^ref Ro-Rn-

It should be noted that this method is equivalent to adding the predefined corrective factor ΔSINR ^ref R _m / Rn to the SINR of each receiver R _n less complex than the best performing receiver, and to selecting the least complex receiver among those whose corrected SINR is greater than to the SINR of the most powerful receiver.

We give in the following an example of predefined gains for the four receivers considered, in the particular case where the terminal has two antennas: the mono-antenna Rake receiver, the two-antenna Rake receiver with combination of optimal type combining (noted below OC-Rake), the T-LMMSE receiver (mono-antenna) and the MT-LMMSE receiver with two antennas.

We assume that in the embodiment considered, the implementation of the different receivers is such that their relative consumption is established as

OC-Rake receiver consumption = 2.0 x Consumption of the mono-antenna Rake receiver

T-LMMSE receiver consumption = 1.7 x Consumption of the mono-antenna Rake receiver

MT-LMMSE receiver consumption = 3.5 x Single-antenna Rake receiver consumption

In the embodiment considered and for a performance / consumption ratio favoring performance, the predefined gains may for example be fixed at (in linear values)

ΔMT-LMMSE / OC-Rake = 0.4 ΔMT-LMMSE / T-LMMSE = 0.5 ΔMT-LMMSE / Rake = 0.5

Δoc-Rake / T-LMMSE = 0.1 Δoc-Rake / Rake = 0.5

Δτ-LMMSE / Rake ⁼ 0.5

While to favor the economy of consumption, we will be able to choose, for example

ΔMT-LMMSE / OC-Rake = 1-7

ΔMT-LMMSE / T-LMMSE = 2.0

ΔMT-LMMSE / Rake = 2.2

Δoc-Rake / T-LMMSE = 0.4 Δoc-Rake / Rake = 2.0

Δτ-LMMSE / Rake = 2.0 In a second embodiment, the selected receiver is the least complex among those achieving a target performance.

If no receiver reaches the target SINR, then the selection is made according to the first mode described above.

It will be noted that in order to limit the consumption of the selection process, the receivers can be tested in the increasing order of their complexity, the selection process ending as soon as one of the receivers reaches the target SINR. Moreover, the coefficients of the filters used by the different detectors can be calculated as and when they are tested, in order to calculate the filters of a complex receiver only if it is necessary to achieve the required performances.

Once the most appropriate configuration has been determined, the calculation means 20 configure (step E50) the receiver 10 according to this configuration to process the received signal in order to demodulate the transmitted symbols, the unnecessary reception channels and all the other detectors. as the selected detector being disabled.

As part of the WCDMA standard, those skilled in the art will understand that it is not useful to activate the selection process more than once per slot.

It may however be activated with a longer periodicity, for example once per frame, or only once every ten frames as in the particular embodiment described. In addition, it will be noted that it is more economical in terms of consumption to obtain the output SINR of a receiver when a more complex receiver is active. For example, it is inexpensive to predict the output SINR of the T-LMMSE receiver (mono-antenna) when an MT-LMMSE receiver is active since the necessary filters are already calculated. Thus, if the selection process including more complex receivers than the active receiver is activated with a given periodicity, the performance of the less complex receivers that the active receiver can be monitored with a shorter periodicity, for example at each slot, so to switch back to a less complex receiver as soon as the active receiver significant gain compared to a more economical receiver in terms of consumption.

Finally, it will be noted that only a part of the receivers implanted in the communication device may be the subject of a performance prediction, the other part being able to intervene in the selection process by means of a measurement of their effective performance. For example, in the case where the available receivers are a mono-antenna Rake and a T-LMMSE receiver, the Rake can be activated in order to measure its performances, while the performances of the T-LMMSE receiver will be predicted.

In summary, the invention consists in implementing, for the selection of the receiver, or more precisely for its configuration, a step of predicting the absolute performance of at least a portion of the receivers implanted in the terminal, this performance prediction being synonymous with absolute performance estimation, performed without activating any of the associated detectors, from the estimation of relevant parameters of the environment such as the transmission channel, as well as configuration parameters of the different receivers such as the coefficients of the filters set This selection method can be implemented regardless of the detectors implanted in the terminal as soon as a mathematical expression of their performance is available as a function of measurable environmental parameters. For example, this method can be implemented for the iterative detector (non-linear) mono-antenna described in the document E. Hardouin and C. Laot, "Iterative channel equalization for the multicode DS-CDMA downlink", VTC 2003- Spring, April 2003, calculating the output SINR of this detector from the mathematical expression of the variance of the output interference provided in the aforementioned document, using a method similar to that described in this document for that this output SINR is a function of measurable parameters of the environment.

We have previously described a method of performance prediction adapted to the class of linear multi-antenna detectors for the downlink of DS-CDMA systems, which includes in particular mono or multi-antenna RAKE receivers and linear EQs. mono or multi-antenna chip level. In this description, we have used the SINR as a performance criterion, but we can also use other performance criteria, such as the mean squared error, provided that we have a mathematical expression of this criterion. function of measurable parameters of the environment.

Furthermore, it should be noted that it is possible to use other performance criteria from the signal to interference and noise ratios (SINR). For example, it is possible to estimate the raw bit error rate (i.e., before channel decoding when the system implements a channel decoding method), denoted BER, corresponding to the SINR predicted using of the formula known in the case of QPSK modulation

where erfc (x) designates the complementary error function.

More generally, those skilled in the art will understand that the bit error rate or the frame error rate can be obtained by using tables giving the value of the criterion corresponding to a given SINR.

In the foregoing description, the four variables J1, 32, J3, J4 are estimated without implementing the receiver for this estimate.

In one variant of the invention, at least one of these variables is measured at the output of the receiver implemented in the associated configuration.

This variant is particularly advantageous for measuring the absolute performance of the receiver in the current configuration.

Finally, it will be noted that the invention can also be implemented to select the most appropriate between several configurations of a given type of detector, such as the type of weighting involved in the Rake receivers, or the length (L _p ) an equalizer at the chip level.

Claims

1. Communication device adapted to receive a signal from a transmission channel and delivered by at least one antenna (Aq), said device comprising a receiver (10) configurable according to a first configuration and at least a second configuration, said receiver (10) being adapted to obtain samples (y _n ^(q) ) of said signal and to process said samples, this device being characterized in that it further comprises: - means (20) for estimating, using at least a parameter of said channel, and without implementing said receiver (10) in said first configuration, a first variable (J) representative of an absolute performance at the output of said receiver (10), for processing said samples (y _n ^(q) ) in said first configuration; means (20) for obtaining, for each of said at least one second configuration, a second variable (J) representative of an absolute performance at the output of said receiver (10) for processing said samples (y _n ^{(C | ))} in said at least a second configuration;

means (20) for selecting one of said configurations according to said first and at least one second variables (J); and

means for configuring said receiver (10) according to said selected configuration to process samples (y _n ^(q) ) of said signal.

2. Device according to claim 1, characterized in that said means (20) for obtaining are adapted to estimate at least one said second variable (J), using at least one parameter of said channel, and without implementing said receiver (10) in said at least one second configuration.

3. Device according to claim 1 or 2, characterized in that said means (20) for obtaining are adapted to obtain at least one said second variable (J) by measuring an absolute performance of said receiver (10), said receiver (10) ) being implemented according to said second configuration.

4. Communication device according to any one of claims 1 to 3, characterized in that said selection means (20) select said configuration taking into account said absolute performance and energy consumption of said receiver (10) in each said first and at least one second configuration.

5. Communication device according to claim 4, characterized in that said selection means (20) select said configuration for which said absolute performance is maximum and the performance gain obtained with respect to another configuration of the consumer receiver (10). lower is greater than a threshold.

6. Communication device according to claim 4, characterized in that said selection means (20) select said configuration for which said absolute performance is greater than a target performance and said consumption is minimal.

7. Communication device according to any one of claims 1 to 6, characterized in that said first variable (J) estimated, respectively said at least one second variable (J) obtained, representative of an absolute performance output of said receiver (10), is selected from a ratio (SINR) signal over interference and noise, an average squared error or a bit error rate at the output of said receiver (10).

The communication device according to claim 7, wherein said first configuration or said at least one second configuration allows said receiver (10) to be configured as a linear receiver used in a downlink established with a base station in a DS-CDMA system. , characterized in that said ratio (SINR) signal on interference and noise is calculated from;

- parameters (h) of said transmission channel;

coefficients (p) of said configuration; a spatio-temporal correlation matrix (RST) of said received signal;

- the spreading factor (SF _CO de) associated with the spreading code of interest; power (P _COd e) associated with the spreading code of interest; and

- the total power (PBS) transmitted by said base station.

9. Communication device according to claim 8, characterized in that said signal to interference ratio and noise is calculated using the ratio (p) between the transmission power of a pilot channel (Pcpic _h ) transmitted by said station base and the total power

(PBS) transmitted by said base station.

10. Communication device according to any one of claims 1 to 9, wherein said signal is received by at least two antennas (Aq) characterized in that said receiver is at least configurable in a first configuration and three second configurations, these first and second configurations being different and chosen from the following four configurations:

- RAKE mono-antenna type receiver;

- RAKE type multi-antenna receiver;

- mono linear antenna equalizer type receiver; and

- receiver type linear equalizer multi-antennas.

11. A method of configuring a receiver (10) configurable in a first configuration and at least a second configuration in a communication device adapted to receive a signal from a transmission channel and delivered by at least one antenna (Aq) said receiver (10) being adapted to obtain and process samples (y _n ^(q) ) of said signal, said method being characterized in that it comprises:

a step (E30) of estimating, using at least one parameter of said channel, and without implementing the receiver (10) in said first configuration, a first variable (J) representative of an absolute output performance said receiver (10) for processing said samples (y _n ^(q) ) in said first configuration;

a step (E30) for obtaining, for each of said at least one second configuration, a variable (J) representative of an absolute performance at the output of said receiver (10), for processing said samples in said at least one second configuration; a step (E40) of selecting one of said configurations as a function of said first and at least one second variables (J); and

a step (E50) of configuring said receiver according to said selected configuration to process samples (y _n ^(q) ) of said signal.

Computer program comprising instructions for executing the steps of the configuration method according to claim 11 when said program is executed by a computer constituting a communication device.

A computer-readable recording medium on which a computer program is recorded including instructions for performing the steps of the configuration method according to claim 11.