US20190173588A1 - Dc current cancellation scheme for an optical receiver - Google Patents
Dc current cancellation scheme for an optical receiver Download PDFInfo
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- US20190173588A1 US20190173588A1 US15/833,314 US201715833314A US2019173588A1 US 20190173588 A1 US20190173588 A1 US 20190173588A1 US 201715833314 A US201715833314 A US 201715833314A US 2019173588 A1 US2019173588 A1 US 2019173588A1
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/60—Receivers
- H04B10/66—Non-coherent receivers, e.g. using direct detection
- H04B10/69—Electrical arrangements in the receiver
- H04B10/691—Arrangements for optimizing the photodetector in the receiver
- H04B10/6911—Photodiode bias control, e.g. for compensating temperature variations
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/60—Receivers
- H04B10/66—Non-coherent receivers, e.g. using direct detection
- H04B10/67—Optical arrangements in the receiver
- H04B10/671—Optical arrangements in the receiver for controlling the input optical signal
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/60—Receivers
- H04B10/66—Non-coherent receivers, e.g. using direct detection
- H04B10/69—Electrical arrangements in the receiver
- H04B10/693—Arrangements for optimizing the preamplifier in the receiver
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/60—Receivers
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/60—Receivers
- H04B10/66—Non-coherent receivers, e.g. using direct detection
- H04B10/69—Electrical arrangements in the receiver
Definitions
- the present invention relates to an optical receiver, and in particular to an optical receiver with DC current cancellation.
- a typical optical receiver front-end is composed of a photo diode (PD) 1 followed by a trans-impedance amplifier (TIA) 2 and main amplifiers (MAs) 3 as shown in FIG. 1 .
- the PD 1 receives a transmitted optical signal 4 and generates a current 6 proportional to the received optical power of the received optical signal 4 .
- the ratio between PD output current 6 to the input optical power of the optical signal 4 is the photo diode responsivity (R).
- the TIA 2 converts the PD current 6 to voltage, which is then amplified by MAs 3 to the desired signal level for the decision circuitry 7 .
- the modulation depth of the optical signal 4 is defined by its extinction ratio, which is the ratio between optical power for symbol one (P 1 ) and optical power for symbol zero (P 0 ).
- the transmitted optical signal 4 has poor extinction ratio and translates into a small modulated current with a large DC current at the output of the photo diode 1 .
- the large DC current saturates the receiver front-end (TIA 2 and MAs 3 ) and significantly degrades the gain and bandwidth performance. Consequently, cancelling photo diode DC current in high data rate receivers is desired for proper receiver operation, i.e. to have zero average modulated PD current 6 .
- PD DC current is expressed as
- I DC R ⁇ ( P LO +P S ), (1)
- Equation (1) shows that the photo diode output DC current of the PD 1 in coherent optical communication links depends on the local laser power and the optical received power. For example, a photo diode 1 with responsivity (R) of 1 A/W results in 4 mA DC current at 6 dBm local laser power input. Such a large DC current is more than enough to saturate the receiver front-end and severely degrades its performance. Thus, it is very important to have DC current cancellation circuitry in front of the TIA 2 of coherent optical communication links.
- FIG. 2 shows a conventional way to AC couple receiver photo diodes 11 to the a front-end TIA 12 using passive AC coupling circuitry.
- An AC coupling capacitor (C C ) is inserted between the photo diode 11 and the front-end TIA 12 to block the DC current; however, it bypasses the modulated AC current to the TIA 12 .
- a biasing resistor (R C ) is used to bias the photo diode anode voltage to be reverse biased, and provides an alternative path for the photo diode DC current I DC .
- the biasing resistor R C with the AC coupling capacitor C C forms a high pass filter section in the RF signal path and its cutoff frequency (FC) is calculated as,
- the required TIA low cutoff frequency (FC) is around 100 kHz which requires either large AC coupling capacitor C C or huge biasing resistor R C .
- a capacitor C C with a capacitance of at least 1.6 pF with a resistor R C with a resistance of at least 1 M ⁇ are required to achieve cutoff frequency of 100 kHz.
- this technique suffers from two main drawbacks: 1) C C parasitic capacitance, and 2) photo diode biasing.
- the bottom plate ground parasitic capacitance of the coupling capacitor C C is around 10% of its value and degrades the front-end TIA bandwidth, which is defined by its input node capacitance.
- there is a maximum coupling capacitor (C C ) that can be used without degrading the TIA bandwidth.
- the biasing voltage across the photo diode 11 is defined by the following equation:
- V BIAS V PD ⁇ V B ⁇ I DC ⁇ R C , (3)
- V BIAS is the reverse biasing voltage across the photo diode PN junction.
- High photo diode reverse biasing voltage is required to obtain good photo diode responsivity and low PN junction capacitance.
- equation (3) shows that V BIAS depends on PD average current and leads to different PD biasing for different received optical power.
- a large R C value impedes receiving high optical power levels as the DC current will be large and the voltage drop across R C will be huge.
- an I DC of 10 ⁇ A leads to a 10V drop on a 1 M ⁇ resistor R C , which is not practical.
- the situation in coherent optical receivers is much worse as the photo diode DC current is around 1 mA and requires an R C of less than 1 k ⁇ for less than 1 V drop across the biasing resistor R C .
- An object of the present invention is to overcome the shortcomings of the prior art by providing a DC current cancellation loop for use with a fully differential front-end TIA structure.
- an optical receiver comprising:
- a first photodetector for converting a first input optical signal into a first PD current comprising a first AC component and a first DC component
- TIA transimpendance amplifier
- a first DC cancellation loop including an input and an output between the first PD and the TIA for cancelling the first DC component, the first DC cancellation loop comprising:
- a first trans-conductance cell capable of drawing in the first DC component, such that the first DC cancellation loop maintains a first DC voltage value of the first input of the TIA the same as a first reference voltage (V REF 1), which represents an actual TIA input voltage for a zero DC current condition;
- FIG. 1 illustrates a schematic diagram of a conventional NRZ optical receiver
- FIG. 2 illustrates a schematic diagram of a conventional differential optical receiver
- FIG. 3 illustrates a schematic diagram of a fully differential optical receiver in accordance with an embodiment of the present invention
- FIG. 4 illustrates a photodetector and differential TIA structure of the optical receiver of FIG. 3 ;
- FIG. 5 illustrates an embodiment of a transconductance cell in accordance with the present invention.
- FIG. 6 illustrates another embodiment of a photodetector and TIA structure of an optical receiver of the present invention.
- FIG. 3 illustrates an embodiment of an electronic component in a packaged optical receiver 30 in accordance with an embodiment of the present invention.
- An input signal enters at a port 31 .
- a small portion of the input optical signal e.g. less than 5%, may be split off and sent to a monitor photodiode 32 , which generates an electrical signal that may be used to monitor properties of the input optical signal, such as its power content.
- the power of the input optical signal can be monitored using different hardware.
- the remainder of the input optical signal may be sent through a variable optical attenuator 33 , which can adjust the signal intensity.
- a polarization beam splitter (PBS) 34 splits the remainder of the input optical signal into x-polarized (X-Pol) and y-polarized (Y-Pol) components.
- the X-Pol component is sent to a 90° hybrid mixer 36
- the Y-Pol component is sent to a 90° hybrid mixer 37 .
- a local oscillator 38 provides a signal that is split by a beam splitter 39 , and components of which are sent to each of 90° hybrid mixers 36 and 37 .
- the 90° hybrid mixers 36 and 37 are optical components that each generate two phase differentiated optical signals, the XI and XQ signals and the YI and YQ signals, respectively.
- each of the four phase differentiated signals are converted to electrical signals by respective optical receivers 40 , including photodetectors 41 and transimpedance amplifiers (TIA) 42 .
- the electrical signals are then provided at four respective output terminals, which may be single-sided signals referenced to a common ground or may be differential signals.
- FIG. 4 illustrates an embodiment of the optical receiver 40 including an active AC coupling circuitry disposed between the photo diode (PD) 41 and the front-end differential TIA 42 .
- An analog DC cancellation loop 45 a and 45 b is located prior to each input of the TIA 42 , and draws the photo diode DC current I DC , whereby only the AC signal I AC is coupled to the TIA 42 .
- Two different analog cancellation loops 45 a and 45 b are used at both TIA inputs.
- Each cancellation loop 45 a and 45 b comprises a low pass filter section 46 a and 46 b , e.g.
- Photo diode DC current I DC is drawn from the input node of the TIA 42 in the G M cell, such that the analog loops 45 a and 45 b maintain the DC voltage value of the TIA input node to be the same as a reference voltage (V REF ).
- This reference voltage V REF represent the actual TIA input node voltage for a zero DC current condition, which means that on average there is no current flowing into each of the differential TIA inputs.
- the term “zero” DC current means very little current, e.g. less than 100 uA or, ideally, less than a value that would significantly impact other TIA performance characteristics, such as linearity and bandwidth. Accordingly, the DC cancellation loops 45 a and 45 b greatly preserve the linearity of the TIA 42 and the optical receiver 40 , because if the DC current is not canceled, it would flow into the TIA 42 and change the bias point, output signal common mode, and may even completely saturate the TIA 42 , rendering it useless.
- a method of generating V REF includes using a replica TIA 47 , also shown in FIG. 4 .
- the replica TIA 47 includes floating inputs and outputs and substantially the same structure as the main TIA 42 , in order to generate a correct V REF voltage, corresponding to a zero DC input current condition.
- Using a replica TIA 47 provides a simple way to track the main TIA 42 across process, voltage and temperature (PVT) variations, and to automatically generate a correct V REF in all cases.
- the trade-off of using the replica TIA 47 is additional power dissipation. In order to minimize the additional power dissipation from the replica TIA 47 , it may be scaled down.
- the replica TIA 47 may use a fraction, e.g. 1 ⁇ 4, of the bias current of the main TIA 42 , flowing through resistors RF that are 4 ⁇ larger than the resistors in the main TIA 42 . Accordingly, the same V REF voltage is generated, while burning only 1 ⁇ 4 of the power of the main TIA 42 . Scaling will reduce the quality of the PVT tracking of the replica TIA 47 , so the scaling factor has to be selected carefully, to achieve the perfect balance between PVT tracking and power dissipation. It should be noted that V REF may be generated in a variety of other ways, without using a replica TIA. For example, a custom analog circuit may be constructed to simultaneously achieve the goals of correct V REF generation, with PVT tracking, and minimum power dissipation.
- the cancellation loops 45 a or 45 b should not affect or attenuate the received high speed signal.
- the loops 45 a or 45 b should track any variation in the photo diode DC current I DC and completely cancel it. This implies that the speed of the analog cancellation loops 45 a and 45 b are bonded by two main upper and lower limits, which are the lowest frequency component of the received data (upper limit) and the fastest variation of the photo diode DC current I DC (lower limit).
- the gain-bandwidth product of this analog loop 45 a or 45 b is calculated as,
- each output depends on both inputs due to the high common mode rejection of the fully differential TIA 42 .
- TIA output voltages V OUTP , V OUIN ) are expressed as,
- V OUTP - R F 2 ⁇ ( I P - I N ) , 6 ⁇ a
- V OUTN R F 2 ⁇ ( I P - I N ) , 6 ⁇ b
- Equation (6) implies that both V OUTP and V OUTN depend on I P and I N with the same weight and opposite effect.
- the prior art cannot be employed with a fully differential TIA because the two cancellation loops 45 a and 45 b will be strongly coupled and affected by each other.
- each output V OUTP and V OUTN (positive or negative) depends only on the corresponding input current I AC which makes the two cancellation loops 45 a and 45 b decoupled and the cancellation performed correctly.
- the proposed AC coupling scheme offers better isolation between the DC cancellation loops 45 a and 45 b , in particular with the fully differential TIA 42 , because the sensing operation is performed at the input of the TIA 42 .
- the trans-conductance cell G M draws a current (I OUT ) proportional to the DC component of the differential input (V INP ⁇ V INN ).
- the AC trans-conductance of the proposed GM cell is expressed as,
- A is the gain of the amplifier 51 and g m is the trans-conductance of the differential pair (T 1 , T 2 ) each transistor having a resistor R connected thereto.
- N is a scaling factor for the output emitter-degenerated current mirror 52 , where the output bipolar transistor 55 is made N longer and the degeneration resistor 56 is N times smaller. N may or may not be an integer.
- the scaling factor N enables the trans-conductance cell to operate with a smaller bias current from bias current source I B , and thereby to reduce power dissipation.
- the bias current source I B is connected to both of the transistors T 1 and T 2 of the differential pair.
- the output I OUT is connected to the second transistor T 2 via an output transistor 55 and the output resistor 56 R/N.
- the capacitor C C of the R C loop pass filter section 46 a and 46 b may be implemented using a miller capacitor between the input and output of the voltage amplifier 51 to boost its value by the voltage amplifier gain, which helps in reducing the size of the implemented R-C section 46 a and 46 b .
- the loop pass filter cutoff frequency is expressed as,
- CMRR common mode rejection ratio
- Coherent receivers often have challenging CMRR requirements, and a fully differential TIA easily meets and exceeds most CMRR specs.
- fully differential TIA front ends have better power supply and ground noise rejection.
- the output of a fully differential TIA is fully compatible with the differential amplifier stages, so there is no need to convert the signal from single-ended to differential.
- the present invention may be used in non-coherent systems, since large DC currents can result from poor extinction ratio of the optical signal or if APDs are used there could be a very significant dark current.
- an optical receiver front-end is composed of a photo diode (PD) 61 followed by a trans-impedance amplifier (TIA) 62 and main amplifiers (MAs) 63 .
- the PD 61 receives a transmitted optical signal 64 and generates a current 66 proportional to the received optical power of the received optical signal 64 .
- the ratio between PD output current 6 to the input optical power of the optical signal 64 is the photo diode responsivity (R).
- the TIA 62 converts the PD current 66 to voltage, which is and then amplified by MAs 63 to the desired signal level.
- An active AC coupling circuitry is disposed between the photo diode (PD) 61 and the TIA 62 .
- An analog DC cancellation loop 67 is located prior to each input of the TIA 62 , and draws the photo diode DC current I DC , whereby only the AC signal I AC is coupled to the TIA 62 .
- the cancellation loop 67 comprises a trans-conductance (GM) cell 68 , and a low pass filter section 69 , e.g. comprised of a cancellation resistor R C and a cancellation capacitor C C .
- the input and output of the cancellation loop 67 is connected to the input of the TIA 62 , i.e.
- both the input and output of the loop 67 are between the PD 61 and the TIA 62 , as shown in FIG. 6 .
- Photo diode DC current I DC is drawn from the input node of the TIA 62 in the GM cell 68 , such that the analog loop 67 maintains the DC voltage value of the TIA input node to be the same as a reference voltage (V REF ).
- This reference voltage V REF is an input node voltage of a replica TIA, which represent the actual TIA input node voltage for a zero DC current condition.
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Abstract
Description
- TECHNICAL FIELD
- The present invention relates to an optical receiver, and in particular to an optical receiver with DC current cancellation.
- A typical optical receiver front-end is composed of a photo diode (PD) 1 followed by a trans-impedance amplifier (TIA) 2 and main amplifiers (MAs) 3 as shown in
FIG. 1 . ThePD 1 receives a transmittedoptical signal 4 and generates a current 6 proportional to the received optical power of the receivedoptical signal 4. The ratio betweenPD output current 6 to the input optical power of theoptical signal 4 is the photo diode responsivity (R). TheTIA 2 converts thePD current 6 to voltage, which is then amplified byMAs 3 to the desired signal level for thedecision circuitry 7. For NRZ modulation, the modulation depth of theoptical signal 4 is defined by its extinction ratio, which is the ratio between optical power for symbol one (P1) and optical power for symbol zero (P0). In high data rate receivers, the transmittedoptical signal 4 has poor extinction ratio and translates into a small modulated current with a large DC current at the output of thephoto diode 1. The large DC current saturates the receiver front-end (TIA 2 and MAs 3) and significantly degrades the gain and bandwidth performance. Consequently, cancelling photo diode DC current in high data rate receivers is desired for proper receiver operation, i.e. to have zero average modulatedPD current 6. - Moreover, for coherent optical communication links, mixing local laser power and the received modulated
optical signal 4 using thephoto diode 1 results in very large DC current. PD DC current is expressed as, -
I DC =R×(P LO +P S), (1) - where PLO is the local optical laser power and PS is the received optical signal power. Equation (1) shows that the photo diode output DC current of the
PD 1 in coherent optical communication links depends on the local laser power and the optical received power. For example, aphoto diode 1 with responsivity (R) of 1 A/W results in 4 mA DC current at 6 dBm local laser power input. Such a large DC current is more than enough to saturate the receiver front-end and severely degrades its performance. Thus, it is very important to have DC current cancellation circuitry in front of theTIA 2 of coherent optical communication links. -
FIG. 2 shows a conventional way to AC couplereceiver photo diodes 11 to the a front-end TIA 12 using passive AC coupling circuitry. An AC coupling capacitor (CC) is inserted between thephoto diode 11 and the front-end TIA 12 to block the DC current; however, it bypasses the modulated AC current to theTIA 12. A biasing resistor (RC) is used to bias the photo diode anode voltage to be reverse biased, and provides an alternative path for the photo diode DC current IDC. The biasing resistor RC with the AC coupling capacitor CC forms a high pass filter section in the RF signal path and its cutoff frequency (FC) is calculated as, -
- Typically, the required TIA low cutoff frequency (FC) is around 100 kHz which requires either large AC coupling capacitor CC or huge biasing resistor RC. As an example, a capacitor CC with a capacitance of at least 1.6 pF with a resistor RC with a resistance of at least 1 MΩ are required to achieve cutoff frequency of 100 kHz. Yet, this technique suffers from two main drawbacks: 1) CC parasitic capacitance, and 2) photo diode biasing. For bulk silicon technologies, the bottom plate ground parasitic capacitance of the coupling capacitor CC is around 10% of its value and degrades the front-end TIA bandwidth, which is defined by its input node capacitance. Thus, there is a maximum coupling capacitor (CC) that can be used without degrading the TIA bandwidth. On the other hand, the biasing voltage across the
photo diode 11 is defined by the following equation: -
V BIAS =V PD −V B −I DC ×R C, (3) - where VBIAS is the reverse biasing voltage across the photo diode PN junction. High photo diode reverse biasing voltage is required to obtain good photo diode responsivity and low PN junction capacitance. However, equation (3) shows that VBIAS depends on PD average current and leads to different PD biasing for different received optical power. Furthermore, a large RC value impedes receiving high optical power levels as the DC current will be large and the voltage drop across RC will be huge. As a numerical example, an I DC of 10 μA leads to a 10V drop on a 1 MΩ resistor RC, which is not practical. Moreover, the situation in coherent optical receivers is much worse as the photo diode DC current is around 1 mA and requires an RC of less than 1 kΩ for less than 1 V drop across the biasing resistor RC.
- An object of the present invention is to overcome the shortcomings of the prior art by providing a DC current cancellation loop for use with a fully differential front-end TIA structure.
- Accordingly, the present invention relates to an optical receiver comprising:
- a first photodetector (PD) for converting a first input optical signal into a first PD current comprising a first AC component and a first DC component;
- a transimpendance amplifier (TIA) for converting the first AC component into a first voltage signal; and
- a first DC cancellation loop including an input and an output between the first PD and the TIA for cancelling the first DC component, the first DC cancellation loop comprising:
- a first input and a first output connected to an input of the TIA;
- a first trans-conductance cell (GM), capable of drawing in the first DC component, such that the first DC cancellation loop maintains a first DC voltage value of the first input of the TIA the same as a first reference voltage (VREF1), which represents an actual TIA input voltage for a zero DC current condition; and
- a first low pass filter.
- The invention will be described in greater detail with reference to the accompanying drawings which represent preferred embodiments thereof, wherein:
-
FIG. 1 illustrates a schematic diagram of a conventional NRZ optical receiver; -
FIG. 2 illustrates a schematic diagram of a conventional differential optical receiver; -
FIG. 3 illustrates a schematic diagram of a fully differential optical receiver in accordance with an embodiment of the present invention; -
FIG. 4 illustrates a photodetector and differential TIA structure of the optical receiver ofFIG. 3 ; -
FIG. 5 illustrates an embodiment of a transconductance cell in accordance with the present invention; and -
FIG. 6 illustrates another embodiment of a photodetector and TIA structure of an optical receiver of the present invention. - While the present teachings are described in conjunction with various embodiments and examples, it is not intended that the present teachings be limited to such embodiments. On the contrary, the present teachings encompass various alternatives and equivalents, as will be appreciated by those of skill in the art.
-
FIG. 3 illustrates an embodiment of an electronic component in a packagedoptical receiver 30 in accordance with an embodiment of the present invention. An input signal enters at aport 31. A small portion of the input optical signal, e.g. less than 5%, may be split off and sent to amonitor photodiode 32, which generates an electrical signal that may be used to monitor properties of the input optical signal, such as its power content. In other embodiments, the power of the input optical signal can be monitored using different hardware. The remainder of the input optical signal may be sent through a variableoptical attenuator 33, which can adjust the signal intensity. A polarization beam splitter (PBS) 34 splits the remainder of the input optical signal into x-polarized (X-Pol) and y-polarized (Y-Pol) components. The X-Pol component is sent to a 90°hybrid mixer 36, and the Y-Pol component is sent to a 90°hybrid mixer 37. Simultaneously, alocal oscillator 38 provides a signal that is split by abeam splitter 39, and components of which are sent to each of 90°hybrid mixers hybrid mixers optical receivers 40, includingphotodetectors 41 and transimpedance amplifiers (TIA) 42. The electrical signals are then provided at four respective output terminals, which may be single-sided signals referenced to a common ground or may be differential signals. -
FIG. 4 illustrates an embodiment of theoptical receiver 40 including an active AC coupling circuitry disposed between the photo diode (PD) 41 and the front-end differential TIA 42. An analogDC cancellation loop TIA 42, and draws the photo diode DC current IDC, whereby only the AC signal IAC is coupled to theTIA 42. Two differentanalog cancellation loops cancellation loop pass filter section cancellation loop TIA input 42, i.e. both the input and output of theloops PD 41 and theTIA 42, as shown inFIG. 4 . Photo diode DC current IDC is drawn from the input node of theTIA 42 in the GM cell, such that theanalog loops DC cancellation loops TIA 42 and theoptical receiver 40, because if the DC current is not canceled, it would flow into theTIA 42 and change the bias point, output signal common mode, and may even completely saturate theTIA 42, rendering it useless. - A method of generating VREF includes using a
replica TIA 47, also shown inFIG. 4 . Thereplica TIA 47 includes floating inputs and outputs and substantially the same structure as themain TIA 42, in order to generate a correct VREF voltage, corresponding to a zero DC input current condition. Using areplica TIA 47 provides a simple way to track themain TIA 42 across process, voltage and temperature (PVT) variations, and to automatically generate a correct VREF in all cases. The trade-off of using thereplica TIA 47 is additional power dissipation. In order to minimize the additional power dissipation from thereplica TIA 47, it may be scaled down. For example, thereplica TIA 47 may use a fraction, e.g. ¼, of the bias current of themain TIA 42, flowing through resistors RF that are 4× larger than the resistors in themain TIA 42. Accordingly, the same VREF voltage is generated, while burning only ¼ of the power of themain TIA 42. Scaling will reduce the quality of the PVT tracking of thereplica TIA 47, so the scaling factor has to be selected carefully, to achieve the perfect balance between PVT tracking and power dissipation. It should be noted that VREF may be generated in a variety of other ways, without using a replica TIA. For example, a custom analog circuit may be constructed to simultaneously achieve the goals of correct VREF generation, with PVT tracking, and minimum power dissipation. - There are two important specifications that are advantageous from the
DC cancellation loops cancellation loops loops analog cancellation loops analog loop -
- where RF is the TIA feedback resistor and AO is the TIA feed-forward amplifier DC gain. The closed loop response of the
loops - 1) more convenient in realizing the low
pass filter section - 2) suitable for fully differential TIA topologies unlike prior art.
- The actual realization of the R-C section of the low pass filters 46 a and 46 b is more convenient in the proposed architecture than implementing passive AC coupling circuitry at the TIA input as in prior art because of two reasons: 1) there is no upper limit on the maximum value of the resistor RC as no DC current flows in it, and 2) CC parasitic capacitance doesn't harm the TIA bandwidth as it is placed away from the RF signal path between the GM cell input to the ground.
- Furthermore, the proposed architecture is more suitable for fully differential front-end TIA architecture than the prior art. In fully
differential TIAs 42, each output depends on both inputs due to the high common mode rejection of the fullydifferential TIA 42. Assuming an ideal differential amplifier AO employed in the TIA 42 (common mode gain=0, differential mode gain=∞), TIA output voltages (VOUTP, VOUIN) are expressed as, -
- where IP and IN are the input positive and negative currents of the
differential TIA 42, respectively. Equation (6) implies that both VOUTP and V OUTN depend on IP and IN with the same weight and opposite effect. Thus, the prior art cannot be employed with a fully differential TIA because the twocancellation loops cancellation loops DC cancellation loops differential TIA 42, because the sensing operation is performed at the input of theTIA 42. - One way to implement the trans-conductance (GM) cell with the
loop pass filter FIG. 5 for the proposed activeAC coupling circuitry 40. The trans-conductance cell GM draws a current (IOUT) proportional to the DC component of the differential input (VINP−VINN). The GM cell comprises avoltage amplifier 51 followed by a differential pair (T1, T2) of transistors. A current steering is performed linearly in the differential pair T1 and T2 using the differential input DC voltage (VINP−VINN) such that the output current equals IB/2*N at VINP=VINN and its maximum value is IB*N. The AC trans-conductance of the proposed GM cell is expressed as, -
GN=A×gm×N1 - where A is the gain of the
amplifier 51 and gm is the trans-conductance of the differential pair (T1, T2) each transistor having a resistor R connected thereto. N is a scaling factor for the output emitter-degeneratedcurrent mirror 52, where the outputbipolar transistor 55 is made N longer and thedegeneration resistor 56 is N times smaller. N may or may not be an integer. The scaling factor N enables the trans-conductance cell to operate with a smaller bias current from bias current source IB, and thereby to reduce power dissipation. The bias current source IB is connected to both of the transistors T1 and T2 of the differential pair. The output IOUT is connected to the second transistor T2 via anoutput transistor 55 and the output resistor 56 R/N. - Accordingly, to determine the value for N, the first step is to determine the value for IOUT or VREF, based on system requirements, i.e. how much DC current IDC needs to be cancelled. For example, if IOUT is 4 mA, it is undesirable to burn another 4 mA in the bias current IB. Accordingly, a scaling factor of, e.g. pick N=40, is selected, whereby IB=IOUT/N=100 uA, which adds only a small number to the overall power dissipation. In practice, N should not be too large, but typically does not have to be, because at some point, it does not make sense to push N to much higher values, as the goal of reducing power dissipation has typically already been achieved. Accordingly, N may be in the range from N=1 to N=1000, preferably N=4 to N=40, but it can be any another number, equal or greater than 1, N>1. Formally, there is no reason why N cannot be less than 1, but may be wasteful.
- The capacitor CC of the RC loop
pass filter section voltage amplifier 51 to boost its value by the voltage amplifier gain, which helps in reducing the size of the implementedR-C section -
- There are several reasons why a fully differential TIA front end is better than the most commonly used single-ended. First, the fully differential TIA front end has an excellent CMRR (common mode rejection ratio). Coherent receivers often have challenging CMRR requirements, and a fully differential TIA easily meets and exceeds most CMRR specs. Second, fully differential TIA front ends have better power supply and ground noise rejection. Third, the output of a fully differential TIA is fully compatible with the differential amplifier stages, so there is no need to convert the signal from single-ended to differential.
- However, the present invention may be used in non-coherent systems, since large DC currents can result from poor extinction ratio of the optical signal or if APDs are used there could be a very significant dark current.
- With reference to
FIG. 6 , an optical receiver front-end is composed of a photo diode (PD) 61 followed by a trans-impedance amplifier (TIA) 62 and main amplifiers (MAs) 63. ThePD 61 receives a transmittedoptical signal 64 and generates a current 66 proportional to the received optical power of the receivedoptical signal 64. The ratio between PD output current 6 to the input optical power of theoptical signal 64 is the photo diode responsivity (R). TheTIA 62 converts the PD current 66 to voltage, which is and then amplified byMAs 63 to the desired signal level. - An active AC coupling circuitry is disposed between the photo diode (PD) 61 and the
TIA 62. An analogDC cancellation loop 67 is located prior to each input of theTIA 62, and draws the photo diode DC current IDC, whereby only the AC signal IAC is coupled to theTIA 62. Thecancellation loop 67 comprises a trans-conductance (GM)cell 68, and a lowpass filter section 69, e.g. comprised of a cancellation resistor RC and a cancellation capacitor CC. The input and output of thecancellation loop 67 is connected to the input of theTIA 62, i.e. both the input and output of theloop 67 are between thePD 61 and theTIA 62, as shown inFIG. 6 . Photo diode DC current IDC is drawn from the input node of theTIA 62 in theGM cell 68, such that theanalog loop 67 maintains the DC voltage value of the TIA input node to be the same as a reference voltage (VREF). This reference voltage VREF is an input node voltage of a replica TIA, which represent the actual TIA input node voltage for a zero DC current condition. - The foregoing description of one or more embodiments of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not by this detailed description, but rather by the claims appended hereto.
Claims (11)
[[G M =A×g m ×N 1 ]]G M =A·g m ·N
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