US11693085B2 - FMCW radar with interference signal suppression - Google Patents
FMCW radar with interference signal suppression Download PDFInfo
- Publication number
- US11693085B2 US11693085B2 US16/656,726 US201916656726A US11693085B2 US 11693085 B2 US11693085 B2 US 11693085B2 US 201916656726 A US201916656726 A US 201916656726A US 11693085 B2 US11693085 B2 US 11693085B2
- Authority
- US
- United States
- Prior art keywords
- spectrum
- radar
- signal
- spectral lines
- baseband signal
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active, expires
Links
Images
Classifications
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/06—Systems determining position data of a target
- G01S13/08—Systems for measuring distance only
- G01S13/32—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
- G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
- G01S7/023—Interference mitigation, e.g. reducing or avoiding non-intentional interference with other HF-transmitters, base station transmitters for mobile communication or other radar systems, e.g. using electro-magnetic interference [EMI] reduction techniques
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/0209—Systems with very large relative bandwidth, i.e. larger than 10 %, e.g. baseband, pulse, carrier-free, ultrawideband
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/06—Systems determining position data of a target
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/06—Systems determining position data of a target
- G01S13/08—Systems for measuring distance only
- G01S13/32—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
- G01S13/34—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated using transmission of continuous, frequency-modulated waves while heterodyning the received signal, or a signal derived therefrom, with a locally-generated signal related to the contemporaneously transmitted signal
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
- G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
- G01S7/35—Details of non-pulse systems
- G01S7/352—Receivers
- G01S7/354—Extracting wanted echo-signals
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
- G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
- G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/06—Systems determining position data of a target
- G01S13/08—Systems for measuring distance only
- G01S13/32—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
- G01S13/34—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated using transmission of continuous, frequency-modulated waves while heterodyning the received signal, or a signal derived therefrom, with a locally-generated signal related to the contemporaneously transmitted signal
- G01S13/343—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated using transmission of continuous, frequency-modulated waves while heterodyning the received signal, or a signal derived therefrom, with a locally-generated signal related to the contemporaneously transmitted signal using sawtooth modulation
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
- G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
- G01S7/35—Details of non-pulse systems
- G01S7/352—Receivers
- G01S7/358—Receivers using I/Q processing
Definitions
- the present description relates to the field of radar sensors, in particular to signal processing methods which are used in radar sensors and which enable disturbing interference to be suppressed.
- Radar sensors are used in a multiplicity of applications for detecting objects, wherein the detecting usually comprises measuring distances and speeds of the detected objects.
- ADAS Advanced driver assistance systems
- cruise control e.g., Adaptive Cruise Control (ACC) or Radar Cruise Control
- ACC Adaptive Cruise Control
- Radar Cruise Control Adaptive Cruise Control
- Such systems can automatically adapt the speed of an automobile in order thus to maintain a safe distance from other automobiles ahead (and also other objects and pedestrians).
- Further applications in the automotive field are e.g. blind spot detection, lane change assist and the like.
- radar sensors will play an important part for the control of autonomous vehicles.
- a radar signal emitted by a first radar sensor (incorporated into a first vehicle) can be picked up by the receiving antenna of a second radar sensor (incorporated into a second vehicle).
- the first radar signal can interfere with an echo of the second radar signal and thereby impair the operation of the second radar sensor
- the method includes calculating a first spectrum, which represents a spectrum of a segment of a complex baseband signal (complex-valued baseband signal).
- the segment is assignable to a specific chirp of a chirp sequence contained in a first RF radar signal.
- the method further includes estimating a second spectrum, which represents a spectrum of an interference signal contained in the complex baseband signal, based on a portion of the first spectrum which is assigned to negative frequencies.
- the method includes calculating a first spectrum, which represents a spectrum of a segment of a baseband signal.
- the segment is assigned to a specific chirp of a chirp sequence contained in a first RF radar signal.
- the method further includes identifying spectral lines which can be assigned to a radar echo, and determining a second spectrum, which represents an estimated value for the spectrum of an interference signal contained in the baseband signal, based on the first spectrum. In this case, those spectral lines which can be assigned to a radar echo are disregarded.
- the radar device includes a radar transceiver having an oscillator and a receiving channel.
- the oscillator is configured to generate a first RF radar signal containing a chirp sequence.
- the receiving channel is configured to generate a complex baseband signal including a multiplicity of segments, wherein each segment is respectively assigned to a chirp of the chirp sequence.
- the device further includes a computing unit configured to calculate a first spectrum, which represents a spectrum of a segment of the complex baseband signal, and to estimate a second spectrum, which represents a spectrum of an interference signal contained in the complex baseband signal, based on a portion of the first spectrum which is assigned to negative frequencies.
- the radar device includes a radar transceiver having an oscillator and having a receiving channel.
- the oscillator is configured to generate a first RF radar signal containing a chirp sequence.
- the receiving channel is configured to generate a baseband signal, wherein the baseband signal includes a multiplicity of segments and each segment is assigned to a specific chirp of the chirp sequence.
- the device further includes a computing unit configured to calculate a first spectrum, which represents a spectrum of a segment of the baseband signal, to identify spectral lines which can be assigned to a radar echo, and to determine a second spectrum, which represents an estimated value for the spectrum of an interference signal contained in the baseband signal, based on the first spectrum. In this case, those spectral lines which can be assigned to a radar echo are disregarded.
- FIG. 1 is a schematic diagram for illustrating the functional principle of an FMCW radar system for distance and/or speed measurement.
- FIG. 2 comprises two timing diagrams for illustrating the frequency modulation (FM) of the RF signal generated by the FMCW system.
- FM frequency modulation
- FIG. 3 is a block diagram for illustrating the fundamental structure of an FMCW radar system.
- FIG. 4 is a schematic diagram for illustrating an example of how interference signals can be picked up by the receiving antenna of a radar sensor.
- FIG. 5 is a circuit diagram for illustrating a simplified example of a radar transceiver and of a further radar transceiver that causes interference.
- FIG. 6 shows, in a timing diagram (frequency versus time), one example of an emitted radar signal having a plurality of sequences of chirps, wherein each sequence has a specific number of chirps which are used for a measurement.
- FIG. 7 shows a timing diagram of a transmission signal of a radar sensor and an interference-causing transmission signal (interference signal) of a further radar sensor (interferer), wherein the signal profiles (frequency versus time) of these signals partly overlap.
- FIG. 8 shows a timing diagram of an exemplary signal profile of a radar signal (after mixing into baseband) including a radar echo from a radar target and an interference signal (interference)
- FIG. 9 illustrates by way of example the digital signal processing of radar signals during range Doppler analysis.
- FIG. 10 illustrates a modification of the example from FIG. 5 , wherein an IQ mixer is used in the receiving channel in order to obtain a complex baseband signal.
- FIG. 11 schematically illustrates the spectrum of a real radar signal and the spectrum of a complex radar signal in baseband.
- FIG. 12 schematically illustrates the estimation of the absolute value spectrum of interfering interference signals.
- FIG. 13 schematically illustrates the estimation of the phase spectrum of interfering interference signals.
- FIG. 14 illustrates the cancellation (in the frequency domain) of the interfering interference signals in the baseband signal.
- FIG. 15 is a flow diagram for summarizing the approach described here for cancelling the interference in the baseband radar signal.
- FIGS. 16 to 18 illustrate the interference signal estimation and reduction in the absolute value spectrum for real baseband signals.
- FIGS. 19 to 21 illustrate the interference signal estimation and reduction in the phase spectrum for real baseband signals.
- FIG. 22 illustrates the detection of zeros in the phase spectrum for real baseband signals.
- FIG. 1 illustrates, in a schematic diagram, the application of a frequency-modulated continuous-wave radar system—usually referred to as FMCW: radar system—as sensor for the measurement of distances and speeds of objects, which are usually referred to as radar targets.
- the radar device 1 comprises separate transmitting (TX) and receiving (RX) antennas 5 and 6 respectively (bistatic or pseudo-monostatic radar configuration). It should be noted, however, that a single antenna can also be used, which serves simultaneously as transmitting antenna and as receiving antenna (monostatic radar configuration).
- the transmitting antenna 5 emits a continuous RF signal s RF (t), which is frequency-modulated for example with a type of sawtooth signal (periodic, linear frequency ramp).
- FIG. 1 shows a simplified example; in practice, radar sensors are systems comprising a plurality of transmitting (TX) and receiving (RX) channels in order also to be able to determine the angle of incidence (Direction of Arrival, DoA) of the backscattered/reflected signal y RF (t) and thus to be able to localize the radar target T more accurately.
- TX transmitting
- RX receiving
- FIG. 2 illustrates by way of example the abovementioned frequency modulation of the signal s RF (t),
- the emitted RF signal s RF (t) is composed of a set of “chirps”, that is to say that the signal s RF (t) comprises a sequence of sinusoidal signal profiles (waveforms) having a rising frequency (up-chirp) or a falling frequency (down-chirp).
- the instantaneous frequency f(t) of a chirp beginning at a start frequency f START rises linearly within a time period T RAMP to a stop frequency f STOP (see lower diagram in FIG. 2 ).
- Such chirps are also referred to as linear frequency ramps.
- FIG. 2 illustrates three identical linear frequency ramps. It should be noted, however, that the parameters f START , f STOP , T RAMP and also the pause between the individual frequency ramps can vary. The frequency variation also need not necessarily be linear (linear chirp). Depending on the implementation, transmission signals with exponential or hyperbolic frequency variation (exponential or hyperbolic chirps, respectively) can also be used, for example.
- FIG. 3 is a block diagram which illustrates one possible structure of a radar device 1 (radar sensor) by way of example. Accordingly, at least one transmitting antenna 5 (TX antenna) and at least one receiving antenna 6 (RX antenna) are connected to an RE frontend 10 which is integrated in a chip and which can include all those circuit components which are required for the RF signal processing. Said circuit components comprise for example a local oscillator (LO), RF power amplifiers, low-noise amplifiers (LNAs), directional couplers (e.g. rat race couplers, circulators, etc.) and mixers for the down-conversion of the RF signals into baseband or an intermediate frequency band (IF band).
- LO local oscillator
- LNAs low-noise amplifiers
- directional couplers e.g. rat race couplers, circulators, etc.
- mixers for the down-conversion of the RF signals into baseband or an intermediate frequency band (IF band).
- the RE front end 10 can be integrated in a chip, which is usually referred to as a monolithic microwave integrated circuit (MMIC).
- MMIC monolithic microwave integrated circuit
- baseband and IF band are those signals on the basis of which the detection of radar targets is carried out.
- the example illustrated shows a bistatic (or pseudo-monostatic) radar system comprising separate RX and TX antennas.
- a directional coupler e.g. a circulator
- radar systems in practice usually comprise a plurality of transmitting and receiving channels having a plurality of transmitting and receiving antennas, respectively, which makes it possible, inter alia, to measure the direction (DoA) from which the radar echoes are received.
- DoA direction
- the individual TX channels and RX channels are usually constructed identically or similarly in each case.
- the RF signals emitted via the TX antenna 5 can lie e.g. in the range of approximately 20 GHz to 100 GHz (e.g. around 77 GHz in some applications).
- the RF signal received by the RX antenna 6 comprises the radar echoes (chirp echo signals), i.e. those signal components which are backscattered at one or at a plurality of radar targets.
- the received RF signal y RF (t) is e.g. down-converted to baseband (or an IF band) and processed further in baseband by means of analog signal processing (see FIG. 3 , analog baseband signal processing chain 20 ).
- the analog signal processing mentioned substantially comprises filtering and, if appropriate, amplification of the baseband signal.
- the baseband signal is finally digitized (see FIG. 3 , analog-to-digital converter 30 ) and processed further in the digital domain.
- the digital signal processing chain can be realized at least partly as software which can be executed on a processor, for example a microcontroller or a digital signal processor (see FIG. 3 , computing unit 40 ).
- the overall system is generally controlled by means of a system controller 50 , which can likewise be implemented at least partly as software which is executed on a processor such as e.g. a microcontroller.
- the RF frontend 10 and the analog baseband signal processing chain 20 (optionally also the analog-to-digital converter 30 and the computing unit 40 ) can be totally integrated in a single MMIC (i.e. an RF semiconductor chip). Alternatively, the individual components can also be distributed among a plurality of integrated circuits.
- FIG. 4 illustrates a simple example for illustrating how an interferer can interfere with the received radar echoes.
- FIG. 7 illustrates a road with three lanes and four vehicles V 1 , V 2 , V 3 and V 4 . At least the vehicles V 1 and V 4 are equipped with radar sensors.
- the radar sensor of the vehicle V 1 emits an RF radar signal s RF (t) and the received RF radar signal y RF (t) includes the radar echoes from the vehicles V 2 and V 3 ahead and also from the oncoming vehicle V 4 .
- the RF radar signal y RF (t) received by the radar sensor of the vehicle V 1 includes a radar signal (interference signal) that was generated by the radar sensor of the oncoming vehicle V 4 .
- the radar sensor of the vehicle V 4 is an interferer for the radar sensor of the vehicle V 1 .
- the signal y RF (t) received by the radar sensor of the vehicle V 1 can be written as follows:
- Equation (2) therefore represents the sum of the radar echoes caused by U different radar targets T i , wherein A T,i denotes the damping of the emitted radar signal and ⁇ t T,i denotes the round trip delay time (RTDT) for a specific radar target T i .
- Equation (3) equally represents the sum of the interference signals caused by V interferers.
- the amplitude of the received interference signal component y RF,I (t) may be significantly higher than the amplitude of the echo signal component y RF,T (t).
- FIG. 5 illustrates one exemplary implementation of a radar transceiver 1 in accordance with the example from FIG. 3 in greater detail.
- the present example illustrates in particular the RF frontend 10 of the radar transceiver 1 and the RE frontend 10 ′ of a different (interfering) radar sensor 1 ′.
- FIG. 5 illustrates a simplified circuit diagram in order to show the fundamental structure of the RE frontend 10 with one transmitting channel (TX channel) and one receiving channel (RX channel).
- TX channel transmitting channel
- RX channel receiving channel
- the RF frontend 10 comprises a local oscillator 101 (LO), which generates an RF oscillator signal s LO (t).
- the RF oscillator signal s LO (t) is frequency-modulated during operation, as described above with reference to FIG. 2 , and is also referred to as LO signal.
- the LO signal usually lies in the SHF (Super High Frequency, centimeter-wave) or in the EHT (Extremely High Frequency, millimeter-wave) band, e.g. in the interval of 76 GHZ to 81 GHz in some automotive applications.
- the LO signal s LO (t) is processed both in the transmission signal path TX 1 (in the TX channel) and in the reception signal path RX 1 (in the RX channel).
- the transmission signal s RF (t) (cf. FIG. 2 ), emitted by the TX antenna 5 , is generated by amplifying the LO signal s LO (t), for example by means of the RF power amplifier 102 , and is thus merely an amplified and possibly phase-shifted version of the LO signal s LO (t).
- the output of the amplifier 102 can be coupled to the TX antenna 5 (in the case of a bistatic or pseudo-monostatic radar configuration),
- the reception signal y RF (t) received by the RX antenna 6 is fed to the receiver circuit in the RX channel and thus directly or indirectly to the RF port of the mixer 104 .
- the RF reception signal y RF (t) (antenna signal) is preamplified by means of the amplifier 103 (gain g).
- the mixer 104 thus receives the amplified RE reception signal g ⁇ y RF (t).
- the amplifier 103 can be e.g. an LNA.
- the LO signal s LO (t) is fed to the reference port of the mixer 104 , such that the mixer 104 down-converts the (preamplified) RF reception signal y RF (t) to baseband.
- the down-converted baseband signal (mixer output signal) is designated by y BB (t).
- Said baseband signal y BB (t) is firstly processed further in analog fashion, wherein the analog baseband signal processing chain 20 substantially brings about amplification and (e.g. bandpass or low-pass) filleting in order to suppress undesired sidebands and image frequencies.
- the resulting analog output signal which is fed to an analog-to-digital converter (see FIG. 3 , ADC 30 ), is designated by y(t).
- Methods for the digital further processing of the digitized output signal are known per se (for example range Doppler analysis) and therefore will not be discussed in further detail here.
- the mixer 104 down-converters the preamplified RF reception signal g ⁇ y RF (t) (i.e. the amplified antenna signal) to baseband.
- the mixing can take place in one stage (that is to say from the RE band directly to baseband) or via one or more intermediate stages (that is to say from the RF band to an intermediate frequency band and further to baseband).
- the reception mixer 104 effectively comprises a plurality of individual mixer stages connected in series. In view of the example shown in FIG.
- FIG. 5 furthermore shows a portion (the TX channel of the RF frontend 10 ′) of a further radar sensor 1 ′, which constitutes an interferer for the radar sensor 1 .
- the RF frontend 10 ′ of the radar sensor 1 ′ includes a further local oscillator 101 ′, which generates an LO signal s LO ′(t), which is amplified by the amplifier 102 ′.
- the amplified LO signal is emitted as RF radar signal s RF,0 ′(t) via the antenna 5 ′ of the radar sensor 1 ′ (cf. equation (3)).
- This RF radar signal s RF,0 ′(t) contributes to the interference signal component y RF,I (t) received by the antenna 6 of the other radar sensor 1 and causes the interference mentioned.
- FIG. 6 schematically illustrates one example of an scheme such as is usually used during the frequency modulation of the LO signal s LO (t) in FMCW radar sensors.
- a sequence of chirps is generated for each measurement.
- the first sequence contains only 16 chirps; in practice, however, a sequence will have significantly more chirps, for example 128 or 256 chirps.
- a number corresponding to a power of two allows the use of efficient FFT (Fast Fourier Transform) algorithms during the subsequent digital signal processing (e.g. during the range Doppler analysis). There may be a pause between the individual sequences.
- FFT Fast Fourier Transform
- FIGS. 7 and 8 illustrate, on the basis of one example, how an interferer can interfere with the radar echoes contained in the RF signal y RF (t) received by the radar sensor 1 .
- FIG. 7 shows, in a diagram (frequency versus time), a chirp emitted by the radar signal 1 and having a chirp duration of 60 ⁇ s.
- the start frequency of the emitted signal s RF (t) is approximately 76250 MHz and the stop frequency is approximately 76600 MHz.
- An interference signal y RF,I (t) generated by a different radar sensor includes an up-chirp having a start frequency of approximately 76100 MHz, a stop frequency of approximately 76580 MHz and a chirp duration of 30 ⁇ s, and a succeeding down-chirp that starts at the stop frequency of the preceding chirp and ends at the start frequency of the preceding chirp and has a chirp duration of 10 ⁇ s.
- the bandwidth of the baseband signal of the radar sensor is substantially determined by the baseband signal processing chain 20 and is indicated by the dashed lines in FIG. 7 .
- FIG. 8 shows one exemplary signal profile of the (preprocessed) baseband signal y(t) of the radar sensor 1 .
- the signal components on account of the interference have a significant amplitude in that time interval in which the frequency of the interference signal lies within the bandwidth B of the radar sensor (see FIGS. 7 and 8 ).
- the interference occurs three times during the chirp duration of 60 ⁇ s, namely at approximately 7 ⁇ s, 28 ⁇ s and 42 ⁇ s.
- the power of the interference signal can be higher than the power of the radar echoes from real targets.
- the interference signals and the transmission signal of the radar sensor 1 considered are uncorrelated, for which reason the interference can be regarded as noise and thus increases the noise floor.
- FIG. 9 illustrates, on the basis of one example, the analog signal processing in a radar sensor through to the digitization of the baseband signal representing the chirp echo signals.
- Diagram (a) from FIG. 9 shows part of a chirp sequence comprising M linear chirps.
- the solid line represents the signal profile (waveform frequency versus time) of the outgoing RF radar signal s RF (t) and the dashed line represents the corresponding signal profile of the incoming radar signal y RF (t), which comprises (if present) the chirp echoes.
- the frequency of the outgoing radar signal beginning at a start frequency f START , rises linearly up to a stop frequency f STOP (chirp No. 0) and then falls back to the start frequency f START , rises again up to the stop frequency f STOP (chirp No. 0, and so on.
- a chirp sequence comprises a multiplicity of chirps; in the present case, the number of chirps in a sequence is designated by M.
- a sequence can also include chirps having different parameters (start and stop frequency, duration and modulation pause).
- start and stop frequency, duration and modulation pause can be e.g. equal to the stop frequency of the previous chirp or to the start frequency of the subsequent chirp (or equal to some other frequency).
- the chirp duration can lie in the range of from a few microseconds to a few milliseconds, for example in the range of 20 ⁇ s to 2 ms. The actual values can also be larger or smaller depending on the application.
- the incoming RF radar signal y RF (t) i.e. that which is received by the RX antenna
- s RF (t) i.e. that which is emitted by the TX antenna
- This time difference ⁇ t corresponds to the signal propagation time from the TX antenna, to the radar target and back to the RX antenna and is also referred to as the Round Trip Delay Time (RTDT).
- RTDT Round Trip Delay Time
- the time difference ⁇ t results in a corresponding frequency difference ⁇ f.
- This frequency difference ⁇ f can be determined by mixing the incoming (and possibly preamplified) radar signal y RF (t) with the LO signal s LO (t) of the radar sensor (see FIG. 5 , mixer 104 ), digitizing the resulting baseband signal y(t) and then carrying out a digital spectral analysis. The frequency difference ⁇ f then appears as a beat frequency in the spectrum of the digitized baseband signal y[n].
- an additional Doppler shift f D in the incoming signal on account of the Doppler effect can influence the distance measurement since the Doppler shift f D is added to the frequency difference ⁇ f explained above.
- the Doppler shift can be estimated/calculated from the outgoing and incoming radar signals and can be taken into account in the measurement, whereas in some applications the Doppler shift may be negligible for the distance measurement. That may be the case e.g.
- the Doppler shift can be eliminated by determining the distance on the basis of an up-chirp and a down-chirp during the distance measurement. Theoretically, the actual distance d T can be calculated as a mean value of the distance values obtained from a measurement with up-chirps and a further measurement with down-chirps. The Doppler shift is eliminated by the averaging.
- FMCW radar sensors determine the target information distance, speed, DoA) by emitting a sequence of chirps (see FIG. 9 , diagram (a)) and mixing the (delayed) Times from the radar targets with a “copy” of the emitted signal (cf. FIG. 5 , mixer 104 ).
- the resulting baseband signal y(t) is illustrated in diagram (b) in FIG. 9 .
- This baseband signal y(t) can be subdivided into a plurality of segments, wherein each segment of the baseband signal y(t) is assigned to a specific chirp of the chirp sequence.
- the target information mentioned can be extracted from the spectrum of the abovementioned segments of the baseband signal y(t) which contain the chirp echoes generated by one or more radar targets.
- a range Doppler map is obtained by means of a two-stage Fourier transformation, for example. Range Doppler maps can be used as a basis for various methods for detection, identification and classification of radar targets. The result of the first Fourier transformation stage is referred to as a range map.
- the interference signal suppression methods described here can be carried out in the spectra of the abovementioned segments of the baseband signal which are contained in such a range map.
- the calculations required for determining the range Doppler maps are carried out by a digital computing unit such as e.g. a signal processor (cf. FIG. 5 , DSP 40 ).
- a digital computing unit such as e.g. a signal processor (cf. FIG. 5 , DSP 40 ).
- other computing units can also be used to carry out the required calculations.
- the calculations can be carried out by various software and hardware units (software and hardware entities) or combinations thereof.
- the term computing unit is understood here to mean any desired combination of software and hardware which is able and configured to carry out the calculations described in association with the exemplary embodiments explained here.
- the calculation of a range Doppler map includes two stages, wherein a plurality of Fourier transformations are calculated (e.g. by means of an FFT algorithm) in each stage.
- the baseband signal y(t) (cf. FIG. 5 ) is sampled such that N ⁇ M samples are obtained for a chirp sequence having M chirps, i.e. M segments with N samples in each case. That is to say that the sampling time interval T SAMPLE is chosen such that each of the M segments (chirp echoes in baseband) is represented by a sequence of N samples.
- T SAMPLE is chosen such that each of the M segments (chirp echoes in baseband) is represented by a sequence of N samples.
- said M segments each associated with N samples can be arranged in a two-dimensional array Y[n, m] (radar data array).
- Each column of the array Y[n, m] represents one of the M considered segments of the baseband signal y(t), and the n-th row of the array Y[n, m] contains the n-th sample of the M chirps.
- the column index m corresponds to the number of the chirp in a chirp sequence.
- a first FFT (usually referred to as range FFT) is applied to each chirp.
- the Fourier transformation is calculated for each column of the array Y[n, m].
- the array Y[n, m] is Fourier-transformed along the fast time axis, and the result obtained is a two-dimensional array R[k, m] of spectra, which is referred to as a range map, wherein each of the M columns of the range map contains in each case N (complex-valued) spectral values.
- the “fast” time axis becomes the frequency axis;
- the row index k of the range map R[k, m] corresponds to a discrete frequency and is thus also referred to as a frequency bin.
- Each discrete frequency corresponds to a distance in accordance with equation 4, for which reason the frequency axis is also referred to as a distance axis (Range Axis).
- the range map R[k, m] is illustrated in diagram (c) in FIG. 9 .
- a radar echo caused by a radar target results in a local maximum (Peak) at a specific frequency index/frequency bin.
- Said local maximum usually appears in all the columns of the range map R[k, m], i.e. in the spectra of all considered segments of the baseband signal y(t) which can be assigned to the chirps of a chirp sequence.
- the associated frequency index k (e.g. in accordance with equation 4) can be converted into a distance value.
- Each row of the range map R[k, m] includes M spectral values of a specific frequency bin, wherein each frequency bin corresponds to a specific distance d Ti of a specific radar target T i .
- the Fourier transformation of the spectral values in a specific frequency bin makes it possible to determine the associated Doppler shift f D , corresponding to a speed of the radar target.
- the two-dimensional array R[k, m] (the range map) is Fourier-transformed row by row, i.e. along the “slow” time axis.
- the “slow” time axis becomes the Doppler frequency axis.
- the associated discrete Doppler frequency values respectively correspond to a specific speed.
- the Doppler frequency axis can accordingly be converted into a speed axis.
- Each local maximum (each peak) in the range Doppler map X[k, l] indicates a potential radar target.
- the row index k (on the range axis) assigned to a local maximum represents the distance of the target
- the column index 1 (on the speed axis) assigned to the local maximum represents the speed of the target.
- a range map and a range Doppler map X a [k, l] for each RX channel, wherein a denotes the number of the antenna and of the associated RX channel.
- the range Doppler maps X a [k, l] can be “stacked” to form a three-dimensional array.
- the output data Y a [m, n] can be regarded as a three-dimensional array. The latter is sometimes referred to as a radar data cube.
- the radar data cubes, the resulting range maps R a [k, m] or the range Doppler maps X a [k, l] can be used as input data for various further signal processing methods.
- various peak detection algorithms are known for detecting, in the range maps R a [n, m] or the range Doppler maps X a [k, l], local maxima (peaks) caused by an object (radar target) in the “field of view” of the radar sensor.
- Other algorithms serve e.g. for calculating the (azimuth) angle of a radar target or for classifying detected radar targets (e.g. whether a radar target is a pedestrian).
- the spectral values in a range map or a range Doppler map contain noise.
- the detectability of the abovementioned local maxima and the reliability of the detection depend on the noise floor of the radar system.
- Various noise sources can contribute to the noise floor, in particular the phase noise of the local oscillator (see FIG. 4 , LO 101 ).
- the interference effects on account of other, interfering radar sensors as discussed further above can also adversely influence the detection of radar targets and the robustness and reliability of the measurement results.
- the interference mentioned can at least temporarily increase the noise floor to such a great extent that detection of radar targets becomes impossible or at least susceptible to errors.
- FIG. 10 shows an RE frontend 10 of a radar sensor having an RX channel RX 1 and a TX channel TX 1 .
- the real part y BB (t) is also referred to as the in-phase component, and the imaginary part y BB ′(t) as the quadrature component.
- IQ mixer IQ demodulator
- the analog baseband signal processing chain has to be duplicated, i.e. the signal processing chain 20 for the real part and a corresponding signal processing chain 20 ′ for the imaginary part.
- the output signals y(t) and y′(t) are digitized by means of the analog-to-digital converter unit 30 (having two channels).
- the local oscillator 101 is configured to feed, in addition to the “normal” LO signal s LO (t), also an LO signal s LO ′(t) orthogonal thereto (phase-shifted by 90°), wherein in the RX channel the LO signal s LO ′(t) is fed to the reference input of the mixer 104 and the corresponding LO signal s LO ′(t) is fed to the reference input of the mixer 104 ′.
- the subsequent digital signal processing with the use of an IQ mixer is not substantially different than with the use of a “normal” mixer as in the example from FIG. 5 .
- the radar Doppler analysis in the frequency domain as summarized above can also be carried out with complex-valued signals. Radar sensors with IQ mixers in the receiving channel are known per se and will therefore not be discussed in greater detail here.
- FIG. 11 illustrates the spectrum of a radar signal in baseband with complex demodulation (by means of an IQ mixer, see FIG. 9 ) in comparison with the spectrum of a radar signal in baseband with real demodulation (see FIG. 5 ).
- Spectra of real signals are always symmetrical, i.e.
- and arg ⁇ Y[k] ⁇ ⁇ arg ⁇ Y[ ⁇ k] ⁇ , wherein in this example Y[k] is the spectrum of a real baseband signal y[n] (cf. FIG. 5 ).
- the spectra Y*[k] of complex signals y*[n] are not symmetrical.
- each radar echo at a target leads to a local maximum at positive frequency f 1 , f 2 and a corresponding local maximum at negative frequency ⁇ f 1 , ⁇ f 2 .
- the signal component y T *[n] radar echoes at real targets
- the signal component y I *[n] disurbing interference
- the spectrum of the (complex) signal component y T *[n] is designated by Y T *[k]
- the spectrum of the (likewise complex) signal component y I *[n] is designated by Y I *[k].
- Y *[ k ] Y T *[ k ]+ Y I *[ k ].
- k denotes the frequency index
- f k ⁇ f
- ⁇ f denotes the frequency resolution in the present example.
- the (discrete) spectrum Y*[k] can represent for example a column of a range map R[k, m] provided that the range map R[k, m] was calculated on the basis of a complex baseband signal y*[n] (having M segments/chirps).
- the signal component y T *[n] has spectral lines only at positive frequencies, which spectral lines can represent in each case a real radar target, i.e.
- zero is a theoretical value that does not take account of noise.
- Theoretical work has shown that the signal component y I *[n] has a symmetrical absolute value spectrum, i.e.
- Various parameter estimation methods known per se can be used for this purpose, for example the method of least mean squares (LMS method) or the like.
- LMS method least mean squares
- the phase spectrum in accordance with equation 10 can also be extrapolated for positive frequencies (k>0).
- a spectrum Y*[k] comprises an even number of complex-valued spectral lines; the frequency index k in this case ranges from ⁇ N/2 to N/2 ⁇ 1.
- the interference signal component can be eliminated from the complex radar signal y*[n], which after all includes radar echoes and interference signals, by means of subtraction (cancelling out).
- This procedure is carried out in the frequency domain and is illustrated graphically in FIGS. 12 to 14 , wherein absolute value spectrum and phase spectrum of the interference signal component y I *[n] are estimated separately.
- the approach described below relates to interference signal suppression in the frequency domain for a complex baseband signal y*[n].
- the spectrum Y*[k] comprises a first portion (left-hand side of the spectrum), which is assigned to negative frequencies, and a second portion (right-hand side of the spectrum), which is assigned to positive frequencies.
- diagram (a) the right-hand portion of the spectrum is illustrated as a dashed line.
- the left-hand portion of the spectrum Y*[k] does not contain echo signals generated from real radar targets, but rather only noise and interference.
- the aforementioned mirroring is often also referred to as “flipping”. In some programming languages there are even specific commands for this operation, such as e.g. “fliplr” (“flip from left to right”).
- of the currently considered segment of the baseband signal y*[n] is a suitable estimation for the absolute value spectrum
- of the currently considered segment of the baseband signal y*[n] is a suitable estimation for the absolute value spectrum
- of the interference signal component y I *[n] is illustrated in diagram (b) in FIG. 12 .
- diagram (b) represents only the absolute value spectrum
- phase spectrum arg ⁇ Y I *[k] ⁇ can be modelled as a second degree polynomial (see equations 10 and 11). The parameters of this model can be estimated from the left-hand portion (assigned to negative frequencies) of the phase spectrum arg ⁇ Y*[k] ⁇ (i.e.
- FIG. 13 Diagram (a) from FIG. 13 shows the phase spectrum arg ⁇ Y*[k] ⁇ , wherein the right-hand portion of the phase spectrum arg ⁇ Y*[k] ⁇ (which can also be influenced by real radar echoes) is illustrated in a dashed manner.
- Diagram (b) from FIG. 13 shows the extrapolated phase spectrum arg ⁇ Y* 1 [k] ⁇ of the interference signal component y I *[n].
- the interference signal component y I *[n] in the frequency domain can be implemented separately for each segment of the baseband signal y*[n].
- the interference signal components can thus be cancelled column by column, with a range map R[k, m] for each column, wherein a range map R[k, m] can be determined for each chirp sequence and each receiving channel.
- FIG. 15 visualizes the example with the aid of a flowchart.
- the method uses a complex baseband signal y*[n] of a radar transceiver with IQ mixer in the receiving channel (see FIG. 10 ).
- Said baseband signal y*[n] comprises a multiplicity of segments, wherein each segment corresponds to a chirp of a chirp sequence contained in the emitted RF radar signal.
- the samples of the baseband signal y*[n] can be organized as a matrix, wherein each column of the matrix includes a segment.
- the method comprises calculating the spectrum Y*[k] (first spectrum) of a segment of the complex baseband signal y*(t) (see FIG. 15 , step S 1 ). This calculation can be carried out in the course of the calculation—described further above—of a range map.
- a range map contains, in the columns, the spectra of (temporally directly) successive segments of the complex baseband signal y*[n].
- step S 2 involves estimating the spectrum Y* 1 [k] (second spectrum) of the interference signal component y I *[n] (generated by interference) contained in the baseband signal segment considered; this estimation is based on that portion of the first spectrum Y*[k] which is assigned to negative frequencies (i.e. Y*[k] for k ⁇ 0).
- step S 3 involves cancelling the interference signal component y I *[n] contained in the complex baseband signal segment y*[n] in the frequency domain.
- the estimation of the second spectrum Y I *[k] is done separately for the absolute value spectrum
- is obtained by mirroring the left-hand portion (negative frequencies) of the absolute value of the first spectrum
- the phase spectrum arg ⁇ Y I *[k] ⁇ is obtained by means of a model-based extrapolation of the left-hand portion (negative frequencies) of the phase spectrum arg ⁇ Y*[k] ⁇ (see equations 10 and 11).
- the method described above presupposes a complex baseband signal y*[n], for which an RF frontend with an IQ mixer is required.
- a description is given below of a modification of the approach described above, which can also be applied to a real baseband signal y[n] and thus also functions for RF frontends with a simple mixer.
- the spectra of real signals are always symmetrical with respect to the zero hertz line (cf. FIG. 11 and the associated explanations). That is to say that the absolute value spectrum
- FIG. 16 A local maximum (Peak) representing a real radar echo is evident approximately at 11 MHz.
- the local maxima are detected in the calculated absolute value spectrum
- This detection can be carried out, as illustrated by way of example in FIG. 16 , by means of comparison with a threshold value Y TH , that is to say that those frequency indices k p (frequency bins, correspond respectively to a frequency f p ) for which
- k p frequency bins k p (and possibly adjacent bins) that are assignable to a real radar echo are not taken into account in the estimation of the interference signal spectrum Y I [k].
- the result of the threshold value comparison may be a set P of frequency indices assigned to real radar echoes.
- the estimation of the interference signal spectrum Y I [k] is based on the calculated spectrum Y[k] of the real baseband signal (e.g. a column of a range map), wherein those frequency ranges (frequency bins) which can be assigned to a radar echo are disregarded. That is to say that, for the frequency bins k ⁇ k p , the absolute value spectrum is approximated as follows:
- Various interpolation methods known per se are applicable here, for example an interpolation by means of cubic splines. However, other methods known per se are also applicable.
- can be seen in FIG. 17 .
- FIG. 18 shows the difference
- An estimated value for the phase spectrum arg ⁇ Y I [k] ⁇ can be ascertained by means of a piecewise linear interpolation in the previously calculated phase spectrum arg ⁇ Y[k] ⁇ , for k ⁇ k p .
- the value in the phase spectrum arg ⁇ Y[k] ⁇ may be unreliable. Accordingly, during the calculation of the estimated value, those frequency ranges k p (frequency bins) which can be assigned to a radar echo and those frequency bins k z which can be assigned to zeros can be disregarded.
- FIG. 19 shows by way of example the phase spectrum arg ⁇ Y[k] ⁇ associated with FIG. 16 ; FIG.
- phase spectrum arg ⁇ Y I [k] ⁇ can be calculated as a regression line on the basis of the phase spectrum arg ⁇ Y[k] ⁇ (e.g. column of a range map) for k ⁇ k p and k ⁇ k z .
- spectral lines Y[k p ] and possibly Y[k z ] are disregarded for the calculation of the estimated value for arg ⁇ Y[k] ⁇ .
- FIG. 21 shows the difference arg ⁇ Y[k] ⁇ Y I [k] ⁇ .
- FIG. 22 illustrates by way of example a possibility for identifying those frequency bins k z which can be assigned to a zero.
- the angle differences that is to say arg ⁇ Y[k] ⁇ arg ⁇ Y[k ⁇ 1] ⁇ , are represented on the ordinate axis. Since sudden phase changes occur in the case of zeroes, local minima and maxima become evident in the case of the zeroes and can likewise be detected by means of a comparison with a threshold value ⁇ THL and ⁇ THU , respectively.
- the frequency bins k z at which a zero is detected can be disregarded in the approximation of the phase spectrum arg ⁇ Y I [k] ⁇ .
- can also be set to zero.
- adjacent frequency bins e.g. k z ⁇ 2, k z ⁇ 1, k z , k z +1 and k z +2
- FIG. 22 points are also depicted as “outside the threshold values” even though they lie between the threshold values ⁇ THL to ⁇ THU but adjoin a frequency bin whose phase value lies outside. How many frequency bins around a zero are “sorted out” in this way is dependent on the actual implementation and may also be dependent on the numerical accuracy of the calculations.
- computing unit is understood to mean any functional unit (entity), which can comprise software and hardware, which is suitable and configured for carrying out the method steps described here.
- functional unit entity
- efficient hardware structures are also known alongside software algorithms.
- range maps and range Doppler maps mentioned here need not necessarily be represented as a two-dimensional data structure. The actual structure used may deviate from the structure described here, depending on the implementation.
Abstract
Description
In equations (1) to (3) above, the signal components yRF,T(t) and yRF,I(t) of the received signal yRF(t) correspond to the radar echoes from real radar targets Ti or to the interference signals. In practice, a plurality of radar echoes and a plurality of interferers may be present. Equation (2) therefore represents the sum of the radar echoes caused by U different radar targets Ti, wherein AT,i denotes the damping of the emitted radar signal and ΔtT,i denotes the round trip delay time (RTDT) for a specific radar target Ti. Equation (3) equally represents the sum of the interference signals caused by V interferers. In this case, AI,k denotes the damping of the interference signal sRF,k′(t) emitted by an interferer and ΔtI,k denotes the associated signal propagation time (for each interferer k=0, 1, . . . , V−1). It should be noted that the radar signal sRF(t) emitted by the vehicle V1 and the interference signal sRF,0′(t) emitted by the vehicle V4 (index k=0 for vehicle V4) will generally have different chirp sequences having different chirp parameters (start/stop frequency, chirp duration, repetition rate, etc.). Furthermore, the amplitude of the received interference signal component yRF,I(t) may be significantly higher than the amplitude of the echo signal component yRF,T(t).
d Ti =c·Δt/2=c·Δf·T CHIRP/(2·B) (4)
y*[n]=y T*[n]=+y I*[n] (5)
wherein yT*[n] denotes the signal component on account of echoes at real radar targets and yI*[n] denotes the signal component on account of interference of interference signals.
Y*[k]=Y T*[k]+Y I*[k]. (6)
In
|Y T*[k]|≈0 for k<0. (7)
Of course, zero is a theoretical value that does not take account of noise. Theoretical work has shown that the signal component yI*[n] has a symmetrical absolute value spectrum, i.e.
|Y I*[k]|=|Y I*[−k] (8)
This has the consequence that the spectrum of the interference signals can be “extracted” (estimated) directly from the overall spectrum Y*[k]. The following equation
denotes a sufficient accurate estimation for the absolute value spectrum |YI*[k]| of the signal component yI*[n] which represents interference signals.
arg{Y I*[k]}=c 1 k 2 +c 2 k+c 3, (10)
wherein c1, c2 and c3 are constant parameters that can be calculated (estimated) from the spectrum Y*[k] for k<0. Various parameter estimation methods known per se can be used for this purpose, for example the method of least mean squares (LMS method) or the like. For positive frequencies, the phase spectrum in accordance with
Y* corr[k]=Y*[k]−Y I*[k]. (12)
As mentioned, the interference signal component yI*[n] in the frequency domain can be implemented separately for each segment of the baseband signal y*[n]. The interference signal components can thus be cancelled column by column, with a range map R[k, m] for each column, wherein a range map R[k, m] can be determined for each chirp sequence and each receiving channel.
Y corr[k]=Y[k]−Y I[k]. (13)
As already explained, the spectra of real signals are always symmetrical with respect to the zero hertz line (cf.
P={k p−2,k p−1,k p, k p+1,k p+2}.
|Y I[k]|≈|Y[k]|, for k∉P, (14)
wherein the “gaps” in the case of the frequency bins k∈P are closed by means of interpolation. Various interpolation methods known per se are applicable here, for example an interpolation by means of cubic splines. However, other methods known per se are also applicable. One example of the estimated absolute value spectrum |YI[k]| can be seen in
Claims (21)
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
DE102018126034.2 | 2018-10-19 | ||
DE102018126034.2A DE102018126034A1 (en) | 2018-10-19 | 2018-10-19 | FMCW RADAR WITH INTERFERENCE CANCELLATION |
Publications (2)
Publication Number | Publication Date |
---|---|
US20200124699A1 US20200124699A1 (en) | 2020-04-23 |
US11693085B2 true US11693085B2 (en) | 2023-07-04 |
Family
ID=70278891
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US16/656,726 Active 2041-06-13 US11693085B2 (en) | 2018-10-19 | 2019-10-18 | FMCW radar with interference signal suppression |
Country Status (4)
Country | Link |
---|---|
US (1) | US11693085B2 (en) |
JP (1) | JP2020067455A (en) |
CN (1) | CN111077514A (en) |
DE (1) | DE102018126034A1 (en) |
Families Citing this family (15)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
DE102018102816B3 (en) * | 2018-02-08 | 2019-07-04 | Infineon Technologies Ag | RADAR WITH PHASE CORRECTION |
TWI669522B (en) | 2018-06-28 | 2019-08-21 | 立積電子股份有限公司 | Doppler signal processing device and signal processing method |
US11885874B2 (en) * | 2018-12-19 | 2024-01-30 | Semiconductor Components Industries, Llc | Acoustic distance measuring circuit and method for low frequency modulated (LFM) chirp signals |
JP7188319B2 (en) * | 2019-08-07 | 2022-12-13 | 株式会社Soken | drive |
EP3816665A1 (en) * | 2019-11-04 | 2021-05-05 | Nxp B.V. | Method of interference suppression in a fmcw radar system |
US11852719B2 (en) * | 2020-11-23 | 2023-12-26 | Qualcomm Incorporated | Coordinated interference cleaning with known interferer location and timing |
CN112698293B (en) * | 2020-12-21 | 2022-11-08 | 广州极飞科技股份有限公司 | Radar signal processing method and device and aircraft |
TWI802001B (en) * | 2021-04-13 | 2023-05-11 | 立積電子股份有限公司 | Doppler signal processing device and doppler signal processing method |
JP2022180111A (en) * | 2021-05-24 | 2022-12-06 | ソニーセミコンダクタソリューションズ株式会社 | Information processing device and information processing method |
EP4105675B1 (en) * | 2021-06-15 | 2024-04-03 | GM Cruise Holdings LLC | Interference mitigation in a fmcw radar system |
TWI771103B (en) * | 2021-07-14 | 2022-07-11 | 立積電子股份有限公司 | Radar apparatus and signal receiving method thereof |
US20230273296A1 (en) * | 2022-01-28 | 2023-08-31 | Qualcomm Incorporated | Noise estimation with signal ramps for radar |
DE102022123720A1 (en) | 2022-09-16 | 2024-03-21 | Bayerische Motoren Werke Aktiengesellschaft | Method for the holistically optimized operation of a radar system and associated radar system, motor vehicle and server device |
DE102022123718A1 (en) | 2022-09-16 | 2024-03-21 | Bayerische Motoren Werke Aktiengesellschaft | Method for operating a radar system in fault situations, radar system and motor vehicle equipped therewith |
CN116068502B (en) * | 2023-04-06 | 2023-06-16 | 中国人民解放军空军预警学院 | Multi-domain combined anti-composite interference method, device and system |
Citations (10)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6704438B1 (en) * | 2000-05-08 | 2004-03-09 | Aloka Co., Ltd. | Apparatus and method for improving the signal to noise ratio on ultrasound images using coded waveforms |
US20060125682A1 (en) * | 2004-12-15 | 2006-06-15 | Kelly Thomas M Jr | System and method for reducing a radar interference signal |
EP3173812A1 (en) * | 2015-11-24 | 2017-05-31 | Autoliv Development AB | A vehicle radar system arranged for reducing interference |
US9806914B1 (en) * | 2016-04-25 | 2017-10-31 | Uhnder, Inc. | Successive signal interference mitigation |
EP3244229A1 (en) * | 2016-05-13 | 2017-11-15 | Autoliv Development AB | A vehicle radar system arranged for interference reduction |
US20180074168A1 (en) * | 2015-04-15 | 2018-03-15 | Texas Instruments Incorporated | Noise mitigation in radar systems |
US9952312B2 (en) * | 2015-07-06 | 2018-04-24 | Navico Holding As | Radar interference mitigation |
US10078131B2 (en) * | 2015-09-15 | 2018-09-18 | Texas Instruments Incorporated | Method and apparatus for FMCW radar processing |
US20190129026A1 (en) * | 2015-06-04 | 2019-05-02 | Chikayoshi Sumi | Measurement and imaging instruments and beamforming method |
EP3489710A1 (en) * | 2017-11-23 | 2019-05-29 | Veoneer Sweden AB | Radar interference suppression |
-
2018
- 2018-10-19 DE DE102018126034.2A patent/DE102018126034A1/en active Pending
-
2019
- 2019-10-18 US US16/656,726 patent/US11693085B2/en active Active
- 2019-10-18 JP JP2019190794A patent/JP2020067455A/en active Pending
- 2019-10-21 CN CN201911000469.4A patent/CN111077514A/en active Pending
Patent Citations (10)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6704438B1 (en) * | 2000-05-08 | 2004-03-09 | Aloka Co., Ltd. | Apparatus and method for improving the signal to noise ratio on ultrasound images using coded waveforms |
US20060125682A1 (en) * | 2004-12-15 | 2006-06-15 | Kelly Thomas M Jr | System and method for reducing a radar interference signal |
US20180074168A1 (en) * | 2015-04-15 | 2018-03-15 | Texas Instruments Incorporated | Noise mitigation in radar systems |
US20190129026A1 (en) * | 2015-06-04 | 2019-05-02 | Chikayoshi Sumi | Measurement and imaging instruments and beamforming method |
US9952312B2 (en) * | 2015-07-06 | 2018-04-24 | Navico Holding As | Radar interference mitigation |
US10078131B2 (en) * | 2015-09-15 | 2018-09-18 | Texas Instruments Incorporated | Method and apparatus for FMCW radar processing |
EP3173812A1 (en) * | 2015-11-24 | 2017-05-31 | Autoliv Development AB | A vehicle radar system arranged for reducing interference |
US9806914B1 (en) * | 2016-04-25 | 2017-10-31 | Uhnder, Inc. | Successive signal interference mitigation |
EP3244229A1 (en) * | 2016-05-13 | 2017-11-15 | Autoliv Development AB | A vehicle radar system arranged for interference reduction |
EP3489710A1 (en) * | 2017-11-23 | 2019-05-29 | Veoneer Sweden AB | Radar interference suppression |
Non-Patent Citations (2)
Title |
---|
J. Smith, III, "Mathematics of the Discrete Fourier Transform (DFT), with Audio Applications," second edition; section with the header "Positive and Negative Frequencies"; W3K Publishing; ISBN 978-0-9745607-4-8; published 2007; section posted on the author's internet page at Stanford University. (Year: 2007). * |
Murali, Sriram, et al. "Interference Detection in FMCW Radar Using A Complex Baseband Oversampled Receiver", 2018 IEEE Radar Conference (RadarConf18), Apr. 23-27, 2018, pp. 1567-1572. |
Also Published As
Publication number | Publication date |
---|---|
US20200124699A1 (en) | 2020-04-23 |
DE102018126034A1 (en) | 2020-04-23 |
CN111077514A (en) | 2020-04-28 |
JP2020067455A (en) | 2020-04-30 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US11693085B2 (en) | FMCW radar with interference signal suppression | |
US11016171B2 (en) | Radar sensing with phase correction | |
US10969463B2 (en) | Radar sensing with interference suppression | |
US11693106B2 (en) | Multiple input multiple output (MIMO) frequency-modulated continuous-wave (FMCW) radar system | |
US11215692B2 (en) | FMCW radar with additional AM for interference detection | |
US11592520B2 (en) | FMCW radar with interfering signal suppression in the time domain | |
US11209523B2 (en) | FMCW radar with interference signal rejection | |
US11885903B2 (en) | FMCW radar with interference signal suppression using artificial neural network | |
US10234541B2 (en) | FMCW radar device | |
US9835723B2 (en) | Radar ambiguity resolving detector | |
US7403153B2 (en) | System and method for reducing a radar interference signal | |
US20160377711A1 (en) | Radar signal processing for automated vehicles | |
JP4168475B2 (en) | Distance ambiguity removal method and apparatus applied to frequency shift keying continuous wave radar | |
US20230341511A1 (en) | Detection of interference-induced perturbations in fmcw radar systems | |
US11789114B2 (en) | FMCW radar with frequency hopping | |
US6833808B2 (en) | Signal processing | |
US8760340B2 (en) | Processing radar return signals to detect targets | |
US20240111020A1 (en) | Near-range interference mitigation for automotive radar system | |
US11391832B2 (en) | Phase doppler radar | |
US20230016890A1 (en) | Correction of phase deviations in the analog frontend of radar systems |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
FEPP | Fee payment procedure |
Free format text: ENTITY STATUS SET TO UNDISCOUNTED (ORIGINAL EVENT CODE: BIG.); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY |
|
AS | Assignment |
Owner name: INFINEON TECHNOLOGIES AG, GERMANY Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:MEISSNER, PAUL;MELZER, ALEXANDER;TOTH, MATE ANDRAS;REEL/FRAME:050921/0854 Effective date: 20191031 |
|
STPP | Information on status: patent application and granting procedure in general |
Free format text: NON FINAL ACTION MAILED |
|
STPP | Information on status: patent application and granting procedure in general |
Free format text: RESPONSE TO NON-FINAL OFFICE ACTION ENTERED AND FORWARDED TO EXAMINER |
|
STPP | Information on status: patent application and granting procedure in general |
Free format text: FINAL REJECTION MAILED |
|
STPP | Information on status: patent application and granting procedure in general |
Free format text: RESPONSE AFTER FINAL ACTION FORWARDED TO EXAMINER |
|
STPP | Information on status: patent application and granting procedure in general |
Free format text: NON FINAL ACTION MAILED |
|
STPP | Information on status: patent application and granting procedure in general |
Free format text: RESPONSE TO NON-FINAL OFFICE ACTION ENTERED AND FORWARDED TO EXAMINER |
|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |