TWI766236B - Pulse detector and blood pressure detector thereof - Google Patents

Abstract

A pulse detector and a blood pressure detector thereof are provided. The pulse detector is used for detecting pulsations and includes a microwave sensor, an amplitude demodulator connected with the microwave sensor and a zero filter connected between the microwave sensor and the amplitude demodulator. The microwave sensor generates an electric field and senses the variation of the electric field which is interfered by the pulsations so as to obtain sensing signals corresponding to the pulsations. The amplitude demodulator demodulates the amplitude of the sensing signals to detect the pulsations according to the amplitude variation of the sensing signals. The zero filter adds zero into the sensing signals.

Description

Pulse wave detector and blood pressure detector thereof

The present invention relates to a pulse measurement technology, in particular to a pulse wave detector and a blood pressure detector thereof.

At present, there are many ways to measure the pulsatile state, such as photoplethysmography (PPG). The blood pressure measurement technology is usually mainly based on the measurement method of the pressure pulse belt. However, for long-term monitoring, there are also measurement technology without pressure pulse belt. Generally, the electrocardiogram and PPG measure the pulse arrival time, use The pulse transmission time is measured by dual PPG, and the pulse transmission time is measured by dual radar. The blood pressure value is calculated at the time measured through the measurement.

However, these measurement methods are still not suitable for long-term measurement of users who are in an active state. For example, the electrode patch will make the user feel uncomfortable, and the sensor needs to be close to the body surface. Moreover, the measurement equipment used in the current technology has high cost, which is not conducive to popularization and use.

In view of this, an embodiment of the present invention provides a pulse wave detector including a microwave sensor, an amplitude demodulator coupled to the microwave sensor, and a pulse wave detector coupled between the microwave sensor and the amplitude demodulator Zero filter to detect pulsating state. The microwave sensor emits an electric field and senses that the electric field is disturbed by the pulse wave state to obtain a sensing signal. The amplitude demodulator demodulates the amplitude of the sensing signal, so as to facilitate detecting the pulsation state according to the change of the amplitude of the sensing signal. The zero-point filter increases the sensing signal Add a frequency response zero.

An embodiment of the present invention also provides a blood pressure detector, which includes the above-mentioned two pulse wave detectors and a data analysis device. The data analysis device synchronously receives the signals of the two pulse wave detectors to obtain the pulse transit time of the blood, and estimates the blood pressure value according to the pulse transit time.

To sum up, according to the pulse wave detector and the blood pressure detector according to the embodiments of the present invention, for the sensing signal of the microwave sensor, the amplitude demodulator can be used to detect the pulse state, and the zero point filter can be used To improve sensitivity, it can reduce hardware cost and simplify signal processing complexity.

100: Pulse detector

120: Microwave sensor

130: Amplitude demodulator

140: Zero filter

150: Amplifier

200: Sensing unit

310, 310': blood vessels

320: Blood

330,330': muscle

400: Substrate

401, 402, 403: Through hole

410: Rift Ring

411: Gap

412: T-section

421, 422: Microstrip line

4211, 4221: T-shaped transverse end

4212, 4222: T-shaped center

430: Complementary Slit Ring

431: Gap

432: L-shaped part

450: Rift Ring

451: Gap

452: L-shaped part

461: Microstrip line

462: Microstrip line

4611, 4621: T-shaped transverse end

4612, 4622: T-shaped center

470: Complementary Slit Ring

471: Gap

472: convex part

473: Ribbon

500: Data Analysis Device

600: Signal Generator

700: Power divider

E: electric field

PTT: Pulse Transit Time

S1, S1': sensing signal

S2, S2': sensing signal

Z, Z': frequency response zero

f,f': frequency

w1,w2: waveform

G1~G9: Spacing

L1~L13: length

W1~W10: Width

D1: Diameter

R _muscle ,R _sub : resistance

C _g ,C _res ,C _sub C _air-gap ,C _muscle ,C _n ,C _CSRR : Capacitance

C _var : variable capacitance

C _b : bypass capacitor

L _res ,L _sub ,L _via ,L _b ,L _CSRR : Inductance

P: port

V _b : bias voltage

1 is a block diagram of a pulse wave detector according to a first embodiment of the present invention.

FIG. 2 is a schematic diagram of the use state of the pulse wave detector according to some embodiments of the present invention.

3 is a schematic diagram of a sensing signal according to some embodiments of the present invention.

4 is a block diagram of a pulse wave detector according to a second embodiment of the present invention.

FIG. 5 is a front view of the first embodiment of the sensing unit according to the second embodiment of the present invention.

FIG. 6 is an equivalent circuit diagram of the first embodiment of the sensing unit according to the second embodiment of the present invention.

FIG. 7 is a schematic diagram of the use state of the blood pressure detector according to some embodiments of the present invention.

8 is a correlation diagram between the pulse transit time measured by the blood pressure detector and the systolic blood pressure actually measured by the blood pressure measurement according to the first implementation of the sensing unit according to the second embodiment of the present invention .

[FIG. 9] A first embodiment of the sensing unit according to the second embodiment of the present invention is used as a blood pressure Correlation plot between the pulse transit time measured by the detector and the actual diastolic blood pressure measured by the blood pressure meter.

10 is a statistical diagram of the Brand-Altman analysis of the estimated systolic blood pressure obtained by the first implementation aspect of the sensing unit according to the second embodiment of the present invention as a blood pressure detector.

11 is a statistical diagram of the Brand-Altman analysis of the estimated diastolic blood pressure obtained by the first implementation of the sensing unit according to the second embodiment of the present invention as a blood pressure detector.

12 is a front view of the substrate of the second embodiment of the sensing unit according to the second embodiment of the present invention.

13 is a rear view of the substrate of the second embodiment of the sensing unit according to the second embodiment of the present invention.

14 is a schematic diagram of a microwave sensor of the second embodiment of the sensing unit according to the second embodiment of the present invention.

[ FIG. 15 ] and [ FIG. 16 ] are respectively the odd-even mode equivalent circuit diagrams of the microwave sensor of the second embodiment of the sensing unit according to the second embodiment of the present invention.

17 is a front view of the second embodiment of the sensing unit according to the second embodiment of the present invention.

FIG. 18 is a schematic diagram of the zero point filter of the second embodiment of the sensing unit according to the second embodiment of the present invention.

19 is a frequency response diagram of the second embodiment of the sensing unit according to the second embodiment of the present invention under different bias voltages.

20 is a schematic diagram of the zero point adjustment of the frequency response of the second embodiment of the sensing unit according to the second embodiment of the present invention.

21 is a block diagram of a blood pressure detector according to a second embodiment of the present invention.

FIG. 22 is a correlation diagram between the pulse transit time measured by the second embodiment of the sensing unit as a blood pressure detector and the systolic blood pressure actually measured by the blood pressure meter according to the second embodiment of the present invention. .

23 is a correlation between the pulse transit time measured by the second embodiment of the sensing unit according to the second embodiment of the present invention as a blood pressure detector and the diastolic blood pressure measured by a commercially available blood pressure meter picture.

24 is a statistical diagram of the Brand-Altman analysis of the estimated systolic blood pressure obtained by the second implementation of the sensing unit according to the second embodiment of the present invention as a blood pressure detector.

25 is a statistical diagram of the Brand-Altman analysis of the predicted diastolic blood pressure obtained by the second implementation of the sensing unit according to the second embodiment of the present invention as a blood pressure detector.

Referring to FIG. 1 and FIG. 2 together, FIG. 1 is a schematic block diagram of a pulse wave detector 100 according to a first embodiment of the present invention, and FIG. 2 is a schematic diagram of a use state of the pulse wave detector 100 according to some embodiments of the present invention. The pulse wave detector 100 includes a microwave sensor 120 and an amplitude demodulator 130 . The pulse wave detector 100 can be used to detect the pulsation state of an object (eg, solid or fluid), such as heartbeat, vibration, and the like. The data analysis device 500 is coupled to the pulse wave detector 100, and can further analyze and apply the detection result. The data analysis device 500 can be, for example, a computing device such as a single chip, a microprocessor, a personal computer, a smart phone, a tablet computer, a web server, or the like. The microwave sensor 120 serves as the sensing unit 200, and the sensing unit 200 is disposed or close to the target to be detected. The microwave sensor 120 may be, for example, a split-ring resonator (SRR). As an example of an implementation, please refer to FIG. 5 and FIG. 12 for its structure, which will be further described later.

Here, the pulsation when the blood 320 of the human body flows is used as the object to be detected Mark as an example. The sensing unit 200 is disposed or close to the body surface of the human body, for example, it can be attached to the skin or fixed on clothes. The blood vessel 310 may contract and dilate the vessel wall due to the pulsation of the blood 320 (compared to the blood vessel 310'). Correspondingly, the muscle 330 surrounding the blood vessel 310 also changes in contraction and relaxation (compared to the muscle 330'). The microwave sensor 120 is configured to emit an electric field E and obtain a sensing signal by sensing that the electric field E is disturbed by a pulsating state. Specifically, the microwave sensor 120 has a resonant cavity, and the electric field E formed by the resonant cavity is directly or indirectly affected by the target to be detected and changes. That is to say, because the blood 320, the blood vessel 310 and the surrounding muscles 330 are all conductive materials, they will affect the distribution of the electric field E, causing the resonant frequency to change. Therefore, the pulsation state can be detected by detecting the variation range of the resonant frequency. . In this example, the blood 320 and the blood vessel 310 are in the lower layer of the muscle 330 and are far from the microwave sensor 120 , so the disturbance induced by the electric field E mainly comes from the contraction and relaxation changes of the muscle 330 .

Referring to FIG. 3 , it is a schematic diagram of a sensing signal according to some embodiments of the present invention. As shown in the left side of FIG. 3 , the sensing signal S1 shows the measurement result that the blood vessel 310 and the muscle 330 are in a contracted state; the sensing signal S1' shows the measurement result that the blood vessel 310 and the muscle 330 are in a diastolic state. It can be seen that the sensing signals S1 and S1' are not only different in frequency, but also different in amplitude. Since the cost of the instrument for identifying the amplitude is lower than that of identifying the frequency, and it can have higher sensitivity, the embodiment of the present invention adopts the amplitude to identify the pulse wave change. The amplitude demodulator 130 is coupled to the microwave sensor 120 to demodulate the amplitudes of the sensing signals S1 and S1', so as to detect the pulsation state (here, the blood 320) according to the amplitude changes of the sensing signals S1 and S1'. pulsation). As shown on the right side of FIG. 3, it is the output result of the sensing signals S1, S1' passing through the amplitude demodulator 130, and the change of the waveform w1 shown can correspond to the change of the blood flow 320 caused by the heartbeat. Here, the amplitude demodulator 130 may be, for example, an envelope detector (Envelope Detector).

Referring to FIG. 4 , it is a block diagram of the pulse wave detector 100 according to the second embodiment of the present invention. The difference from the aforementioned first embodiment is that the sensing unit 200 includes, in addition to the microwave sensor 120, a null filter 140, which is coupled between the microwave sensor 120 and the amplitude demodulator 130 for use in The frequency response zero point (Zero) is added to the sensing signals S1 and S1'. The null filter 140 may be, for example, a Complementary Split-Ring Resonator (CSRR), a Bandstop Filter, or a Notch Filter. Here, the zero-point filter 140 is an example of the implementation of the complementary slit ring resonator. Please refer to FIG. 5 and FIG. 12 for its structure, which will be further described later. Referring back to FIG. 3 , as shown on the left side of FIG. 3 , the sensing signal S2 is the result of adding the frequency response zero Z to the sensing signal S1 ; the sensing signal S2 ′ is the result of adding the frequency response zero Z to the sensing signal S1 ′ result. It can be seen that due to the increase of the frequency response zero point Z, the slope change between the frequency response poles (Pole) P, P' and the frequency response zero point Z is steeper than that of the first embodiment, achieving the effect of improving the sensitivity. From the right side of Figure 3, it can be seen that the change of the waveform w2 is more obvious.

Referring to FIG. 5 , it is a front view of the first embodiment of the sensing unit 200 according to the second embodiment of the present invention. Here, the sensing unit 200 may be fabricated using a printed circuit board fabrication technique, which includes a substrate 400 , a slit ring 410 , two microstrip lines 421 , 422 and a complementary slit ring 430 . The slit ring 410 and the two microstrip lines 421 and 422 are made of metal. The reverse side of the substrate 400 is a metal ground plane, and a complementary slit ring 430 is formed by the metal missing portion. The slot ring 410 , the microstrip line 421 and part of the microstrip line 422 constitute the microwave sensor 120 , and the other part of the microstrip line 422 and the complementary slot ring 430 constitute the null filter 140 . The slit ring 410 and the two microstrip lines 421 and 422 are located on the front side of the substrate 400 , and the complementary slit ring 430 is located on the reverse side of the substrate 400 . Complementary slit rings 430 are represented in phantom in FIG. 5 .

The slit ring 410 is located between the two microstrip lines 421 and 422 , and the slit ring 410 is separated from the two microstrip lines 421 and 422 by a distance G1 respectively. Here, the two microstrip lines 421 and 422 are T-shaped, but the embodiment of the present invention is not limited to this. The T-shaped transverse ends 4211 and 4221 of the two microstrip lines 421 and 422 are close to the slit ring 410 . The end of the T-shaped central portion 4212 of the microstrip line 421 is used for feeding high frequency signals. The T-shaped central portions 4212, 4222 taper from broad to narrow at the ends. The wide band width is denoted as W1, and the narrow band width is denoted as W2. Here, the slit ring 410 is roughly in the shape of a rectangular ring (the length of the long side and the width of the short side are denoted as L1 and L2, respectively, and the width of the line is denoted as W3), but the embodiment of the present invention is not limited to this shape, and the slit ring 410 is not limited to this shape. There is a gap 411 (the gap between the gaps 411 is marked as G2). The slit ring 410 has two T-shaped portions 412 on both sides of the notch 411 with opposite lateral ends (the length of the lateral ends is denoted as L3).

The complementary slit ring 430 also has a shape similar to that of the slit ring 410, and is also roughly shaped as a rectangular ring (the length of the four sides is marked as L4, and the width of the line is marked as W4), but the embodiment of the present invention is not limited to this shape, and has a gap 431 (the gap 431 spacing is marked as G3). The complementary slit ring 430 has two inwardly extending L-shaped portions 432 on both sides of the notch 431 (the extension length is denoted as L5). Here, the rear section of the T-shaped central portion 4222 of the microstrip line 422 traverses the complementary slit ring 430 and passes through the position corresponding to the gap 431 . As shown in FIG. 5 , the slit ring 410 and the complementary slit ring 430 are sequentially arranged along the extending direction of the microstrip lines 421 and 422 . As an example, the values of various parameters can be listed in Table 1.

。電容C_CSRR和電感L_CSRR為互補裂隙環430的共振電容和電感值。電感L為微帶線422的等效電感。 Referring to FIG. 6 , it is an equivalent circuit diagram of the first embodiment of the sensing unit 200 according to the second embodiment of the present invention. The resistances R _muscle and R _sub are the loss resistances of the muscle 330 and the substrate 400 , respectively. The capacitance C _sub is the parasitic capacitance of the substrate 400 . The capacitance C _g is the coupling capacitance between the slit ring 410 and the microstrip lines 421 and 422 respectively. The capacitance C _res and the inductance L _res are the resonant capacitance and inductance values of the slit ring 410 . The capacitance C _air-gap is the equivalent capacitance between the sensing unit 200 and the muscle 330 . The capacitance C _muscle is the equivalent capacitance of the muscle 330 . The total capacitance C _total can be expressed as C _res + (C _air-gap ∥C _muscle ). Therefore, the resonance frequency is

. Capacitance C _CSRR and inductance L _CSRR are the resonant capacitance and inductance values of complementary slit ring 430 . The inductance L is the equivalent inductance of the microstrip line 422 .

Referring to FIG. 7 , it is a schematic diagram of the use state of the blood pressure detector according to some embodiments of the present invention. The blood pressure detector can utilize the aforementioned two pulse wave detectors 100 to perform blood pressure detection. Specifically, the sensing units 200 of the two pulse wave detectors 100 are respectively disposed at positions adjacent to the carotid artery and the radial artery to measure the pulsation of the blood 320 in these two parts. However, the embodiment of the present invention is not limited to the two measurement positions, and the sensing units 200 of the two pulse wave detectors 100 can be placed in the proper position of the same closed cycle (Closed Circulation). As shown on the left side of Figure 7, the upper waveform is the measurement result of the carotid artery, and the lower waveform is the measurement result of the radial artery. The data analysis device 500 can find out the pulse transit time (PuL2e Transit Time) PTT from the difference between the two measurement results. After the pulse transit time PTT is found, the blood pressure value can be estimated according to some conversion equations between the pulse transit time PTT and the blood pressure (eg, equation 1 and equation 2). Equation 1 is a formula for calculating systolic blood pressure (SBP). Formula 2 is a calculation formula of diastolic blood pressure (DBP). Among them, a _s , b _s , a _d , and b _d are calibration constants. However, the embodiments of the present invention are not limited to these calculation formulas. For example, the natural logarithm (Ln) in formula 1 and formula 2 can be changed to a common logarithm (Log). Alternatively, other correlation formulas between pulse transit time PTT and blood pressure can be obtained through regression analysis.

SBP=a _s Ln(PTT)+b _s (Equation 1)

DBP=a _d Ln(PTT)+b _d (Formula 2)

Referring to FIG. 8 , it is the time between the pulse transit time PTT measured by the blood pressure detector and the systolic blood pressure actually measured by the blood pressure measurement according to the first implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. the correlation diagram. It can be seen that there is a significant correlation between PTT and systolic blood pressure (correlation coefficient R=-0.79). Among them, the circle symbol represents the measurement results in the static state, and the cross symbol represents the measurement results in the recovery state after exercise, with a total of 56 data points.

Referring to FIG. 9 , it is the difference between the pulse transit time PTT measured by the blood pressure detector and the diastolic blood pressure actually measured by the blood pressure measurement according to the first implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. the correlation diagram. It can also be seen that there is a significant correlation between pulse transit time PTT and diastolic blood pressure (correlation coefficient R=-0.82). Among them, the circle symbol represents the measurement results in the static state, and the cross symbol represents the measurement results in the recovery state after exercise, with a total of 56 data points.

Referring to FIG. 10 and FIG. 11 , respectively, the first embodiment of the sensing unit 200 according to the second embodiment of the present invention is used as the estimated value of systolic blood pressure and the estimated value of diastolic blood pressure obtained by the blood pressure detector. Statistical plot of Bland-Altman Analyses. Convert the pulse transit time PTT to the estimated value of systolic blood pressure according to the aforementioned formula 1, and measure it with a commercially available blood pressure meter. The obtained systolic blood pressure reference value is shown in Figure 10. Similarly, according to the aforementioned formula 2, the pulse transit time PTT is converted into a diastolic blood pressure estimated value, and the diastolic blood pressure reference value measured by a commercially available blood pressure meter, the statistics are shown in FIG. 11 . It can be seen that, as shown in Figures 10 and 11, the values all fall within the confidence interval of the mean ±1.96 standard deviation.

12 and 13 are respectively a front view and a rear view of the substrate 400 of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. The main difference from the first embodiment is that, in the second embodiment, the null filter 140 can also provide the function of microwave sensing in addition to the function of increasing the frequency response null Z. Through the through hole 401 , the sensing area of the microwave sensor 120 is led to the same side and the same area as the sensing range of the null filter 140 , so that the sensing effect can be improved.

12 and 13 are referred to together. The slit ring 450, the microstrip line 461 and part of the microstrip line 462 constitute the microwave sensor 120 (also referred to as a slit ring resonator), and the other part of the microstrip line 462 and the complementary slit ring 470 constitute the null filter 140 ( Also here is the complementary slit ring resonator). The slit ring 450 and the two microstrip lines 461 and 462 are located on the front side of the substrate 400 , and the complementary slit ring 470 is located on the reverse side of the substrate 400 . The slit ring 450 and the two microstrip lines 461 and 462 are made of metal. The reverse side of the substrate 400 is a metal ground plane, and a complementary slit ring 470 is formed by the missing metal portion. Complementary slit rings 470 are shown in phantom in FIG. 12 , and in FIG. 13 the aforementioned front-side elements are not shown.

The slit ring 450 is located between the two microstrip lines 461 and 462, and the slit ring 450 is separated from the two microstrip lines 461 and 462 by a distance G4, respectively. Here, the two microstrip lines 461 and 462 are T-shaped and have T-shaped lateral end portions 4611 and 4621 and T-shaped central portions 4612 and 4622 respectively, but the embodiment of the present invention is not limited thereto. The T-shaped transverse ends 4611, 4621 of the microstrip lines 461, 462 are close to the slit ring 450 (the line width of the T-shaped lateral ends 4611, 4621 is marked as W5). The T-shaped transverse end portions 4611 and 4621 each have an extension section on both sides, respectively extending along the periphery of the slit ring 450 . The distance between the T-shaped lateral ends 4611, 4621 is denoted as G5. The end of the T-shaped central portion 4612 of the microstrip line 461 is used for receiving high frequency signals. The T-shaped central portions 4612, 4622 taper from broad to narrow at the ends. The broadband width is denoted as W6. The length of the T-shaped central portion 4612 of the microstrip line 461 is denoted as L6; the length of the T-shaped central portion 4622 of the microstrip line 462 is denoted as L7. Here, the slit ring 450 is roughly in the shape of a rectangular ring (the length of the long side and the width of the short side are denoted as L8 and L9, respectively, and the width of the line is denoted as W7), but the embodiment of the present invention is not limited to this shape, and the slit ring 450 is not limited to this shape. There is a gap 451 (the gap between the gaps 451 is marked as G6). Here, the slit ring 450 has two outwardly extending L-shaped portions 452 on both sides of the notch 451 (the length of the extension is denoted as L10).

As shown in FIG. 13 , the complementary slit ring 470 also has an appearance similar to that of the slit ring 450 , and is also roughly shaped as a rectangular ring (the length of the long side and the width of the short side are marked as L11 and L12 respectively, and the width of the line is marked as W8 ), but this The embodiment of the invention is not limited to this shape, and has a notch 471 (the distance between the notch 471 is marked as G7). The complementary slit ring 470 has a protruding portion 472 on the opposite side of the notch 471, and the width of the protruding portion 472 is denoted as W9. Inside the protruding portion 472 are two band-shaped portions 473 (the length is denoted as L13, the width is denoted as W10, the distance is denoted as G8, and the distance between the band-shaped portion 473 and the convex portion 472 is denoted as G9). Each strip portion 473 is respectively coupled to the L-shaped portion 452 of the slit ring 450 through the through hole 401 (the diameter of the through hole 401 is denoted as D1 ). Here, the strip portion 473 is made of metal. As shown in FIG. 12 , the T-shaped central portion 4622 of the microstrip line 462 is meandering across the complementary slit ring 470 . The slit ring 450 and the complementary slit ring 470 are respectively located on two sides of the substrate 400 , staggered up and down, and coupled through the through holes 401 . Through the through hole 401, the sensing area of the slit ring 450 is led to the opposite side of the substrate 400 (ie, one side with the complementary slit ring 470), and the sensing area That is, taking the staggered position between the slit ring 450 and the complementary slit ring 470 as the center, it expands to a range. As an example, the values of each parameter can be listed in Table 2.

FIG. 14 is a schematic diagram of the microwave sensor 120 of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. Referring to FIG. 12 and FIG. 14 in combination, the slit ring 450 uses a coplanar waveguide (CPW) technology, and is coupled to the complementary slit ring 470 through the through hole 401 . The slit ring 450 can be regarded as a slit ring resonator integrated with a transmission line and a step impedance. The inductance L _via is the equivalent inductance of the through hole 401 . Since the slit ring 450 has a symmetrical structure, the analysis can be performed according to the equivalent circuit of the odd-even mode. Referring to FIG. 15 and FIG. 16 , which are respectively the odd-even mode equivalent circuit diagrams of the microwave sensor 120 of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. According to the transmission line theory, if the input impedance observed from port P is infinite, the odd-even mode resonance can be expressed as Equation 3 and Equation 4, respectively. Wherein Z ₁ and Z ₂ are the characteristic impedances of the transmission lines coupled to the front and back sides of the through hole 401 respectively, and θ ₁ and θ ₂ are the electrical lengths, respectively. As an example, Z ₁ is 60.8 Ω and Z ₂ is 54.5 Ω; at 2.67 GHz, θ ₁ is 83.4°, θ ₂ is 5.3°; L _via is 0.146 nH.

Z ₁ tanθ ₁ =Z ₂ cotθ ₂ -ωL _via (Equation 3)

Z ₁ cotθ ₁ =Z ₂ cotθ ₂ +ωL _via (Equation 4)

Since the complementary slit ring 470 is a complementary structure of the slit ring 450 , the analysis principle of the odd-even mode resonance is the same as that described above, and will not be repeated here. It should be noted that, in some embodiments, the pulse wave detector 100 further includes a variable capacitor C _var coupled to the zero filter 140 , so as to change the zero filter according to the capacitance value of the variable capacitor C _var 140 frequency response zero Z. Referring to FIG. 17 , it is a front view of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. The variable capacitor C _var can be realized by a varactor diode. Depending on the magnitude of the bias voltage V _b given to the varactor diode, the capacitance value of the variable capacitor C _var can be controlled, thereby changing the zero point Z of the frequency response. To isolate the bias voltage V _b , an inductance L _b is added between the input terminal of the bias voltage V _b and the variable capacitor C _var . To avoid short circuit of the variable capacitor _Cvar , a bypass capacitor _{Cb is} added. The variable capacitor C _var is coupled to the ground plane on the reverse side of the substrate 400 through the through hole 402 ; the bypass capacitor C _b is coupled to the ground plane on the reverse side of the substrate 400 through the through hole 403 . The through holes 402 and 403 are located on the inner side and the outer side of the complementary slit ring 470 respectively. Through the through holes 402 and 403 , the variable capacitor C _var , the bypass capacitor C _b and the inductor L _b can be arranged in the available space on the front surface of the substrate 400 . As an example, the change range of the capacitance value of the variable capacitor C _var is 1.3pF~22.62pF, the capacitance value of the bypass capacitor C _b is 0.68pF, the inductance value of the inductor L _b is 25nH, and the frequency response zero point Z controllable range It is 2.2GHz~2.7GHz.

Referring to FIG. 18 , it is a schematic diagram of the zero point filter 140 of the second implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Since the complementary slit ring 470 is coupled to the variable capacitor C _var and the bypass capacitor C _b , the size of the complementary slit ring 470 can be reduced compared to the pure complementary slit ring 470 if the same resonance frequency is used. In addition, since the strip portion 473 and the surrounding ground plane also generate the capacitance C _n , the resonance frequency will also decrease, but since the capacitance value of the variable capacitance C _var can be adjusted, it can compensate for the resonance frequency caused by the capacitance C _n declining impact.

Referring to FIG. 19 , it is a frequency response diagram under different bias voltages V _b of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. It can be seen that the zero point Z of the frequency response can be changed by applying the bias voltage V _b of different voltage values (here, 0V~8V is taken as an example).

Referring to FIG. 20 , it is a schematic diagram of adjusting the frequency response zero point Z of the second implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Comparing the left side of Figure 3 with Figure 19, it can be seen that if there is no frequency response zero point Z is fixed, the closer to the frequency response zero point Z, the less obvious the amplitude change (as shown on the left side of Figure 3); Move from frequency f to frequency f') to adjust the frequency response zero point Z, as shown in Figure 19, by adjusting from the frequency response zero point Z to the frequency response zero point Z', the amplitude change can be maintained to a considerable extent.

In some embodiments, the variable capacitor C _var can also be coupled to the microwave sensor 120 to change the frequency response pole of the microwave sensor 120 , and can also respond to the frequency offset, so that the amplitude change can be maintained at a comparable value degree.

Referring to FIG. 21 , it is a block diagram of a blood pressure detector according to the second embodiment of the present invention. As mentioned above, two pulse wave detectors 100 are also used for blood pressure detection, and the detected signals are further analyzed by the data analysis device 500 . Each constituent element is further described here. In addition to the aforementioned sensing unit 200 and the amplitude demodulator 130 , the pulse wave detector 100 may further include an amplifier 150 . The amplifier 150 is coupled after the sensing unit 200 to amplify the signal sensed by the sensing unit 200, so that the subsequent amplitude demodulator 130 can better process the signal. In some embodiments, the amplifier 150 may be, for example, a Low Noise Amplifier (LNA). Here, if the sensing unit 200 has the aforementioned variable capacitor C _var , the blood pressure detector can provide a bias voltage V _b to adjust the capacitance value of the variable capacitor C _var . The blood pressure detector further includes a signal generator 600 and a power divider 700 . The signal generator 600 is used for generating high-frequency signals, and the power divider 700 is used for outputting the high-frequency signals from the two output ports and inputting the high-frequency signals to the microwave sensors 120 of the two sensing units 200 respectively, so as to generate a resonance effect. Here, the power divider 700 may be, for example, a Wilkinson power divider. In some embodiments, the frequency of the high frequency signal is 2.4GHz˜2.5GHz.

Referring to FIG. 22 , it is the time between the pulse transit time PTT measured by the blood pressure detector and the systolic blood pressure actually measured by the blood pressure measurement according to the second implementation of the sensing unit 200 according to the second embodiment of the present invention. the correlation diagram. It can be seen that there is a significant correlation between PTT and systolic blood pressure (correlation coefficient R=-0.79). Among them, the circle symbol represents the measurement result in the static state, and the cross symbol represents the measurement result in the recovery state after exercise, with a total of 155 data points.

Referring to FIG. 23 , it is the second embodiment of the sensing unit 200 according to the second embodiment of the present invention as the difference between the pulse transit time PTT measured by the blood pressure detector and the diastolic blood pressure measured by the commercially available blood pressure meter. correlation diagram between. It can also be seen that there is a significant correlation between pulse transit time PTT and diastolic blood pressure (correlation coefficient R=-0.72). Among them, the circle symbol represents the measurement result in the static state, and the cross symbol represents the measurement result in the recovery state after exercise, with a total of 155 data points.

Referring to FIG. 24 and FIG. 25 , respectively, the second implementation of the sensing unit 200 according to the second embodiment of the present invention is used as the predicted value of systolic blood pressure and the predicted value of diastolic blood pressure obtained by the blood pressure detector. Statistical graph of the De-Altman analysis. According to the aforementioned formula 1, the pulse transit time PTT is converted into a systolic blood pressure estimated value, and the systolic blood pressure reference value measured by a commercially available blood pressure meter, the statistics are shown in Figure 24. Similarly, according to the aforementioned formula 2, the pulse transit time PTT is converted into a diastolic blood pressure estimated value, and the diastolic blood pressure reference value measured by a commercially available blood pressure meter, the statistics are shown in FIG. 25 . It can be seen that, as shown in Figures 24 and 25, the values all fall within the confidence interval of the mean ±1.96 standard deviation.

To sum up, according to the pulse wave detector and the blood pressure detector according to the embodiments of the present invention, for the sensing signal of the microwave sensor, the amplitude demodulator can be used to detect the pulse state, which can reduce the hardware cost , while simplifying the signal processing complexity. According to the pulse wave detector of some embodiments, the microwave sensor is coupled to the zero point filter to increase the frequency response zero point in the sensing signal, which can improve the sensing sensitivity. According to some embodiments, the zero point filter is further coupled with a variable capacitor, so that the frequency response zero point can be adjusted, so that the state of different frequency offsets still maintains a considerable change in amplitude, and the sensing sensitivity is maintained. In some embodiments, the variable capacitor can be changed to be coupled to the microwave sensor to adjust the frequency response pole, and the sensing sensitivity can also be maintained. According to some embodiments, the microwave sensor and the zero-point filter are located on two sides of the substrate, and are coupled through through holes, so that the sensing areas of the microwave sensor and the zero-point filter overlap to increase the induction effect.

100: Pulse detector

120: Microwave sensor

130: Amplitude demodulator

200: Sensing unit

500: Data Analysis Device

Claims (9)

A pulse wave detector is used to detect a pulse state, the pulse wave detector comprises: a microwave sensor, which emits an electric field and senses that the electric field is disturbed by the pulsation state to obtain a sensing signal; an amplitude a demodulator, coupled to the microwave sensor, to demodulate the amplitude of the sensing signal, so as to detect the pulsation state according to the change of the amplitude of the sensing signal; and a zero-point filter coupled to the Between the microwave sensor and the amplitude demodulator, a frequency response zero is added to the sensing signal. The pulse wave detector of claim 1, wherein the zero point filter is a complementary slit ring resonator. The pulse wave detector of claim 1 further comprises a variable capacitor coupled to the zero point filter, so as to change the zero point of the frequency response of the zero point filter according to the capacitance value of the variable capacitor. The pulse wave detector as claimed in claim 1, wherein the microwave sensor is a slit ring resonator. The pulse wave detector of claim 1, wherein the amplitude demodulator is an envelope detector. The pulse wave detector of claim 1 further comprises a variable capacitor coupled to the microwave sensor, so as to change a frequency response pole of the microwave sensor according to the capacitance value of the variable capacitor. The pulse wave detector according to claim 1, wherein the microwave sensor is a slit ring resonator, the pulse wave detector further comprises a complementary slit ring resonator, and the complementary slit ring resonator is coupled to the Between the slit ring resonator and the amplitude demodulator, the complementary slit ring resonator includes a complementary slit ring, the slit ring resonator includes a slit ring, and the complementary slit ring and the slit ring are placed on a substrate separately the two sides of the device are coupled through at least one through hole. The pulse wave detector of claim 7, wherein a microstrip line traverses the complementary slit ring relative to the other side of the complementary slit ring, and the microstrip line is coupled to the slit ring resonator. A blood pressure detector, comprising two pulse wave detectors as described in any one of claims 1 to 8 and a data analysis device, the data analysis device receives the signals of the two pulse wave detectors to obtain A pulse transit time of blood, and a blood pressure value is estimated according to the pulse transit time.

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