TWI622279B - Communication device and method for beamforming - Google Patents

Abstract

A method for beamforming comprising the following steps. Determine the optimal full-digital precoder matrix for the communication device and fix the RF precoder matrix. The best all-bit precoder matrix, the baseband precoding array, and the RF precoding array are operated using a least squares method to obtain an optimized baseband precoder matrix, and the fundamental frequency precoder is optimized The matrix updates the fundamental frequency precoder matrix. Fixed baseband precoder matrix. Iteratively computes the best all-digital precoder matrix, the fundamental precoding array, and the RF precoding array to obtain an optimized RF precoder matrix, and optimizes the RF precoder matrix to update the RF precoding. The matrix is such that the beamformer processes the RF signals based on the updated RF precoder matrix.

Description

Communication device and method for beamforming

The present invention relates to a communication device and a method for beamforming, and more particularly to a communication device suitable for millimeter wave communication and a method for beamforming applied to the communication device.

With the development of wireless communication technology, today's wireless communication systems have been able to achieve high transmission rate requirements. However, commercial spectrum resources are limited resources, and as mobile terminals and their demand for data transmission increase, the problem of tight commercial spectrum resources is gradually emerging. In response to the problem of tight commercial spectrum resources, the industry has proposed corresponding solutions, one of which is millimeter-wave large-scale multi-input multi-output (MIMO) communication technology, which uses multi-antenna antenna array transmission. And receiving millimeter wave RF signals. However, the frequency band for transmitting the millimeter-wave RF signal is different from that of the conventional wireless communication system (for example, Long Term Evolution Advanced (LTE-A), wireless local area network communication system, etc.), and the channel characteristics are also The difference is that the design of the millimeter wave transmission system is different from the design of the conventional wireless communication system. On the other hand, the architecture of large-scale multi-input and multi-output communication is still limited by the huge amount of computation required.

It is an object of the present invention to provide a communication device and a method for beamforming that can effectively reduce computational complexity, increase the speed at which a particular value of a phase shifter unit is obtained, and maintain high spectral efficiency and low residual error.

One aspect of the present invention is to provide a communication device suitable for performing millimeter-wave large-scale multi-input multi-output (MIMO) communication, and the communication device includes a transceiver unit and a processing unit . The transceiver unit is configured to wirelessly communicate with a remote entity, the transceiver unit including a baseband processing circuit, a plurality of radio frequency links, a beamformer, and an antenna array. The baseband processing circuit is configured to process the plurality of data streams to convert the data stream into a plurality of baseband signals, the baseband processing circuit corresponding to the baseband precoder matrix. The RF link is coupled to the baseband processing circuit for converting the baseband signal output by the baseband processing circuit into a plurality of RF signals. The beamformer is coupled to the RF link for adjusting the phase of the RF signals. The beamformer corresponds to the RF precoder matrix. The antenna array is used to transmit the RF signals. The processing unit is configured to: determine an optimal full-digital precoder matrix of the communication device, and fix the RF precoder matrix; use the least squares method for the best all-digital precoder matrix, the baseband precoding array, and the RF The precoding array is operated to obtain an optimized baseband precoder matrix, and the baseband precoder matrix is updated to optimize the baseband precoder matrix; the fixed fundamental precoder matrix; and the best The full-digit precoder matrix, the baseband precoding array, and the RF precoding array are iteratively operated to obtain an optimized RF precoder matrix, and the RF precoder matrix is updated to update the RF precoder matrix. Beamforming The RF signals are processed according to the updated RF precoder matrix.

According to an embodiment of the invention, the processing unit performs an iterative operation on the optimal all-digital precoder matrix, the fundamental precoding array, and the radio frequency precoding array by an iteration of less than or equal to 5.

According to a further embodiment of the invention, the beamformer comprises a plurality of phase shifter unit groups and a plurality of combiner units. The phase shifter unit groups are used to adjust the phases of the RF signals according to the plurality of phase values. The combiner unit is coupled to the phase shifter unit groups for combining the RF signals output by the phase shifter unit groups.

According to still another embodiment of the present invention, the number of resolution bits of the phase values is less than or equal to 5.

According to a further embodiment of the present invention, the number of the data streams is less than or equal to the number of the radio frequency links, and the number of the radio frequency links is smaller than the number of antennas in the antenna array. .

According to still another embodiment of the present invention, the radio frequency links are up-converted according to the carrier signals, and the frequency range of the carrier signals is between 10 GHz and 300 GHz.

According to still another embodiment of the present invention, the antenna array is a uniform planar antenna array.

Another aspect of the present invention is to provide a method for beamforming, which is performed in a communication device suitable for millimeter wave large-scale multi-input multi-output communication and having a fundamental frequency processing Circuit and beamformer, the baseband processing circuit and the beam The device respectively corresponds to the baseband precoder matrix and the radio frequency precoder matrix, and the method comprises: determining an optimal full digital precoder matrix of the communication device, and fixing the RF precoder matrix; using the least square method to optimize the total The digital precoder matrix, the baseband precoding array and the RF precoding array are operated to obtain an optimized baseband precoder matrix, and the fundamental frequency precoder matrix is optimized to update the fundamental frequency precoder matrix. a fixed baseband precoder matrix; and iterative operations on the best all-digital precoder matrix, the baseband precoding array, and the RF precoding array to obtain an optimized RF precoder matrix, which is optimal The RF precoder matrix updates the RF precoder matrix such that the beamformer processes the RF signals based on the updated RF precoder matrix.

According to an embodiment of the invention, the number of iterations of performing the iterative operation on the optimal all-digital precoder matrix, the baseband precoding array and the radio frequency precoding array is less than or equal to 5.

According to still another embodiment of the present invention, the method further includes: generating a plurality of phase values, such that the beamformer adjusts the phases of the RF signals according to the phase values, wherein the number of resolution bits of the phase values is less than Or equal to 5.

100‧‧‧Wireless communication system

110, 120, 200, 300, 700‧‧‧ communication equipment

130‧‧‧Wireless channel

210, 340‧‧‧ fundamental frequency processing circuit

220(1)~220(M), 330(1)~330(N)‧‧‧RF link

230, 320‧‧‧ Beamformer

232(1)~232(M), 322(1)~322(N)‧‧‧ phase shifter unit

234(1)~234(M ' ), 324(1)~324(N)‧‧‧ combiner unit

240, 310‧‧‧ antenna array

242(1)~242(M ' ), 312(1)~312(N ' )‧‧‧ antenna

400‧‧‧ method

402, 404, 406, 408, 410, 412, 414, 416‧ ‧ steps

710‧‧‧Processing unit

720‧‧‧ memory unit

730‧‧‧ transceiver unit

For a more complete understanding of the embodiments and the advantages thereof, reference is made to the following description in conjunction with the drawings in which: FIG. 1 is a schematic diagram of a communication system in accordance with an embodiment of the present invention; 2 is a schematic diagram of a communication device according to an embodiment of the present invention; [FIG. 3] is a schematic diagram of a communication device according to an embodiment of the present invention; [FIG. 4] is a method for beamforming according to an embodiment of the present invention. [Fig. 5] is a simulation result of the relationship between the signal-to-noise ratio and the spectral efficiency of the embodiment and the comparative example of the present invention; [Fig. 6] is the relationship between the iteration number and the average residual error of the embodiment and the comparative example of the present invention. The simulation results; and [Fig. 7] are schematic views of a communication device in accordance with an embodiment of the present invention.

Embodiments of the invention are discussed in detail below. However, it will be appreciated that the embodiments provide many applicable concepts that can be implemented in a wide variety of specific content. The examples discussed and disclosed are illustrative only and are not intended to limit the scope of the invention.

In the following description, , Representing a collection of real and plural numbers, respectively. {. } represents the expectation operator, (.) ^T , (.) ^H , (.) ^* , (.) ^† represent transpose matrix, conjugate transposed matrix, complex conjugate (complex conjugate) Number and pseudo inverse matrix, And ∥. ∥ _F represents the Kronecker product operation and the Frobenius norm, respectively, and minimize f (.) is the solution that minimizes the value of the f (.) function. "subject to" A stands for "Restricted by" condition A, and "for μ =1 ,...,U " represents a case where μ is any positive integer from 1 to U. In addition, Represents an identity matrix of size N _S × N _S , [. _m,n represents the nth row element of the mth column in the matrix, vec(.) represents the vectorization of the matrix, Re (.) is the real part of the complex number, and arg(.) is the principal argument of the complex number. , Represents a complex Gaussian distribution with a variance value of σ ² and an average of 0, and j is defined as .

Please refer to FIG. 1. FIG. 1 is a schematic diagram of a communication system 100 in accordance with some embodiments of the present invention. The communication system 100 is a millimeter wave large-scale multi-input multi-output (MIMO) communication system including communication devices 110, 120 and a wireless channel 130, and the communication devices 110, 120 are communicatively connected via a wireless channel 130. Each communication device 110, 120 can be used as a signal transmission end, a signal receiving end or a signal transmission/reception end. Herein, a communication device (eg, communication device 110, 120) may represent a variety of different implementations including, but not limited to, for example, user equipment (UE), mobile station (MS), notebook computer, A mobile device such as a mobile phone and a fixed device such as a base station (BS), an evolved base station (evolved NodeB; eNB), a computer device, a server device, and a workstation. In addition, the communication device can communicate wirelessly with the remote entity in a mobile environment or in a fixed environment. In some embodiments, communication system 100 supports other communication technologies, such as Long-term Evolution Advanced (LTE-Advanced), wireless local area network (WLAN) communication technologies, and/or the like. Similar wireless communication technology.

2 is a schematic diagram of a communication device 200 in accordance with an embodiment of the present invention. The communication device 200 has at least a millimeter wave large-scale multi-input multi-output communication function, which may be the communication device 110, 120 of FIG. 1 or other communication device suitable for millimeter wave signal transmission. As shown in FIG. 2, the communication device 200 includes a baseband processing circuit 210, radio frequency links 220(1) to 220(M), a beamformer 230, and an antenna array 240, but is not limited thereto. In some embodiments, the communication device 200 also has an RF signal receiving function.

The baseband processing circuit 210 is configured to process the data stream, such as channel coding, symbol mapping, cyclic prefix, and/or other processing to convert the data stream into a base stream. The frequency signal is sent to the corresponding RF link 220(1)~220(M) respectively. The baseband processing circuit 210 also includes a digital-to-analog converter (DAC) for converting the baseband signal of the digital format to the fundamental frequency signal of the analog format. In some embodiments, the number M of radio frequency links 220(1)~220(M) is greater than or equal to the number of data streams.

The RF links 220(1) to 220(M) are respectively used for up-converting the baseband signal to convert the baseband signal into an RF signal. Each of the RF links 220(1)-220(M) includes an oscillator and a mixer, wherein the oscillator is used to generate a carrier signal, and the mixer is used to mix the baseband signal and the carrier signal into an RF signal. The frequency range of the carrier signal generated by the oscillator may be between 10 GHz and 300 GHz, but is not limited thereto. In some embodiments, the radio frequency links 220(1)-220(M) share the same oscillator. In addition, each RF Both ends of the links 220(1) to 220(M) may include filters for limiting the transmission of the fundamental frequency signals and the RF signals of the specific frequency components.

The beamformer 230 is configured to perform beamforming processing on the RF signal. As shown in FIG. 2, the beamformer 230 includes phase shifter unit groups 232(1) to 232(M) and combiner units 234(1) to 234(M ' ), wherein the phase shifter unit group 232(1) ~232(M) is used to adjust the phase of the RF signal based on a plurality of phase values, that is, to provide a desired phase difference to the RF signal outputted by the RF link 220(1)~220(M), thereby obtaining a phase shift. RF signal, and each combiner unit 234(1)~234(M ' ) is used to combine the phase shift RF signals output by the phase shifter unit group 232(1)~232(M), and the same phase shift The RF signals output by the unit groups 232(1) to 232(M) are combined by the combiner units 234(1) to 234(M ' ), respectively. In addition, the beamformer 230 may further include a plurality of power amplifiers for increasing the intensity of the phase-shifted RF signals, respectively.

The antenna array 240 includes antennas 242(1)-242(M ' ) for transmitting phase-shifted RF signals. The antenna array 240 can be a tunable antenna array, and the beamformer 230 can adjust the beam pattern and phase accordingly depending on the configuration of the antenna array 240. That is, the antennas 242(1)-242(M ' ) can generate multiple beams in all directions or in various field types by operating the beamformer 230. In some embodiments, the number M ' of antennas 242(1)-242(M ' ) is greater than the number M of radio frequency links 220(1)-220(M).

FIG. 3 is a schematic diagram of a communication device 300 in accordance with an embodiment of the present invention. The communication device 300 has at least a millimeter wave large-scale multi-input multi-output communication function, which may be the communication device 110, 120 of FIG. 1 or other suitable for Communication equipment for millimeter wave signal transmission. As shown in FIG. 3, the communication device 300 includes an antenna array 310, a beamformer 320, radio frequency links 330(1)-330(N), and a baseband processing circuit 340, but is not limited thereto. In some embodiments, the communication device 300 also has an RF signal transmission function.

The antenna array 310 includes antennas 312(1)-312(N ' ) for receiving radio frequency signals. The antenna array 310 can be a tunable antenna array, and the antennas 312(1)-312(N ' ) in the antenna array 310 can be generated in multiple directions or in various field types by operating the beamformer 230. Beam. The antenna array 310 can be used to receive millimeter wave signals having a frequency range between 10 GHz and 300 GHz.

The beamformer 320 is configured to adjust the beam pattern and phase according to the configuration of the antenna array 310. As shown in FIG. 3, beamformer 320 includes phase shifter unit groups 322(1)-322(N ' ) and combiner units 324(1)-324(N), wherein phase shifter unit group 322(1) ~322(N ' ) is used to restore the phase of the RF signal based on the plurality of phase values, that is, to provide the required phase difference of the RF signal received by the antenna array 310, thereby obtaining a phase-shifted RF signal, and each The combiner units 324(1)-324(N) are used to combine the phase-shifted RF signals output by the phase shifter unit groups 322(1)-322(N ' ), and the same phase shifter unit group 322 ( 1) The RF signals outputted by ~322(N ' ) are combined by the combiner units 324(1)~324(N). In addition, the beamformer 320 can further include a plurality of power amplifiers for increasing the strength of the RF signals received by the antenna array 310, respectively.

The RF links 330(1) to 330(N) are respectively used for down-converting the baseband signal to convert the RF signal into a baseband signal. Each of the RF links 330(1)-330(N) includes an oscillator and a mixer, wherein the oscillator is used to generate a carrier signal, and the mixer is used to mix the RF signal and the carrier signal. The frequency range of the carrier signal generated by the oscillator may be between 10 GHz and 300 GHz, but is not limited thereto. In some embodiments, the radio frequency links 330(1)-330(N) share the same oscillator. The signal obtained by the mixer can be subjected to low-pass filtering to remove the high-frequency components of the signal output by the mixer, and the unremoved signal component is the fundamental frequency signal. In addition, the input of each of the RF links 330(1)-330(N) may include a filter for limiting the passage of RF signals of specific frequency components. In some embodiments, the number N of radio frequency links 330(1)-330(N) is less than the number N ' of antennas 312(1)-312(N ' ).

The baseband processing circuit 340 is configured to process the baseband signals output by the RF links 330(1)-330(N), such as timing adjustment, frequency compensation, channel estimation, baseband signal equalization, baseband signal bonding, and / or other suitable processing to convert the baseband signal into a stream of data. The baseband processing circuit 340 also includes an analog-to-digital converter (ADC) for converting the fundamental frequency signal of the analog format into a baseband signal of the digital format. In some embodiments, the number of data streams obtained by the baseband processing circuit 340 is less than or equal to the number N of radio links 330(1)-330(N).

The following paragraphs illustrate beamforming using the communication system for the communication device 200 of FIG. 2 and the communication device 300 of FIG. 3 as an example. The relevant configuration of the communication devices 200, 300 is first defined. The number of data streams processed by the communication device 200 and the number of data streams obtained by the communication device 300 are N _S , and the RF links 220 (1) to 220 (M) and the RF links 330 (1) to 330 (N) The number is with And the number of antennas 242(1)~242(M ' ) and antennas 312(1)~312(N ' ) are N _t and N _r , respectively, where N _s < N _t and N _S < N _r . In the communication device 200, the data stream is represented by a vector s (s ), The weight matrix of beamformer 230 is , where F _D The fundamental frequency precoder matrix of the fundamental frequency processing circuit 210, F _RF A matrix of radio frequency precoders representing beamformer 230. In this way, the RF signal transmitted by the communication device 200 can be expressed as a vector x. . In addition, the normalized transmission power limit of the communication device 200 is expressed as . In the communication device 300, the weight matrix of the beamformer 320 is , where W _D Group represents a baseband processing circuit 340 frequency-binding matrix, W _RF A matrix of RF combiners representing beamformer 320. The RF signal transmitted by the communication device 200 is transmitted to the communication device 300 via the block attenuation channel massive MIMO channel, and the data stream vector y obtained by the baseband processing circuit 340 of the communication device 300 is expressed as: Where ρ is the average received power, H For the block attenuation channel large-scale MIMO channel coefficient matrix, z is an additive white Gaussian noise (AWGN) vector, and all elements in z are independent and identically distributed and have an average value of 0. And the number of variations is The plural normal distribution.

Next, using the SV channel (Saleh-Valenzuela channel) model, the channel coefficient matrix H is expressed as: Where N _cl is the number of clusters, each cluster has N _ray conduction paths, and a _r (.), a _t (.) are the receive and transmit array response vectors, respectively, β _il , , , , Represented as the complex gain, the azimuthal arrival angle (AOA), the elevational arrival angle, and the angle of departure (AOD) of the lth conduction path in the i- th cluster, respectively. ) and face the angle of departure. In addition, the complex gain β _{il has} an average of 0 and a variance of The complex normal distribution is a model, where Is the average power of the ith cluster, and To ensure Normalized vector. Array response vector for N _h antennas in the horizontal direction and N _v antenna uniform planar arrays in the vertical direction ( τ {t , r}; that is, receive array response vector Or transmission array response vector Can be expressed as follows: Where λ and d represent the wavelength and antenna spacing of the RF signal, respectively, and 1 α N _h and 1 β N _v .

Given a given channel coefficient matrix H, this paper considers the problem of hybrid beamforming designs for maximizing spectral efficiency. The antenna efficiency is expressed by the following formula: The amplitude and phase of the fundamental precoder matrix F _D and the baseband combiner matrix W _D can be adjusted, but only the phase values can be changed for the RF precoder matrix F _RF and the RF combiner matrix W _RF . In addition, the value of each phase shifter unit can be quantized to belong to the set An element, where B is the number of resolution bits of the phase shifter unit, and A uniform quantization step for the phase shifter unit. Therefore, the element selection of the RF precoder matrix F _RF and the RF combiner matrix W _RF is limited, ie [ F _RF ] _m,n F and [ W _RF ] _m,n F , where , ω = e ^{j △} . Under the condition of using a finite resolution phase shifter unit and the total power limit of the transmission, the problem of this hybrid beamforming design can be expressed as follows:

However, the problem P 1 involves the joint operation of four matrix variables (ie, F _RF , F _D , W _{RF ,} and W _D ) and the RF precoder matrix F _RF and RF combination shown in equations (5c) and (5d). The non-convex constraint of the elements of the matrix W _RF , it is too difficult and complicated to directly solve the problem P 1 . In order to solve this difficulty, the problem P 1 can be divided into two optimization problems, namely, a hybrid precoding problem and a hybrid combining problem, respectively, to simplify the joint hybrid precoding and combining design. In general, the hybrid precoding problem is similar to the mathematical form of the hybrid combining problem, and the hybrid precoding problem also includes power limitation considerations. Therefore, this paper focuses on solving the hybrid precoding problem, and the solution can also be applied. Solve the problem of hybrid integration.

The above hybrid precoding problem can be expressed as follows: Wherein F _opt is the optimal bit full precoder matrix, N _s front row comprising matrix V, the matrix V may be by channel coefficient matrix H is singular value decomposition (singular value decomposition), i.e. channel coefficient matrix and the matrix V relationship between H is ^H = UΣV H. However, the problem P 2 still involves the joint operation of two matrix variables (ie, F _RF and F _D ) and the non-convex limitation of the elements of the RF precoder matrix F _RF shown in equation (5c), so the problem P is directly solved. 2 is still too difficult and complicated.

Problem P 2 involves the fundamental precoder matrix F _D and the RF precoder matrix F _RF , so the second frequency iterative step based on the alternating minimization operation can be used to simultaneously obtain the fundamental precoder matrix F _D and the RF precoder Matrix F _RF .

In detail, in the first phase, the optimal baseband precoder matrix F _D is obtained with a fixed RF precoder matrix F _RF . Without considering the limitation shown in (5b), the problem P 2 can be further simplified as: It becomes a linear least squares problem, and the optimal fundamental precoder matrix F _D is

Then, in the second stage, assuming that the fundamental precoder matrix F _D is fixed, the radio frequency precoder matrix F _RF can be obtained by solving the problem P 4 :

However, in the above-described second-order iterative step based on the alternating minimization operation, iteratively updating the combinatorial complexity of the radio frequency precoder matrix F _RF is a major obstacle. In detail, under a given finite set F , the set of all possible RF beamforming is also limited, so all possible combinations can be computed using the exhaustive search method to get the problem P 4 The best solution. However, question P 4 is involved in The search for a possible combination is still too large.

In view of this, an iterative method is proposed to solve the problem P 4, which is described in the following paragraphs.

Before starting to explain the iterative method, first define the vector Vector function And matrix ,among them , K = N _t × N _s , and The number of elements in the RF precoder matrix F _RF . The objective function of equation (8) can be expressed as And the problem P 4 can be equivalent as follows:

It should be noted that the problem P 5 is a multivariate (ie ~ The discrete optimization problem, so it still requires a huge amount of computation. For problem P5 , this can be done by solving a series of single variable optimization problems. Specifically, in the iterative operation, the variable is fixed first. ~ ( U -1) variables in the middle, and only optimize the remaining one. That is to say, in each iterative operation, the angle is updated by solving the following problem. :

The above method can monotonically decrement the objective function. Although non-convex limiting formula (10b) as shown in the still to solve the problem P 6 cause difficulties, but this can be difficult to optimize variables do to reduce the use of search method. Meanwhile, if the non-convex limit shown by equation (10b) is negligible, the solution of the problem P 6 can be parsed by a special form: in a given vector Minimize Variable The best value

It should be noted that the optimum value obtained by equation (11) It is not guaranteed to belong to set B. Therefore, you need to re-optimize the value Quantize processing as shown below: among them The quantized optimal value, and Q (.) is a quantization function that makes the variable equal to the element with the closest value in set B. In other words, the best quantization coefficient Can be selected according to the following formula:

The above iterative method is further illustrated by the method 400 for beamforming in accordance with an embodiment of the present invention. As shown in FIG. 4, method 400 includes steps 402, 404, 406, 408, 410, 412, 414, 416. In step 402, the optimized full digital precoder matrix F _opt and the base frequency precoder matrix F _D are input. In step 404, the optimized full digit precoder matrix F _{opt is} vectorized into ,definition For the matrix G = [ G _kμ ], and initialize the variables And the initialization variable μ, q =1.

In step 406, the optimal value is determined according to equation (11). And quantify the optimal value according to equation (12) Optimum value for quantification And quantify the best value To update the best value .

In step 408, it is determined whether the variable μ is greater than or equal to U. If no, proceed to step 410; if yes, proceed to step 412.

In step 410, the variable μ is incremented by 1, even if μ = μ +1. After step 410 ends, return to step 406.

In step 412, it is determined whether the variable q is greater than or equal to Q. If no, proceed to step 414; if yes, proceed to step 416.

In step 414, the variable q is incremented by 1, and the variable μ is initialized to 1, even if q = q +1 and μ = μ +1. After the end of step 414, the process returns to step 406.

In step 416, the resulting variable is used. To establish the RF precoder matrix F _RF and output the RF precoder matrix F _RF .

Method 400 can effectively reduce computational complexity. In detail, the method 400 involves a KU complex multiplication operation in each phase value update, so the method 400 requires a KU ² complex multiplication operation as a whole. In contrast, if you use the exhaustive search method and the MO-AltMin algorithm to solve the problem P 4, you need 2 ^KU ( K +1) U times and (4 KU +10 U ) T respectively. Sub-multiple multiplication, where T is the number of iterations required for the MO-alternating minimization method to reach the stop condition.

FIG. 5 is a signal-to-noise ratio (SNR ratio) and spectral efficiency of an embodiment and a comparative example (including an MO-AltMin algorithm and an orthogonal matching pursuit (OMP) algorithm) according to the present invention. The simulation result of the relationship is that the simulation environment is a millimeter-wave massive MIMO communication system, where N _r , N _t , , , N _s are 36, 144, 3, 3, 3 respectively, and the number of clusters N _cl is 5, and both the transmitting end and the receiving end are uniform square planar arrays with antenna spacing of half wavelength (ie, N _h = N _v ), each The number of conduction paths of a cluster is N _ray is 10, and the average power of each cluster For 1, the positive azimuth arrival angle, the positive azimuth arrival angle, the positive azimuth departure angle, and the frontal exit angle are all Laplacian distributions having a uniformly distributed flat angle and an angular spread (AS) of 10°. The embodiment of the present invention and each of the comparative examples include the simulation results when the resolution bit number B of the phase shifter unit is 2, 3, and 5, respectively, and the comparative example further includes the resolution bit number of the phase shifter unit. The simulation result when B is ∞ (infinity), and each RF beamforming phase is quantized according to equation (13). In addition, FIG. 5 has an optimum digital solution to facilitate comparison between the embodiment of the present invention and each comparative example. Each simulation result is an average of more than 1000 independent channel implementations.

In Figure 5, the signal-to-noise ratio is defined as ρ/ . As can be seen from FIG. 5, when the resolution bit number B and the signal-to-noise ratio of the same phase shifter unit are the same, the spectral efficiency of the embodiment of the present invention is significantly higher than that of the comparative examples. In addition, in the embodiment of the present invention, the spectral efficiency when the resolution bit number B of the phase shifter unit is only 2 is equivalent to the spectral efficiency of the OMP algorithm when the resolution bit number B of the phase shifter unit is ,. In the embodiment of the present invention, the spectral efficiency when the resolution bit number B of the phase shifter unit is 5 is close to the spectral efficiency of the unquantized MO-AltMin algorithm (the number of resolution bits of the phase shifter unit B is ∞) .

Further, FIG. 6 in the present embodiment of the invention the phase shifter unit resolution the number of bits B are 3, 5 and MO-AltMin algorithm when the phase shifter unit B is the number of bits of resolution ∞ The simulation results of the relationship between the number of iterations and the average residual error, where the residual errors are ∥ F _opt - F _RF F _D ∥ ² and ∥ W _opt - W _RF W _D ∥ ² . As can be seen from FIG. 6, the embodiment of the present invention reduces the average residual error when the resolution bit number B of the phase shifter unit increases. In particular, when the resolution bit number B is 5, the average residual error of the embodiment of the present invention converges to approximately 0.52, which is close to the unquantized MO-AltMin algorithm (the number of resolution bits of the phase shifter unit) B is the average residual error of ∞). Moreover, regardless of the size of the resolution bit number B , the embodiment of the present invention can obtain a specific value of the phase shifter unit without exceeding the fifth iteration operation.

It can be seen from the comparison of the spectral efficiency of the above embodiments of the present invention and the comparative example that the spectral efficiency of the embodiment of the present invention using the limited number of resolution bits B is equivalent to the spectrum of the comparative example using the infinite number of resolution bits B. Efficiency, and embodiments of the present invention can quickly converge residual errors. Therefore, the embodiment of the present invention can effectively reduce the computational complexity, increase the speed of obtaining a specific value of the phase shifter unit, and maintain high spectral efficiency and low residual error.

Please refer to FIG. 7. FIG. 7 is a schematic diagram of a communication device 700 according to an embodiment of the present invention. The communication device 700 can be the communication device 110, 120 in the wireless communication system of FIG. 1, the communication device 200 of FIG. 2, or the communication device 300 of FIG. 3, which can be used to perform the method 400. The communication device 700 includes a processing unit 710, a memory unit 720, and a transceiver unit 730. The processing unit 710 can be, for example, a conventional processor, A digital signal processor (DSP), a microprocessor, an application-specific integrated circuit (ASIC), etc., but is not limited thereto. Method 400 can be edited as a code, and the edited code can be stored in memory unit 720. When the communication device 700 is in communication with the remote entity, the processing unit 710 can perform a corresponding operation by reading and executing the code stored in the memory unit 720, including performing the method 400.

The memory unit 720 can be any data storage device that can be read and executed by the processing device 710. The memory unit 720 can be, for example, a subscriber identity module (SIM), a read-only memory (ROM), an erasable programmable read only memory (EPROM), and an electronic erasable memory. Programmable read-only memory (EEPROM), random access memory (RAM), CD-ROM, magnetic tape, hard disk, solid state drive (solid state disk; SSD), flash memory or other data storage device suitable for storing code, but not limited to this. The transceiver unit 730 can be a wireless transceiver that wirelessly communicates with the remote entity based on the results of the processing by the processing unit 710. For example, if the communication device 700 is the transmission end of the communication system, the transceiver unit 730 may include the baseband processing circuit 210, the RF link 220(1)~220(M), and the beamformer as shown in FIG. 2. 230 and the antenna array 240; if the communication device 700 is the receiving end of the communication system, the transceiver unit 730 may include the antenna array 310, the beamformer 320, and the radio frequency links 330(1)-330 (N) as shown in FIG. And a baseband processing circuit 340.

Although the present invention has been disclosed in the above embodiments, it is not intended to limit the present invention, and any one of ordinary skill in the art can make some changes and refinements without departing from the spirit and scope of the present invention. The scope of the invention is defined by the scope of the appended claims.

Claims (10)

A communication device suitable for performing millimeter-wave large-scale multi-input multi-output (MIMO) communication, the communication device comprising: a transceiver unit for wirelessly communicating with a remote entity, the transceiver The unit includes: a baseband processing circuit for processing the plurality of data streams to convert the data streams into a plurality of baseband signals, wherein the baseband processing circuit corresponds to a baseband precoder matrix The plurality of radio frequency links are coupled to the baseband processing circuit, wherein the radio frequency links are used to convert the baseband signals output by the baseband processing circuit into a plurality of radio frequency signals; and a beamformer coupled to the radio frequency signals a beamformer for adjusting a phase of the RF signals, the beamformer corresponding to a matrix of RF precoders, and an antenna array for transmitting the RF signals; and a processing unit for performing The following operations: determining one of the best all-digital precoder matrix of the communication device, and fixing the RF precoder matrix; using the least squares method for the best all-digital precoder Array, the baseband precoding array and the radio frequency precoding array perform operations to obtain an optimized baseband precoder matrix, and update the baseband precoder matrix with the optimized baseband precoder matrix Fixing the baseband precoder matrix; Performing an iterative operation on the optimal all-digital precoder matrix, the baseband precoding array, and the RF precoding array to obtain an optimized RF precoder matrix, and optimizing the RF precoder matrix The RF precoder matrix is updated such that the beamformer processes the RF signals according to the updated RF precoder matrix. The communication device of claim 1, wherein the processing unit performs an iterative operation on the optimal all-digital precoder matrix, the fundamental precoding array, and the radio frequency precoding array by an iteration of less than or equal to 5 . The communication device of claim 1, wherein the beamformer comprises: a plurality of phase shifter unit groups for adjusting phases of the RF signals according to a plurality of phase values; and a plurality of combiners The unit is coupled to the phase shifter unit groups, and the combiner units are respectively configured to combine the RF signals output by the phase shifter unit groups. The communication device of claim 3, wherein the number of resolution bits of the phase values is less than or equal to 5. The communication device of claim 1, wherein the number of the data streams is less than or equal to the number of the radio frequency links, and the number of the radio frequency links is smaller than the antenna in the antenna array. Number. The communication device of claim 1, wherein the radio frequency links respectively upconvert the baseband signals according to a carrier signal, and the carrier signals have a frequency range between 10 GHz and 300 GHz. The communication device of claim 1, wherein the antenna array is a uniform planar antenna array. A method for beamforming is performed in a communication device suitable for performing millimeter-wave large-scale multi-input multi-output communication and having a fundamental frequency processing circuit and a beamformer, the fundamental frequency The processing circuit and the beamformer respectively correspond to a baseband precoder matrix and a radio frequency precoder matrix, the method comprising: determining an optimal full digital precoder matrix of the communication device, and fixing the RF precoder a matrix; the best all-bit precoder matrix, the baseband precoding array, and the radio frequency precoding array are operated using a least squares method to obtain an optimized fundamental frequency precoder matrix, and the best Updating the baseband precoder matrix by the baseband precoder matrix; fixing the baseband precoder matrix; Performing an iterative operation on the optimal all-digital precoder matrix, the baseband precoding array, and the RF precoding array to obtain an optimized RF precoder matrix, and optimizing the RF precoder matrix The RF precoder matrix is updated such that the beamformer processes the RF signals according to the updated RF precoder matrix. The method of claim 8, wherein the iterative operation of the optimal all-digital precoder matrix, the fundamental precoding array, and the radio frequency precoding array is less than or equal to five. The method of claim 8, further comprising: generating a plurality of phase values, so that the beamformer adjusts phases of the RF signals according to the phase values, wherein the phase values of the phase values are resolved. The number is less than or equal to 5.