TWI489772B - Close-loop class-d audio amplifier and control method thereof - Google Patents

Description

Closed-circuit D-class audio amplifier and control method thereof

The invention relates to a class D power amplifier, in particular to a closed-loop fixed frequency class D power amplifier and a control method thereof.

At present, there are many different types of power amplifiers, such as Class A, Class B, Class AB, Class D, etc. Among them, class D power amplifiers are different from other power amplifiers in that they are basically a switching power amplifier or a pulse modulation power amplifier. In this class D power amplifier The device is either fully turned on or completely turned off, greatly reducing the power consumption of the output device. The audio signal is used to modulate the pulse width modulation (PWM) carrier signal, and the carrier signal drives the output power level to obtain the high frequency PWM side. Wave, and finally output the audio signal to the load through the low-pass filter. At present, people design Class D power amplifiers to pay more attention to increasing their power density while reducing costs. In terms of circuit structure, a class D power amplifier can be regarded as a conventional inverter, which inputs a direct current power source and generates an amplified audio signal based on a reference signal. Therefore, all traditional closed-loop control methods, such as transient voltage mode feedback control or voltage-current double feedback loop control, can be used in Class D power amplifiers. A typical device used in this control method is the error amplifier. Usually such a circuit structure is not only complicated but also costly. MPS' patent-protected Analog-to-Availability Modulation (AAM) technology is a relatively simple method for implementing closed-loop control of Class D power amplifiers.

As shown in Fig. 1, in the "AAM" circuit configuration, resistors R1 and R2 form a voltage divider, and node A provides a bias voltage equal to 1/2 Vcc to the non-inverting input terminal Pin of the comparator. The comparator has an internal hysteresis. Comparator in section The input signal of point B is compared with 1/2Vcc±dV, where dV represents the comparator hysteresis voltage, and the output PWM wave of the comparator is alternately turned on by the driving circuit to control the transistors M1 and M2, and the source of the transistor M1 is connected to the output node. C, and the source of the transistor M2 is grounded. The wave of the transistor M1 is connected to the power source Vcc, and the wave of the transistor M2 is connected to the output node C. The transistors M1 and M2 function as switches which form part of the output stage of the class D power amplifier circuit to produce a square wave output at the output node C when the output stage is used in the switching mode. The waveform SW at the output node C is filtered by the filter L composed of the inductor L and the capacitor C1, and after the DC blocking capacitor C2 is restored, the amplified audio signal is sent to a load (such as a speaker). At the same time, the SW signal is charged/discharged to the capacitor Cint through the resistor Rf, thereby achieving adaptive control.

As shown in Fig. 2, when the output PWM square wave of the output node C is at a high level, the capacitor Cint is charged, and Nin, that is, the voltage on the node B rises until reaching the upper limit of the hysteresis, thereby turning off the upper transistor. M1 opens the lower transistor M2, causing the output PWM square wave to go low. When the output PWM square wave of the output node C is low, Cint discharges, and the voltage on Nin drops until the lower limit of the hysteresis is reached, thereby turning off the lower transistor M2 and turning on the upper transistor M1, causing the output PWM square wave to become High level. The high frequency PWM square wave of node C is repeatedly generated in this way, and the filter then filters the high frequency carrier signal to obtain a certain gain amplified output audio signal. The feedback network feeds back the signal of node C to the inverting input of the comparator, thereby controlling the output audio signal to follow the input audio signal and achieve a certain amplification gain. "AAM" technology allows different gains to be set for the amplifier, Good control results can be achieved in both single-ended and bridge configurations. However, the circuit using "AAM" technology has a large change in switching frequency during operation, and its spectrum is easily overlapped with signal bands such as FM/AM, thus affecting the reception quality of these signals, resulting in this control technology in automotive electronics and audio broadcasting. The limitations of other applications.

The technical problem to be solved by the present invention is to provide a closed-loop control method for a class D power amplifier, so that the switching frequency of the amplifier is kept constant, so that it is easy to avoid the band of some important signals, and the control circuit has a simple structure, and some devices can accumulate. Physicalization.

According to an aspect of the present invention, a closed-loop class D power amplifier is provided, comprising: an input stage for receiving an input signal, the input stage comprising a comparator and a triangular wave generator or a rectangular wave generator, the comparator receiving the Inputting a high frequency triangular wave generated by the triangular wave generator or the high frequency rectangular wave generated by the rectangular wave generator, and outputting a first signal; an output stage connected to the input stage, responsive to the first signal Generating a second signal; a filter coupled to the output node of the output stage for filtering the second signal to obtain an output signal; and a feedback network coupled to the output node of the output stage and said Between the input nodes of the input stage, the feedback network performs noise shaping on the second signal to obtain a feedback signal, and negatively feeds the feedback signal to the input stage to subtract from the input signal The feedback signal.

According to the above power amplifier, preferably, the non-inverting input of the comparator loads the high frequency triangular wave, and the inverting input loads the input signal And the feedback signal, the comparator obtaining the first signal by comparing the high frequency triangular wave with a signal obtained by subtracting the feedback signal from the input signal.

According to the above power amplifier, preferably, the non-inverting input of the comparator is loaded with a DC bias voltage, and the inverting input is loaded with the input signal, the feedback signal and the high frequency rectangular wave, the comparison The first signal is obtained by comparing the DC bias voltage with a signal obtained by subtracting the feedback signal from the input signal and then superimposing the high frequency rectangular wave.

According to the above power amplifier, preferably, the feedback network includes a capacitor, the first end of the capacitor being simultaneously connected to the input signal, the inverting input of the amplifier, and the second signal, the second end Ground or connect to DC bias voltage.

According to the power amplifier described above, preferably, the feedback network is connected to a feedback resistor between the first end of the capacitor and the second signal.

According to the power amplifier described above, preferably, the high frequency triangular wave or the high frequency rectangular wave has a larger charge and discharge capability to the capacitance than the second signal.

According to the power amplifier described above, preferably, the input stage has one of the comparators; the output stage includes: a driving circuit connected to the comparator; and a half bridge switching circuit connected to the driving circuit, according to The driving signals generated by the driving circuit are alternately turned on to correspondingly generate a second signal.

According to the power amplifier described above, preferably, the input stage has a The output stage includes: a driving circuit connected to the comparator; two half bridge switching circuits connected to the driving circuit, alternately conducting according to a driving signal generated by the driving circuit, to correspondingly generate Two second signals and one of the second signals is fed back.

According to the above power amplifier, preferably, the input stage has two of the comparators; the output stage includes: a driving circuit respectively connected to the two comparators; two half bridge switching circuits, and the The driving circuits are respectively connected, and are alternately turned on according to a driving signal generated by the driving circuit to generate two of the second signals, and are respectively fed back to the corresponding comparators.

According to another aspect of the present invention, a control method of a closed-loop class D power amplifier is provided, comprising the following steps: a first step of receiving an input signal and a high frequency rectangular wave generated by a high frequency triangular wave or rectangular wave generator generated by a triangular wave generator And outputting a first signal; a second step of generating a second signal in response to the first signal; a third step of filtering the second signal to obtain an output signal; and a fourth step of, the second signal A feedback signal is obtained after noise shaping, and the feedback signal is negatively fed back to an input receiving the input signal to subtract the feedback signal from the input signal.

According to the above control method, preferably, in the second step, two of the second signals are generated in response to the first signal; and in a fourth step, at least one of the two second signals is subjected to noise Negative feedback to the input after shaping.

The class D power amplifier circuit of the present invention replaces the hysteresis loop in the "AAM" circuit structure by using a comparator with a hysteresis loop small enough to be negligible. Device. At the input of the comparator, the low frequency portion of the power stage output signal is cancelled out with the audio input signal, and the high frequency portion is fed to the comparator to obtain a modulated output audio signal at the output.

A fixed-frequency triangular wave signal can be connected to the non-inverting input of the comparator, and the feedback signal and the amplifier input signal obtained from the output stage are connected to the inverting input terminal.

The bias voltage Vdd can also be connected to the non-inverting input of the comparator, and a square wave signal of a fixed frequency, a feedback signal obtained from the output stage, and an amplifier input signal are connected to the inverting input terminal.

The present invention will be further described in detail below in conjunction with the drawings and specific embodiments. For the sake of clarity and conciseness, elements similar to those of FIG. 1 are denoted by like reference numerals.

Figure 3 shows an embodiment of a system according to the invention comprising a class D power amplifier and a load. A high frequency triangular wave with a Vcc/2 DC offset and a frequency of several hundred kilohertz is applied to the noninverting input Pin(A) of the comparator. The inverting input terminal Nin(B) of the comparator is grounded through a capacitor Cint. The input audio signal is charged/discharged to the capacitor Cint through the capacitor Cin and the resistor Ri. The output PWM wave of the comparator is alternately turned on by the driving circuit control transistors M1 and M2, the source of the transistor M1 is connected to the output node C, and the source of the transistor M2 is grounded. The wave of the transistor M1 is connected to the power source Vcc, and the wave of the transistor M2 is connected to the output node C. The transistors M1 and M2 function as switches which form part of the output stage of the class D power amplifier circuit to produce a square wave output at the output node C when the output stage is used in the switching mode. The waveform SW at the output node C is filtered by the filter L composed of the inductor L and the capacitor C1, and after the DC blocking capacitor C2 is restored, the amplified audio signal is sent to a load (such as a speaker). At the same time, the SW signal is charged/discharged to the capacitor Cint through the resistor Rf. Ideally, the charge/discharge effect of the audio signal on the capacitor Cint and the low-frequency component Vsw_low of the SW signal at the output node C can exactly cancel the charge/discharge effect of Cint, that is, the comparator inverting input Nin is at Vsw_low. In conjunction with the audio input, the voltage following the non-inverting input of the comparator is maintained. That is, when the comparator outputs a high level, the transistor M1 is turned on, the transistor M2 is turned off, and the comparator inverting input terminal voltage V _Nin is compared with the sum of the comparator noninverting input terminal Pin voltage and the hysteresis voltage V _Pin +dV ( Where dV represents the comparator hysteresis voltage), at this time, the SW signal is at a high level. After the feedback, the Nin terminal voltage rises until it is greater than the voltage V _Pin +dV, the comparator output jumps to a low level, and the transistor M1 turns off. When the crystal M2 is turned on, the voltage of the inverting input terminal V _{Nin is} compared with V _Pin -dV, and the feedback of the low level signal of the SW causes the voltage of the Nin terminal to drop until it is less than the voltage V _Pin -dV, and the output of the comparator jumps to a high level, resulting in a jump. The SW signal is high, so loop back, as shown in Figure 5. Therefore, in order to achieve this fixed-frequency feedback control and ensure stable operation of the system, the Nin terminal voltage rise and fall speed should be smaller than the change slope of the given triangle wave at the Pin terminal. Therefore, when designing Cint capacitors and Rf feedback resistors, consider the rate of change of the triangular wave given by the Pin terminal. The high frequency component of the SW signal is obtained by the modulation of the high frequency triangular carrier of the noninverting input pin Pin to obtain the gate drive signals of the transistors M1 and M2. The gain of the amplifier is determined by the ratio of the resistors Rf and Ri.

Fig. 4 and Fig. 5 show the main operational waveforms of the circuit of Fig. 3. The switch control method employed by the present invention can be clearly seen from Fig. 5. Therefore, by feedback modulation, when the input audio signal changes, the system will adaptively modulate the SW signal, so that the Nin terminal voltage of the inverting input terminal always follows the non-inverting input terminal Pin, thereby achieving the purpose of controlling the output. Of course, the switching frequency does not remain absolutely constant, but makes small changes in the range of a few hundred hertz. The cause of this small change is the small audio sinusoidal signal present at the inverting input of the comparator, which also charges/discharges Cint, causing a change in the Nin voltage. However, since this change is very small relative to the switching frequency, this control method can still be regarded as fixed frequency control.

Figure 6 is a view showing the basic structure of the control circuit similar to that of Figure 1 according to another embodiment of the present invention, except that the triangular wave applied to Pin in Figure 1 is applied to the inverting input terminal Nin via a resistor Rs. Replaced by a square wave with a 1/2Vcc DC offset. The square wave's charge/discharge effect on the capacitor Cint is similar to the triangular wave directly connected to the non-inverting input pin of the comparator. The non-inverting input pin is directly connected to the 1/2Vcc DC offset. In the previous embodiment, the slope of the given triangular wave is always greater than the slope of the Nin terminal of the inverting input terminal, that is, the slope of the high-frequency charging and discharging ripple on the integrating capacitor Cint, and the charging and discharging effect of the triangular wave on the capacitor Cint is stronger than the feedback signal. The charge and discharge effect of the SW signal on the capacitor Cint. Similarly, it is required that the charge and discharge effect of the given square wave on the capacitor Cint in this embodiment is also stronger than the charge and discharge effect of the feedback signal SW signal on the capacitor Cint.

Fig. 7 and Fig. 8 show the main operational waveforms of the circuit of Fig. 6. Referring to Figure 8, a work cycle can be divided into five phases: the first phase (t0-t1): at time t0, the square wave changes from high to low, SW The signal and the square wave signal simultaneously discharge the capacitor Cint, and the voltage Vcint on Cint continues to decrease; the second phase (t1-t2): at time t1, Vcint<1/2Vcc, the comparator output is inverted, and the SW signal is changed from low to high. . The SW signal charges Cint while the square wave continues to discharge Cint. Since the effect of the discharge is more intense, Vcint drops at a smaller rate; the third stage (t2-t3): at time t2, the square wave changes from low to high, and the SW signal and the square wave simultaneously charge Cint, Vcint Rising; fourth stage (t3-t4): At time t3, Vcint>1/2Vcc, the comparator output flips again, the SW signal goes from high to low, the SW signal discharges Cint and the square wave continues to charge Cint, due to The effect of charging is more intense, so Vcint rises at a small rate; the fifth stage (t4-t5): at time t4, the square wave changes from high to low, and the SW signal and the square wave simultaneously discharge Cint, and Vcint drops. .

As described above, to achieve the constant frequency feedback control of this example, it is required that the Nin terminal voltage continues to decrease when the second stage SW signal changes from low to high, and continues to rise in the fourth stage. That is, it is required to design the feedback resistor Rf, the square wave voltage V _{square wave} with 1/2Vcc DC offset, the SW signal V _sw and the resistance Rs to satisfy the following formula:

That is, it is required that the charge and discharge effect of the given square wave on the capacitor Cint in this embodiment is stronger than the charge and discharge effect of the feedback signal SW signal on the capacitor Cint. Because the square wave has a better charge/discharge effect on the capacitor than the feedback SW The charging/discharging effect of the signal, so although there are several hundred hertz changes, the frequency of the SW signal is generally fixed, determined by the frequency of the square wave.

Similar to the single-ended (SE) class D power amplifier mentioned above, the invention can also be applied to a bridged load (BTL) power amplifier. The inherent differential output structure of the bridge topology eliminates harmonic distortion and DC offset. Figure 9 is a diagram showing another embodiment of the present invention. The H-bridge has two half-bridge switching circuits, typically powered by a single power supply Vcc. For a given Vcc, the differential output mode of the H-bridge circuit provides a maximum output signal amplitude. Double in single-ended mode and four times the output power. Only one comparator is used here, the output of which controls the transistors S1, S2, S3 and S4 via the drive circuit, so that S1, S4 and S2, S3 are alternately turned on, thereby obtaining SW1 and SW2 signals of opposite phases. SW1 and SW2 are respectively sent to the two ends of the load through L1, C1, L2, and C2 rectification filters. It is only necessary to feed back one of the SW1 and SW2 for the feedback control loop. In Fig. 9, SW2 is used as the feedback control signal. Figure 10(a) shows the audio input and audio output waveforms of the circuit of Figure 9. Section 10(a) shows the partial amplification of the operating waveform. It can be seen that the SW1 signal can be obtained by inverting the SW2 signal.

Another embodiment of the present invention in a bridge amplifier system is shown in Fig. 11, in which each half bridge has its own dedicated comparator so that the two drive circuits can be independently controlled. The switching frequency of each bridge arm is set by the same external triangular wave applied to node A, so their frequencies are the same. Except that there is no DC blocking capacitor (see capacitor C2 in Figure 3 and Figure 6), the control circuit structure of each bridge arm is the same as that of the single-ended amplifier, and the gain can also be calculated by Rf/Ri. The input audio signal is a differential signal The numbers are opposite in phase and are applied to the inverting input terminals B1 and B2 of the two comparators, respectively, and compared with the triangular wave. Figure 12(a) shows the audio input and audio output waveforms for the circuit in Figure 11, and Figure 12(b) shows a partial enlargement of the operating waveform.

Those skilled in the art will appreciate that the drive circuits shown in Figures 3, 6, 9, and 11 can be implemented by a gate drive circuit or other circuit capable of performing the same function. In addition, the number of driving circuits shown in FIGS. 3, 6, 9 and 11 is merely illustrative, indicating that the number of driving circuits is such that they can be controlled by the respective comparators in the above figures. And driving the corresponding transistor, it is not intended to limit the number of driving circuits to be equal to the number of blocks representing the driving circuit in the above figure.

Referring to FIG. 13, a control method of a closed-loop class D power amplifier according to the present invention includes the following steps: a first step of receiving an input signal and a high frequency rectangular wave generated by a high frequency triangular wave or rectangular wave generator generated by a triangular wave generator And outputting the first signal; the second step, generating the second signal, that is, the SW signal, in response to the first signal; the third step of filtering the SW signal to obtain an output signal; and the fourth step of performing noise shaping on the SW signal A feedback signal is then obtained and the feedback signal is negatively fed back to the input receiving the input signal to subtract the feedback signal from the input signal. Compared with the well-known "AAM" drive, the present invention only needs to increase a DC bias of 1/2 Vcc. Set the power supply and a suitable triangle or square wave. After these parts are integrated, the closed-loop fixed-frequency control of the class D power amplifier can be realized quite simply.

Ri, Rf‧‧‧ resistance

Pin‧‧‧ non-inverting input

CINT, CIN, C1‧‧‧ capacitor

Nin‧‧‧ Inverting input

M1, M2‧‧‧ transistor

SW‧‧‧ waveform

L‧‧‧Inductance

C2‧‧‧ DC blocking capacitor

Vcc, VDD‧‧‧ bias voltage

S1, S2, S3, S4‧‧‧ transistors

Figure 1 shows a schematic diagram of a known "AAM" Class D power amplifier drive circuit.

Figure 2 shows the operating waveforms of a known "AAM" Class D power amplifier.

Figure 3 is a circuit diagram showing the operation of a single-ended class D power amplifier in accordance with a first embodiment of the present invention.

Fig. 4 is a view showing the waveforms of the audio input signal and the audio output signal of the circuit of Fig. 3.

Fig. 5 is a view showing the waveforms of the SW signal of the circuit of Fig. 3, the signal of the non-inverting input terminal Pin, and the signal of the inverting input terminal Nin.

Figure 6 is a circuit diagram showing the operation of a single-ended class D power amplifier in accordance with a second embodiment of the present invention.

Fig. 7 shows the waveforms of the audio input signal and the audio output signal of the circuit of Fig. 6.

Fig. 8 is a view showing the waveforms of the square wave, the SW signal, the signal of the non-inverting input terminal Pin, and the signal of the inverting input terminal Nin of the circuit of Fig. 6.

Figure 9 is a circuit diagram showing the application of a bridge type D power amplifier in accordance with a third embodiment of the present invention.

Fig. 10A and Fig. 10B show partial waveforms of the circuit of Fig. 9.

Figure 11 is a circuit diagram showing the application of a bridge type D power amplifier in accordance with a fourth embodiment of the present invention.

Fig. 12A and Fig. 12B show partial waveforms of the circuit of Fig. 11.

Figure 13 is a diagram showing the control of a closed-loop Class D power amplifier according to the present invention. Flow chart of the method.

Ri, Rf‧‧‧ resistance

Pin‧‧‧ non-inverting input

CINT, CIN, C1‧‧‧ capacitor

Nin‧‧‧ Inverting input

M1, M2‧‧‧ transistor

SW‧‧‧ waveform

L‧‧‧Inductance

C2‧‧‧ DC blocking capacitor

Vcc, VDD‧‧‧ bias voltage

Claims (9)

A closed-loop class D power amplifier comprising: an input stage for receiving an input signal, the input stage comprising a comparator and a triangular wave generator or a rectangular wave generator, the comparator receiving the input signal and the triangular wave generator Generating a high frequency triangular wave or a high frequency rectangular wave generated by the rectangular wave generator and outputting a first signal; an output stage connected to the input stage, generating a second signal in response to the first signal; a filter, An output node coupled to the output stage for filtering the second signal to obtain an output signal; and a feedback network coupled between the output node of the output stage and the input node of the input stage, The feedback network performs noise shaping on the second signal to obtain a feedback signal, and negatively feeds the feedback signal to the input stage to subtract the feedback signal from the input signal; The feedback network includes a capacitor, the first end of the capacitor being simultaneously connected to the input signal, the inverting input of the comparator, and the second signal, and the second end Ground or DC bias voltage. The power amplifier of claim 1, wherein the non-inverting input of the comparator loads the high frequency triangular wave, and the inverting input loads the input signal and the feedback signal, the comparator The first signal is obtained by comparing the high frequency triangular wave with a signal obtained by subtracting the feedback signal from the input signal. The power amplifier of claim 1, wherein the non-inverting input of the comparator is loaded with a DC bias voltage, and the inverting input terminal is loaded. Inputting the input signal, the feedback signal, and the high frequency rectangular wave, the comparator superposing the high frequency rectangular wave by subtracting the feedback signal from the DC bias voltage and the input signal The resulting signals are compared to obtain the first signal. The power amplifier of claim 1, wherein the feedback network further comprises a feedback resistor coupled between the first end of the capacitor and the second signal. The power amplifier according to claim 1, wherein the high frequency triangular wave or the high frequency rectangular wave has a larger charge and discharge capability than the second signal to the capacitor. The power amplifier of claim 1, wherein: the input stage has one of the comparators; the output stage comprises: a driving circuit connected to the comparator; and a half bridge switching circuit connected to The driving circuit is alternately turned on according to a driving signal generated by the driving circuit to correspondingly generate one of the second signals. The power amplifier of claim 1, wherein: the input stage has one of the comparators; the output stage comprises: a driving circuit connected to the comparator; and two half bridge switching circuits connected To the driving circuit, alternately conducting according to a driving signal generated by the driving circuit to correspondingly generate two of the second signals and feeding back one of the second signals. The power amplifier of claim 1, wherein: the input stage has two of the comparators; the output stage comprises: a driving circuit, respectively connected to the two comparators; and two half bridges The switching circuit is respectively connected to the driving circuit, and is alternately turned on according to a driving signal generated by the driving circuit to generate two of the second signals, and respectively fed back to the corresponding comparators. A control method of a closed-loop class D power amplifier includes the following steps: a first step of receiving an input signal and a high-frequency triangular wave generated by a triangular wave generator or a high-frequency rectangular wave generated by a rectangular wave generator, and outputting a first signal; Step: generating a second signal in response to the first signal; a third step of filtering the second signal to obtain an output signal; and a fourth step of performing noise shaping on the second signal to obtain a feedback signal, And negatively feeding back the feedback signal to an input end of the input signal to subtract the feedback signal from the input signal, wherein the feedback network includes a capacitor, and the first end of the capacitor is simultaneously connected To the input signal, the inverting input of the comparator, and the second signal, the second terminal is grounded or connected to a DC bias voltage.