RU2530322C1 - Method for radio reception of high-speed information of space radio line, and device for its implementation - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: method for radio reception of high-speed information of a space radio line, in which an emitted reference signal is received when a radio wave is leaving ionospheric formation; reception of the emitted reference signal throughout the radio line, in the area of insignificant fading due to Doppler effect; reception in a working session of signals of a spectrum with two side components of the reference signal and the high-speed signal; if a decoder shows status information, compensation of parasitic shifts of spectrum components of the signal due to Doppler effect is performed by variation of phases of components of frequency decomposition by a distortion compensation operator, error-control decoding of a high-speed signal and transmission of the received information to an information recipient.

EFFECT: compensation of determined distortions caused by Doppler effect to reduce signal loss.

2 cl, 15 dwg

Description

The invention relates to space technology, in particular to a method for radio reception of high-speed information of a space radio line and a device for its implementation.

State of the art

Known methods for radio reception of information of a space radio link (IPC H04L 27/22, H04B 1/10, H04L 27/10, H04B 1/06):

A method of obtaining information about signal quality in a receiver [5], RU 2113061 published in 1998;

A method of obtaining information about signal quality in a receiver [6], RU 2216871 published in 2003;

Method and device for codeless signal reception of satellite navigation systems [10], RU 2363099 published in 2007;

A way to eliminate the influence of tropospheric and ionospheric measurement errors in single-frequency receivers of satellite navigation [11], RU 2237257, published in 1982;

A method for transmitting and receiving quadrature amplitude modulation signals, a system for its implementation, a computer-readable medium and applying the method for synchronizing the reception of quadrature amplitude modulation signals [12], RU 2286025, was published in 2005.

The principle of construction and design of devices for radio reception of high-speed information are described in the patent-associated literature, in particular in monographs:

Aces G.I. and others. Interference immunity of radio systems with complex signals. Radio and Communication, 1985;

THEM. Teplyakov, B.I. Roshchin, A.I. Fomin, V.A. Weitzel. Radio transmission systems. Moscow, Radio and Communication, 1982;

Kalashnikov N.I. Communication systems through satellite. "Communication", 1969, as well as in the descriptions of devices (IPC H04B 1/06), in particular:

Radio reception device for high-speed information of a space radio line ”[1], RU 116293 published in 2011;

Digital information radio receiver ”[15], RU 2371845 published in 2008;

A frequency shift device [16], SU No. 824401, published in 1979.

In patents RU 2113061 [5] and RU 2216871 [6], according to the statistics of phase errors of phase-modulated digital symbols, an average estimate of the deterioration of coherent reception is generated depending on the signal-to-noise ratio.

Patent RU 2113061 [5] protects a method for obtaining signal quality information in which a plurality of M-position phase-modulated digital symbols are received by a receiver, a corresponding phase estimate for each phase-modulated digital symbol is generated in the receiver. From the phase estimate, the corresponding phase error signal is determined at the receiver. The method contains signs of:

obtaining information about the quality of the signal at the receiver, at which the receiver receives a plurality of M-position phase-modulated digital symbols;

generating an appropriate phase estimate for each phase-modulated digital symbol in the receiver;

determine at the receiver, from the phase estimate, the phase error signal.

The method has low instrumental accuracy at high signal-to-noise ratios; it requires an excessively large number of characters to evaluate signal quality. The efficiency of use is reduced when using decoders other than convolutional ones, or when using discrete message signals in the transmission system without error-correcting coding, since the method proposes using the angular distance as a measure of the reliability of the decision made.

Patent RU 2216871 [6] protects a method for obtaining information about signal quality at a receiver. The invention relates to the exchange of discrete information over communication channels using digital phase modulation (FM) in discrete information transmission systems.

The technical solution provides high instrumental accuracy at all values of the input signal-to-noise ratio. For the resulting signal-to-noise ratio of the received signal (M-position FM digital signals), a change in the second φ ₂ shifted by φ ₃ = const of the phase estimate φ _{1 of the} phase-modulated digital symbols is generated. According to the calculated first and second estimates, the phases of the digital symbols make a decision on the correspondence of the first and second m-bit code. Find the Hamming distance between the first and second m-bit codes. The resulting Hamming distance of the first and second m-bit codes is summarized. The information obtained is considered to be associated with an equivalent deterioration in the noise immunity of coherent reception by a value in terms of signal / noise and a decrease in noise immunity of coherent reception in a value in terms of signal / noise.

Signs of RU 2216871, coinciding with the essential features of the claimed invention:

generating a phase estimate at the receiver;

determining a phase error signal in the receiver;

receive deterioration in coherent reception from the signal-to-noise ratio.

The efficiency of use is reduced when using decoders other than convolutional ones, or when using discrete message signals in the transmission system without error-correcting coding, since the method proposes to use the angular distance as a measure of the reliability of the decision made. The method has low instrumental accuracy at high signal-to-noise ratios; it requires an excessively large number of characters to evaluate signal quality. The method takes into account the noise components of the radio line and the receiving device and does not take into account the noise components of the Doppler effect, causing deep fading of the received high-speed signal, giving a signal-to-noise ratio below the threshold.

In the Method RU 2363099 [10] the following features apply:

carrier frequency tracking;

measurement of Doppler frequencies of channels by filters with multi-bit digital low-pass filters, PLL system.

Signs of RU 2363099, coinciding with the essential features of the claimed invention:

carrier frequency tracking;

measurement of Doppler frequencies by the PLL system.

The applicant knows the reason that hinders the technical result: the Doppler shift of the spectrum components of the high-speed broadband signal is different from the Doppler shift of the spectral components in the radio frequency band.

The Method RU 2237257 [11] eliminates the influence of tropospheric and ionospheric measurement errors in single-frequency satellite navigation receivers by taking into account vertical tropospheric and ionospheric delays.

The applicant is aware of the reason that impedes the achievement of a technical result - a vertical delay in the ionosphere, which does not make it possible to obtain estimates of the stray phase shift of the spectrum of the received signal.

In the Method RU 2286025 [12] apply the transmission and reception of quadrature amplitude modulation signals, the synchronization of signal reception.

The Method RU 2286025 contains the following features:

types of modulation of satellite communications (p. 4, 30 place of description);

use of noise-resistant coding (p. 4, 35);

carrier frequency recovery (p. 6, 5);

loss of ability to work at low signal-to-noise ratios due to lack of synchronization (p. 6, 20);

dependence of the demodulation threshold on the type of modulation and the type of noise-resistant coding (p. 6, 20);

transmitting (transmitting side) and receiving signals (at the receiving side) over the communication channel (p. 7, 30). Moreover, the receiving side, the input of which is connected to the communication channel, contains conventional amplifiers, filters, and converters to intermediate frequencies that are not shown in the figures, but are assumed to be available (p. 8, 50);

use of direct Fourier transform (p. 9, 15), (p. 15, 30);

the use of phase-locked loop for the selection of clock frequencies (p. 9, 40);

application of calculators for finding the difference of signals (p. 11, 5);

the selection of the intermediate frequency adjustment signal (p. 11, 20);

frequency generation from the meander signal (p. 11, 40);

finding the sum of signals (p. 11, 45);

the selection of low-frequency components of the total signal (p. 11, 50);

analog-to-digital conversions that convert the components of the received signal into digital clock samples (p. 14, 45);

The actions of the method of transmitting and receiving signals are realized, if necessary, not only in hardware but also in software, since the processed signal is sampled, digitized and converted into binary samples. The samples are processed by the computer processor in accordance with the program of the functioning algorithm (p. 17, 45);

using the inverse Fourier transform (p. 21, 35);

conversion of m-level samples into a binary sequence of bit symbols of a clock frequency (p. 22, 50).

Analogue RU 2286025 is the closest analogue prototype of the proposed method. The following features coincide with the essential features of the claimed invention:

receiving signals when transmitting and receiving signals, consisting of a transmitting side and a receiving side connected by a communication channel. The receiving device contains the usual means for any receiver amplification, filtering and conversion to an intermediate frequency, which are not shown in the figures, but are assumed to be available;

use of error-correcting coding;

the use of phase-locked loop;

use of direct Fourier transform;

using the inverse Fourier transform;

the use of calculators to find the difference of signals;

conversion of m-level samples into a binary sequence of bit symbols of the clock frequency.

The features of the method are implemented not only in hardware but also in software, the samples are processed by the computer processor.

The technical result is a decrease in the demodulation threshold, by providing a low threshold for synchronization on the carrier frequency. The result is achieved by supplementing the pack of M m-level symbols, by adjusting the synchronization frequency, which makes it possible to closely approach the Shannon threshold. Shannon's formula:

, $$C=B{\log }_2\frac{P_{\mathrm{from}}+P_w}{P_w}$$

,

Where:

C is the speed of transmission and reception of information,

B is the width of the frequency spectrum of the transmitted signal in the communication channel,

P _with - signal power at the input of the receiver,

P _W - the power of the noise brought to the input of the receiver in the frequency band B.

Calculation of the radio link taking this formula into account implies the achievement of a given reception speed at a signal-to-noise ratio above the acceptable threshold under conditions of compensation for the Doppler shift of the carrier frequency.

The ratio threshold determines the maximum power value.

The reason that prevents the technical result from being obtained is that the Doppler shifts of the spectrum components of the broadband signal differ from the Doppler shift of the carrier, the Doppler shifts of the spectrum components create signal fading and a decrease in the signal-to-noise ratio below the threshold.

Among the analogues of the claimed device include the following technical solutions.

RU 2371845 Digital information radio receiver [15] (published on 10.27.2009, author G. Meleshkov, patent holder of the Federal State Unitary Enterprise “RNII KP”) can be used in radio systems with a phase modulation method for receiving blocks of digital information via communication channels.

Achievable technical result - accounting and partial compensation of stray phase signal offset from the Doppler effect. The device contains a band-pass filter, a matched filter, a balanced modulator, a demodulator, a memory block of digital samples of the amplitude phase detector signal, a decimal symbol block, two block decoders, a signal block receiving chopper, a phase signal processing block, a frequency band shift block, an information receiver, one of the inputs of the balanced modulator is connected to the output of the noise-like signal generator.

Compensation of the stray phase signal offset is carried out more fully in technical solution RU 116293 Radio reception device for high-speed information of a space radio link [1], which is the closest analogue.

A device for receiving high-speed information of a space radio link contains: a band-pass filter (PF), a matched filter for receiving a radio signal from one side (SFOB), a first balanced modulator (BM), a first demodulator (DM), a first block decoder (DK), an information receiver (PI) , memory block counts signal recipient _(P PAM), the apparatus parasitic phase offset compensation signal (SPS CC), the unit shift compensation parasitic spectral components (CC CCC), a matched filter receiving a radio signal from one side (SFDB) in a second balanced modulator (BM), a second demodulator (DM), a second block decoder (DK), a PC of the control unit (PC), a PC bus interface (interface), software (software), and the device input is the PF input, the output of which is connected to SFOB, BM, DM, DK, PI connected in series, the second output of the DM is connected to the PAM _P , the output of which is connected to the SFS CC and the CCC CC, the outputs of which are connected to the second PI input, the BM output is connected to the second input of the reference CCCC signal, input SFOB connected to the input SFDB, the output of which is connected to the input of the connection connected in series, to the second BM, the second DM, the second DC, the output of which is connected to the third input of the PI, the software is connected to the PC, which is connected by the interface to the first DC, the second DC, PI, CC.

However, the known technical solution does not fully compensate for the deterministic distortions caused by the Doppler effect (eliminating deep fading and distortion of the signal passing through the ionosphere) in order to reduce signal loss.

Disclosure of invention

The inventive method

The technical result of the proposed method is to compensate for the deterministic distortions caused by the Doppler effect (elimination of deep fading and distortion of the signal passed through the ionosphere) in order to reduce signal loss.

The technical result is achieved by the fact that radio reception of high-speed information is performed after entering into communication and receiving narrowband signals at a carrier frequency. In the proposed method, reconfiguration of receiving a signal with two lateral, a signal from one lateral, with compensation for stray phase signal displacement and compensation for stray shifts of spectral components is applied. In the radio line, with the Doppler effect, zones are formed at different ranges: signal fading, signal distortion zones and zones where the influence of the Doppler effect is not manifested. Signal fading zones and signal spectrum distortion zones are detected by the decoder when status information is generated. If the receiver has the ability to receive a signal in several configurations, then the reception configuration is the one in which the decoder transmitted the received signal to the information consumer (no status information was found).

The method of radio reception of high-speed information of a space radio line, which consists in the following:

a) receiving the emitted reference signal when the radio wave exits the ionospheric formation (wave propagation time t _S = t ₁ ), including:

entering into communication on a carrier frequency;

synchronization of the clock frequency of the reception of bits, synchronization of reception and reception of the control frequency of the reference signal;

translation of the control frequency into the frequency domain by the fast Fourier transform;

determine in the phase spectrum the total discrepancy φ _Pi1 , i: = 0 ... N-1 of the control frequency (from the ionosphere and the Doppler effect), write the data to the memory;

estimation of the discrepancy between the control frequency and the Doppler effect (prediction)

, i:=0…N-1; $${\varphi }_{Pi\mathrm{eleven}}=({\Omega }_{iD}\cdot t_S)=i\cdot \frac{n_S\cdot 2\pi +{\varphi }_K}{(N_K-N_0)}$$

, i: = 0 ... N-1;

estimation of the difference in the control frequency from the ionosphere φ _ПИИОН : = mod _2π (φ _Пi1 -φ _Пi11 ), for i: = 0 ... N-1 write data to the memory;

b) receiving the emitted reference signal in the continuation of the radio line, in the area of slight fading from the Doppler effect (predicted wave propagation time t _S = t ₂ , containing:

entering into communication on a carrier frequency;

synchronization of the clock frequency of the reception of bits, synchronization of reception and reception of the control frequency of the reference signal;

translation of the control frequency into the frequency domain by the fast Fourier transform;

determination in the phase spectrum of the complete discrepancy φ _Pi2 , i: = 0 ... N-1 of the control frequency (from the ionosphere) and recording data in memory;

estimation of the initial distortions of the reference signal from the Doppler effect in the case of minor distortions by the Doppler effect at t _S = t ₁ after passing through the ionosphere,

φ _ПИИОН : = mod _2π (φ _Пi1 -φ _Пi11 ), i: = 0 ... N-1; writing data to memory;

c) reception in the working session, the propagation time of the signal t _S , spectrum signals with two lateral reference signal s _arr [λ, t] and high-speed signal s [λ, φ (t), t] ,, including:

noise-resistant decoding of the signal s [λ, φ (t), t] and transmission of the received information to the recipient of information;

d) if the operation decoder c) showed the status information, then the following actions are performed:

receiving signals s _arr [λ, t] and s [λ, φ (t), t] in the spectrum of one side and recording signals s _arr [λ, t] and s [λ, φ (t), t];

noise-resistant decoding of the signal s [λ, φ (t), t] and transmission of the received information to the recipient of information;

d) if the operation decoder d) showed the status information, then compensation for the stray offset of the phase signal level, including:

phase estimate φ ₁ at the beginning of interval dimensional phase data memory samples s [λ, φ (t) , t];

estimation of phase φ _{2 at the} end of the measuring interval;

determine the compensated frequency Ω = (φ ₂ -φ ₁ + 2πn) / T,

where n is the number of phase jumps by 2π;

an estimate of the errors in the system, which are divided into small (normal) and large n = 1, 2, ... (abnormal), and the error value is estimated only for normal errors;

mixing compensation is achieved by shifting the frequency band:

e _ol (t _i ) = mod _2π [e _c (t _i ) + e _g (t _i )],

Where:

e _c (t _i ) - samples of the signal from the memory block s [λ, φ (t), t],

e _g (t _i ) - readings of stray displacement;

transform the samples of the phase signal into bits of information by the decisive rule (the conversion of m-level samples into a binary sequence of bit symbols of the clock frequency);

perform noise-resistant decoding of a high-speed signal and the received information is transmitted to the recipient of information;

f) if the operation decoder e) showed the status information, then the stray shift of the spectral components of the recorded high-speed signal s [λ, φ (t), t] is performed using the data of the reference signal s _arr [λ, t] in the working session and the initial values of the reference signal recorded in paragraph b), including:

translation into the frequency domain by the fast Fourier transform of the signals s [λ, φ (t), t] and s _arr [λ, t] e;

determination of the complete discrepancy between the phases of the signal and the control frequency during the signal propagation time t _S ;

determination in the frequency grid i: = 0 ... N-1 of spurious discrepancies in the signal spectrum from the distortion data of the reference signal from the Doppler effect, from small initial distortions φ _Пi11 after passing through the ionosphere t _S = t ₂ to t _S

φ _Пi : = mod _2π (φ _Пi2 -φ _Пi1 ), for i: = 0 ... N-1;

compensation for spurious shifts of the spectral components of the signal s [λ, φ (t), t] from the Doppler effect by changing the phases of the components of the frequency decomposition by the distortion compensation operator:

Ф _i : = mod _2π (arg (Y _i ) -φ _Пi1 ), for i: = 0 ... N-1;

decisive transformation of phase signal samples into bits of information;

performing error-correcting decoding of a high-speed signal and transmitting the received information to the information recipient;

g) if the decoder of operation e) showed the status information, then the stray shift of the spectral components of the recorded high-speed signal s [λ, φ (t), t] is performed using the data of the reference signal s _arr [λ, t] in the working session and the initial values exemplary signal recorded in paragraph b), including:

determination of spurious discrepancies in the frequency grid i: = 0 ... N-1 of the signal spectrum from the distortion data of the reference signal from the Doppler effect, from large initial distortions φ _Pi1 after passing through the ionosphere t _S = t ₂ to t _S

φ _Пi : = mod _2π (φ _Пi2 -φ _ПиИОН ) for i: = 0 ... N-1;

compensation for spurious shifts of the spectral components of the signal s [λ, φ (t), t] of the Doppler effect by changing the phases of the components of the frequency decomposition by the distortion compensation operator:

F _{i: = mod 2π (arg (} Y i) -φ _Pi1) for i: = 0 ... N-1;

decisive transformation of phase signal samples into bits of information;

noise-resistant decoding of a high-speed signal and transmission of the received information to the recipient of information.

The inventive device

The technical result of the claimed device is to compensate for the deterministic distortions caused by the Doppler effect (elimination of deep fading and distortion of the signal passed through the ionosphere), in order to reduce signal loss.

The technical result is achieved by the fact that radio reception of high-speed information is performed after entering into communication and receiving narrowband signals at a carrier frequency. In the proposed device, the reconfiguration of signal reception in real time in the reception frequency band is applied:

two side;

with one side;

with one side and memorized phase signal.

Depending on the Doppler frequency, the period of formation of deep interference fading and signal distortion from a parasitic determinate shift of the spectral components is different, the wave exit time from the ionospheric formation can be much less than the period or close to half the period when the maximum spurious effect from the Doppler shift is formed. In the radio line, with the Doppler effect, zones are formed at different ranges: signal fading, signal distortion zones and zones where the influence of the Doppler effect is not manifested. Signal fading zones and signal spectrum distortion zones are detected by the decoder when status information is generated. If the receiver has the ability to receive a signal in several configurations, then the reception configuration is the one in which the decoder transmitted the received signal to the PI (no status information was found).

A device for receiving high-speed information of a space radio link comprising:

- band-pass filter (PF);

- An agreed filter for receiving a radio signal from one side (SFOB);

- the first and second respectively balanced modulator (BM);

- the first and second, respectively, demodulator (DM);

- recipient of information (PI);

- memory block of phase samples of the receiver signal (PAM _P );

- a device for compensating for stray phase signal displacement (CC SPS);

- a coordinated filter for receiving a radio signal with two side (SPSB);

- a processor configured to form commands:

- recording samples of phase signals and exemplary messages when receiving information signal respectively in the storage unit samples a signal recipient _(P PAM);

- inclusion of a block of the device for compensation of parasitic shift of spectral components (CC SSS);

- control the first and second, respectively, block decoder (DC);

- selection of electrical circuits for FPGA, RAM, ROM, microprocessors, made in the form of boot modules for spacecraft signals with an a priori known structure and the formation of a boot module for reception in a communication session;

- display the status of the reception of information of a space radio link at the consumer of information (PI),

moreover, the input of the band-pass filter (PF) is the input of the device, and the output is connected to the inputs of a matched filter for receiving a radio signal from one side (SFB) and a matched filter for receiving a radio signal with two side (SFB), the first balanced modulator (BM), the input of which is connected to the output of the matched filter for receiving the radio signal from one side (SOPF), and the output is connected to the input of the first demodulator (DM) and the first input of the device for compensating the stray shift of spectral components (CC CC), the second input of which is connected to ode phase signal samples of the storage unit of the recipient (AMP _R) and the second input device compensation of interference offset phase signal (CC SPS), and the parasitic displacement of the phase of the signal output of the compensation unit (MC SPS) connected to the output compensation device parasitic shift spectral components (CC CCC) and the second input of the recipient of information (PI), and the first input of the device for compensating for spurious phase shift signal (CC SPS) is connected to the output of the processor, the input-output of which is connected to the inputs and outputs: Twa shift compensation parasitic spectral components (CC CCC), phase counts memory block signal recipient _(P PAM), the first decoder (DC) and a second decoder (DW), respectively, the block decoder (DC), the recipient information (PI); the first and third inputs of the information recipient (PI) are respectively connected to the outputs of the first (DC) and second decoders (DC), and the input of the first decoder (DC) is connected to the output of the first demodulator (DM), and the input of the second decoder (DC) is connected to the output the second demodulator (DM), the input of the second demodulator (DM) is connected to the output of the second balanced modulator (BM), the input of which is connected to the output of the matched radio signal reception filter with two side ones (SFDB).

Brief Description of the Drawings

The features and essence of the claimed group of inventions are explained in the following detailed description, illustrated by drawings (see Figure 1 - Figure 15), which shows the following.

Figure 1. Scheme of a device for receiving high-speed information of a space radio link;

Figure 2. The scheme of the device for compensation of parasitic bias of the phase signal of the CC SFS;

Figure 3. The scheme of the device for compensation of the parasitic shift of the spectral components of the CC SSS;

Figure 4. Spectral representation of a modulating sinusoid;

Figure 5. Vector representation of the components of the fast Fourier transform;

6. The scheme of the device for compensation of the parasitic shift of the spectral components of the CC SSS;

7. Fading from the Doppler effect;

Fig. 8. Signal modulation meander;

Fig.9. Square wave distortion by Doppler;

Figure 10. The signal of the component vectors;

11. Signal freeze

Fig. 12. The meander distortion by the phase difference of the component frequencies;

Fig.13. Example configuration search;

Fig.14. Lunar station satellite communications network;

Fig.15. A graphical representation of the sequence of operations.

Figure 1 shows a diagram of a device for receiving high-speed information of a space radio link, containing:

1 - band-pass filter (PF);

2 - a consistent filter for receiving a radio signal from one side (SFOB);

3 - coordinated reception of a radio signal from a radio signal with two side (SFDB);

4, 5 - the first and second, respectively, balanced modulator (BM);

6, 7 - the first and second, respectively, demodulator (DM);

8, 9 - the first and second, respectively, decoders (DC);

10 - recipient of information (PI);

11 - memory block phase samples of the receiver signal (PAM _P );

12 is a device for compensating for stray phase signal displacement (CC SPS);

13 is a device for compensating for stray shift of spectral components (CC SSS);

14 - processor, configured to generate commands:

- recording samples of the phase signals of the message and the reference signal when receiving information, respectively, in the memory block of the samples of the signal of the recipient (PAM _P );

- inclusion of a block of the device for compensation of parasitic shift of spectral components (CC SSS);

- control the first and second, respectively, block decoder (DC);

- selection of electrical circuits for FPGA, RAM, ROM, microprocessors, made in the form of boot modules for spacecraft signals with an a priori known structure and the formation of a boot module for reception in a communication session;

- display the status of the reception of information of the space radio link with the consumer of information (PI).

A device for receiving high-speed information of a space radio link containing: a band-pass filter (PF, 1), a matched filter for receiving a radio signal from one side (SFOB, 2), a matched filter for receiving a radio signal with two side (SPSB, 3), the first and second respectively balanced modulator (BM , 4, 5), the first and second, respectively, demodulator (DM, 6, 7), the receiver of information (PI, 10), the memory block of the phase samples of the receiver signal (PAM _P , 11), a device for compensating for stray phase signal offset (CC SFS, 12), a processor (10) made with the ability to form teams:

- recording samples of the phase signals of the message and the reference signal when receiving information, respectively, in the memory block of the samples of the signal of the recipient (PAM _P , 11);

- inclusion of a block of the device for compensation of parasitic shift of spectral components (CC SSS, 13);

- control the first and second, respectively, block decoder (DK, 8, 9);

- selection of electrical circuits for FPGA, RAM, ROM, microprocessors, made in the form of boot modules for spacecraft signals with an a priori known structure and the formation of a boot module for reception in a communication session;

- displaying the state of receiving information of a space radio link at the consumer of information (PI, 10);

moreover, the input of the band-pass filter (PF, 1) is the input of the device, and the output is connected to the inputs of the matched filter for receiving a radio signal from one side (SFOB, 2) and the matched filter for receiving a radio signal with two side (SFDB, 3), the first balanced modulator (BM , 4), the input of which is connected to the output of the matched filter for receiving the radio signal from one side (SOBF, 2), and the output is connected to the input of the first demodulator (DM, 6) and the first input of the device for compensating the stray shift of spectral components (CC SSS, 13) whose second input is The key to the output phase signal samples of the storage unit of the recipient (PAM _II, 11) and the second input device compensation of interference offset phase signal (CC SPS 12), wherein the compensation device output parasitic offset phase signal (CC SPS, 12) connected to the output compensation device spurious shift of the spectral components (CC SSS, 13) and the second input of the recipient of information (PI, 10), and the first input of the parasitic bias compensation device for the phase signal (CC SFS, 12) is connected to the output of the processor (14), the input-output of which is connected the inputs-outputs: compensation device parasitic shift spectral components (CC CCC, 13) of the storage unit of phase samples addressee signals (PAM _II, 11), the first (DC 8) and second (DC) respectively of the decoder block (DC 8 9), the recipient of information (PI); the first and third inputs of the information recipient (PI, 10) are respectively connected to the outputs of the first (DK, 8) and second decoders (DK, 9), and the input of the first decoder (DK, 8) is connected to the output of the first demodulator (DM, 6), and the input of the second decoder (DK, 9) is connected to the output of the second demodulator (DM, 7), the input of the second demodulator (DM, 7) is connected to the output of the second balanced modulator (BM, 5), the input of which is connected to the output of the matched filter for receiving the radio signal with two lateral (SFDB, 3).

Figure 2 shows a diagram of a device for compensating for stray bias of a phase signal (CC SPS, 12), where it is indicated:

10 - recipient of information (PI);

11 - memory block phase reports of the recipient signal (PAM _P );

12 is a device for compensating for stray phase signal displacement (CC SPS);

14 - processor;

15 - calculator stray offset signal (UPU);

16 - digital signal offset compensation device (CCU);

17 - the first decision block character (RBS);

18 - the third block decoder (DK).

The signal from the processor (14) is fed to the first input of the SPS CC (12), which is connected to the first inputs of the IPS (15), the CMS (16), and the second input of the DC (18); the output of the PAM _{P is} connected to the second input of the IPN (15) and the third input of the CCU (16); the second input of the central control unit (16) is connected to the output of the IPN (15); the output of the CCU (16) is connected to the input of the first RBS (17), the output of which is connected to the first input of the third block decoder (DC), the output of which is the output of the SFS CC (12) and connected to the PI input (10).

Figure 3 shows a diagram of a device for compensating for stray shift of spectral components (CC SSS, 13), where it is indicated:

4 - the first balanced modulator (BM);

10 - recipient of information (PI);

11 - memory block phase samples of the receiver signal (PAM _P );

14 - processor;

19 - calculator distortion reference signal (VIS);

20 - block distortion compensation (BKI);

21 is a block of samples of the corrected phase signal (BFS);

22 - the second decisive block character (RBS);

23 - the third block decoder (DK);

24 - memory block samples of a reference signal (PAM _OS );

25 - the third demodulator (DM).

The signal from the output of the first balanced modulator (4) goes to the first input of the CCC CC (13), which is connected to the input of the third demodulator (DM, 25), the output of which is connected to the PAM _OS input (24), the PAM _OS input-output _is connected to the input - VIS output (19), VIS output (19) is connected to the second input of the BKI (20), the output of the BKI (20) is connected to the second input of the BFS (21), and the third input of the BKI (20) is the third input of the CCC CC (13) which is connected to the output of PAM _P (11); the first input of the BKI (20) is connected to the second input of the CCS CC (13), which is connected to the output of the processor (14) and the first input of the BFS (21), the output of the BFS (21) is connected to the input of the second RBS (22), the output of which is connected to the input of the third decoder (DK, 23), the output of which is the output of the CC CCC (13), which is connected to the input of the PI (10).

Figure 4 shows a spectral representation of a modulating sinusoid.

Figure 5 shows a vector representation of the components of the fast Fourier transform.

The following notation is used in the text:

SSPD - means of communication and data transmission;

"Message" - a high-frequency or low-frequency signal of the receiving device in a temporary form or spectral form, carrying information transmitted to the consumer (software);

"Master signal" - model signal s _arr [λ, t], high frequency or low frequency signal receiving apparatus in the temporary shape or the spectral form, used to determine the divergence components of the spectrum of the Doppler effect, is transmitted in the radio link in the sum with the radio signal messages s [λ, φ (t), t] and the synchronization signal s ₁ [λ, t];

s _fs [λ, t] - characteristics of the transmitted message — signal in the block (initial displacement φ _g and parameter Ω _{g of the} linear law of the change in time displacement);

e _with (t _i ) - samples of the phase signal in the transmitted data block.

Figure 6 shows a diagram of a device for compensating for stray shift of spectral components (CC CCC) 13, which contains:

4 - the first balanced modulator (BM);

10 - recipient of information (PI);

11 - memory block phase samples of the receiver signal (PAM _P );

19 - calculator distortion reference signal (VIS);

20 - block distortion compensation (BKI);

21 is a block of samples of the corrected phase signal (BFS);

22 - the second decisive block character (RBS);

23 - the third decoder (DK);

24 - memory side of samples of a reference signal (PAM _OS );

25 - the third demodulator (DM);

26 is the processor.

The signal from the output of the first balanced modulator (4) is fed to the first input of the CCC CC (13), which is connected to the input of the third demodulator (DM, 25), the output of which is connected to the PAM _OS input (24), the PAM _OS input-output _is connected to the input - VIS output (19), VIS output (19) is connected to the second input of the CCI (20), the third input of the CCI (20) is the third input of the CCC CC (13) and connected to the output of the PAM _P (11), and the first input of the CCI ( 20) connected to the second input of the CCC CC (13), which is connected to the output of the processor (14), the first input of the BFS (21) and the second input of the third decoder DC (23); the second input of the BFS (21) is connected to the output of the BCI (20), and the output of the BFS (21) is connected to the input of the second RBS (22), the output of which is connected to the first input of the third decoder (DK, 23), the output of which is the output of the CCC CC ( 13), which is connected to the input of the PI (10).

Figure 7 shows the built-in computers in the Delphi 3 Standart system signal fading patterns, which at different reception ranges from the Doppler effect are periodic. In Fig.7 "a" and "b" shows the side of the radio signal, Fig.7 "c" - the total signal of the side, Fig.7 "g" - fading, spurious decrease in amplitude to zero at the time of reception.

On Fig-11 in the distortion and fading patterns used signal conversion schemes in the receiving devices on the vector diagrams of complex signals [4, p. 27, 31].

On Fig shows the modulation signal meander. If there is no phase shift, then φ _r = 0 (Fig.8 "b").

Figure 9 shows the distortion of the meander by Doppler, the phase shift of the component vectors. There is no phase shift (Fig. 9 “a”): φ _r = 0.

Figure 10 shows the signal of the component vectors, where: φ _r = 0.

Figure 11 shows the fading of the signal at the output of the demodulator, the phase shift of the components is π, φ _r = π.

On Fig shows the distortion of the meander by the phase difference of the component frequencies.

Figure 13 shows an example of a search for configurations in a device during reception, depending on the reception range 0-D ₁ , D ₂ , D ₃ , D ₄ , D ₅ .

On Fig shows an example implementation of the claimed invention during the exploration of the moon and the flight of the expedition to Mars.

On Fig is a graphical representation of the sequence of blocks of operations when implementing the proposed technical solution.

The implementation of the invention

The principle of operation of the claimed method of radio reception of high-speed information of a space radio link and a device for its implementation is as follows.

The radio reception device for high-speed information of a space radio link shown in FIG. 1 performs communication into a carrier frequency, synchronizes a clock frequency of receiving bits, synchronizes reception and reception of a message in real time. Blocks PF (1), SFDB (3), the second: BM (5), DM (7), DC (9) receive a signal with two side. Information from DC 9 goes to the third input of the recipient of information (PI, 10). Blocks PF (1), SFOB (2), the first blocks BM (4), DM (6), DC (8) receive a signal from one side. Information from the first DC (8) is fed to the first input of the PI (10). The matched filters for receiving the radio signal SFOB (2) and SFDB (3) respectively filter the radio signal with one and two side frequencies. The first DM (6) and the second DM (7) perform analog-to-digital conversion of the phase signal. Reception BM, DM, DC is built in the traditional way using optimal reception schemes and high-performance noise-resistant coding schemes.

The status information of the data block of the first decoder (DK, 8), if errors are detected by the first decoder (DK, 8) and are not corrected, is fed to the first input of the information recipient (PI, 10) from the input-output of which goes to the input-output of the processor (14 ) From the input-output of the processor (14), the samples of the phase signals of the message and the reference signal when receiving information, respectively, go to the input-output of the memory block of the samples of the recipient signal (PAM _P , 11) and are recorded in the PAM _P (11). The indicated samples of the signals from the output of the PAM _P (11) are fed to the second input of the device for compensating the parasitic bias of the phase signal (CC SFS, 12) and the second input of the device for compensating the parasitic signal of spectral components (CC SSS, 13). Figure 2 shows an example of the execution of the scheme of the Criminal Code SPS (12). According to the processor command (14), arriving at the first input of the SPS CC (12), the parameters of the stray phase signal level shift are calculated according to the PAM _P data (11) s [λ, φ (t), t] and compensate them in the phase samples of the signal e _c (t _i ), according to which the first decisive block character RBS (17) determines the bits of the data block ("0" or "1"). An error-free information block received by error-correcting decoding is transmitted to the information recipient (PI, 10).

The status information from the third decoder (DK, 18) of the SFS CC unit (12) is fed to the output of the SPS CC (12) connected to the input of the information consumer (PI, 10), the input-output of which is connected to the input-output of the processor (Figure 1 ) The output of the processor (14) receives the command "Enabled" (Figure 1) to the input-output of the PAM _P (11), from the output of which the command goes to the second input of the CC SPS unit (12), which proceeds to the signal processing.

Figure 3 and Figure 6 show the implementation options of the CC CCC scheme (13)

CCC CC (13), at the command of the processor (14), calculates (Fig. 3) the stray shift of the spectral components in the fast Fourier transform (FFT) of the received signal, compensates them in the spectrum and, by converting it to the time domain, obtains samples of the phase signal of the data block. To determine the distortion, a reference signal is used.

In the calculation of spurious distortions, the data of the PAM _OS (24) and PAM _P (11) are used. The spurious distortion of the phases of the spectral components is determined by the difference in the spectral components obtained by the FFT, the received s _sample [λ, t] and the emitted (undistorted by the Doppler effect) reference signal. In the signal spectrum, s [λ, φ (t), t] compensate for spurious phase distortion of the spectral components and the inverse FFT transform produces a vector of time-domain signal samples.

The samples of the corrected signal arrive at the decisive device, which forms a sequence of bits of the data block (characters "0" and "1"). The decoder (DK, 23) performs noise-resistant decoding, the received block of information is transmitted by the PI (10).

The VIS (19), BKI (20), and BFS (21) devices use the fast Fourier transform (FFT) to compensate for distortions. The FFT in the frequency domain has a frequency grid. A grid of N frequencies is formed by frequencies ω ₀ + ω (i), where ω ₀ = 2π · f ₀ , ω (i) = i · Ω, i is the frequency number in the grid, i: = 0, 1, 2, ... N- one. The grid frequency step Ω is unchanged. The difference in neighboring frequencies is the same, the Doppler shift of the difference in neighboring frequencies is the same, we denote it by Ω _D. With the Doppler effect, a grid of difference frequencies i · Ω _D and a grid of discrepancies i · (Ω _D -Ω) are formed on the frequency axis for i: = 0 ... N-1.

The Doppler effect ascertains the linearity of the Doppler frequency shift from the velocity component of a satellite or spacecraft, directed at the point of its location along the tangent to the wave path, curved in the case of inhomogeneous propagation medium.

The linearity property is manifested in a change in the distance between the grid frequencies and a change in each grid frequency with a coefficient a _D from the Doppler effect. We take two grid frequencies f ₁ = k ₁ · Ω, f ₂ = k ₂ · Ω, k ₁ , k ₂ are integers, the difference frequency F ₁ = f ₂ -f ₁ , when there is no Doppler effect. The frequencies with the Doppler effect f _1D , f _2D , the difference frequency F _1D = f _2D -f _1D lies on the frequency axis, where the coefficient of linear frequency change from the Doppler effect a _D , the Doppler effect for points of the frequency axis allows you to write 2πf _1D = a _D 2πf ₁ , 2πf _2D = a _D 2πf ₂ , the discrepancy in the rotational frequencies Ω _rO = 2π (F _2D -F _1D ) = 2π (f _2D -f _1D -f ₂ + f ₁ ) forms a discrepancy in the vibration vectors.

The device uses an exemplary signal with frequencies from a frequency grid. The harmonic frequencies of the reference signal (OS) are called “adjustable” f ₁ and “control” f ₂ . The OS is transmitted via a radio link, is extracted from the input signal from the output of the first BM (4) and goes to the input of the first demodulator (DM, 6), from the second output (DM 6) it goes to the first input of the PAM _P (11), where it is entered into the memory block PAM samples of the reference signal. OS frequencies do not go beyond the spectrum of the information signal. Difference frequency F ₁ = f ₂ -f ₁ when there is no Doppler effect.

The difference frequency F _1D = f _2D -f _1D with the Doppler effect.

The discrepancy Ω _KP = 2π (f _2D -f _1D -f ₂ + f ₁ ).

The discrepancy gives a parasitic phase shift of frequencies, causing signal distortion, depending on the time t _S.

An exemplary signal from the output of the first BM (4) is transferred by the amplitude phase first DM (6) to the low-frequency region. In the device, the local oscillator frequency is taken in the sum of the frequencies ω ₀ + 2πf ₁ . In the region of zero frequencies, a combination of the frequency and phase of the “tunable” frequency with the local oscillator is achieved by the self-tuning system, and an OS frequency difference signal is generated.

The first transformation by the first DM (6) is taken for free space in the absence of Doppler shift, when there is no reception delay, t _S ≈ 0, a frequency oscillation F _{1 is formed} . The second conversion is performed in a communication session, where the frequency fluctuation F _{1D is} formed from the Doppler effect. Digital samples of the phases of the oscillation of the frequencies F ₁ and F _1D of the OS block are stored in the PAM _P (11) reports of the model signal.

The operator for calculating the distortion of the reference signal in the VIS 19 determines the stray phase shifts of the spectral components of the OS for the time from radiation to reception t _S.

VIS 19 translates the fast Fourier transform (FFT) into the frequency domain of the frequency fluctuations F ₁ and F _1D , which determines the discrepancy of the control frequency.

.FFT transformation operators are known, for example [6]: cfft (Y), icfft (F), transformation vectors Y, F. F: = cfft (Y) with respect to the arguments xi: = i · Δ, i: = 0 ... N-1 , $$\Delta :=\frac{T}{N}$$

.

и фазы Φ_i:=arg(Y_i).Vector Y form: modules $$M_i:=|\; |\; |Y_i|\; |\; |$$

and phases Φ _i : = arg (Y _i ).

. Обратное преобразование БПФ выполняет оператор Y:=icfft(F), F - вектор Фурье спектра. В частотной области разложение по частотам $$F_i=(i+1)\cdot \frac{1}{T}$$

, где i:=0…N-1. Числа комплексной формы:Modules are the values of the samples of the amplitudes in the data block of the phase detector, the phases are the samples of the phases of the phase detector in the data block. Sample vectors with the number of elements N = 2 ^{n are} accepted for processing, the missing elements are supplemented with zeros, samples at equal intervals. As a result of the direct transformation, the Fourier vector of the spectrum F is obtained from the vector Y. The components of the vector F: phases ФF _i : = arg (F _i ) and the modules $$M{F}_i:=|\; |\; |F_i|\; |\; |$$

. The inverse FFT transform is performed by the operator Y: = icfft (F), F is the Fourier vector of the spectrum. In the frequency domain, frequency decomposition $$F_i=(i+\mathrm{one})\cdot \frac{\mathrm{one}}{T}$$

where i: = 0 ... N-1. Complex numbers:

- Y: = 19.785j + 0.1;

- Im (Y) = 19.785;

- Re (Y) = 0.1;

- | Z | = 23;

- arg (Z) = 0.1;

- J is a complex unit.

Frequency control phase discrepancy φ _n = (Ω _1D · t _S) for the time t _S in the spectral decomposition of the radian measure find the expression:

φ _П = n _С · 2π + φ _К , where n _C is the number of integer vibrations (2π), φ _К is the phase of frequency discrepancy during the session.

The discrepancy n _C = N _MPA -N _MP , N _MP is the maximum number of the module in the spectrum of the emitted signal (recorded in the SAM of the sample signal reports 14), N _MPA is the maximum number of the module in the session.

The modules and phases of the spectral decomposition of N _MP and N _{MPA of the} three variants “a”, “b”, “c” in Figure 4 determine the total phase difference for time t _{S of the} control frequency. Option "A" Figure 5 shows the module of the resulting vector and its phase components of Figure 4. Option B, FFT decomposition spectrum, vectors of spectral components are shown. The number of components of the spectral decomposition by the fast Fourier transform i: = 0 ... N-1.

The resulting vector is represented by the sum of N-1 frequency decomposition vectors; Fig. 5 shows the modules of the decomposition vectors. To determine the phase of the resulting vector, we will summarize a part of the component vectors that sufficiently fully reflect the length of the resulting vector, for example, take the sum of the numbers of the “complex form” that form the extremum.

максимален, берется N_MPA=i, фаза φ_К=ΦF_i:=arg(F_i), модули остальных составляющих равны нулю.Option "a" - the oscillation period contains an integer number of oscillation periods of the spectral decomposition, ie the oscillation frequency coincides with the frequency of the grid i frequency decomposition of the FFT, module $$M{F}_i:=|\; |\; |F_i|\; |\; |$$

maximum, N _MPA = i is taken, phase φ _К = ΦF _i : = arg (F _i ), the modules of the remaining components are equal to zero.

максимален, берется N_MPA=i. Образуются модули соседних составляющих, убывающие по дальности расположения, фазы составляющих ФF_i:=arg(F_i). Определение фазы φ_К делается по сумме группы составляющих «контрольной» частоты, образующих экстремум, числами «комплексной формы».Option “b” - the oscillation frequency differs from the grid frequencies by less than 0.5 steps, for example by 0.25 steps, the module $$M{F}_i:=|\; |\; |F_i|\; |\; |$$

maximum, N _MPA = i is taken. Modules of neighboring components are formed, decreasing in their range, phases of the components ФF _i : = arg (F _i ). Determination of phase φ _K is the sum of the group comprising the "control" frequency generators extremum numbers "complex shape".

Option “c” - the oscillation frequency differs from the grid frequencies by 0.5 steps. Module MF _i : = 0. Equal modules of neighboring components are formed, the remaining modules decrease in terms of location, component phase, ФF _i : = arg (F _i ). The resulting vector is on the verge of jumping, either left or right by 0.5 steps, we believe that it did not jump, remained in the middle of N _MPA = i, where the maximum modulus is zero. The determination of the phase φ _K is made by the sum of the group of components of the "control" frequency, forming the extremum, by the numbers of the "complex form". The total phase difference of the control frequency for time t _S is

φ _PK = n _C · 2π + φ _K

. При Δφ=φ_ПК, Δt=t_S получим $$\Omega =\frac{{\varphi }_{ПК}}{t_S}$$

.We write the frequency discrepancy causing the phase discrepancy Δφ over time Δt as a frequency $$\Omega =\frac{\Delta \varphi }{\Delta t}$$

. When Δφ = φ _PC , Δt = t _S we get $$\Omega =\frac{{\varphi }_{P\mathrm{TO}}}{t_S}$$

.

.Substituting φ _PC , we determine the discrepancy by the grid step $${\Omega }_r=\frac{n_C\cdot 2\pi +{\varphi }_{\mathrm{TO}}}{t_S\cdot (N_{\mathrm{TO}}-N_0)}$$

.

Grid of frequency discrepancies Ω _iD = i · Ω _r , i: = 0 ... N-1

The total phase difference φ _Пi (spurious displacements) during the time t _{S of the} spectral components Ω _{iD is} determined using the formula for the grid of frequency differences

, i:=0…N-1 $${\varphi }_{Pi}=({\Omega }_{iD}\cdot t_S)=i\cdot \frac{n_S\cdot 2\pi +{\varphi }_{\mathrm{TO}}}{(N_{\mathrm{TO}}-N_0)}$$

, i: = 0 ... N-1

, i:=0…N-1. Фазы - отсчеты фазового сигнала ПАМ отчетов сигнала получателя 12. Модули отсчетов полагаем единице, M_i=1. Получаем вектор F с составляющими: фазы Φ_Fi:=arg(F_i) и модули $$M_{Fi}:=\left|F_i\right|$$

, i:=0…N-1.BKI 20 uses the vector φ _{Пi of the} parasitic phase displacement of the VIS 19 block, translates, with the PPBS 7 command, the operator F: = cfft (Y) the vector Y with the components: phases Φ _i : = arg (Y _i ), modules $$M_i:=|\; |\; |Y_i|\; |\; |$$

, i: = 0 ... N-1. Phases - samples of the phase signal PAM reports of the signal of the recipient 12. We assume the units of samples to unity, M _i = 1. To obtain the vector components F: Phase _{Φ Fi: = arg (F i} ) and modules $$M_{Fi}:=|\; |\; |F_i|\; |\; |$$

, i: = 0 ... N-1.

From the Doppler effect at the time of reception, only the frequency changes, the signal modules from the relative speed of the receiver and transmitter do not change, therefore the modules of the vectors M _Fi , the frequency decomposition with the Doppler effect, are considered unchanged. Frequency changes over time t _{S are} taken into account by the complete phase difference φ _{Пi of the} components of the decomposition frequencies.

Distortion compensation is performed by changing the phases of the components of the frequency decomposition. Distortion Compensation Operator:

Φ _i : = mod _2π (arg (Y _i ) -φ _Пi ) for i: = 0 ... N-1

Φ _Fi phases contain distortion.

The output of the BKI block (20) is the phase vector Ф _Fi and the vector of modules M _Fi i: = 0 ... N-1.

Block samples the corrected phase signal (BFS, 21) uses vectors P _i, M _{Fi, i:} = 0 ... N-1 obtained in the BCH (20) of the second input, converts the frequency-domain signal into a time domain operator inverse transform Y: = icfft (F) where:

vector F - vector of Fourier spectrum data with components;

Φ _i - phase vectors;

M _Fi is the vector of modules.

The output of the BFS (21) is the phase reference vector Φ _Fi = arg (Y _i ), i: = 0 ... N-1.

In the signal conversion operators of the receiving device, the number of samples N does not change, the normalization of the base system is not violated, which corresponds to the requirements of transforms in the FFT [4, p. 224].

The processor unit 14 stores electrical circuits for FPGA, RAM, ROM, microprocessors, made in the form of loading modules for spacecraft signals with an a priori known structure. The processor 14 selects the boot module from the software and includes receive configurations in the communication session.

Figure 6 shows the scheme of the CCC CC, containing a distortion calculator of the reference signal (VIS, 19), a distortion compensation unit (BKI, 20), a sample block of the corrected phase signal (BFS, 21), and a second decimal symbol block (RBS, 22), the third block decoder (DK, 23), the status information output of which is connected to the output of the CC SSS (13), a memory block of samples of the reference signal (PAM _OS , 24), the third demodulator (DM, 25), the input of the reference signal of the CC SSS 13 _-1 connected to the DM input (25), the output of which is connected to the PAM _OS (24), the input-output of which is connected to the VIS input-output (1) 9), the output of which is connected to the second input of the BKI (20), the output of which is connected to the second (signal) input of the BFS (21), the output of which is connected to the input of the second EinSS (22), the output of which is connected to the first input of the third DC (23) , the (signal) output of which is the output of the CCS 13 CC, the third input (signal - message) of the CCC _{13-3 is} connected to the third input of the BCI (20), the output of the BCI (20) is connected to the second input of the BFS (21), the output of the BFS ( 21) connected to the input of the second EinSS (22); the input of CC CCC _{13-2 is} connected to the first input of the BCI (20), the first input of the BFS (21), characterized in that the second input (signal of status information) of the third decoder DC (23) is connected to the second input of the CC CCC, connected to the processor output .

A method for eliminating distortions from the Doppler effect, when passing through the ionosphere radio link (explanation of Fig. 15), ensures the achievement of a technical result, which consists in reducing distortion and signal loss in circuits with noise-resistant coding in a communication session, depending on the Doppler effect and contains:

a) receiving the emitted reference signal when the radio wave exits the ionospheric formation (wave propagation time t _S = t ₁ ), including:

entering into communication on a carrier frequency;

synchronization of the clock frequency of the reception of bits, synchronization of reception and reception of the control frequency of the reference signal;

translation of the control frequency into the frequency domain by the fast Fourier transform;

determine in the phase spectrum the total discrepancy φ _Pi1 , i: = 0 ... N-1 of the control frequency (from the ionosphere and the Doppler effect), write the data to the memory;

estimation of the discrepancy between the control frequency and the Doppler effect (prediction)

, i:=0…N-1; $${\varphi }_{Pi\mathrm{eleven}}=({\Omega }_{iD}\cdot t_S)=i\cdot \frac{n_S\cdot 2\pi +{\varphi }_{\mathrm{TO}}}{(N_{\mathrm{TO}}-N_0)}$$

, i: = 0 ... N-1;

estimation of the difference in the control frequency from the ionosphere φ _ПИИОН : = mod _2π (φ _Пi1 -φ _Пi11 ), for i: = 0 ... N-1; write data to memory;

b) receiving the emitted reference signal in the continuation of the radio line, in the region of slight fading from the Doppler effect (predicted wave propagation time t _S = t ₂ ), containing:

entering into communication on a carrier frequency;

synchronization of the clock frequency of the reception of bits, synchronization of reception and reception of the control frequency of the reference signal;

translation of the control frequency into the frequency domain by the fast Fourier transform;

determining the spectrum of the phase difference φ complete _Pi2, i: = 0 ... N-1 reference frequency (the ionosphere) and writing data into memory;

estimation of the initial distortions of the reference signal from the Doppler effect in the case of minor distortions by the Doppler effect at t _S = t ₁ after passing through the ionosphere,

_Pi1 φ: = mod _2π (φ -φ _{Pi1 Pi2),} i: = 0 ... N-1; writing data to memory;

c) reception in the working session, the propagation time of the signal t _S , spectrum signals with two side reference signal s _arr [λ, t] and high-speed signal s [λ, φ (t), t], including:

noise-resistant decoding of the signal s [λ, φ (t), t] and transmission of the received information to the recipient of information;

d) if the operation decoder c) showed the status information, then the following actions are performed:

receiving signals s _arr [λ, t] and s [λ, φ (t), t] in the spectrum of one side and recording signals s _arr [λ, t] and s [λ, φ (t), t];

noise-resistant decoding of the signal s [λ, φ (t), t] and transmission of the received information to the recipient of information;

d) if the operation decoder d) showed the status information, then compensation for the stray offset of the phase signal level, including:

phase estimate φ ₁ at the beginning of interval dimensional phase data memory samples s [λ, φ (t) , t];

estimation of phase φ _{2 at the} end of the measuring interval;

determine the compensated frequency Ω = (φ ₂ -φ ₁ + 2πn) / T,

where n is the number of phase jumps by 2π;

an estimate of the errors in the system, which are divided into small (normal) and large n = 1, 2, ... (abnormal), and the error value is estimated only for normal errors;

mixing compensation is achieved by shifting the frequency band:

e _ol (t _i ) = mod _2π [e _c (t _i ) + e _g (t _i )

Where:

e _c (t _i ) - samples of the signal from the memory block s [λ, φ (t), t],

e _g (t _i ) - readings of stray displacement;

transform the samples of the phase signal into bits of information by the decisive rule (the conversion of m-level samples into a binary sequence of bit symbols of the clock frequency);

perform noise-resistant decoding of a high-speed signal and the received information is transmitted to the recipient of information;

f) if the operation decoder e) showed the status information, then the stray shift of the spectral components of the recorded high-speed signal s [λ, φ (t), t] is performed using the data of the reference signal s _arr [λ, t] in the working session and the initial values of the reference signal recorded in paragraph b), including:

translation into the frequency domain by the fast Fourier transform of the signals s [λ, φ (t), t] and s _arr [λ, t] e;

determination of the complete discrepancy between the phases of the signal and the control frequency during the signal propagation t _S ;

determination in the frequency grid i: = 0 ... N-1 of spurious discrepancies of the signal spectrum from the distortion data of the reference signal from the Doppler effect, from small initial distortions φ _Pi1 after passing through the ionosphere t _S = t ₂ to t _S

φ _Pi: = mod _2π (φ -φ _{Pi1 Pi2)} for i: = 0 ... N-1;

compensation for spurious shifts of the spectral components of the signal s [λ, φ (t), t] from the Doppler effect by changing the phases of the components of the frequency decomposition by the distortion compensation operator:

Ф _i : = mod _2π (arg (Y _i ) -φ _Пi ) for i: = 0 ... N-1;

decisive transformation of phase signal samples into bits of information;

performing error-correcting decoding of a high-speed signal and transmitting the received information to the information recipient;

g) if the decoder of operation e) showed the status information, then the stray shift of the spectral components of the recorded high-speed signal s [λ, φ (t), t] is performed using the data of the reference signal s _arr [λ, t] in the working session and the initial values exemplary signal recorded in paragraph b), including:

determination of spurious discrepancies in the frequency grid i: = 0 ... N-1 of the signal spectrum from the distortion data of the reference signal from the Doppler effect, from large initial distortions φ _Pi1 after passing through the ionosphere t _S = t ₂ to t _S

φ _Пi : = mod _2π (φ _Пi2 -φ _ПиИОН ), for i: = 0 ... N-1;

compensation for spurious shifts of the spectral components of the signal s [λ, φ (t), t] from the Doppler effect by changing the phases of the components of the frequency decomposition by the distortion compensation operator:

φ _i : = mod _2π (arg (Y _i ) -φ _Пi ) for i: = 0 ... N-1;

decisive transformation of phase signal samples into bits of information;

noise-resistant decoding of a high-speed signal and transmission of the received information to the recipient of information.

These operations can reduce distortion and signal loss in circuits with error-correcting coding in a communication session, depending on the Doppler effect and reception range, with bitwise transmission of numerical information in free space. A lack of action arises after the radio signal passes through the ionosphere. The ionosphere gives spurious phase shifts of the components of the signal spectrum s [λ, φ (t), t] and the spectrum of the reference signal s _obrned [λ, t] and s _arr [λ, t] as a result of these shifts, signal distortions s [λ, φ (t), t].

Changes in the parameters of the emitted radio signal from the propagation conditions of the signal in the ionosphere are known, for radio waves whose frequencies exceed the critical frequencies of the ionospheric layers (the frequency f = 11 GHz refers to such frequencies), the formulas / 2, page 185 / change in the phase of the wave dφ when passing through waves of frequency f in the ionosphere of the path dh:

где $$\sqrt{\epsilon }\approx 1-\mathrm{40,5}\frac{N_{Э}}{f^2}$$

, ε - относительная диэлектрическая проницаемость, N_э - электронная плотность. $$d{\varphi }_{\mathrm{one}}=\frac{2\pi f_{\mathrm{one}}}{c}\sqrt{{\epsilon }_{\mathrm{one}}}dh(\mathrm{one}),$$

Where $$\sqrt{\epsilon }\approx \mathrm{one}-40.5\frac{{\mathrm{<figure-callout\; id="2"\; label="N\; E\; f"\; filenames="00000059.png,00000060.png"\; state="\{\{state\}\}">N\; </figure-callout>}}_E}{f^{}}$$

, ε is the relative dielectric constant, N _e is the electron density.

When the wave of the ionosphere crosses the electron density N _{e sr of the} region Δh in the sample signal from the oscillations f ₁ and f _{2, the} phase shift will change. Ionospheric distortions add spurious phase shifts to the phases of the reference signal, which are transferred to the phase shifts of all spectral components of the reference signal from the Doppler effect. Instead of eliminating distortion, distortion will be amplified in all spectral components of the received signal of the communication session, which is a disadvantage.

The disadvantage is eliminated by the following operations of the claimed method.

Two ratings are entered:

- φ _{Pi11 is an} estimate of the frequency difference of the reference signal from the Doppler effect (predicted estimate of the control frequency), during the transit time of the radio wave ionosphere t _S = t ₁ .

- φ _Pi12 estimate the frequency difference of the reference signal of the communication session (estimate the frequency difference of the ionosphere with the Doppler effect), estimate the frequency difference of the reference signal found during the wave along the radio line (in the region of slight fading from the Doppler effect), t _S = t ₂ .

Blocks: - PF (1), SFDB (3), second blocks: BM (5), DM (7), DC (9) receive a signal with two side ones. Information from the recreation center (9) receives PI (10).

Blocks: PF (1), SFOB (2), the first blocks: BM (4), DM (6), DC (8) receive a signal from one side. Information from the second recreation center (8) receives PI (10).

The matched filters for receiving the radio signal SFOB (2) and SFDB (3) respectively filter the radio signal with one and two side frequencies.

The first and second DMs (6, 7) perform analog-to-digital conversion of the phase signal.

Transmission: BM, DM, DK is built in the traditional way using optimal reception schemes and highly efficient noise-resistant coding schemes. The status information of the data block of the second recreation center (9), if errors are detected by the second decoder of the recreation center (9) and are not corrected, is sent to the consumer of information (PI 10), the input-output of which is connected to the input-output of the processor (14). At the command of the processor (14), the samples of the phase signals of the message and the reference signal are recorded when receiving information, respectively, in the memory device; - samples of signals from the memory are transmitted to the CC SFS (12) and CC SSS (13).

The parasitic shift compensation device for the spectral components (CCC CC, 13), by the processor command (14), calculates the parasitic shift of the spectral components in the fast Fourier transform (FFT) of the received signal, compensates them in the spectrum and, by converting them to the time domain, obtains phase signal samples of the data block.

To determine the distortion, a reference signal is used. In calculations of spurious distortions, the data of the PAM _OS (24) and PAM _P (11) are used.

The spurious distortion of the phases of the spectral components is determined by the difference in the spectral components obtained by the FFT, the received s _sample [λ, t] and the emitted (undistorted by the Doppler effect) reference signal.

In the signal spectrum, s [λ, φ (t), t] compensate for spurious phase distortion of the spectral components and the inverse FFT transform produces a vector of time-domain signal samples. The samples of the corrected signal arrive at the decisive device, which forms a sequence of bits of the data block (characters "0" and "1"). The decoder performs error-correcting decoding, the received block of information is transmitted to the consumer of information (PI, 10).

The devices used a reference signal (OS) with frequencies from the frequency grid of the FFT transforms. OS harmonic frequencies are called “adjustable” f ₁ and “control” f ₂ .

The OS is transmitted via a radio link, is extracted from the input signal by the first BM (4), then it is converted into the first DM (6) and entered into the memory block PAM _OS (24) of samples of the reference signal. OS frequencies do not go beyond the spectrum of the information signal. Difference frequency F ₁ = f ₂ -f ₁ when there is no Doppler effect. The difference frequency F _1D = f _2D -f _1D with the Doppler effect. The discrepancy Ω _KP = 2π (f _2D -f _1D -f ₂ + f ₁ ). The discrepancy gives a parasitic phase shift of frequencies, causing signal distortion, depending on the time t _S.

The devices BKI 20 and BFS 21 used the fast Fourier transform (FFT) to compensate for distortion. The FFT in the frequency domain forms a grid of N frequencies, with frequencies ω ₀ + ω (i), where ω ₀ = 2π · f ₀ , ω (i) = i · Ω, i is the frequency number in the grid, i = 0, 1, 2, ... N-1. The grid frequency step Ω is unchanged. The difference in neighboring frequencies is the same, the Doppler shift of the difference in neighboring frequencies is the same, we denote it by Ω _D. With the Doppler effect, a grid of difference frequencies i · Ω _D and a grid of discrepancies i · (Ω _D -Ω) are formed on the frequency axis for i = 0, 1, 2 ... N-1.

. Вектор Y образуют: модули $$M_i:=\left|Y_i\right|$$

, и фазы Ф_i:=arg(Y_i). Модули - значения отсчетов амплитуд в блоке данных фазового детектора, фазы - отсчеты фаз фазового детектора в блоке данных. Принимаются к обработке векторы отсчетов с числом элементов N=2ⁿ, недостающие элементы дополняются нулями, отсчеты через равные промежутки. В результате прямого преобразования, из вектора Y получается вектор Фурье спектра F. Составляющие вектора F: фазы ФF_i:=arg(F_i) и модули $$M{F}_i:=\left|F_i\right|$$

. Обратное преобразование БПФ выполняет оператор Y:=icfft(F), F - вектор Фурье спектра. В частотной области разложение по частотам $$F_i=\left(i+1\right)\cdot \frac{1}{T}$$

, где i:=0…N-1. Числа комплексной формы: Y, Im(Y), Re(Y), $$\left|Z\right|$$

, arg(Z), J.The FFT transform operators are applied [1]: cfft (Y), icfft (F), transformation vectors Y, F. F: = cfft (Y) with respect to the arguments xi: = i · Δ, i: = 0 ... N-1, $$\Delta :=\frac{T}{N}$$

. Vector Y form: modules $$M_i:=|\; |\; |Y_i|\; |\; |$$

, and phases Φ _i : = arg (Y _i ). Modules are the values of the samples of the amplitudes in the data block of the phase detector, the phases are the samples of the phases of the phase detector in the data block. Sample vectors with the number of elements N = 2 ^{n are} accepted for processing, the missing elements are supplemented with zeros, samples at equal intervals. As a result of the direct transformation, the Fourier vector of the spectrum F is obtained from the vector Y. The components of the vector F: phases ФF _i : = arg (F _i ) and the modules $$M{F}_i:=|\; |\; |F_i|\; |\; |$$

. The inverse FFT transform is performed by the operator Y: = icfft (F), F is the Fourier vector of the spectrum. In the frequency domain, frequency decomposition $$F_i=\left(i+\mathrm{one}\right)\cdot \frac{\mathrm{one}}{T}$$

where i: = 0 ... N-1. Complex numbers: Y, Im (Y), Re (Y), $$|\; |\; |Z|\; |\; |$$

, arg (Z), J.

The total phase difference φ _Пi (spurious displacements) during the time t _{S of the} spectral components Ω _iD is determined using the formula for the grid of frequency differences

, i:=0…N-1 $${\varphi }_{Pi}=({\Omega }_{iD}\cdot t_S)=i\cdot \frac{n_S\cdot 2\pi +{\varphi }_{\mathrm{TO}}}{(N_{\mathrm{TO}}-N_0)}$$

, i: = 0 ... N-1

the discrepancy of the spectral components for the reference signal passing through the ionosphere, t _S = t ₂ is determined, the discrepancies from the Doppler effect are obtained.

Discrepancies from the Doppler effect can form fading and distortion of signals. In the distortion and fading patterns in Figs. 8-11, signal conversion schemes in the receiving devices on the vector diagrams of complex signals are used [4, p. 27, 31]. The fading patterns of Fig. 7 are built by a computer in the Delphi 3 Standart system. Signal fading at different reception ranges from the Doppler effect is periodic.

The discrepancy between the two coherent frequencies of harmonic oscillations with the frequency ratio f ₂ = kf ₁ [2, p. 185] form the discrepancy f _r = Δf _and -Δf, where Δf = f ₂ -f ₁ is the difference frequency in free space before the ionosphere passes, Δf _and - difference frequency after passing through the ionosphere. The discrepancy also arises from the Doppler effect. The difference frequencies Δf and Δf _d , where the difference Δf = f ₂ -f ₁ when there is no Doppler effect; Δf _d , there was a Doppler effect, form a discrepancy f _r = Δf _d -Δf. The discrepancy f _r forms a phase shift of the oscillations φ _r , during the time t of the signal propagation before reception, φ _r = mod _2π  _2π · f _r · t. The values of φ _r can be different, including zero and π, with a period of circular discrepancy frequency Ω _D = 2π · f _r .

The discrepancy between the frequencies and the change in the waveform can be seen when the carrier frequency is modulated by the meander, the case of radiation emission in one side frequency band, ω ₀ + Ω = 2π · f _H , ω ₀ + kΩ = 2π · f _B , k = 3.

When receiving a signal in the absence of Doppler offsets, the shape of the received signal is shown in Figure 4 "c". Dimension of frequency values Hz (hereinafter).

Example, the carrier frequency f ₀ = 11000 × 10 ⁶ , the components f _H = 11000 × 10 ⁶ + 5 × 10 ⁶ , f _B = 11000 × 10 ⁶ + 15 × 10 ⁶ , the difference frequency Δf = f _B -f _H = 10 * 10 ⁶ . Doppler shift of the carrier frequency 50 × 10 ³ , f _0Д = 11000 × 10 ⁶ + 50 × 10 ³ .

The discrepancy in the coherently radiated frequencies leads to the formation of a phase shift φ _r = π, we obtain t _S = 11 ms. For 11 ms, the wave travels 3.3 thousand km. The oscillation phase of figure "b" Fig.9 has changed relative to "b" Fig.8. The divergence of the component frequencies of the meander at a reception range of 3.3 thousand km gave a distortion in Fig.9 "c".

Mirror frequencies. Change in the signal value (fading) from the Doppler frequency shift when receiving a radio signal with vibrational components of the upper side frequency band ω ₀ + Ω = f _B and the lower side frequency band ω ₀ -Ω = f _H.

Carrier frequency f ₀ = 11000 × 10 ⁶

Modulating frequency Ω = 15 × 10 ⁶

The difference frequency of the component frequencies Δf is Δf = f _B -f _H = 30 × 10 ⁶ .

Doppler effect. Doppler shift of the carrier frequency 50 × 10 ³ .

The difference frequency Δf _D = f _VD -f _ND = 30 × 10 ⁶ +136.36

The discrepancy between the frequencies f _r = Δf _D -Δf = 136.36.

Figure 10 shows the signal of the demodulator when the phase shift of the frequency components of the signal of the meander was not.

The discrepancy in the frequencies f _r leads to the formation of a phase shift φ _r in the plane Pl (t = 0 + t _S ) versus time φ _r = mod _2π [2πf _r t _S ]. We determine t _S , with a phase shift π, we obtain t _S = 3.7 ms. For 3.7 ms, the wave travels the path of 1.1 thousand km.

With a phase shift of π, the signal amplitude decreased and became equal to zero, Fig. 11, the signal fading.

Signal fading - a parasitic decrease in amplitude to zero at the time of reception is illustrated in Fig. 7 "d", Fig. 12 shows the meander distortion by the phase difference of the component frequencies, where the frequency offset during modeling was taken into account by the coefficient Δω / ω = 0.015. The graphs shown in Fig.7 and Fig.12, built on a computer in the Delphi 3 Standart system.

In the radio line, with the Doppler effect, zones are formed at different ranges: signal fading, signal distortion zones and zones where the influence of the Doppler effect is not manifested. Signal fading zones and signal spectrum distortion zones are detected by the decoder when status information is generated. If the receiver has the ability to receive a signal in several configurations, then the reception configuration is the one in which the decoder transmitted the received signal to the PI (no status information was found).

Depending on the Doppler frequency, the period of formation of deep interference fading and signal distortion from a parasitic determinate shift of the spectral components is different, the wave exit time from the ionospheric formation can be much less than the period or close to half the period when the maximum spurious effect from the Doppler shift is formed.

To implement the invention, a change of configurations is used. An example of a configuration change in the device during reception, from the reception range D, is shown in Fig.13.

The first configuration “a” is real-time reception, a radio signal with two side signals. Zones 0-D ₁ , D ₂ -D ₃ , D ₄ -D _{5 are} marked with slight fading, where distortion of the signal from the CPIS is not observed, and zones D ₁ , -D ₂ , D ₃ , - D ₄ are signal fading zones, where the ratio P _С / P _Ш decreases and reaches a threshold value due to a decrease in the power of the received signal; Reconfiguration of reception according to the status information of the decoder at a range of D ₁ -D ₂ , D ₃ -D ₄ .

The second configuration “b” - real-time reception, a radio signal from one side. Distortion from the Doppler effect is not observed at a range of up to D ₁ , from D ₂ to D ₃ . Distortions are detected by the status information of the decoder at a range of D ₁ -D ₂ . Reconfiguration of reception according to the status information of the decoder at a range from D ₁ to D ₂ .

The third configuration “c” is the compensation of the parasitic displacement of the phase signal φ _g , the processing of the phase signal from the PAM memory block, the radio signal from one side. Distortion from the Doppler effect is not observed at a range of up to D ₁ , from D ₂ to D ₃ . Distortions are detected by the status information of the decoder at a range of D ₁ -D ₂ . Reconfiguration of reception according to the status information of the decoder at a range from D ₁ to D ₂ .

The fourth configuration “g” is the compensation of spurious shifts of the spectral components from the Doppler effect in the phase signal of the PAM memory block, a radio signal from one side. Implementation of the reception is depicted when, at the reception range within the estimated range, when decoding the data block, no errors were found in the information or all errors were corrected.

The influence of the ionosphere and Doppler shift of received radio frequencies on the quality of reception of transmitted information in the space segment is characteristic of the Earth surrounded by the ionosphere. The Earth’s ionosphere leads to loss of power, gives a change in wave polarization and fading, introduces errors in determining the angular coordinates of spacecraft and satellites [2, p. 222]. The use of geostationary Earth satellites in a satellite communication network for relaying signals eliminates the Doppler frequency shift on the transmission line from the radiation source from the Earth, thereby eliminating the influence of the Doppler effect in the ionosphere. The use of a geostationary satellite in the communication channel has the disadvantage of an unstable spatial position. Another option to eliminate the Doppler frequency shift in the Earth’s ionosphere is to use the moon. Located at a distance of 384,405 km from Earth, the Moon satellite without the ionosphere and this drawback does not occur when using the space communications station on the Moon.

Scientific studies of the Moon, Mars, and Venus led to the creation of a program for the practical use of Lunar Stations in astronautics (Fig. 14 shows the Lunar Station of the satellite communications network, data from the TV channel RUSSIA K), the introduction of which is scheduled for 2019.

The identified sources of information known from the prior art do not describe technical solutions that are completely identical to the method and device proposed by the authors, the distinguishing features are not known for the achieved technical result, which allows us to conclude that the group of inventions meets the patentability conditions “industrial applicability”, “inventive step” "And" novelty. "

Bibliography

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4. Trakhtman A.M. Introduction to the generalized spectral theory of signals. Moscow, "Soviet Radio", 1972.

5. A method of obtaining information about the quality of the signal in the receiver, RU 2113061, 10.06.1998.

6. A method of obtaining information about the quality of the signal in the receiver, RU 2216871, 20.11.2003.

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Claims (2)

1. The method of radio reception of high-speed information of a space radio line, which consists in the following:
a) reception of the emitted reference signal when the radio wave leaves the ionospheric formation (wave propagation time

), including:
entering into communication on a carrier frequency;
synchronization of the clock frequency of the reception of bits, synchronization of reception and reception of the control frequency of the reference signal;
translation of the control frequency into the frequency domain by the fast Fourier transform;
determine the total discrepancy in the phase spectrum

control frequency (from the ionosphere and the Doppler effect), write data to memory;
estimation of the discrepancy between the control frequency and the Doppler effect (prediction)

estimation of the discrepancy of the control frequency from the ionosphere

for

write data to memory;
b) reception of the emitted reference signal in the continuation of the radio line, in the region of slight fading from the Doppler effect (predicted wave propagation time

) containing:
entering into communication on a carrier frequency;
synchronization of the clock frequency of the reception of bits, synchronization of reception and reception of the control frequency of the reference signal;
translation of the control frequency into the frequency domain by the fast Fourier transform;
determination of the complete discrepancy in the phase spectrum

control frequency (from the ionosphere) and writing data to memory;
estimation of the initial distortions of the reference signal from the Doppler effect in the case of slight distortions by the Doppler effect at

after passing through the ionosphere,

writing data to memory;
c) reception in a working session, signal propagation time t _S , spectrum signals with two side reference signal

and high speed signal

including:
noiseless signal decoding

and transmitting the received information to the recipient of the information;
d) if the operation decoder c) showed the status information, then the following actions are performed:
receiving signals

and

in the spectrum of one side and recording signals

and

noiseless signal decoding

and transmitting the received information to the recipient of the information;
d) if the operation decoder d) showed the status information, then compensation for the stray offset of the phase signal level, including:
estimation of phase φ ₁ at the beginning of the measuring interval according to the data of the phase samples

estimation of phase φ _{2 at the} end of the measuring interval;
determine the compensated frequency

where n is the number of phase jumps by 2π;
an estimate of the errors in the system, which are divided into small (normal) and large n = 1, 2, ... (abnormal), and the error value is estimated only for normal errors;
mixing compensation is achieved by shifting the frequency band:

Where:

- samples of the signal from the memory block

- counts of parasitic displacement;
transform the samples of the phase signal into bits of information by the decisive rule (the conversion of m-level samples into a binary sequence of bit symbols of the clock frequency);
perform noise-resistant decoding of a high-speed signal and the received information is transmitted to the recipient of information;
e) if the operation decoder e) showed the status information, then compensation for the stray shift of the spectral components of the recorded high-speed signal

using reference signal data

in the working session and the initial values of the reference signal,
recorded in paragraph b), including:
frequency domain translation by fast Fourier transform of signals

and

determination of the complete discrepancy between the phases of the signal and the control frequency during the signal propagation t _S ;
grid definition

spurious discrepancies of the signal spectrum from the distortion data of the reference signal from the Doppler effect, from small initial distortions

after passing through the ionosphere t _S = t ₂ to t _S

for

compensation for spurious shifts of the spectral components of the signal

from the Doppler effect by changing the phases of the components of the frequency decomposition by the distortion compensation operator:

for

decisive transformation of phase signal samples into bits of information;
decisive transformation of phase signal samples into bits of information;
performing error-correcting decoding of a high-speed signal and transmitting the received information to the information recipient;
g) if the decoder of operation e) showed the status information, then compensation for the stray shift of the spectral components of the recorded high-speed signal

using reference signal data

in the working session and the initial values of the reference signal recorded in paragraph b), including:
determination of spurious discrepancies in the frequency grid

spectrum of the signal according to the distortion data of the reference signal from the Doppler effect, from large initial distortions

after passing through the ionosphere t _S = t ₂ to t _S

for

compensation for spurious shifts of the spectral components of the signal

Doppler effect by changing the phases of the components of the frequency decomposition by the distortion compensation operator:

for

decisive transformation of phase signal samples into bits of information;
noise-resistant decoding of a high-speed signal and transmission of the received information to the recipient of information. 2. A device for receiving high-speed information from a space radio link, comprising:
- band-pass filter (PF);
- An agreed filter for receiving a radio signal from one side (SFOB);
- the first and second respectively balanced modulator (BM);
- the first and second, respectively, demodulator (DM);
- recipient of information (PI);
- memory block of phase samples of the receiver signal (PAMP);
- a device for compensating for stray phase signal displacement (CC SPS);
- a coordinated filter for receiving a radio signal with two side (SPSB);
- a processor configured to form commands:
- recording samples of the phase signals of the message and the reference signal when receiving information, respectively, in the memory block of the samples of the signal of the recipient (PAM _p );
- inclusion of a block of the device for compensation of parasitic shift of spectral components (CC SSS);
- control the first and second, respectively, block decoder (DC);
- selection of electrical circuits for FPGA, RAM, ROM, microprocessors, made in the form of boot modules for spacecraft signals with an a priori known structure and the formation of a boot module for reception in a communication session;
displaying the state of receiving information of a space radio link at the consumer of information (PI),
moreover, the input of the band-pass filter (PF) is the input of the device, and the output is connected to the inputs of a matched filter for receiving a radio signal from one side (SFB) and a matched filter for receiving a radio signal with two side (SFB), the first balanced modulator (BM), the input of which is connected to the output a matched filter for receiving a radio signal from one side (SOPF), and the output is connected to the input of the first demodulator (DM) and the first input of the device for compensating for stray shift of spectral components (CC SSS), the second input of which is connected to the output to the memory block of the phase samples of the recipient signal (PAMp) and the second input of the device for compensating the stray phase shift signal (UK SPS), and the output of the device for compensating stray phase signal (UK SPS) is connected to the output of the device for compensating stray shift of spectral components (UK SSS) and the second input of the recipient of information (PI), and the first input of the device for compensating for stray phase signal offset (CC SPS) is connected to the output of the processor, the input-output of which is connected to the inputs and outputs of: devices and compensation for the stray shift of the spectral components (CCC CC), the memory block of the phase samples of the recipient signal (PAMP), the first decoder (DC) and the second decoder (DC), respectively, a block decoder (DC), information receiver (PI); the first and third inputs of the information recipient (PI) are respectively connected to the outputs of the first (DC) and second decoders (DC), and the input of the first decoder (DC) is connected to the output of the first demodulator (DM), and the input of the second decoder (DC) is connected to the output the second demodulator (DM), the input of the second demodulator (DM) is connected to the output of the second balanced modulator (BM), the input of which is connected to the output of the matched radio signal reception filter with two side ones (SFDB).

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