RU2402167C1 - Detection and demodulation of data for wireless communication systems - Google Patents

Abstract

FIELD: information technologies.

SUBSTANCE: methods are disclosed for detection and demodulation of data transfers in systems of wireless communication. In one aspect, detector with control by decision detects transfers of data in received signal, using received symbols of data and also received symbols of pilot-signal. Detector with control by decision may be intended to do differential detection in frequency domain or coherent detection in time domain, and may be used in modulation with multiple carriers (for instance, OFDM). In another aspect, adaptive threshold is used to detect received data transfers. Threshold may be detected for each hypothetically received transfer of data. Threshold may be calculated, for instance on the basis of signal and noise energy of hypothetical data transfer.

EFFECT: increased efficiency of detection.

37 cl, 17 dwg, 1 tbl

Description

FIELD OF THE INVENTION

The present invention generally relates to data transmission, and more particularly to methods for detecting and demodulating data transmissions in wireless communication systems.

State of the art

In a wireless communication system, the transmitted data is usually processed (e.g., encoded and modulated), and then upconverted to a radio frequency (RF) signal of the carrier to generate an RF modulated signal that is more suitable for transmission over a wireless channel. The RF modulated signal is then transmitted from the transmitter and can arrive at the receiver through multiple propagation paths in the wireless channel. The characteristics of propagation paths usually change over time due to many factors, such as, for example, attenuation, multipath propagation and external interference. Consequently, the RF modulated signal may be influenced by various channel conditions (conditions) (for example, various attenuation effects and multipath effects) and may be associated with various composite benefits through the operating bandwidth of the system.

To achieve high efficiency, a pilot signal (i.e., a reference signal) is often transmitted by a transmitter to assist the receiver in performing a variety of functions. A pilot signal is typically generated based on known symbols and processed in a known manner. The pilot signal can be used by the receiver to estimate the transmission channel, capture synchronization and frequency, coherent demodulation, and so on.

It is often desirable or necessary to detect the presence of data transmissions in a received signal. Data transmission detection is typically achieved by processing the pilot for each hypothetical received data transmission. If the pilot energy is greater than a certain threshold, then the hypothetical data transmission is further processed (for example, demodulated and decoded). In this case, in order to determine whether the data transmission was decoded correctly or with errors, they usually rely on an error-detecting code such as a cyclic redundancy code (CRC).

In some wireless communication systems, detection based on pilot alone is not enough. This may occur, for example, when operating with a low received signal-to-noise ratio (SNR). In addition, an error detection code may not be available when used to confirm the correctness of the received data transmission.

Therefore, in the prior art for such wireless communication systems, there is a need for methods for detecting and demodulating data transmissions.

SUMMARY OF THE INVENTION

Methods for detecting and demodulating data transmissions in wireless communication systems are proposed. In one aspect, a decision-driven detector is provided for detecting data transmissions in a received signal. This detector uses the received data symbols as well as the received pilot symbols to perform detection, and is thus capable of providing improved detection performance. A detector with control by decision may be designed to operate in the frequency domain or in the time domain. For systems using multi-carrier modulation (e.g., orthogonal frequency division multiplexing (OFDM) - OFDM), the detector may be designed to perform differential detection in the frequency domain or coherent detection in the time domain, both of which are described in detail below.

In another aspect, an adaptive threshold is used to perform detection of received data transmissions. A threshold may be determined for each hypothetical received data transmission. The threshold can be calculated, for example, based on the total energy of the received signal (i.e., signal plus noise plus interference) of a hypothetical data transmission. Using an adaptive threshold can provide robust detection parameters in many operating environments, such as an unlicensed frequency range in which various sources of interference can be present.

Various aspects and embodiments of the invention are described in detail below. For example, receiver structures for various transmission schemes are also described herein.

Brief Description of the Drawings

The signs, nature, and advantages of the present invention will become more apparent from the detailed description formulated below, considered in conjunction with the drawings, in which the same reference numbers respectively throughout the text coincide and in which:

figure 1 depicts a wireless communication system;

figa and 2B depict the usual Protocol data modules (TIRs, PDU) for channels 1 and 2, respectively;

figa depicts a block diagram of a transmitter unit;

3B illustrates an OFDM symbol;

4 is a block diagram of a receiver unit;

5 depicts a correlation detector;

6 depicts an embodiment of a correlation detector;

7 depicts a data detector / demodulator, which includes a data demodulator and a decision-controlled detector;

figa depicts an embodiment of a data demodulator;

figv depicts a detector with decision control, which performs differential detection in the frequency domain;

figs depicts a decision-controlled detector that performs coherent detection in the time domain;

Fig.9 depicts a block diagram of an access point and user terminal;

figa and 10B depict the usual transmission channel 1 and 2, respectively; and

11A and 11B depict receiver processing for channels 1 and 2, respectively.

Detailed description

The word "ordinary" is used here to mean "serving as an example, model, or illustration." Any embodiment described herein as “normal” need not be construed as preferred or advantageous in relation to other embodiments.

FIG. 1 depicts a wireless communication system 100 that includes a plurality of access points (APs) 110 that communicate with a plurality of user terminals (PTs, UTs) 120. (For simplicity, only one access point is shown in FIG. 1. ) An access point may also be called a base station or some other terminology. Each user terminal may be a fixed or mobile terminal and may also be called an access terminal, a mobile station, a remote station, user equipment (software, UE), a wireless device, or some other terminology. Each user terminal can at any given moment exchange information with one or, possibly, a plurality of access points on the downlink and / or uplink. A downlink (i.e., a forward link) refers to transmission from an access point to a user terminal, and an uplink (i.e., a reverse link) refers to transmission from a user terminal to an access point.

The methods described herein for detecting and demodulating data transmission can be used in various wireless communication systems. For example, these methods can be used in systems that use (1) one or many antennas for data transmission, and one or many antennas for receiving data, (2) various modulation methods (for example, code division multiple access) ( CDMA, CDMA, OFDM, and so on), and (3) one or a plurality of frequency bands for the downlink and uplink.

For clarity, methods for a conventional wireless communication system are described in detail below. In this system, the receiver is equipped with multiple (T) antennas for receiving data, and the transmitter may be equipped with one or multiple antennas. The system further utilizes orthogonal frequency division multiplexing (OFDM), which effectively partitions the overall system bandwidth into multiple (N) orthogonal subbands. In OFDM, the data or pilot that will be transmitted in each subband is first modulated (i.e., the symbol is converted) using a specific modulation scheme. Zero-valued signals provide subbands not used for data / pilot transmission. For each OFDM symbol period, modulation symbols and zero signal values for all N subbands are converted to the time domain using the inverse fast Fourier transform (IFFT) to obtain a transformed symbol that contains N samples of the time domain. To combat intersymbol interference (ISI), part of each transformed symbol is often repeated to form the corresponding OFDM symbol, which is then transmitted wirelessly. The period of the OFDM symbol (or simply the symbol period) corresponds to the duration of one OFDM symbol, which is the smallest transmission element for the system. In one specific project, the system bandwidth is 20 MHz, N = 64, the subbands are assigned indices from -32 to +31, the duration of each converted symbol is 3.2 picoseconds (ps), the cyclic prefix is 800 nanoseconds (ns), and the duration each OFDM symbol is 4.0 microseconds (μs).

For clarity, two specific transmission schemes and two receiver structures are described below. The first transmission scheme is used for transport channel 1 (or simply channel 1 or KAN1, CH1) and has the following characteristics: (1) transmissions on channel 1 do not contain time compensation in the transmitter and reach arbitrary durations in the receiver, and (2) each transmission Channel 1 includes multiple OFDM symbols for data and pilot. The second transmission scheme is used for transport channel 2 (or simply channel 2 or KAN2, CH2) and has the following characteristics: (1) transmissions on channel 2 contain time compensation in the transmitter and achieve time synchronization within the time interval in the receiver, and (2 ), each transmission on channel 2 includes a single OFDM symbol for both data and pilot. Slow and fast random access channels with characteristics similar to those of channels 1 and 2 are described in US patent application No. 60/432440.

2A depicts a conventional Protocol Data Module (TIR) 210 that can be used for channel 1 (TIR CAN1). TIR KAN1 210 contains a reference part 220, which is divided by time using a time division signal (TDM) with a part of the message KAN1 230. The reference part 220 includes P OFDM pilot symbols 222, where P can be any integer: unit or more. The OFDM pilot symbols are used to facilitate the capture and detection of transmission of CAN1, as well as to assist in coherent demodulation of a portion of the CAN1 message. Part of the message KAN1 230 includes D characters 232 data OFDM, where D can be any integer: unit or more. OFDM pilot symbols and OFDM data symbols may be generated as described below.

2B shows a conventional TIR 250 that can be used for channel 2 (TIR CAN2). TIR KAN2 250 comprises a support portion 260, which is a subband multiplexed with KAN2 message portion 270. The reference portion 260 comprises a pilot symbol set, which is transmitted in one set of subbands (shown as shaded subbands in FIG. 2B). Part 270 of the CAN2 message contains a group of data symbols that is transmitted in another set of subbands. Data symbols are generated by encoding, interleaving, and transforming the symbols of the 280 CAN2 message. The frequency-domain-multiplexed pilot symbols and data symbols are processed to generate the TIR CAN2 250 time domain, as described below.

In the embodiment shown in FIG. 2B, the pilot subbands and data subbands are interlaced such that each data subband is framed on both sides by pilot subbands. Pilot symbols transmitted in pilot subbands may be used to estimate channel responses for data subbands and for coherent demodulation. Also, without departing from the scope of the invention, other subband multiplexing schemes may be implemented. For example, each group of Q data subbands may be framed on both sides with pilot subbands, where Q may be any positive integer.

FIG. 3A depicts a block diagram of an embodiment of a transmitter unit 300 that can perform transmission processing for channels 1 and 2 described above. A transmitter unit 300 that can be implemented within an access point or user terminal includes a transmit processor 310 ( TX) data, an additional TX spatial processor 330, and one OFDM modulator 340 for each transmit antenna.

Within the TX data processor 310, a CRC generator 312 receives data for the message CAN1 or CAN2 and (optionally) generates a CRC value for the message. Encoder 314 then encodes the message data and the CRC value (if any) in accordance with a specific coding scheme, providing code bits. Then, interleaver 316 interleaves (i.e., reorders) the code bits based on a specific interleaving scheme to provide frequency and possibly time diversity. Then, the symbol conversion unit 318 converts the interleaved data in accordance with a specific modulation scheme, providing modulation symbols, which are also called data symbols and are denoted as s (k) .

A multiplexer (MUX) 320 receives and multiplexes data symbols with pilot symbols in the manner specified for the processed messages KAN1 or KAN2. For the embodiment shown in FIG. 2A, the TIR CAN1 contains P OFDM pilot symbols followed by D OFDM data symbols. For reporting CAN1, the multiplexer 320 provides a set of pilot symbols { p ₁ (k) } for each of the P OFDM pilot symbols, and then data symbols for each of the D OFDM data symbols. For the embodiment shown in FIG. 2B, TIR CAN2 contains L + 1 pilot symbols alternating with L data symbols. For CAN2, the multiplexer 320 provides a set of L + 1 pilot symbols { p ₂ ( k )} multiplexed with a group of L data symbols. In either case, multiplexer 320 provides a stream of multiplexed pilot symbols and data symbols.

The table depicts a specific embodiment of two sets of pilot symbols, { p ₁ ( k )} and { p ₂ ( k )}, for the supporting parts KAN1 and KAN2. In this embodiment, only 52 of the 64 full subbands are used for data and pilot transmissions, and the other 12 subbands (with zero entries in the table) are not used. In an embodiment, pilot symbols are quadrature phase shift keying (QPSK) symbols. 52 pilot symbols for the reference part of CAN1 are selected so that the signal generated based on these pilot symbols has a minimum spread of peak to average values. This feature allows the OFDM pilot symbol to be transmitted at a higher power level, which can provide improved performance.

Table
Pilot symbols for CAN1 and CAN2 Sub-range Index Symbol
pilot signal CAN1
p _one ( k ) Pilot symbol
KAN2
p ₂ ( k ) Subdiap Zone Index Symbol
pilot-
signal CAN1
p _one ( k ) Symbol
pilot signal
KAN2
p ₂ ( k ) Sub-range Index Symbol
pilot signal
CAN1
p _one ( k ) Symbol
pilot signal
KAN2
p ₂ ( k ) Subdiap Zone Index Symbol
pilot-
signal CAN1
p _one ( k ) Symbol
pilot signal CAN2
p ₂ ( k ) -32 0 0 -16 -1 + j data 0 0 0 16 -1 + j data -31 0 0 -fifteen 1-j 1 + j one 1-j -1-j 17 -1 + j 1-j -thirty 0 0 -fourteen 1 + j data 2 -1-j data eighteen 1-j data -29 0 0 -13 1-j 1 + j 3 -1-j -1-j 19 1 + j -1-j -28 0 0 -12 1-j data four -1-j data twenty -1 + j data -27 0 0 -eleven -1-j 1 + j 5 -1 + j 1 + j 21 1 + j -1-j -26 -1-j -1 + j -10 -1-j data 6 1 + j data 22 -1 + j data -25 -1 + j -1 + j -9 1-j 1-j 7 -1-j -1-j 23 1 + j -1-j -24 -1 + j data -8 -1-j data 8 -1 + j data 24 -1 + j data -23 -1 + j -1-j -7 1 + j -1 + j 9 -1-j 1-j 25 1-j -1 + j -22 1-j data -6 -1 + j data 10 -1-j data 26 -1-j 1-j -21 1-j -1-j -5 -1-j -1-j eleven 1 + j 1 + j 27 0 0 -twenty 1 + j data -four -1 + j data 12 1-j data 28 0 0 -19 -1-j -1-j -3 -1 + j -1 + j 13 -1 + j 1-j 29th 0 0 -eighteen -1 + j data -2 1-j data fourteen -1-j data thirty 0 0 -17 1 + j 1 + j -one -1 + j -1 + j fifteen 1 + j -1 + j 31 0 0

If multiple antennas are available, an additional TX spatial processor 330 may be used to perform spatial processing on the multiplexed pilot symbols and data symbols. For example, a TX spatial processor 330 may perform spatial processing for: (1) controlling or generating a position of a main lobe of an antenna pattern for transmitting symbols on a separate spatial channel of a MIMO channel, (2) transmitting diversity for transmitting symbols over multiple antennas and subbands so that to achieve diversity, or (3) spatial multiplexing for transmitting symbols over multiple spatial channels. Spatial processing for all of these transmission modes is described in detail in provisional application US No. 60 / 421,309.

TX spatial processor 330 provides one transmit symbol stream for each antenna. The transmitted symbols are simply multiplexed data symbols and pilot symbols if spatial processing is not performed. Each transmitted symbol stream is provided with a corresponding OFDM modulator 340. Within each OFDM modulator 340, an inverse Fast Fourier Transform (FFT) block 342 converts each sequence of N transmitted symbols into a transformed time-domain symbol consisting of N time-domain samples, where N represents the total number of subbands. For each transformed symbol, the cyclic prefix generator 344 repeats a portion of the transformed symbol in order to generate a corresponding OFDM symbol consisting of M samples. The cyclic prefix generator 344 provides an OFDM symbol stream with a transmitter 346 (TMTR), which converts the OFDM symbol stream into one or more analog signals and further amplifies, filters, and upconverts the analog signal (s) in order to generate an RF modulated signal, which is then transmitted from the associated antenna 350.

3B illustrates an OFDM symbol, which is composed of two parts: a cyclic prefix and a transformed symbol. In an embodiment N = 64, the cyclic prefix contains 16 samples and each OFDM symbol contains M = 80 samples. The cyclic prefix is a copy of the last 16 samples (i.e., a cyclic continuation) of the transformed character and is inserted before the transformed character. The cyclic prefix ensures that the OFDM symbol retains its orthogonality properties in the presence of delay due to multipath propagation.

Fig. 10A depicts a conventional transmission on channel 1. The time schedule for channel 1 is divided into intervals KAN1, and each interval KAN1 has a certain duration (for example, P + D periods of the OFDM symbol). In an embodiment, one TIR of CAN1 may be transmitted for each slot of CAN1.

The time and frequency characteristics of user terminals A and B are fixed with respect to similar characteristics of the system. This can be achieved by receiving a transmission (for example, the interrogator pilot), which is carrier or embedded for synchronization information. The user terminals then establish their synchronization based on the received synchronization information. However, the synchronization of each user terminal may be impaired (or delayed) with respect to system synchronization, the magnitude of the violations usually corresponding to the propagation delay for transmission containing synchronization information. If both user terminals and the system receive their synchronizing information from a common time source (for example, the Global Positioning System - GPS), then there may not be any synchronization disturbances between these objects.

10A, user terminals A and B (e.g., randomly) select two different KAN1 intervals (e.g., intervals 3 and 1, respectively) in order to transmit their TIR KAN1. Since user terminals A and B are associated with various timing outages and different propagation delays, their TIR CAN1 reach the access point with different delays (called two-way delays (forward and reverse delays on the signal, or RTD) with respect to the boundaries of the intervals of the access point CAN1.

Fig. 10B depicts a normal transmission on channel 2. The time schedule for channel 2 is divided into KAN2 intervals, with each KAN2 interval having a specific duration (for example, one period of an OFDM symbol). In an embodiment, one TIR of CAN1 may be transmitted for each slot of CAN1. One TIR CAN2 can be transmitted for each CAN2 interval.

10B, the temporal characteristics of user terminals A and B are fixed with respect to similar characteristics of the system and these terminals additionally have information about their RTD, which can be determined by the access point (for example, when accessing the system) and transmitted back to the user terminals. The user terminals can then adjust their transmitted synchronization to calculate their RTD so that their TIR KAN2 reach a time match at the access point with the selected boundaries of the KAN2 interval.

In FIG. 10B, user terminals A and B (eg, randomly) select intervals 3 and 1 of CAN2, respectively, in order to transmit their TIRs to CAN2. Since the transmissions of the user terminals A and B are time compensated, the TIRs of the CAN2 reach the access point approximately matched to the boundaries of the selected intervals of the CAN2, as shown in FIG. 10B.

Figure 4 depicts a block diagram of an embodiment of a receiver unit 400 that can perform the above processing of received data for channels 1 and 2. Receiver unit 400, which can also be implemented within an access point or user terminal, includes one receiver ( RCVR) 410 for each of the T receive antennas 408, a data detector / demodulator 420, and a received (RX) data processor 450.

Each antenna 408 receives RF modulated signals transmitted by the transmitter unit and supplies the corresponding receiver 410 with the received signals. Each receiver 410 adapts (eg, amplifies, filters, and downconverts) its received signal and digitizes the adapted signal to provide samples denoted as x _i ( n ).

The data detector / demodulator 420 includes a data demodulator 430 and a detector 440 that receive and process samples from all receivers 410 in order to detect and demodulate data transmissions on channels 1 and 2. Processing by block 420 is described in more detail below. Block 420 provides recovery of data symbols denoted as ŝ (k) , which are estimates of the transmitted data symbols s (k) . Within the RX data processor 450, the reconstructed data symbols are back-converted by the symbol-back-de-energized unit 452, de-interleaved by the de-interleaved unit 454, and decoded by a decoder 456 to provide decoded data for messages CAN1 and CAN2. If the recovered message includes a CRC value, then the CRC check unit 458 checks the message with a CRC value to determine if it was decoded correctly or in error.

11A depicts received data processing for channel 1 that does not contain time compensation. Turning back to Fig. 10A, we see that although the transmitter units try to transmit at certain intervals of CAN1, the transmission of CAN1 does not contain time compensation, and the resulting behavior of channel 1 is similar to the behavior of a channel that is not divided into time intervals. In this case, referring back to FIG. 11A, we will see that the receiver unit can use a tracking correlation detector to detect transmissions for CAN1, each of which can be received starting from any sampling period.

A correlation detector, which can function in the time domain, monitors the entire time range in which the TIR KAN1 can be received, one sampling period at a time. The detection window indicates the period of time in which samples for one TIR KAN1 should be processed by the detector. This detection window can be initialized to the beginning of the first CAN1 interval and then slide forward one sampling period at a time. For each sampling period that corresponds to the hypothesis, the correlation detector processes the samples within the detection window, determining the indicator for the hypothetical received TIR KAN1, starting from this sampling period. If the indicator exceeds the threshold KAN1, then the TIR KAN1 is additionally decoded to restore the message KAN1. The indicator may relate to signal energy or some other parameters. The threshold KAN1 can be fixed or adaptive (for example, dynamically determined based on samples within the detection window).

(n) линию задержки/буфер 530. Для МДП КАН1, показанного на Фиг.2A, накапливающий сумматор 520 символов выполняет накопление P символов пилот-сигнала ОЧРК, причем накопление выполняется на выборочной основе, обеспечивая накопление символов пилот-сигнала ОЧРК, имеющих М выборок. Линия задержки/буфер 530 обеспечивает хранение для N из М выборок и фактически отбрасывает M-N выборок для циклического префикса. Эти N выборок для преобразованного символа являются соответствующими накопленному символу пилот-сигнала ОЧРК.FIG. 5 is a block diagram of a correlation detector 440a, which is one embodiment of the detector 440 of FIG. 4. Samples x _i ( n ) for each of the T receiving antennas are provided with a corresponding antenna processor 510. Within each processor 510, the accumulating adder 520 symbols receives and accumulates samples for the current hypothesis and supplies accumulated samples

( n ) a delay line / buffer 530. For the TIR CAN1 shown in FIG. 2A, the accumulating symbol adder 520 performs the accumulation of P OFDM pilot symbols, the accumulation being performed on a sample basis, providing the accumulation of OFDM pilot symbols having M samples . The delay line / buffer 530 provides storage for N of the M samples and effectively discards the MN samples for the cyclic prefix. These N samples for the transformed symbol correspond to the accumulated OFDM pilot symbol.

The signal detector 540 then determines a metric for the accumulated OFDM pilot symbol. In an embodiment and as described below, the metric refers to the energy of the signal N samples for the accumulated OFDM pilot symbol. However, without departing from the scope of the invention, another indicator may also be used. The adaptive threshold calculation unit 550 determines an adaptive threshold value Y _i ( n ) for use in determining whether transmission of CAN1 has actually been received. An adder 560 summarizes threshold values for all T antennas to provide an aggregate threshold value Y _tot ( n ) that is further scaled with a scale factor S ₁ by a multiplier 562 to obtain a final threshold value Y ( n ). Adder 564 summarizes the metrics for all T antennas, providing the final metric E ( n ), which is then compared to the final threshold Y ( n ) using comparator 570. The detector output will indicate that if E ( n )> Y ( n ), then TIR KAN1 was accepted, otherwise TAN KAN1 was not accepted.

(n) выборок для накопленных символов пилот-сигнала ОЧРК. Выборки

(n) могут быть выражены следующим образом: FIG. 6 is a block diagram of a correlation detector 440b, which is one embodiment of the detector 440a of FIG. 5. Samples x _i ( n ) for each receiving antenna are provided with a storage symbol adder 520, which is implemented based on P-1 delay elements 522 and P-1 adders 524. Each delay element 522 provides one OFDM delay symbol (i.e., M samples) of delay. P-1 adders 524 accumulate P OFDM pilot symbols based on samples, and the last adder provides

( n ) samples for accumulated OFDM pilot symbols. Selections

( n ) can be expressed as follows:

(n)=

, где i∈{1 ... T}(1)

( n ) =

, where i ∈ {1 ... T} (1)

(n) снабжается линия/буфер 530 задержки, которые реализуются на базе N-1 элементов 532 задержки, каждый из которых обеспечивает один период выборки задержки.Selections

( n ) a delay line / buffer 530 is provided that is implemented based on the N-1 delay elements 532, each of which provides one delay sample period.

(j), где j∈ {0 ... N-1}. Для получения {

(j)} набор символов пилот-сигнала {p ₁(k)} для поддиапазонов пилот-сигнала и нулевые значения сигналов для неиспользованных поддиапазонов (например, как показано в таблице), преобразуются во временную область, используя N-точечный ОБПФ, получая N выборок пилот-сигнала,

(0) через

(N-1), которые после этого сопрягаются и поставляются N умножителям 542. Каждый умножитель 542 перемножает эту выборку

(n-j) с сопряженной ей выборкой пилот-сигнала

(j) и поставляет результат на сумматор 544. Сумматор 544 суммирует результаты от всех N умножителей 542 и предоставляет суммированный результат блоку 546. Блок 546 определяет квадрат величины суммированного результата, который предоставляется в качестве значения показателя E _i(n). Значение показателя для каждой антенны может быть выражено следующим образом:The signal detector 540 correlates the accumulated OFDM pilot symbol with a known OFDM pilot symbol and determines a value of an indicator E _i ( n ) for the accumulated OFDM pilot symbol. Each of the N samples for the accumulated OFDM pilot symbol is provided with a corresponding multiplier 542, which also receives a corresponding conjugate pilot sample

( j ), where j ∈ {0 ... N-1}. For getting {

( j )} pilot symbol set { p ₁ ( k )} for pilot subbands and zero signal values for unused subbands (for example, as shown in the table) are converted to the time domain using an N-point IFFT, obtaining N pilot samples

(0) through

(N-1), which are then mated and delivered to N multipliers 542. Each multiplier 542 multiplies this sample

( nj ) with its associated pilot sample

( j ) and delivers the result to adder 544. Adder 544 sums the results from all N multipliers 542 and provides the summed result to block 546. Block 546 determines the square of the summed result, which is provided as the value of E _i ( n ). The value of the indicator for each antenna can be expressed as follows:

Adder 564 receives and summarizes metric values for all T antennas to provide a final metric value E ( n ), which can be expressed as follows:

(3) E ( n ) =

(3)

The threshold calculation unit 550 determines an adaptive threshold for use in detecting MIS KAN1 for the current hypothesis. Each of the N samples for the accumulated OFDM pilot symbol is provided to a corresponding block 552, which determines the square of the sample size. After that, the adder 554 sums the squares of the values from all N blocks 552 to provide a threshold value Y _i ( n ). An adder 560 receives and sums the threshold values for all T antennas to provide an aggregate threshold value Y _tot ( n ), which can be expressed as follows:

(4) Y _tot ( n ) =

(four)

After that, the multiplier 562 scales the total threshold value with a scale factor S ₁ , providing a final threshold value, which can be set in the form Y ( n ) = S ₁ · Y _tot ( n ).

The comparator 570 compares the final value of the indicator E ( n ) with the final threshold value Y ( n ) and provides the output signal of the detector D ( n ), which can be expressed as follows:

(5) D ( n ) =

(5)

If TIR KAN1 is detected, then the synchronization of the OFDM symbol is established at the moment of detecting TIR KAN1 (that is, to the specific value n at which TIR KAN1 was detected).

The scale factor S ₁ is a positive constant selected to ensure: (1) a certain probability of missed detection, which is the probability of not detecting the TIR CAN1 that was transmitted, and (2) a certain probability of false alarms, which is the probability of a false indication that TIR KAN1 was accepted despite the fact that it was not transmitted. It is desirable that the probability of missed detection be less than the error rate of the message (MER) so that the MER is due to the received signal-to-noise ratio (SNR) and other parameters, and not the detector. MER can be defined for channel 1, for example, by 1 percent or less. The output of the detector can be used to determine whether the received TIR KAN1 was actually processed in order to recover the transmitted message KAN1. Determining whether the message KAN1 is decoded correctly or with an error can be based on the CRC value contained in the message.

For a given MIS KAN1, this may become possible for a correlation detector declaring multiple detections. This is because the detection can be declared noise in one or more OFDM symbols and in the signal in other OFDM symbols for the detected MIS KAN1. For example, with P = 2, the first detection may occur with noise in the first OFDM symbol and with a signal in the second OFDM symbol, and the second detection with a large total value of the indicator will occur when the second OFDM signal symbol later reaches one period of the OFDM symbol. Thus, at P> 1, the detector can be controlled by continuing to detect for TIR KAN1 additional P-1 periods of OFDM symbols, finding for TIR the largest total value indicator. After that, the synchronization of the OFDM symbol is established by the detection of the highest total value indicator, and RTDs are also calculated based on the time associated with this detection.

The detection processing can be performed independently of the message processing, that is, the detection processing can continue normally, regardless of whether the TIRs of CAN1 are actually detected. Thus, if TIR CAS1 is initially detected in nj sample periods with the total value of the indicator E ( nj ), and later another TIR CAN1 in n sample periods with the final value of the indicator E ( n ) is found, where E ( n )> E ( nj ) and j is smaller than the size of the detection window, then the current processing of the message for TIR CAN1 detected in nj sample periods can be stopped, and TIR CAN1 detected in n sampling periods can be processed instead.

11B depicts received data processing for channel 2, which comprise time compensation. Turning back to FIG. 10B, we see that the transmitter blocks transmit at certain intervals of CAN2 and the transmission of CAN2 contain time compensation, reaching the receiver unit at selected boundaries of the CAN2 interval. In this case, referring back to Fig. 11B, we see that the receiver unit can detect the transmission of CAN2 in each interval of CAN2 (instead of each sampling period), and the detection window can move from interval to interval. For each KAN2 interval that corresponds to the hypothesis, a decision-controlled detector processes the samples taken within the detection window, determining the indicator of the hypothetical MIS KAN2 adopted in this interval. If the indicator exceeds the threshold KAN2, then TIR KAN2 is considered adopted.

FIG. 7 is a block diagram of an embodiment of a demodulator / data detector 420c, which may also be used for block 420 of FIG. 4. The data demodulator / detector 420c includes a data demodulator 430c used to perform coherent demodulation, and a decision-controlled detector 440c used to detect MIS CAN2. Samples for each of the T receiving antennas are provided to a corresponding antenna demodulator 710 located within the data demodulator 430c and a corresponding decision-controlled detector 750 located within the detector 440c.

Each antenna demodulator 710 simultaneously performs coherent demodulation of one received OFDM symbol for one antenna. For each received OFDM symbol, the fast Fourier transform (FFT) block 712 receives samples x _i ( n ) for the OFDM symbol, removes the cyclic prefix to obtain the transformed symbol, and performs fast Fourier transform (FFT) of the transformed symbol, providing N received symbols r _i ( k ), which include the received data symbols r _{i, d} ( k ) and the received pilot symbols r _{i, p} ( k ). After that, the channel estimator 720 estimates the response of the data subband channel based on the received pilot symbols r _{i, p} ( k ). A demodulator 730 performs coherent demodulation of the received data symbols r _{i, d} ( k ) with channel estimates to provide recovered data symbols ŝ _i ( k ).

The accumulating symbol adder 740 receives and accumulates the reconstructed data symbols from demodulators 710a through 710t for T receiving antennas and provides the recovered symbols ŝ ( k ). Thereafter, the RX data processor 450 processes the reconstructed symbols ŝ ( k ) as described above with respect to FIG. 4, providing decoded data. In an embodiment, the CAN2 message does not include a CRC, and the CRC check is not performed by the data processor RX. TX data processor 310 then processes the decoded data to provide remodulated symbols c ( k ), which are estimates of the transmitted data symbols s ( k ). Processing by the processor 310 includes encoding, interleaving, and symbol conversion, as described above with respect to FIG. 3A. Processing by RX processor 450 of data is often referred to simply as “decoding,” and processing by TX data processor 310 is often referred to as “re-encoding.”

Each decision-controlled detector 750 simultaneously performs detection for one received OFDM symbol. For each received OFDM symbol, the FFT block 752 receives samples x _i ( n ) for the OFDM symbols and performs FFT on the corresponding transformed symbol to provide N received symbols r _i ( k ). FFT blocks 712 and 752 are typically implemented by a single FFT block, but are depicted as two blocks in FIG. 7 for clarity.

The signal detector 760 then processes the received pilot symbols and data symbols with their expected symbols to provide an indicator E _i ' ( n ) for the OFDM symbols being processed. The adaptive threshold calculation unit 770 determines the value of the adaptive threshold Y _i ' ( n ) used to determine whether the TIR CAN2 has actually been received. An adder 780 summarizes the threshold values for all T antennas, providing an aggregate threshold value Y _tot ' ( n ), which is further scaled with a scale factor S ₂ by a multiplier 782 in order to obtain a final threshold value Y' ( n ). Adder 784 summarizes the metrics for all T antennas, providing a final metric E ' ( n ), which is then compared with the final threshold Y' ( n ) using comparator 790. The detector output will indicate that if E ' ( n )> Y ' ( n ), then TIR CAN2 was accepted, and otherwise CAN2 TIR was not accepted.

Fig. 8A depicts a block diagram of a data demodulator 430d, which is one embodiment of the data demodulator 430c shown in Fig. 7. Samples x _i ( n ) for each receive antenna are converted by FFT unit 712 to provide N received symbols r _i ( k ) for each transformed symbol. For the embodiment shown in the table, N received symbols include 28 received pilot symbols for 28 pilot subbands, 24 received data symbols for 24 data subbands, and 12 additional symbols for 12 unused subbands. For simplicity, the following description is for the embodiment shown in FIG. 2B, wherein N received symbols include L + 1 received pilot symbols for L + 1 pilot subbands and L received data symbols for L subbands data, with each data subband framed on both sides by pilot subbands, and the subband index k for the pilot subbands and data subbands is defined as k ∈ K, where K = {1 ... 49}.

Coherent demodulation of each of the L data subbands is performed by first generating a channel response estimate for the data subbands using two pilot subbands framing the data subband. The channel estimate ĥ _i ( k ) for the k-th data subband can be adopted by adding the channel estimates for the two framing pilot subbands, which can be expressed as follows:

ĥ _i ( k ) = ĥ _i ( k- 1) + ĥ _i ( k +1) = p ₂ ^* ( k -1) r _i ( k -1) + p ₂ ^* ( k +1) r _i ( k +1)

where k ∈ K _d and i ∈ {1 ... T}, (6)

where p ₂ ( k ) is the pilot symbol transmitted in the k-th subband for channel 2 and K _d represents a set of data subbands, that is, K _d ∈ {2, 4, ... 2L}.

The reconstructed data symbols ŝ _i ( k ) for each data subband in this case can be expressed as follows:

ŝ _i ( k ) = ĥ _i ^* ( k ) · r _i ( k ), where k ∈ K _d and i ∈ {1 ... T} (7)

The reconstructed data symbols for all T receiving antennas for each data subband can then be found in the form:

где k∈K _d (8) ŝ ( k ) =

where k ∈ K _d (8)

In Fig. 8A, the channel estimation shown in equation (6) is performed by L + 1 by a multiplier 722 and L by adders 724. Each multiplier 722 multiplies a received symbol for a corresponding pilot subband with a conjugate known pilot symbol for that subband to provide channel estimation for the pilot subband. In such a case, each adder 724 summarizes the channel estimates for the two pilot subbands framing the corresponding data subband to provide a channel estimate for that data subband. Channel estimates for L data subbands can also be obtained based on interpolation or some other methods without departing from the scope of the invention.

для этого поддиапазона, чтобы обеспечить восстановленный символ данных ŝ _i(k) для поддиапазона данных. Типовое накопление для всех T приемных антенн, как показано в уравнении (8), выполняется L сумматорами 742. Каждый сумматор 742 принимает и суммирует T восстановленных символов данных ŝ(k) для T приемных антенн для связанного поддиапазона данных, чтобы обеспечить восстановленный символ ŝ(k) для этого поддиапазона.The coherent demodulation shown in equation (7) is performed by L multipliers 732. Each multiplier 732 multiplies the received symbol r _i ( k ) for the corresponding data subband with an associated channel estimate

for this subband to provide a reconstructed data symbol ŝ _i ( k ) for the data subband. Typical accumulation for all T receive antennas, as shown in equation (8), is performed by L adders 742. Each adder 742 receives and sums T reconstructed data symbols ŝ ( k ) for T receive antennas for an associated data subband to provide a reconstructed symbol ŝ ( k ) for this subband.

As noted above, subband multiplexing may be such that each group of Q data subbands is framed on both sides by pilot subbands, where Q may be greater than one. If Q> 1, then coherent demodulation can be performed in several ways. In one embodiment, a received pilot symbol for each pilot subband is used as a coherent reference for two adjacent data subbands, and received data symbols for these data subbands can be coherently demodulated based on this received pilot symbol. In such a case, solid decisions can be made and used to remove modulation from the newly discovered data symbols in order to obtain improved channel estimates for the next two data subbands. The demodulation process may begin with the final data subbands (i.e., adjacent to the pilot subbands) and be performed in the direction of the middle data subband. Improved channel estimates for data subbands further farther from the pilot subbands can be obtained since each pair of received data symbols is detected. In another embodiment, the received pilot symbols for each pair of pilot subbands are interpolated in order to obtain a channel estimate for each of the Q data subbands framing these pilot subbands.

A CRC value is often used to determine if a received message has been decoded correctly or with errors. In certain cases, it may be necessary to include the CRC value in the message due to an overhead associated with the CRC value and / or for some other reasons. In this case, another mechanism is needed to determine whether the received message is valid. For the embodiment shown in FIG. 7, the data demodulator 430c and the data processor 450 RX can be controlled to provide a decoded message for each hypothesis, and the detector 440c can be controlled to provide an indication of whether the message was actually received for hypothesis.

FIG. 8B is a block diagram of a decision-controlled detector 440d that performs differential detection in the frequency domain and is one embodiment of the detector 440c of FIG. 7. Samples x _i ( n ) for each receive antenna are converted by FFT block 752 in order to provide N received symbols r _i ( k ) for each transformed symbol.

To determine the value of the indicator E ' ( n ) for each transformed symbol, first, for each receiving antenna, the detection statistics g _i ( n ) is obtained by summing the scalar products generated by the adjacent part of the 2L pilot subbands and data subbands from the real part 2L. The detection statistics g _i ( n ) can be expressed as follows:

z(k)·z*(k+1), где i∈ {1… T} (9a) g _i ( n ) =

z ( k ) · z * ( k +1), where i ∈ {1 ... T} (9a)

Where

z _i ( k ) =

The value index E ' ( n ) for the transformed symbol in this case can be expressed as follows:

, где i∈ {1 … T} (10a) E ' ( n ) =

, where i ∈ {1 ... T} (10a)

In an alternative embodiment, the value indicator E ' ( n ) can be expressed as follows:

, где i∈ {1 … T} (10b) E ' ( n ) =

, where i ∈ {1 ... T} (10b)

In FIG. 8B, the calculation of the detection statistics g _i ( n ) shown in equation (9) is performed by 2L + 1 multipliers 762, 2L multipliers 764, and adder 766. Each multiplier 762 multiplies a received symbol for a corresponding pilot subband or data subband with a conjugate known pilot symbol or a re-modulated symbol for this subband. Each multiplier 764 performs a scalar product of the output from a pair of multipliers 762 for a pair of adjacent pilot sub-band and data sub-band. Then, adder 766 sums the output signals from L multipliers 764 to provide detection statistics g _i ( n ). (For the embodiment shown in equation (10a), block 768 receives g _i ( n ) and provides the real part to adder 784, which sums the real parts g _i ( n ) for all T antennas. Then, the output from adder 784 is input to square by block 786 to provide an indication of the value of E ' ( n ). For the embodiment shown in equation (10b), block 786 may be placed between block 768 and adder 784.

The adaptive threshold calculation unit 770 determines the adaptive threshold Y ' ( n ) to use for each received transformed symbol. Each of the 2L + 1 received symbols r _i ( k ) for the pilot subband and data subband is provided to a corresponding block 772, which determines the square of the symbol value. Then, adder 774 sums the squared values from all 2L + 1 blocks 772 in order to provide threshold values Y _i ' ( n ). An adder 780 receives and sums the threshold values for all T antennas, providing an aggregate threshold value Y ' _tot ( n ), which can be expressed as follows:

(11) Y ' _tot ( n ) =

(eleven)

A multiplier 782 scales the cumulative threshold with a scale factor S _2a , providing a final threshold that can be defined as Y ′ ( n ) = S _2a · Y ′ _tot ( n ). In general, both the threshold value Y ' ( n ) and the value of the indicator E' ( n ) are accumulated according to the duration of the detected TIR. Thus, if the TIR covers many periods of the OFDM symbol, the threshold values and indicator values are first calculated as described above for each of these OFDM symbols and then accumulated, providing the final threshold values and indicator values for the TIR.

The comparator 790 compares the final value of the indicator E ' ( n ) with the final threshold value Y' ( n ) and provides the output signal of the detector D ' ( n ), which can be expressed as follows:

(12) D ' ( n ) =

(12)

If the output signal of the detector D ' ( n ) indicates that the MOS KAN2 is present, then the KAN2 message decoded by the data processor RX is considered reliable, and according to the situation it can be further processed by the controller. Otherwise, CAN2 is rejected.

FIG. 8C is a block diagram of a decision-controlled detector 440e that performs differential detection in the frequency domain and represents one embodiment of the detector 440c of FIG. 7. Samples x _i ( n ) are provided with a delay line / buffer 830, which are implemented based on N-1 delay elements 832, each of which provides one delay sample period.

Detector 440e correlates each received OFDM symbol with its corresponding “reconstructed" OFDM symbol, determining the metric E " ( n ) for the received OFDM symbol. Each of N samples x _i ( n ) for the received OFDM symbol is provided to a respective multiplier 842, which also receives corresponding conjugate reconstructed sample d * ( j ), where j ∈ {0 ... N-1}. To obtain d * ( j ), the pilot symbols p ₂ ( k ) for the pilot subbands (for example, as shown in the table), the modulated c ( k ) symbols for the subbands data, and zero signal values for unused subbands (i.e., N symbols for N full subbands) for the OFDM symbol period, are converted to the time domain by N-point IFFT 830, obtaining N reconstructed samples, from d (0) to d (N-1 ), which are then conjugated and provided to N multipliers 842. The operations performed by the other elements shown in FIG. 8C are similar to those described above with respect to FIG. 6. The value of the indicator E _i " ( n ) for each antenna can be expressed as follows:

, где i∈ {1 … T} (13) E _i " ( n ) =

, where i ∈ {1 ... T} (13)

The final value of the indicator E _i " ( n ) for all T antennas in this case can be expressed as follows:

, (14) E " ( n ) =

, (14)

The threshold Y ″ ( n ) used for comparison with the final value of the metric E ″ ( n ) can be determined similarly to that described above with respect to FIG. 6. In particular, the total threshold value Y ” _tot ( n ) for all T antennas, which can be expressed as follows:

(15) Y ” _tot ( n ) =

(fifteen)

The final threshold value in this case can be set in the following form: Y " ( n ) = S _2b · Y” _tot ( n ).

For the decision-controlled detector, the scale factor S ₂ (which is the coefficient S _2a for the detector 440d shown in FIG. 8B and the coefficient S _2b for the detector 440e shown in FIG. 8C) is a positive constant chosen to provide: ( 1) a certain probability of missed TIR KAN2 detection, and (2) a certain probability of false alarms for the incorrectly declared presence of TIR KAN2. If CAN2 messages are determined so that they do not include CRC values, then determining whether the CAN2 messages are actually present, rely solely on the detector. Erroneous CAN2 messages can be provided to the controller for the following reasons:

- false alarm - noise in the received signal falsely triggers detection; and

- incorrect decoding - the signal starts the detection correctly, but the decoded message KAN2 includes uncorrected and undetected errors.

If channel 2 is used as a random access channel, then a false alarm for TIR CAN2 can cause the system to allocate resources to a non-existent user terminal, which in this case leads to a waste of resources. In this case, it is desirable to choose the scale factor S ₂ in such a way as to minimize the likelihood of a false alarm, since it is undesirable to have noise, often starting a waste of resources.

The probability of incorrect decoding refers to the probability of detection and a higher probability of detection can cause an increase in cases of incorrect decoding. When a case of incorrect decoding occurs, an erroneously decoded CAN2 message is provided to the controller. The controller may be able to verify the accuracy of the CAN2 message in some other ways. For example, if the KAN2 message includes a unique identifier for the user terminal that sent the message, then the controller can visually check whether the unique identifier for the restored KAN2 message is included in the list of valid identifiers. If the unique identifier in the received CAN2 message is defined as valid, then the system can allocate resources to the user terminal associated with this identifier.

When choosing the scale factor S _{2, it} may be necessary to detect as many valid CAN2 messages as possible while maintaining the false alarm and incorrect decoding probabilities below a certain level. You can also change the scale factor S ₂ based on system load. For example, if the system load is low and there are a small number of valid identifiers, then the probability of an erroneous allocation of system resources seems small. In this case, a lower detection threshold may be used. As the system load increases, the detection threshold can be increased to reduce cases of incorrect decoding.

9 is a block diagram of an embodiment of an access point 110x and a user terminal 120x in the system 100. For this embodiment, both the access point 110x and the user terminal 120x are equipped with multiple antennas. In general, both the access point and the user terminal can be equipped with any number of transmit / receive antennas.

In the user terminal 120x, the uplink TX data processor 310 receives and processes the data traffic from the data source 308, as well as other data (e.g., for the messages CAN1 and CAN2) from the controller 360, providing multiplexing of the data and pilot symbols, such as described above with reference to FIG. 3A. TX spatial processor 320 can perform spatial processing of pilot symbols and data symbols, providing a stream of transmitted symbols for each antenna. Each modulator 340 receives and processes a respective stream of transmitted symbols, providing a corresponding modulated uplink signal, which is then transmitted from the corresponding antenna 350.

At the access point 110x T antennas (408a through 408t), the transmitted uplink modulated signals are received from the user terminal, and each antenna delivers the received signal to a respective receiver 410. Each receiver 410 adapts the received signal and further digitizes the adapted signal to provide samples. Then, as described above, the data demodulator / detector 420 performs processing by detecting the messages KAN1 and KAN2. An RX data processor 450 processes the recovered symbols, providing a decoded data stream (which can be provided to the data receiver 452 for storage) and recovered messages KAN1 and KAN2 (which can be provided to the controller 460 for further processing).

The processing for the downlink may be the same or different from the processing for the uplink. Data from data source 468 and service signals (eg, response messages) from controller 460 are processed (eg, encoded, interleaved, and modulated) by TX data processor 470, and can be spatially processed by TX spatial processor 480. Then, symbols transmitted from TX spatial processor 480 are symbols are processed by modulators 410a-410t to generate T modulated downlink signals that are transmitted through antennas 408a-408t.

At user terminal 120x, downlink modulated signals are received by antennas 350, adapted and digitized by receivers 340, and processed by RX spatial processor 370 and RX data processor 380 in a manner complementary to that performed at the access point. Decoded data for the downlink may be provided to the data receiver 382 for storage and / or to the controller 360 for further processing.

Controllers 360 and 460 control the operation of various processing units in a user terminal and access point, respectively. The memory units 362 and 462 store data and program codes used by the controllers 360 and 460, respectively.

For clarity, certain correlation embodiments and decision-controlled detectors, demodulators, and receiver units have been described for specific TIR formats. Also without departing from the scope of the invention, various other embodiments and uses of these detectors are possible. For example, a correlation detector can be used for a channel, whereby transmissions contain time compensation, and a decision-controlled detector can be used for a channel, whereby transmissions do not contain time compensation.

A decision-controlled detector may be implemented in the frequency domain (as shown in FIG. 8B) or in the time domain (as shown in FIG. 8C). In addition, a decision-controlled detector can be used for various TIR formats. For example, a decision-controlled detector can be used for the TIR format, according to which the data and the pilot signal are sub-band multiplexed (as described above for TIR CAN2), the TIR format, according to which the data and the pilot signal contain time division Signals (TDM) (as described above for TIR CAN1), etc. A decision-controlled detector can also be used with or without a pilot signal. In general, a decision-controlled detector uses received frequency-domain data symbols or reconstructed time-domain data samples to detect data transmissions in a received signal. This detector can advantageously be used to detect message errors when CRC or other error detection mechanisms are not available.

Using an adaptive threshold can provide robust detection parameters in many operational scenarios, such as an unlicensed frequency range in which various sources of interference can be present. The threshold can be set based on private statistics for detected transmissions. These statistics may relate to useful signal energy, noise and interference in the transmission, or to some other parameters.

The detectors, demodulators, and receivers described herein can be used for various types of data channels. For example, these blocks can be used for various types of random access channels, such as those described in detail in the aforementioned US patent application No. 60 / 432,440 and provisional patent application US No. 60 / 421,309.

The detectors, demodulators and receivers described here can also be used for various wireless multiple access communication systems. One such system is a multi-input multiple output multiple access (MIMO) wireless communication system described in the aforementioned provisional patent application US No. 60 / 421,309. In general, these systems may or may not use OFDM, or may use some other multi-carrier modulation scheme instead of OFDM, and may or may not use MIMO.

The detectors, demodulators and receivers described herein can be implemented using various means. For example, these blocks may be implemented in hardware, software, or a combination thereof. For hardware execution, detectors and receivers can be implemented within one or more specialized integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), user programmable gate arrays (FPGAs) ), processor, controller, microcontroller, microprocessor, other electronic unit, designed to perform the functions described here, or a combination of the above.

For the case of software execution, signal processing for detectors, demodulators, and receivers can be implemented by modules (for example, procedures, functions, and so on) that perform the functions described here. Program codes may be stored in a memory unit (e.g., memory units 362 and 462 shown in FIG. 9) and executed by a processor (e.g., controllers 360 and 460). The memory unit can be implemented in the processor or be external to the processor, in which case it can be connected with the ability to exchange information to the processor through various means known from the prior art.

The previous description of the disclosed embodiments has been undertaken to provide any person skilled in the art with the possibility of practicing or using the present invention. Various modifications to these embodiments will be apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to limit the embodiments shown here, but should be within the broadest scope consistent with the principles and new features disclosed herein.

Claims (37)

1. A receiver unit in a wireless communication system, comprising
a demodulator configured to process the received data symbols to provide the restored symbols;
a first data processor configured to process the recovered symbols to provide decoded data;
a second data processor configured to process the decoded data to provide remodulated symbols; and
a detector configured to process the received data symbols and remodulated symbols to provide a detector output signal. 2. The receiver unit according to claim 1, in which the received data symbols are intended for hypothetically received data transmission, and the output signal of the detector indicates whether the data transmission is considered accepted. 3. The receiver unit according to claim 1, additionally containing
a threshold calculation unit configured to determine a threshold for use in data transmission, wherein the detector is configured to provide an indicator based on the received data symbols and re-modulated symbols, and the detector output signal is determined based on the indicator and threshold. 4. The receiver unit according to claim 3, in which the threshold calculation unit is configured to determine a threshold based on a plurality of received signals for a plurality of antennas, and the detector is configured to determine an indicator based on the plurality of received signals. 5. The receiver unit according to claim 1, wherein the detector is further configured to process the received pilot symbols to provide an output signal from the detector, the data symbols being transmitted in the data subbands and the pilot symbols being transmitted in the pilot subbands and subbands data are multiplexed with pilot subbands. 6. The receiver unit according to claim 5, in which the data subbands are interlaced with the pilot subbands such that each of the data subbands is framed on both sides by the pilot subbands. 7. The receiver unit according to claim 1, in which the detector is configured to perform coherent detection in the time domain, the received data symbols are obtained based on input samples for a hypothetically received data transmission, and the detector is configured to correlate between input and reconstructed samples obtained based on re-modulated symbols. 8. The receiver unit according to claim 1, in which the detector is configured to perform differential detection in the frequency domain. 9. The receiver unit according to claim 1, in which the detector is configured to process the received pilot symbols to provide an output signal from the detector of multiplying each of the received data symbols with a corresponding symbol from the remodulated symbols to provide a demodulated data symbol, multiplying each of the received symbols a pilot signal with a corresponding symbol from known pilot symbols to provide a demodulated pilot symbol, performing scalar products between demodulated data symbols and demodulated symbols of the pilot signal and accumulation of the results of scalar products. 10. A receiver unit in a wireless communication system, comprising a processor configured to process received data symbols for hypothetically received data transmission and provide re-modulated symbols, which are estimates of the transmitted data symbols; and a detector configured to process the received data symbols and remodulated symbols to provide an output from the detector, which indicates whether the data transmission is considered to be received. 11. The receiver unit of claim 10, wherein the processor is configured to demodulate the received data symbols to provide the recovered symbols, decode the restored symbols to provide decoded data, and re-encode the decoded data to provide remodulated symbols. 12. The receiver unit of claim 10, wherein the processor is further configured to process the received pilot symbols for transmitting data and corresponding known pilot symbols to provide an output signal from the detector. 13. The method of receiving data in a wireless communication system, namely, that
demodulating received data symbols to provide recovered symbols; processing the recovered symbols to provide decoded data; process decoded data to provide remodulated symbols; and detecting the received data symbols and remodulated symbols to provide a detector output signal. 14. The method according to item 13, in which the received data symbols are intended for hypothetically received data transmission, and the output signal of the detector indicates whether the data transmission is considered accepted. 15. The method according to item 13, further comprising determining a threshold for use in data transmission, wherein the detection step includes providing an indicator based on the received data symbols and remodulated symbols, and the detector output signal is determined based on the indicator and threshold. 16. The method of claim 15, wherein the step of determining a threshold includes determining a threshold based on a plurality of received signals for a plurality of antennas, and the detection step includes determining a metric based on a plurality of received signals. 17. The method of claim 13, wherein the detecting step includes processing the received pilot symbols to provide a detector output, the data symbols being transmitted in data subbands and the pilot symbols being transmitted in pilot subbands and data subbands multiplexed with pilot subbands. 18. The method of claim 17, wherein the data subbands alternate with the pilot subbands such that each of the data subbands is framed on both sides by the pilot subbands. 19. The method of claim 13, further comprising receiving received data symbols based on input samples for a hypothetically received data transmission, wherein the detection step includes performing coherent detection in the time domain and the detection step further includes performing a correlation between input and reconstructed samples based on re-modulated symbols. 20. The method according to item 13, in which the detection step includes the implementation of differential detection in the frequency domain. 21. The method according to item 13, in which the detection step includes processing the received symbols of the pilot signal to provide the output signal of the detector,
multiplying each of the received data symbols with a corresponding symbol from re-modulated symbols to provide a demodulated data symbol, multiplying each of the received pilot symbols with a corresponding symbol from known pilot symbols to provide a demodulated pilot symbol, performing scalar products between demodulated symbols data and demodulated symbols of the pilot signal, and the accumulation of the results of scalar products. 22. The method of receiving data in a wireless communication system, namely, that
process the received data symbols for a hypothetically received data transmission and provide remodulated symbols, which are estimates of the transmitted data symbols; and detecting the received data symbols and re-modulated symbols to provide an output signal from the detector, which indicates whether the data transmission is considered to be received. 23. The method of claim 22, wherein the processing step includes demodulating the received data symbols to provide recovered symbols, decoding the recovered symbols to provide decoded data, and re-encoding the decoded data to provide remodulated symbols. 24. The method of claim 22, wherein the processing step includes processing the received pilot symbols for transmitting data and corresponding known pilot symbols to provide an output from the detector. 25. A computer-readable medium having instructions stored thereon for executing a method according to any one of claims 13-24. 26. A receiving device in a wireless communication system, comprising: means for demodulating received data symbols to provide recovered symbols; means for processing the restored symbols to provide decoded data; means for processing decoded data to provide remodulated symbols; and means for detecting received data symbols and remodulated symbols to provide a detector output signal. 27. The device according to p, in which the received data symbols are intended for hypothetically received data transmission, and the output signal of the detector indicates whether the data transmission is considered accepted. 28. The device according to p. 26, further containing means for determining a threshold for use in data transmission, and the means for detecting includes means for providing an indicator based on the received data symbols and re-modulated symbols, and the output signal of the detector is determined based on the indicator and the threshold. 29. The apparatus of claim 28, wherein the means for determining a threshold includes means for determining a threshold based on a plurality of received signals for a plurality of antennas, and the means for detecting includes means for determining an indicator based on a plurality of received signals. 30. The device according to p. 26, in which the means for detecting includes means for processing the received symbols of the pilot signal to provide the output signal of the detector, and the data symbols are transmitted in the data subbands, and the pilot symbols are transmitted in the pilot subbands, and data subbands are multiplexed with pilot subbands. 31. The apparatus of claim 30, wherein the data subbands alternate with the pilot subbands such that each of the data subbands is framed on both sides by the pilot subbands. 32. The device according to p. 26, further comprising a means for obtaining received data characters based on input samples for a hypothetically received data transmission, and the means for detection includes means for performing coherent detection in the time domain and means for correlating between input and reconstructed samples obtained based on remodulated symbols. 33. The device according to p, in which the means for detection includes means for performing differential detection in the frequency domain. 34. The device according to p, in which the means for detection includes
means for processing the received pilot symbols to provide the detector output signal; means for multiplying each of the received data symbols with a corresponding symbol from the modulated symbols to provide a demodulated data symbol; means for multiplying each of the received pilot symbols with a corresponding symbol from the known symbols a pilot signal for providing a demodulated pilot symbol, means for performing scalar products between demodulators GOVERNMENTAL data symbols and demodulated pilot symbols, and
means for accumulating the results of scalar products. 35. A reception device in a wireless communication system, comprising: means for processing received data symbols for hypothetically received data transmission; and means for providing re-modulated symbols, which are estimates of the transmitted data symbols; and means for detecting the received data symbols and re-modulated symbols to provide a detector output signal that indicates whether the data transmission is considered to be received. 36. The apparatus of claim 35, wherein the processing means and the providing means include means for demodulating the received data symbols to provide the restored symbols, means for decoding the restored symbols to provide decoded data, and means for re-encoding the decoded data to provide re-modulated characters. 37. The apparatus of claim 35, wherein the processing means and the providing means include means for processing the received pilot symbols for transmitting data and corresponding known pilot symbols for providing an output signal from the detector.

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