MXPA06007535A - Hybrid spread spectrum radio system - Google Patents

Abstract

Systems and methods are described for hybrid spread spectrum radio systems. A method includes modulating a signal by utilizing a subset of bits from a pseudo-random code generator to control an amplification circuit that provides a gain to the signal. Another method includes:modulating a signal by utilizing a subset of bits from a pseudo-random code generator to control a fast hopping frequency synthesizer;and fast frequency hopping the signal with the fast hopping frequency synthesizer, wherein multiple frequency hops occur within a single data-bit time.

Description

HYBRID EXTENDED SPECTRUM RADIO SYSTEM FIELD OF THE INVENTION The invention relates generally to the field of communications. More particularly, the invention relates to extended spectrum communications. BACKGROUND OF THE INVENTION Conventional radio frequency (RF) digital data transmission schemes are generally highly susceptible to errors caused by multiple path propagation and other interference conditions.

Historically, direct-frequency extended-spectrum techniques have offered reasonably good immunity against the types of long-path reflections (for example, outdoors), where the time distribution (dispersion) of the arrival of individual, successive signal reflections is generally greater than the effective shredding period (inverse of the spread spectrum shredding rate) of the transmitted signals. Outdoor environments often offer delayed dispersion profiles in the range of 3 to 100 μs (typically about 25 μs of maximum RMS [for terrain with mountains]) and thus are in general Ref.: 174154 directed with dispersion speeds or extension or signal of approximately 1 Msegment / second, (especially when used with quadrature modulation schemes shifted in time, such as OQPSK); these are in fact very similar to the existing parameters of the IS-95 CDMA cellular telephone system in wide use throughout the United States and in many other countries. In the case of indoor propagation, however, the dispersion times are much shorter - typical figures are in the range of 10 to 250 ns, with an average RMS dispersion value of 50 ns. Longer values could imply a minimum dispersion ratio of approximately 4 Msegments / second, while shorter values (the worst case) represent speeds of approximately 100 Msegments / second (and, thus, at least 100 MHz). signal bandwidth for standard direct sequence (DS) signals using the conventional binary phase shift key (BPSK) or frequency shift key modulation (FSK) For OQPSK schemes as mentioned above, The minimum bandwidth required is halved, but nevertheless it is still very difficult to handle in overloaded RF bands Obviously, this bandwidth is not reasonably obtainable in any of the Industrial, Scientific and Medical (ISM) bands currently allocated for the transmission of extended spectrum in the United States below 5 GHz (and would require the entire band of 100 MHz available for ISM and Unlicensed National Information Infrastructure applications [U-NII] above 5 GHz), so other techniques must be applied to overcome the problem of multiple paths for indoor wireless connections. A prevalent option is to use the frequency hop, so that by periodic changes of carrier frequency, the signal will jump to frequencies that do not show zeros of multiple trajectories (destructive interferences) coming from the transmitter at the desired reception points. In general, the total received RF energy from many of these data bursts (hops) will be canceled by the zeros (and thus produce bad data packets), but in general a majority will be of satisfactory quality to provide reasonably efficient operation. effective link or connection. However, any complex error correlation and / or interleaving algorithms (and delay inducers) (eg, Reed-Solomon) must be introduced into the link, or numerous packet retransmissions will be required to successfully transport the load. data. In any case, significant levels of latency and concurrent limitations of rate or link speed will inevitably result. Some methodologies have even been developed to build "jump tables" of usable frequencies (low error) in the control software of the system, and thus avoid the frequencies with propagation zeros, but in general according to the signal transmission environment changes with the movements of equipment, personnel and sources of RF interference, the group of "bad" frequencies will need to be constantly updated; even so; statistically and practically, some bad packages will nevertheless always be received. In addition, the use of "smart" jump schemes that on average avoid certain jump channels in a coordinated manner have historically not been used by the Federal Communications Commission (FCC) on the ISM bands, since the channel occupation the average would be non-uniform, and this is biased from the normal long-range random signal frequency statistics, intended for the operation of the frequency hopping system, of the ISM band, resulting in a statistical increase in interference to other users (although through changes in rules by the FCC in the last few months, this prohibition has been somewhat relaxed). In general, however, this scheme although it is usually workable for fixed devices, generally fails in mobile applications or when the RF environment is dynamic, since the positions of the multipath zeros (and thus the groups of "bad" channels) are constantly changing. To date, the requirements of a more robust scheme (fewer data errors) that will work effectively even in severe multipath environments (eg, areas highly reflective to RF) and still avoids the introduction of extremely complex error correction hardware , with substantial latency (delay) in the transmission process and / or the requirement for frequent retransmissions, offering more a solution to the problem of link latencies (which can be particularly significant in high-speed control applications where delays can cause problems of loop stability or circuit for the RF systems in the loop referred to above) have not been fully met. For the operation in the United States, the scheme must additionally comply with the regulations of Part 15 of the FCC for the ISM and U-NII bands by guaranteeing adequately random spectral characteristics of its transmissions at all times. Another essential aspect of modern RF telemetry systems is that of the efficient use of energy. It is desirable to operate many distributed devices, including sensors, alarm systems, RFID tags, and the like from low-cost, compact battery sources for maintenance-free intervals of 1 to about 5 years (or even longer). It is therefore highly desirable to provide a system RF telemetry protocol that achieves reliable data transmission with an absolute minimum of power consumption of the remote device. Another critical need in many systems is to simultaneously operate a large number of RF devices (such as markers or labels, sensors and the like) in a proximal area without significant statistical levels of mutual interference; in the common language, this is the problem of multiple family access that is handled by multiplexing or multiple access techniques of frequency, time or code division (typically referred to as FDMA, TDMA and CDMA for its acronym in English, respectively). However, these methods have so far not been simultaneously employed in a programmably or adaptively coupled or coordinated manner, to provide a useful increase in the allowable number of devices operable in a given area for a specific amount of mutual interference. Yet another need is for an RF signal transmission technique which, even in the presence of multipath and multiple user interference, can support an accurate radiolocation function where the respective locations of the RF devices can be easily detected, such as to track the equipment, containers and personnel. Another key need is for an RF signaling protocol that offers enhanced transmission security against reception, decoding or even detection by unauthorized parties. Finally, there is a need for an RF signaling technique that also provides a high degree of programming capability and signal adaptability to quickly achieve exchanges and exchanges in DS code lengths, frequency and time skip patterns, and interrelationships thereof to effectively address the dynamic signal and device usage conditions (eg, changing multipath conditions and RFI and the functional requirements of the system). What is needed is then a solution that is limited to all these requirements. BRIEF DESCRIPTION OF THE INVENTION There is a need for the following modalities. Of course, the invention is not limited to these modalities. According to a first aspect of the invention, a method comprises: modulating a signal by using a subset of bits from a pseudo-random code generator to control an amplification circuit that provides a gain to the signal. According to a second aspect of the invention, an apparatus comprises: a pseudo-random code generator; and an amplitude controller coupled to the pseudo-random code generator. According to a third aspect of the invention, a method comprises: directly synthesizing a digital signal including: the amplitude modulation of an output channel with a quadrant amplitude multiplier to provide an amplitude signal with vibration. According to a fourth aspect of the invention, an apparatus comprises a direct digital synthesizer that includes a four-quadrant amplitude multiplier, wherein the direct digital synthesizer provides an amplitude signal with vibration. According to a fifth aspect of the invention, a method comprises: directly synthesizing a digital signal that includes: the amplitude modulation of a "in-phase" channel (I) with a first four-quadrant amplitude multiplier; and the amplitude modulation of a "quadrature phase" channel (Q) with a second amplitude multiplier of four quadrants. According to a sixth aspect of the invention, an apparatus comprises: a direct digital synthesizer that includes a first amplitude amplifier of four quadrants and a second multiplier of amplitude of four quadrants, wherein the first multiplier of amplitude of four quadrants and the second amplitude multiplier of four quadrants, are coupled together in parallel. According to a seventh aspect of the invention, a method comprises: modulating a signal by using a subset of bits from a pseudo-random code generator, to control a fast-hop frequency synthesizer; and the rapid jump of frequency of the signal with the fast jump frequency synthesizer, where the multiple frequency jumps occur within a simple time of bit of data. According to an eighth aspect of the invention, an apparatus comprises: a pseudo-random code generator; and a fast-jump frequency synthesizer coupled to the pseudo-random code generator, wherein multiple frequency jumps occur within a simple data bit time. These and other embodiments of the invention will be better appreciated and understood when considered in conjunction with the following description and the appended figures, it should be understood, however, that the following description, while indicating the various embodiments of the invention and numerous specific details of it, is given by way of illustration and not limitation. Many substitutions, modifications, additions and / or rearrangements may be made within the scope of the invention, without departing from the spirit thereof, and the invention includes all such substitutions, modifications, additions and / or rearrangements.i.

BRIEF DESCRIPTION OF THE FIGURES The figures that accompany and that are part of this specification are included to describe certain aspects of the invention. A clearer conception of the invention, and of the components and operation of the systems provided with the invention, will become more readily apparent by reference to the exemplary, and therefore non-limiting, embodiments illustrated in the drawings, wherein the Similar reference numbers (if they appear in more than one view) designate the same elements. The invention may be better understood by reference to one or more of these drawings in combination with the description presented herein. It should be noted that the features illustrated in the drawings are not necessarily drawn to scale. Figure 1 illustrates a schematic block view of a hybrid direct frequency / sequence direct hopper (BPSK-modulated) transmitter, which represents one embodiment of the invention.

Figure 2 illustrates a block schematic view of a hybrid direct frequency / direct sequence hopper (BPSK-modulated) receiver, which represents one embodiment of the invention. Figure 3 illustrates a schematic block view of a hybrid time-spread / direct sequence extended-spectrum transmitter, representing one embodiment of the invention. Figure 4 illustrates a block schematic view of a hybrid time-extended / forward sequence hybrid spectrum receiver, representing one embodiment of the invention. Figure 5 illustrates a schematic block view of an extended-spectrum hybrid, frequency change / time hopping / forward sequence hybrid, representing one embodiment of the invention. Figure 6 illustrates a schematic block view of an extended spectrum receiver, frequency change / time hopping / forward sequence hybrid, representing one embodiment of the invention. Figure 7 illustrates a schematic block view of a hybrid programmable direct digital synthesizer (DDS) transmitter representing one embodiment of the invention. Figure 8 illustrates a schematic block view of a hybrid, interconnected, programmable direct digital synthesizer (DDS) transmitter, representing one embodiment of the invention. Figure 9 illustrates a schematic block view of an RF up converter circuit, representing one embodiment of the invention. Figure 10 illustrates a schematic block view of an alternative RF up-converter circuit, representing an embodiment of the invention. Figure 11 provides a block diagram of an RF transmitter system incorporating an RF signal bias control circuit, which represents one embodiment of the invention. Figure 12 provides a block diagram of an RF receiver system corresponding to the transmitter of the Figure 11, which incorporates the capacity of polarization diversity reception, which represents a modality of the invention. Figure 13 illustrates an indoor RF propagation environment, typical via its characteristic profile of signal delay. Figure 14 depicts a representative, moderately sharp, multi-trajectory cancellation zero, typical of indoor RF environments, with a superimposed group of five hybrid medium-band extended spectrum signals, representing one embodiment of the invention .

DESCRIPTION OF THE INVENTION The invention and the various features and advantageous details thereof are explained more fully with reference to the non-limiting modalities which are illustrated in the appended figures and detailed in the following description. Descriptions of well-known components and processing techniques are omitted so as not to unnecessarily obscure the invention in detail. It should be understood, however, that the detailed description and specific examples, while indicating the preferred embodiments of the invention, will be by way of illustration only and not by way of limitation. Various substitutions, modifications, additions and / or rearrangements within the spirit and scope of the underlying inventive concept will become apparent to those skilled in the art from this detailed description. The United States Patents referred to below describe the modalities that were satisfactory for the purposes for which they are intended. The complete contents of United States Patents Nos. 6,556,942; 5,623,487; 5,521,937; 5,274,665; and 4,550,292 are expressly incorporated by reference herein, for all purposes. The complete contents of the United States Patent Application 09 / 671,636, filed September 27, 2000, in which the shipping rights have been paid, are expressly incorporated by reference herein, for all purposes. The invention may include the specific combination related to the code of the standard direct sequence (DS) modulation of extended spectrum (SS) with "fast" frequency hopping (FFH), where the multiple frequency jumps occur within a time of bit of data, simple. Specifically, the most significant benefit to the rapid frequency jump is that each bit is represented by transmissions in pieces or segment at multiple frequencies. If one or more pieces are corrupted by multiple trajectories or by interference in the RF link, statistically a majority should still be correct. Of course, with the proper error detection, if even a piece is correct, the original data bit can still be recovered correctly. [Standard frequency hop or "slow" (SFH or simply FH), in contrast, transmits at least one and usually several bits of data in each hop interval]. In the invention, alternatively, for example, the hop can be increased or replaced by frequency sweeps or "parasites" of an unconstrained or constrained nature (eg, continuous frequency and / or continuous phase). The environments of propagation of RF in interiors, difficult, typical of offices and large industrial spaces, dictate that highly robust techniques are necessary for the proper functioning of the link. The multiple reflections of the signals cause significant deterioration of conventional signals and the high data error rates expected. The temporal graph of Figure 13 illustrates an indoor RF propagation environment, typical by means of its characteristic profile of signal delay. Note that the extension or dispersion of the RMS delay (the reflections of time signals from the walls, ceiling and floor are still reaching the antenna) is close to 50 ns; without special RF signaling techniques, this will cause major data errors, unless the information rate or rate is very low. Even conventional extended spectrum connections will need more than 20 MHz of bandwidth for successful operation in this way. The effect of multiple trajectories can also be clearly observed in the frequency domain plot of Figure 14, which describes a representative, moderately sharp, multi-trajectory cancellation zero (typical of indoor RF environments) close to the center of the signal band ®, with a superimposed group of five hybrid signals of extended spectrum of bandwidth (® al ©). Due to zero propagation, the signal band ® is essentially blocked, the bands © and © are somewhat attenuated (and distorted due to the spectral tilt), while the bands © and © are largely unaffected. Clearly, by using multiple frequencies, the probability that the data reaches its destination correctly is much greater than in conventional systems. A further significant advantage of the rapid jump aspect of the invention is that of "avoiding" multiple trajectories: this can be appreciated by noting that in a multiple reflection environment as described in Figure 13, if the time between the signal jumps successive is less than among the reflections successfully received, then there will be very little opportunity for subsequent signal arrivals to cause destructive interference when combined with previous trajectories; of course, this may (depending on the particular delay profile of the area) require clearly high jump rates or rates, although this requirement may also be improved to some extent by the presence of the DS component in the full HSS signal. Another aspect of the invention is the high degree of programming capability of the different facets of the HSS signal in the preferred implementations; here, the complete software / firmware address (computer hardware / microprogramming) of the dispersion rate ratio DS (shred), the jump ratio FH, the carrier frequencies and the time / frequency profiles, plus the jump control of time (TH) and / or interrelated or independent polarization, provides a hitherto unobtainable degree of programming capability and signal adaptability to quickly and efficiently achieve exchanges in the DS code lengths, in the jump patterns of frequency and time, the polarization of the signal, and the interrelationships thereof to effectively address the dynamic signal and the conditions of use of the device (for example, multiple trajectories and changing RFI conditions, and the functional requirements of the system ). The excellent security of the signal is another advantage of the invention: since so many signal parameters can change in a fast way, and since successful synchronization requires a priori substantial knowledge of the code relationships between the DS, FH components , TH and polarization of the HSS signal, the unintended listener will experience great difficulty in fully synchronizing the HSS transmission and the decoding of the embedded data. An even higher level of HSS signal security can be achieved by dynamically altering the PN code relationships within the HSS transmitter; this may take the form of rolling code segments, codebook override, or reassignments based on tables of bit pattern relationships. Obviously, the degree of difficulty in the interception of such complex transmission without the required "keys" or code index is extreme; Of course, standard data coding techniques in addition to the above can also be added to provide even greater security of the transmitted data. Another major problem solved by the invention is that of efficient multiple access: in a proximal area, frequently from dozens to hundreds or even thousands of devices (e.g., wireless sensor nodes, RFID tags, alarm units, and the like) are required to interoperate without significant loss of data due to RF interference, multiple trajectories or collisions from other devices in the group. Properly programmed HSS can show superior multi-access performance even to CDMA systems controlled at full power (eg cell phones) due to the concatenation of DS-CDMA techniques with the additional dimensions of the FH and TH modulations. These added signal dimensions allow for greater diversity, access for more users, and / or can substantially facilitate the need for high accuracy in the CDMA energy control function. In addition, the improved flexibility of the invention can support single transmission devices which can not be controlled in energy, and thus are not suitable for operating in a true CDMA-only environment. Since the multiple access interference caused by the uncontrolled energy levels of these units quickly exceeds the tolerable limits for CDMA operation, especially with variant device sites (and, thus, the classic near-far problem for DS systems) , the HSS protocol of the invention offers an effective means of accommodating such uncontrolled, one-way, random transmission units without adversely affecting the multiple system access operations of the entire system. This is mainly achieved by the assignment of specific CDMA codes to such devices, and by constraining their hopping frequency and time sequences to statistically avoid the more conventional (controlled energy) slots or transmission intervals of the units. further, to provide enhanced performance in typical, multi-trajector, wireless, industrial, commercial, and wireless signal propagation environments, the ability to perform a continuous phase synthesis (CP) of the combined DS / FFH signal waveform, allows the most effective use of the RF spectrum (narrower bandwidth control and avoidance of transient spectral "spurious radiations") and simultaneously the potential for faster, more stable receiver synchronization in adverse transmission environments. A further aspect of the invention is the application of amplitude vibration modulation statistically and / or the statistical polarization control of the combined CD-DS / FFH signal by means of pseudorandom polynomial techniques to provide even more immunity to signal cancellations. Induced by multiple trajectories, which typically cause deactivations and high speeds accompanying data errors. The use of amplitude vibration minimizes the effect of successive chunk cancellations due to reflections of destructive signals, closely spaced in low loss RF environments (eg with metal wall). This can be used to disturb or "break" the zeros of signals induced by multiple trajectories, where the signal is effectively canceled by destructive interferences due to unfortunate combinations of multiple reflected signals from various paths. When the vector sum of the multiple components approaches zero, by rapidly changing the amplitude of the transmitted signal (within the quarter of time of the successfully received reflection signals), the cancellation of the multipath signals can at least partially be "undone", leaving a reasonable amount of instantaneous signal amplitude in the receiver to decode. To achieve the benefit of maximum error reduction, regardless of this amplitude vibration technique, the vibration time ratio (change in amplitude) must be at least comparable to the rate or speed of arrival of the successive reflected signals in the designated receiver (and ideally somewhat larger); this, in turn, still requires a clearly high speed or vibration ratio, unless it is combined with other aspects of the invention. The application of the polarization control controlled by the PN code, when properly synchronized between the transmitter and the receiver (s), provides even greater signal diversity and, thus, immunity against zeroes and offsets of signals induced by multiple trajectories ( as well as unauthorized reception of signals). This is achieved by altering the signal wave polarization transmitted as a function of a specified PN code; with different orientations of the electric (E) and magnetic (H) fields of signals, the respective reflection coefficients at the boundaries of the room and other interfaces will also change, thus shifting (or "vibrating") the null depths of trajectories multiple, frequencies and locations. As will be described in the following examples, the programming of the transmitted, effective signal polarization can be achieved by the concurrent control of the relative proportion of the energy sent to two orthogonally polarized antennas, typically one vertical and the other horizontal. Frequently, the concatenation of some, most or all of the aforementioned methodologies, will be required to provide a highly reliable link suitable for commercial, industrial and / or critical military applications in adverse RF environments (eg, multi-path prone and interference). Although there is a plurality of possible implementations at the board level, specific to the modulator and / or hybrid extended spectrum demodulator (HSS), the most practical, compact and most inexpensive way is certainly one that involves the segments of custom IP integrated circuits. , highly integrated, to achieve the desired functionality of the system, the programming capacity, the size and the reasonable consumption characteristics of the energy at a usefully low cost. All the aforementioned features (with the possible exception of antenna polarization control at higher energy levels) can be easily implemented in the form of a monolithic integrated circuit to provide a low cost, highly robust digital transmission device for a plurality of wireless, commercial and military applications. Numerous direct digital synthesizer (DDS) devices have appeared on the market, which are generally suitable to form the core of a practical, hybrid extended spectrum transmission system according to the present invention, although obviously none can simultaneously achieve all of them. the required functions. Perhaps the most commercially available device so far is the Analog Devices AD9854, which includes a continuous-phase RF frequency synthesizer, which can be raised in frequency or swept and / or modulated in downstream phase, as desired. The digitally modulated carrier phase data is internally converted to digitalized waves in parallel phase ("I") and quadrature phase ("Q"). Each is optionally passed through a reverse synchronized response FIR filter (to compensate for the frequency response deviations caused by the signal sampling / digitizing process) and a downstream amplitude control multiplier and finally converted to voltages analogue by a pair of high-speed digital-to-analog converters (DACs). Unfortunately, several features necessary to fully synthesize the HSS format are not included in the AD9854 device (or any other currently commercially available device); these include separate batches of phase modulators upstream of the amplitude I and Q phase converters [read-only memory blocks formatted in sine and cosine (ROM)]; the full-resolution high-amplitude modulators of four quadrants (for example, 14 bits or better) (multipliers) ahead of the DACs, or alternatively, the high-precision two-quadrant multiplication DAC structures; a parallel, wide, fast programming and data interconnection (eg, 32 bits or greater) to support high segment rates and / or skip / chirp velocities; and parameter concealment in fast segment, sufficient, to complement the parallel interconnection speed and thereby achieve reprogramming rates of fast, desired, internal synthesizer parameters (eg> 10 MHz). A moderately fast, practical HSS system implementation achievable with the currently available devices includes: an AD9854; a set of standard clock circuits; an external DSP control device, programmable gate arrays in high-speed field (FPGAs) for parallel data interconnection; two fast external downstream analog multipliers (for example, Analog Devices AD834s); and a final RF output signal combiner, to sum the modulated signals in channel 1 and Q coming from the multipliers. With a 300 MHz version of the AD9854 segment, RF output bandwidths of up to 120 MHz are achievable for upstream I / Q broadband conversion to a convenient RF transmission band; alternatively, a broad bandwidth surface acoustic wave (SAW) IF band pass filter (eg, a 36 MHz width at a center frequency of 70 MHz) can be employed with a frequency synthesized upconverter, separated , to produce the highest overall lapses required to cover the standard ISM of 2.45 and 5.7 GHz, and un-franchised RF transmission bands U-NII of 5.1 and 5.3 GHz. (For the narrowest ISM band of 902- 928 MHz, a simple fixed up converter could be replaced by the separate synthesizer stage). EXAMPLES The specific embodiments of the invention will now be further described by the following, non-limiting examples, which will serve to illustrate in some detail various features. The following examples are included to facilitate an understanding of the ways in which the invention can be practiced. It should be appreciated that the examples that follow represent discovered modalities to function well in the practice of the invention, and thus can be considered to constitute preferred embodiments for the practice of the invention. However, it should be appreciated that many changes can be made in the exemplary embodiments that are described, while obtaining the same or similar results without departing from the spirit and scope of the invention. Accordingly, the examples should not be considered as limiting the scope of the invention.

EXAMPLE 1 With reference to Figures 1-2, an extended frequency hybrid / direct sequence hybrid spectrum (FH / DS) scheme is described. An extended-frequency hybrid hybrid / direct sequence (FH / DS) transmitter described in Figure 1, is configured for the standard modulation of PBSK data. A subset of m bits from the full n bit PN code (n> m) is used in synchronization with the segment sequence to excite an RF 100 synthesizer, to generate the exact RF carrier frequency desired for transmission SS hybrid. Note that the higher-order, additional p-bits in the frequency control word of the RF synthesizer (DDS) are used to specify the RF operating band; the lower order bits from the PN register select the individual hop channels. The precise mapping of the jump frequencies in the RF 100 synthesizer can be handled directly via a ROM map or within an FPGA or equivalent device. The latest implementations are particularly useful when implementing fast frequency hopping or for very high data throughput and / or segment speeds. A described amplitude control 110 can be directly applied (as shown) to modulate or control the final RF output power of the transmitter; additionally or alternatively, oscillated amplitude data, obtained in general as k lines (parallel bits) from the PN generator, can be used to control power levels to orthogonal polarization antennas (for example, horizontal and vertical) ) to provide improved signal diversity in the RF environment, as also described in Figure 11. There are numerous possible implementations of this amplitude control, including (but not limited to) the classical high and low AM modulators. , the pulse width modulation (PWM) control of the supply voltage that feeds the final RF amplifier (s), the Class C or E amplifier stages summed by transformer, and so on. Alternatively, any variations in amplitude incidental to the output signal are often undesirable for many environments, and constant envelope modulation techniques have to be preferred where the highest efficiency of the RF energy amplifier is desired (eg, Classes C, D, E, F or S). In addition, in high multipath scenarios, a constant envelope transmitted HSS signal will allow for improved processing of the signal by the receiver, as explained later in the text describing Figure 12. The architecture of the spread spectrum receiver Hybrid FH / DS, corresponding is shown in Figure 2. Agui, the hybrid input signal in a front end low noise amplifier (LNA) 200, converted down to the desired intermediate frequency (IF) via a balanced mixer 210, amplified and filtered in bandpass in a bandpass amplifier block 220 (BPA), and finally demodulated in a conventional manner via a bit-matched filter, typical or correlator structure 230. As in the previous transmitter, the bits from the n-bit PN code generator (although typically shifted by a fixed number of those in the transmitter to provide the desired IF difference frequency) are used to select the desired hop channels within the selected RF receiving band. A fixed local oscillator signal is used to upwardly convert (preferably in a single-sided ["SSB"] image canceling mode) the output frequency of the synthesizer to the appropriate value to produce the target IF; this programmable, final local oscillator signal FH / DS is modulated by the polynomial code sequence PN, and applied to the balanced mixer 210. This extended or scattered signal is then (when properly synchronized) automatically despread or desdispersed the DS portion of the signal hybrid entry; the jump of the local signal also removes the component FH, leaving a single frequency signal modulated in data, simple, at the input to the chain 220 of the band pass IF amplifier. The synchronization for the system is derived from the data stream, the RF carrier frequency, or a combination thereof. An optional multipath amplitude detector block 240 can provide a relative measurement of the number of disturbances caused by the multiple paths in the received signal. This is typically implemented via the detection of variations in the envelope of the received signal (amplitude), which are used by the set of separate signal processing circuits to estimate the impairments induced by the multiple paths in the input signal and they perform at least one first-order cancellation of the same, to improve the final quality of the data at the receiver's output.

Example 2 With reference to Figures 3-4, an extended time hybrid / direct sequence hybrid spectrum (TH / DS) scheme is described. An extended time hybrid / direct sequence (TH / DS) extended spectrum transmitter is represented by Figure 3. Here, as in the TH / DS pre-transmission scheme, the n-bit 300 main PN code generator also provides a secondary group of m bits to an entry or pattern matching gate 310. When the selected m-bit pattern is recognized, a data gate control unit 330 introduces a burst of data bits within the disperser DS (via an exclusive gate 340 OR (O) that feeds a balanced modulator 350). Simultaneously, a bistable trigger ("T") 360, essentially excited by the coincidence output line, opens a final RF energy amplifier 370 for the prescribed interval (plus the small up and down energy times) to complete the burst transmission jumped over time. The architecture of the extended spectrum receiver TH / DS Hybrid, corresponding is shown in Figure 4. Agui, the TH / DS receiver is similar to the FH / DS receiver unit described at the beginning, except that the bits extracted from the master PN 400 generator excite a match detector 410 as one on the previous transmitter, instead of a frequency synthesizer. A bistable circuit "T" 420 in this case, once the system is synchronized, simply closes the RF input until the desired time lapse occurs, which minimizes the receiver's operational power requirements, as well as the effects of the interference signals not synchronized in time. As described above, an oscillator 430 is modulated by the synchronized, regenerated PN code. The IF signal produced at the output of a mixer stage 440, balanced at reception is downconverted and desdispersed; a data demodulator 450 in the form of bits, then directly extracts the regulated output current. An optional multipath amplitude detector block 460 may provide, as described above, a relative measurement of the number of disturbances caused by the multiple paths in the received signal.

Use 3 Referring now to Figures 5-6, a hybrid extended spectrum frequency hopping / time hopping / direct frequency (FH / TH / DS) scheme is described. The hybrid FH / TH / DS transmitter (Figure 5) and receiver (Figure 6) combine the FH / DS and TH / DS schemes to achieve more complex dispersion or extension distributions, and provide even greater data security than the simplest types; the details of the specific circuit follow those in the previous diagrams. These types of hybrid systems (FH / TH / DS) are probably the most useful for wireless sensor devices, in practical burst mode, for difficult industrial RF environments. Alternatively, the bursts may be synchronized in a periodic manner instead of a pseudo-random manner (thus, obviously, the elimination of the true TH modulation component) when the update rates of the system need to be highly regular, or when desired the standard periodic time lapse. Although these described transmitter implementations employ RF carrier frequency data modulation methods, finals, IF modulation can alternatively be efficiently used, particularly since many highly integrated DDS devices are available for popular IF frequency ranges (eg example, 70 MHz). Of course, the IF modulator in such systems will be followed by a frequency upconversion stage or subsystem to generate the final RF bearer. The dual-stage synthesizer architectures may be required by some DDS implementations of IF output, to provide the full RF output frequency range, when the available standard IF filter bandwidths are exceeded. It should be noted that the various transmitter and receiver block diagrams merely describe the functional arrangements to illustrate the concepts of signal processing; as such, they do not attempt to cover the full range of possible configurations under numerous aspects and / or implementations of the invention.

Example 4 With reference to Figure 7, a hybrid, programmable DDS transmitter capable of generating hybrid spread spectrum signals is described. A frequency accumulator 700 is electrically coupled to an adder 705 and to itself 700 (as feedback). A digital signal multiplexer 710 (MUX) is also electrically coupled to the add-on 705. The add-on 705 is electrically coupled to a phase accumulator 715. The phase accumulator 715 is electrically coupled to: i) itself 715 (as feedback); ii) a phase adder (I) 720; and iii) a phase adder (Q) 725. The phase adder (I) 720 is electrically coupled to a Seno 730 ROM of a phase-to-amplitude converter 731. The phase additive (Q) 725 is electrically coupled to a Cosine ROM 735 of the phase-to-amplitude converter 731. Still with reference to Figure 7, the Seno 730 ROM of the phase-to-amplitude converter 731 is electrically coupled to an amplitude multiplier 740 and a deflection circuit 780. The Cosine ROM 735 of the phase-to-amplitude converter 731 is electrically coupled to an amplitude multiplier 745 and to a bypass circuit 785. The bypass circuit 780 is electrically coupled to a MUX 750. The amplitude amplifier 740 is also electrically coupled to the MUX 750. The bypass circuit 785 is electrically coupled to a MUX 755. The amplitude amplifier 745 is also electrically coupled to the MUX 755. The MUX 750 is electrically coupled to a DAC (I) 760. The DAC (I) 760 is electrically coupled to an LPF 770. The LPF 770 is electrically coupled to an analog output (I) 790. The MUX 755 is electrically coupled to a DAC (Q) 765. The DAC (Q) 765 is electrically coupled to an LPF 775. The LPF 775 is electrically coupled to an analog output (Q) 795. Still with reference to Figure 7, a group of lines used to control the hybrid programmable DDS transmitter is also described. A delta frequency word line 701 is electrically coupled to the frequency accumulator 700. A delta frequency ramp rate clock line 702 is electrically coupled to the frequency accumulator 700. A group of words of frequency control lines 711 is electrically coupled to the MUX 710. A main clock line 716 DDS is electrically coupled to the phase accumulator 715. A phase control word line (I) 721 is electrically coupled to the phase additive (I) 720. A line 726 of The phase control word (Q) is electrically coupled to the phase additive (Q) 725. A line 741 of the amplitude control word (I) is electrically coupled to the amplitude multiplier 740. A word line 746 (Q) . amplitude control is electrically coupled to the amplitude amplifier 745. An amplitude control word on / off line (I1) is electrically coupled to the MUX 750. An amplitude control on / off (Q) 752 word it is electrically coupled to the MUX 755.

Example 5 With reference to Figure 8, a hybrid programmable DDS transmitter with external interconnection is described. The configuration is similar to that described for Figure 7, with the following modifications: the sine ROM 730 is electrically coupled to a FIR compensating filter 810 and a biasing circuit 830. The cosine ROM 735 is electrically coupled to a 820 filter The compensating FIR and the bypass circuit 840. The compensating filter 810 FIR is electrically coupled to a MUX 850. The bypass circuit 830 is also electrically coupled to the MUX 850. The compensating FIR filter 820 is electrically coupled to a MUX 860. The Deviation circuitry 840 is also electrically coupled to MUX 860. MUX 850 is electrically coupled to amplitude multiplier 740. MUX 860 is electrically coupled to amplitude multiplier 745. Still with reference to Figure 8, a group of lines used to control the hybrid programmable DDS transmitter, is similar to that described for Figure 7, with the following additions: a word (I) 811 for controlling the compensator is electrically coupled to the filter 810 of the compensating FIR. A word (Q) 821 of control of the compensator is electrically coupled to the filter 820 of compensating FIR. An on / off line 851 for compensating control is electrically coupled to the MUX 850. A line 852 on / off control (Q) of the compensator is electrically coupled to the MUX 860. A logical parallel interconnection 800 of processor External, high-speed, provides the lines used to program the transmitter described in Figure 8. The 800 parallel logic interconnection of the external, high-speed processor, essentially transfers the programming data of the device parameter from DSP or other controller to the respective hardware subsystems, respectively, to perform the desired specific modulation function (s). To overcome the speed limitations and / or the typical parallel interconnect pin number, some of the internal buffer and perhaps the decoding and / or synchronization logic (between the external interconnection and the synthesis blocks) is highly useful in helping to achieve high DS segment rates (dispersion or extension), frequency hopping / parasites, and even data modulation, particularly when more complex modulation formats are being implemented (such as the slower QAM constellations).

Additional features of the implementations of Figures 7-8 include the ability to generate the parasitic or composite DS / parasite modulations, with amplitude modulation capabilities downstream, via the front end frequency accumulator stage; to it, a sweep word rate and start frequency can be programmed within the frequency control circuitry. The next phase accumulator block with its phase-to-amplitude converter (sine / cosine ROMs) and the amplitude multiplier (modulator) will then send almost sinusoidal waves to the DACs of channel I and Q in the frequency band of the selected IF interval. In addition, the amplitude multipliers can alternatively (and preferably) be four-quadrant multipliers, complete, which in addition to reducing spectral spurious radiation will also allow variations of positive and negative amplitude (oscillations), and the option of purification of binary phase shift, downstream (which can be implemented by means of an inversion of simple amplitude or change of sign).

Example 6 With reference to Figure 9, a typical RF up-converter circuit is described, used in digital data transmitters. The RF up converter is used to convert the analog signal outputs in quadrature (I and Q) (such as from the systems as described in FIGS. 7 and 8) into RF signals, such as those employed to implement the present invention. A master clock 900 may be the same clock indicated in the previous figures or even a dedicated unit, which in conjunction with an RF frequency synthesizer 910 generates the final RF carrier frequency. Two identical carrier frequency signals feed a pair of balanced modulators 920, 930. The balanced modulator 920 mixes the RF carrier frequency signal with the analog intermediate frequency "I" (IF) output from the DDS circuit (described in the Figure 7 or in Figure 8). The balanced modulator 930 mixes the RF carrier frequency signal with the "Q" (IF) intermediate frequency output from the DDS circuit (described in Figure 7 and Figure 8). Still with reference to Figure 9, a pair of signals from the balanced modulators 920, 930 is linearly added in the sum block 940 to produce the quadrature-modulated RF dispersion data signal. A bandpass filter (BPF) 950 eliminates out-of-band energy (including RF harmonics and images) before feeding the signal on the linear RF amplifier 960. An amplitude control signal 970 derived from the system control hardware alters the RF output energy as desired, for power control, amplitude oscillation / modulation, or both. The final output signal then increases an antenna 980.

Example 7 With reference to Figure 10, an alternative but well-known "single sideband" or "image rejection" RF upconverter arrangement is described. The RF up converter is essentially identical to the circuit set of Figure 9, except that the RF frequency synthesizer 1000 provides two quadrature carrier signals 1010, 1020 (0o and -90 °) to the two balanced modulators 1030, 1040, which in turn are powered by the analog IF signals "I" and "Q", from the DDS circuitry (described in Figure 7 and Figure 8), respectively. Since the image frequency component largely bypassed at the 1050 output of the adder, a simple low pass filter (LPF) 1060 can be used in many cases to provide adequate RF spectral purity of the final transmitted signal. The implementations of the corresponding receiver systems, in spite of their fine complexity, can in general architecturally follow the examples of the previous figures.

Example 8 With reference to Figure 11, a typical dual polarization HSS transmitter implementation incorporates a master clock 1100 to generate all required synchronization signals from a common reference source, including the reference signal 1101 of the synthesizer ( typically 10 MHz) and the extended-spectrum segment clock frequency 1102, which drives the PN12 polynomial code generator 1112 of the host system. The code generator outputs a segment current 1114 of n-bit PN code, for signal DS to gate XOR 1115, which binary multiplies the segment current 1114 by the digital data input sequence at 1116. The resulting extended data stream BPSK at the XOR output 1117 is then fed to the balanced modulator 1130 so that the extended spectrum DS modulates the RF carrier frequency signal 1111. Meanwhile, generator 1112 PN, corresponding to the internal state of its internal binary polynomial registers, also outputs an m-bit word 1103, parallel to block 1110 RF frequency synthesizer, which in part controls the output frequency of RF at 1111. The rest of the digital frequency control data is sent from a collective bar 1104 of p-bit width from an external RF band selection data source. The input words at 1103 and 1104 fully specify the final, instantaneous RF carrier frequency generated by the synthesizer 1110, including the desired frequency hop pattern (usually pseudo random). The skipped signal that exists at 1111 is then multiplied by the DS component in the balanced modulator 1130 to generate the HSS signal of composite format FH / DS, at 1131. The unwanted harmonics and other spurious signals are eliminated by LPF 1132 and the signal is divided into two typically equal-level components, identical by the divisor 1133. The two output signals from the splitter (1134, 1135) each feed a gain-controlled RF amplifier (1136, 1137) and in turn a polarized antenna in a unique manner (1140, 1141) to transmit the signals vertically polarized (V) and horizontally polarized (H). A separate J-bit data word 1113 is extracted from the PN generator 1112 and feeds the amplitude control circuit 1120, which in turn digitally adjusts the output level of the RF amplifier 1136 according to the value of the word of J-bit via the signal 1123. The data of 'Jc-bit is also presented to the logical block 1122 of the subtractor, which simply calculates the difference between the word 1113 and a constant equal to 2k. This remaining value, which appears at 1121, represents the rest of the output energy and is input to the amplitude control block 1121. In turn, the control signal at 1124 alters the output of the RF amplifier 1137; in this way, the total combined energy emitted from the antennas V and H remains constant, but the relative proportions assigned to the polarized signals V and H will vary in a pseudo random manner according to the code word 1113 of k-bits. This oscillation of the polarization of the relative transmitted signal is another means of multipath migration and represents a highly useful, additional aspect of the present invention.

Example 9 With reference to Figure 12 (and after the basic implementation of Figure 2), a polarization diversity HSS receiver is described, typical according to one aspect of the invention. The respective polarized HSS signals V and H are received on antennas 1200, 1201 on the left, reinforced by LNAs 1202, 1203, and input to balanced mixers 1204, 1205 of down-conversion. The mixers are each fed with a local oscillator signal 1206 modulated by FH / DS, generated in block 1220 of balanced modulator. The two diversity input signals (V and H) are converted down to the desired intermediate frequency (IF) by means of balanced mixers, amplified, and bandpass filtered in the bandpass amplifier (BPA) blocks 1240, 1242, combined via the input, the weighted or unweighted sum, the maximum ratio combination, and other techniques in block 1250 , and finally demodulated in a conventional manner 1252 according to the type of data modulation employed (for example, BPSK, QPSK, MSK, etc.). The combiner 1250 may also employ a modified form of the feedback directed by the majority logic, through the demodulator 1252 to optimally combine the input signals V and H, separated; alternatively, the demodulator could via two separate channels, immediately decode the data signals V and H before developing the optimized result. Of course, the use of diversity antennas should not be considered in the invention to be constrained to two (V and H), but could easily include additional received channels of both polarizations, to provide spatially diverse reception capacity, for the improved signal reception efficiency, such as current CDMA cellular telephone systems, which utilize multi-finger "tilt" receivers to deal with multipath signals. Conversely, the use of the signal polarizations V and H in the HSS receiver could certainly include combination schemes similar to those of the "tilt" architecture. An additional option in the HSS receiver would be to use a combiner specifically optimized to manipulate the constant envelope signals produced by a specific variety of the HSS transmitter. Since the incidental multipath reflections invariably cause spurious amplitude (AM) modulations in the received signals (although the transmitted signal was nominally of constant amplitude), the detection of these amplitude variations can be explicitly used to trigger corrective circuits that serve to minimize the incidental AM, and thereby reduce the waveform dispersion of received signal caused by the multipath. As in the previous transmitter, the m bits coming from the n-bit PN code generator (although it is typically displaced by a fixed amount of those at the transmitter, to provide the desired IF difference frequency) are used to select the desired hop channels within the selected RF reception band. This is achieved by the operation of a PN 1230 code generator, master essentially identical to that in the HSS transmitter of Figure 11, and extracting a collective m-bit parallel data bus (corresponding to the internal registers of the PN generator ) to excite the RF frequency synthesizer block 1212, with the jump component of the complete code. The serial segment code stream is sent at 1231; it feeds an input of the balanced modulating block DS 1220. The output of the synthesizer at 1213 is mixed at 1214 with the output of a local oscillator 1210, typically fixed, which operates at a shift (IF) of the central frequency of RF input , nominal, to convert the synthesizer up to the desired RF band. The bandpass filter (BPF) 1215 removes the unwanted image component; the filtered output at 1216 is then applied to the balanced modulator 1220 to generate the local oscillator signal modulated in HSS desired (FH / DS) at 1206. This locally scattered or extended signal (when suitably synchronized) then automatically disperse the DS portion of the hybrid input signal; the jump of the local signal also eliminates the FH component, leaving a simple frequency signal, modulated in simple data, at the input to the bandpass amplifier chains 1240, 1242. The synchronization for the system 1254 is derived from the data stream, the RF carrier frequency, or a combination thereof. The synchronization of the Master Clock 1259, the Generator PN 1230, and the Local Oscillator 1210 are achieved through the feedback signals 1255, 1256 and 1258, respectively. An optional multipath amplifier detector block 1260, fed from the IF V and H output 1241, 1243 (via lines 1244, 1245) can provide a relative measurement of the number of perturbations caused by multiple paths in the received signal . This is typically implemented via the detection of variations in the envelope of the received signal (amplitude), which are used by the set of separate signal processing circuits, to estimate the impairments induced by multiple paths in the input signal, and they perform at least one first-order cancellation of the same to improve the final quality of the data at the receiver's output. The final processed output of this multipath estimation block is shown at 1261. An additional feature of the multipath detector is that the dynamic estimates of the received signal quality obtained there may be used not only to adaptively improve the reception of the signal. the input data stream, but also (in an HSS transceiver scenario) to trigger the dynamic optimization of the output HSS transmit signal format (for example, DS, FH, TH and / or polarization parameters) to help Counter the static and dynamic changes in the RF channel environment.

Practical Application of the Invention Practical applications of the invention that have value within the technological field include: wireless sensors and data networks in industrial plants, offices, commercial buildings and warehouses; environmental monitoring systems; Container tracking applications / active / personal and telemetry; wireless local area networks (WLANs); medical telemetry; battlefield / tactical sensors; and secure data transmission for industrial, military, and national security applications, all equally valid for domestic and foreign markets. There are virtually innumerable uses for the invention, all of which do not need to be detailed to you.

Advantages of the invention A hybrid spread spectrum system, which represents one embodiment of the invention, may be inexpensive and advantageous for some of the following reasons. The invention improves the quality and / or reduces the costs compared to the previous procedures. Other advantages may include: superior multipath rejection capabilities, improved data integrity / security, better low probability-of-detection / probability-of-interception (LPD / LPI) probabilities, minor link delay (latent) figures, superior narrow / wide band disturbance resistance, fast synchronization, higher user density, less manual interference between users in a given area or frequency band, increased diversity of statistical signals (for better robustness), superior properties of near reception / FH anus (a major inconvenience of pure DS systems), and at lower occupied bandwidths (fewer spinal "radiations"), compared to most extended-spectrum axial techniques. An additional key use of the HSS protocol is in the combined (or separate) reader / interrogator units to extract data from passive, semi-passive, semi-active, or fully active tags, commonly referred to as RFID tags. The use of the HSS protocols in the tag test units, which typically combine interrogating (illuminating) the RF transmitter and associated receiver devices, will provide higher tag reading efficiencies than standard systems based on DS or FH, due to the greater diversity of signals of the HSS emissions. While virtually all existing tag vectors have null or "dead" points in their RF tag interrogation fields, the HSS signal constantly with statistically release will have much less signal zeros and with this will show a higher efficiency reading efficiency of RF, with less "no readings" of tag. A greater appreciation of the benefits of the hybrid extended spectrum emission signaling technique can be had by examining the typical indoor RF delay profile at 915 MHz of Figure 13. In this environment, the dispersion times of signals are in the range of about 10 to 250 ns, with an average RMS scattering value of about 46 ns. Longer values, corresponding to larger spaces or enclosed areas typical of the manufacturing facilities of industries, could imply a minimum dispersion rate speed of approximately 4 Msegments / seconds, while shorter values (in all cases) found in offices or in smaller rooms explain the dispersion speeds of up to approximately 100 Msegment / second (and thus, at least 100 MHz of the signal bandwidth transmitted in a total manner). Obviously, this latter requirement exceeds the available bandwidth on either the 915 or 2450 MHz of the ISM bands, and requires the full width of any of the three ISM / U-NII bands of 5 GHz, allocated for the transmissions of extended spectrum, not franchised in the United States. Even in the latter case, to operate multiple full band devices in the same space, careful coordination of multiple users would be required (via multiplexing over time or multiple access techniques or division of controlled code into energy [CDMA ]) to prevent significantly large numbers of collisions and a consequential loss of data packets. Compared to conventional direct sequence or frequency hopping extended spectrum schemes, the hybrid technique of the invention offers improved process gain , disturbance margin and multiple access capabilities. In addition, the hybrid technique offers the advantages of the relative freedom of near-far effects of FH compared to conventional DS. Since in the hybrid system the DS component can be of smaller bandwidth for comparable total performance, the IF front endband censorship of the hybrid receiver, can be significantly smaller (and thus, proceed higher selectivity) than in the standard DS implementation, which offers greater rejection based on filter signals adjacent channel, out of band and spurious. In this way, the much higher amplitudes of a nearby transmitter but out of channel, will not cause the blocking of signal signals in the desired, weaker channel (more distant), typical of conventional DS systems. From chapter 2 of the standard text Spread Spectrum Systems wi th Commercial Applications, 3 -, Edition, by Robert C. Dixon, John Wiley & Sons, Inc. 1994, pp. 18-58, the classical equation for the generalized process gain of a standard extended spectrum signal (direct sequence or frequency hop with non-overlapping, non-contiguous channels): Gp = BWRF + Rinfo where Gp is the effective processing gain, BWRF is the bandwidth of the full-spectrum (two-sided) RF signal, and Rinfo is the modulation data rate (pre-extension) or the bandwidth of information, also in the total modulated representation (of two sides). For standard direct sequence systems, the gain is generally the extension code length; in the case of simple frequency hopping systems, the processing gain (long-term average) for contiguous or non-contiguous channel sets is simply equal to the total number of hop channels. If the DS and FH methods are used concurrently, assuming that the bandwidth of the DS signal is small compared to the width of the total available RF band, so that there is a reasonably large number of salt to hybrid channels, The complete hybrid signal process gain is simply the product of the two process gains of the two individual process gains: Gp (FH / DS) = Gp (FH) X Gp (DS) In decibels, the equation becomes: GP (FH / DS) dB = GP (FH) dB + GP (DS) dB = 10 log (number of jump channels) where GP (FH / DS) is the hybrid extended spectrum process gain, GP (FH) is the frequency hopping gain, and GP (DS) is the direct direct sequence gain. At the theoretical limit, for a fixed available bandwidth (eg, 26 MHz for the ISM band of 902-928 MHz) and non-overlapping FH channels, the composite process gain for hybrid DS / FH system can not access the ratio of the total bandwidth to the information rate of proportion. For example, if someone wishes to send a standard data rate of 19.2 kb / second using a simple DS signal that occupies the entire band, then the maximum bandwidth process gain achieved using the QPSK modulation would be 26 x 106 + 19.2 x 103 _ = 1354 _ 31.3 dB. For full-band noise, interference or disturbance, then hybrid DS / FH techniques using groups of channels are overlapped, theoretically they will no longer provide process gain than for the simple DS signal, but in practice it will still be higher than Full-band DS format in the rejection of errors induced by multiple trajectories, in the resolution of effects of near / far interference, and in allowing multiple signals to be sent simultaneously within the confines of the selected band. [However, if the FH channels are overlapped by 50% (for example, by half the DS comminution rate), then the system DS / FH can effectively achieve approximately double (+3 dB) of the process gain of the direct DS version].

In addition, compared to conventional DS implementations, hybrid SS systems will provide equivalent or better link performance, with composite, lower segmentation speeds, slower hardware processing speeds, and overall reduced power consumption of the transmission device in the receptor. Concatenating the time jump coordinated with hybrid DS / FH methods adds several additional modes of operationally enhanced flexibility. In the usual sense, the jump in time itself does not provide process gain, assuming that the receiver's input is closed during the non-transmission intervals; rather, an effective gain or margin of interference for continuous interference signals, is provided in a manner directly proportional to the duty cycle of the signal, for example, the ratio of the transmitter (or receiver) "in" times to the total interval under consideration. Correspondingly, the effective "process gain" for intermittently interference signals (statistically) simply the product of the work cycles of the two signals. The main advantage in adding coordinated or synchronized time hopping in code is improved system performance with multiple devices in a shared RF environment (the "multiple access" scenario.) For example, if there are several hundred sensor devices in a common area (for example, a factory complex, an elevated office building, a boarding terminal yard, etc.) that will all need to send intermittent telemetry data to a central receiver, the use of DS alone with random burst transmissions can only succeed if the product of the number of devices and their RF burst lengths is relatively small compared to unity; for example, the work factor of the system is «100%. Otherwise, a large number of collisions will occur and many data will be lost; frequent transmissions will improve the average transmission reliability of the system, but at the expense of the speed of other significantly lower aggregates and wasted energy of the device. The use of code division multiple access (CDMA) techniques can solve most collision problems, but invariably adds significant complexity (and cost) to the system by the addition of larger infrastructure components (eg, base stations). ) and also forcing the addition of RF receivers, encoders, and complex control hardware and algorithms to close the required RF link power control loop, in each device. The massive complexity of the CDMA procedure is generally low cost only in the mass consumer cell phone market, due to the clear numbers (tens of millions) of units deployed; for general low-power RF telemetry applications (battery-operated), especially those that include simple, cheap, simple sensor and labeling devices, the DS / CDMA general procedure is neither low-energy nor low-cost. Another more intuitive explanation of the advantages of the combined FH / TH / DS signaling method of the invention is shown below. Assume that two independent sensor / telemetry devices have to transmit their data in a common environment (eg, a large factory) at the same time, using the FH / TH / DS technique. Assuming that both are in the same group and have the same DS extension code assigned, each is programmed to select a different portion of that code, to excite its FH and TH circuit board; in this way, each one (statistically) will transmit different frequencies and at different times. Even if the two perform the transmission simultaneously, statistically, they will be at different frequencies; if they use the same jump frequencies, statistically they will use them at different times. Assuming negligible noise levels, the effective probability of a data error due to a collision between two hybrid FH / TH / DS signals can be calculated as the product of: (1) the probability of a collision in the time domain [ example, the product of the randomized individual work cycles of the two devices]; (2) the probability of a frequency collision [approximately the product of the inverse of the lengths of the individual FH control sequences, assuming non-overlapped jump channels]; and (3) the ratio of the cross-correlation energy of the two codes, assuming almost equal signal energies in the receiver, as in the case of energy-controlled CDMA systems. If unequal energies exist, the third term [the separation ratio DS] will be degraded to some extent, by approximately 1 dB or 2.5 dB from the energy deviation received from the ideal (eg equal); in this way, the probability of effective error ratio due to collision of the two signals will be: Pe - Pee = ct * Pcf * Rec where the probability Pe of total bit error, in the absence of significant background noise and multipath effects, is approximately equal to Pec, the error due to collisions alone (for example, a variant of multiple access interference) classic dominant in CDMA cellular telephone systems); this in turn is simply the product of the respective collision rates or ratios, in time and frequency, given by Pc and PCf, with the energy separation rate of effective code Rcc (cross product of energy versus self proportion) of the code DS selected versus its company codes (for example, approximately 24 dB for a Gold code increased by a length of 1024, of order 10, assumed). In usual HSS application, asynchronous code groups such as MLS, Gold and Kasami are preferred; the Walsh codes used in the CDMA cellular telephone systems require completely synchronous (coherent) acceptance to achieve usefully low levels of cross-code formulation, and thus, demand significantly greater system complexity. For example, if the complete polynomial length of each signal is 1024 = 210, and the signal length FH is 64 = 26 and the work factor TH of both is 1/16, then the approximate probability averaged over time of a collision Simple jump between the two HSS transmissions is given by: -2.4, (1/16) • (1/64) • (1/8) »(1/64) * Rc (1/1024)» (1/512) • (10- ^ 4) 7.58 x 10" assuming good energy control the device. Obviously, in code separation it provides adequate isolation even if both devices transmit at the same time (or partially overlapping times) and over the same frequency. [The overlapping time lapse scenario gives two chances for the collision, thus causing a duplication of the nominal factor TH 1/16 for the second device]. For multi-hop transmissions, the expressions quickly become very complex and the detection mode must also be included (eg, standard, majority logic, soft logic, etc.); for a complete system calculation, all user parameters (bit rates, segmentation rates, hopping speeds, bandwidths, extension code correlation characteristics, energy levels in the device, modulation / demodulation methods, etc.) must also be considered. For the case of multiple users, different device, the HSS system will exceed the usual operating levels for DS-CDMA energy controller systems, standards, as described by the standard equation for the probability of multiple access bit errors : Pe (DS) = Y2 erfe [2 / (k-l) * (fb / fc)] 1/2, where fb and fc are the bit and segment rates, respectively. The corresponding equation for the HSS case will also include the terms for the FH access statistics. { PCf) and TH (Pcfc), which are highly dependent on system-specific parameters but will generally decrease the Pe figure by at least an order of magnitude (often more): Pe (Hss) = V * erfe [2 / (k-1) »(fb / fc)] 1/2» PCf »Pct The disturbance margin for the hybrid system is similarly complex but is comparably still greater than that of its standard DS and FH counterparts; in general, the disturbance ratio is simply the gain of the spread spectrum process minus the implementation losses, and the required signal to noise ratio, so that the HSS case is invariably more robust for the usual forms of interference or disturbance . As previously explained, the fast-forward form of the HSS waveform (certain modalities of which are referred to as "FastHSS") may include the specific combination related to the code, of the direct-frequency extended spectrum (DS) modulation standard, with "fast" frequency hopping (FFH), where multiple frequency jumps occur within a single data bit time. Again, the most significant benefit of this waveform is that each bit is represented by segment transmissions at multiple frequencies. If one or more segments (or jumps) are corrupted by multiple trajectories or by interference on the RF link, a majority will be statistically correct. In addition, specifically, for the fast-jump form of the HSS waveform, the improved bit error ratio is achieved in the receiver by comparing the bit-length cross-correlation functions with the cross correlations of bit lengths. sub-bits on a bit by bit basis and using any value that is optimal (for example, it has a higher level of correlation). For example, if an HSS signal has a dispersion or extension length of N segment / bit and H jumps / bit, then there are (N - ^ H) segments / hop. Thus, if one of the jumps experiences interference due to multiple trajectories or perturbation, then the other (N + H) segments are still available for the establishment of the value of that bit. In fact, if each group of xn = (N -T- H) segments, where [n = 1, 2 ... H] can be declared valid versus invalid, and only the valid groups used for cross-correlation, then the The ability of the receiver to correctly decode that particular bit is increased to that extent. Typically, DSSS receivers do a segment by segment determination of a carrier (assuming PSK modulation) or they use the multi-level logic (accumulation of the correlation sum) and use a pseudo bit length integration.

By using FastHSS, the receiver can dynamically optimize (bit by bit) this exchange or barter, while DSSS systems of the existing technique or FH / DS system that jump slower than once per bit, can not use this advantage. Therefore, during scanning, each group of xn segments are cross-correlated with the corresponding portion of the stored PN code. In parallel, the receiver logic performs a standard bit auto-correlation of the entire segment sequence. If the standard auto-correlation function exceeds the detection threshold, sufficiently to recognize a valid data bit result, then the cross-correlation functions of subgroups are ignored. However, if the value of the bit-length auto-correlation is insufficient for the detection of unambiguous data bits, then the cross-correlation values of the subgroup H are evaluated. If the threshold for the bit-length correlation is assumed as a • N, then the cross-correlation thresholds of subgroups would be adjusted to, a • (N -s-H). If i of the cross-correlation values of subgroups are considered sufficiently unambiguous, then those subgroups are concatenated and use a threshold of i • a • (N H), where i < H. Analytically, let Si = cos [(ct >? + H ??) t] • pi? (T) d? At) represent the received signal, where di '(t) represents the data encoded by the source , m? (t) represents the extension or dispersion code and eos [(Cúi + h ??) t] is the hopped carrier signal. Letting the received received waveform be represented as S2 = Si + I + N0, where I represents the sources of interference and does not represent the additive white Gaussian noise (AWGN). In this way, the autorelation function for the complete bit is: where To is the bit period and S? (t) are periodic, and the autocorrelation function used by the subgroups is: In the case where there is no noise or interference, R (t) = Rsubgroups (T), and these sufficiently surpass an unambiguous threshold. However, in cases where at least one of the H frequencies is experiencing interference, one or both of these equations may not meet the requirements for the detection of ambiguous bits. In this case, the cross-correlation functions of individual subgroups are evaluated to determine which meet the criteria: If any of R? (T) meets this criterion and all ambiguous Ri (t) are removed, then the bit can be unambiguously detected by the calculation of: 1? h + J'To / H JS ,, (/) Sa (/ + r) _- 'o- J-iW "where there are J ambiguous subgroups and j = b ± designate the individual ambiguous subgroups.

A more probabilistic, simpler procedure for a more optimal detection of the FastHSS segments involves the use of the majority logic, in which for the H bit jumps, they only need to be correct, slightly more than half. For odd H (for example, H = 2n - 1), the minimum number of correct jumps (or groups of segments) is 2n_1; for H pair (for example, H = 2n), the minimum number required is 2n_1 + 1. If you designate the number of total jumps (groups of segments) that represent a single bit like H, then the total probability of bit error FastHss for difficult binary decision detection, can be expressed as: where Pe is the total bit error probability, H is the total number of jumps or groups of segments per bit, r is the minimum number of correct values per bit (for the decision of the majority logic), C is the symbol for the probabilistic combination (of H things taken x at a time), p is the probability of the sample error (jump) and x is the sum index. For example, if three jumps are used for one bit of data, then two sample decisions (jump) must be correct to avoid an error. In this way, for a p = 10 ~ 2 basic, Pe is 2.98 x 10"4; similarly, for p = 10-3, Pe = 2,998 x 10"6. If instead we use 3 of 5 jumps for the decision, for a p of 10" 3 we get Pe ~ 9.8 x 10 ~ 4 and for p = 10"3, Pe becomes approximately 9.8 x 10" 7. Clearly, the use of multiple jumps and majority decision logic can produce significant performance improvements over existing bit detection procedures. An even greater advantage (> 2 dB) can be obtained in this case by the use of soft decision techniques (multiple levels), well known in bit detection schemes, of maximum probability such as Viterbi decoders. The terms one or one, as used herein, are defined as one or more than one. The term plurality, as used herein, is defined as two or more than two. The term "other", as used herein, is defined at least one second or more. The terms "comprising" (understood, understood), "including" (includes, included) and / or "having" (have, had), as used herein, are defined as open language (eg, which requires what is indicated below, but open to the inclusion of one or several procedures, one or several structures and / or one or several ingredients not specified, include larger quantities .The terms "consisting" (consists, consisted) and / or "comprising" (compounds, compounds), as used herein, close the method, apparatus or composition indicated to the inclusion of procedures, structure (s) and / or ingredient (s) other than those indicated, except by auxiliaries, adjuncts and / or impurities ordinarily associated with them The citation of the term "essentially" together with the terms "consisting" or "comprising" makes the indicated method, the apparatus and / or the unique open composition for inclusion of one or several procedures (s), structure (s), and / or ingredient (s) not specified, which do not materially affect the basic novel characteristics of the composition. The term "coupled", as used herein, is defined as connected although not necessarily directly, and not necessarily mechanically. The term "any" as used herein is defined as all applicable members of a group or at least a subgroup of all applicable members of the group. The term "approximately", as used herein, is defined as at least close to a given value (eg, preferably within 10% of, more preferably within 1% of, and most preferably within 0.1% of from). The term "substantially", as used herein, is broadly but not necessarily fully defined as it is specified. The term "in general" or "generally", as used herein, is defined as at least approaching a given state. The term "deployment" as used herein is defined as the design, construction, shipping, installation and / or operation. The term "means", as used herein, is defined as physical equipment (hardware), microprogramming (firmware) and / or computer software (software) to achieve a result. The term "program" or the phrase "computer program" as used herein is defined as a sequence of instructions designed for execution in a computer system. A program or computer program, can include a subroutine, a function, a procedure, an objective method, a goal implementation, an executable application, a subroutine, a server program (servlet), a source code, an objective code, a shared library / dynamic load library and / or other sequence of instructions designed for execution or in a computer or computer system. All the described embodiments of the invention, detailed herein, can be made and used without undue experimentation in the light of the description. Although the best mode of carrying out the invention contemplated by the inventor or inventors is described, the practice of the invention is not limited thereto. Accordingly, it will be appreciated by those skilled in the art that the invention may be practiced otherwise than as specifically described herein. In addition, the individual components do not need to be combined in the described configurations, but could be formulated in virtually any configuration. In addition, a variation may be made in the steps or in the sequence of steps that make up the methods described herein. Furthermore, although the programmable, hybrid, DDS transmitter system described herein may be a separate module, it will be apparent that the programmable, hybrid DDS transmitter system can be integrated into the system with which it (or they) are associated. In addition, all the described elements and features of each detailed embodiment can be combined with, or replaced by, the elements described and the features of each other modality described, except where such elements or features are mutually exclusive. It will be apparent that various substitutions, modifications, additions and / or rearrangements of the features of the invention can be made without deviating from the spirit and / or scope of the underlying inventive concept. The spirit and / or scope of the underlying inventive concept as defined by the appended claims, and their equivalents, are considered to cover all such substitutions, modifications, additions and / or rearrangements. The appended claims should not be construed as including the limitations of means plus function, unless such limitation is explicitly indicated in a given claim using the phrase "means for" and / or "step for". The subgeneric embodiments of the invention are delineated by the appended independent claims and their equivalents. Specific embodiments of the invention are differentiated by the appended dependent claims and their equivalents.

It is noted that in relation to this date, the best method known to the applicant to carry out the aforementioned invention is that which is clear from the present description of the invention.

Claims (67)

1. CLAIMS Having described the invention as above, the content of the following claims is claimed as property: 1. A method, characterized in that it comprises the modulation of a signal by using a subset of bits from a pseudo-random code generator, to control an amplification circuit that provides a gain for the signal.
2. 2. The method of compliance with the claim 1, characterized in that the signal includes an extended spectrum signal.
3. 3. The method of compliance with the claim 2, characterized in that the extended spectrum signal includes an extended spectrum signal of the direct sequence.
4. 4. The method according to claim 1, characterized in that it also comprises the frequency hopping of the signal.
5. 5. The method according to claim 4, characterized in that multiple frequency jumps occur within a single data bit time.
6. 6. The method according to claim 4, characterized in that the frequency hopping includes the frequency sweep.
7. 7. The method according to claim 1, characterized in that it also comprises the time jump of the signal.
8. The method according to claim 1, characterized in that the modulation of the signal includes the amplitude vibration of the signal.
9. The method according to claim 1, characterized in that it further comprises the modulation of a polarization of the signal.
10. The method according to claim 9, characterized in that the modulation of the polarization of the signal includes the control of the power levels of the power supply, to the orthogonal polarization antennas.
11. The method according to claim 1, characterized in that it further comprises transmitting the signal to a radio frequency label and receiving a transformed version of the signal from the radio frequency label.
12. 12. A computer program, characterized in that it comprises the elements of the program readable in computer or machine translatable to implement the method according to claim 1.
13. 13. The method according to claim 1, characterized in that it also comprises the transmission of the signal .
14. 14. An electronic means, characterized in that it comprises a program for carrying out the method according to claim 1.
15. 15. An apparatus, characterized in that it comprises: a pseudo-random code generator; and an amplitude controller coupled to the pseudo-random code generator.
16. 16. The apparatus according to claim 15, characterized in that it comprises an amplification circuit coupled to the amplitude controller.
17. 17. The apparatus according to claim 15, characterized in that it further comprises a signal attenuator circuit coupled to the amplitude controller.
18. 18. The apparatus according to claim 15, characterized in that it further comprises a matching gate coupled to the pseudo-random code generator and a switch coupled between the matching gate and the amplification circuit.
19. The apparatus according to claim 15, characterized in that it further comprises a fast-hop frequency synthesizer coupled to the pseudo-random code generator.
20. 20. A method, characterized in that it comprises: directly synthesizing a digital signal that includes: the amplitude modulation of an output channel, with an amplitude multiplier of four quadrants, to provide a signal oscillated in amplitude.
21. 21. The method according to claim 20, further characterized in that it comprises the transformation of a channel with an equalizer or compensator.
22. 22. The method according to claim 21, further characterized in that it comprises the programming of the compensator.
23. 23. The method according to claim 20, characterized in that it further comprises the transmission of the digital signal to a radio frequency label and the reception of a transformed version of the digital signal from the radio frequency label.
24. 24. An apparatus, characterized in that it comprises a direct digital synthesizer that includes an amplitude multiplier of four quadrants, wherein the direct digital synthesizer provides a signal oscillated in amplitude.
25. 25. The apparatus according to claim 24, characterized in that it further comprises a first compensator applied to the amplitude multiplier of four quadrants.
26. 26. The apparatus according to claim 25, characterized in that the compensator includes a finite impulse response filter.
27. 27. The apparatus according to claim 26, characterized in that the finite impulse response filter is programmable.
28. 28. The apparatus according to claim 27, characterized in that it further comprises a logical interconnection coupled to the integrated circuit.
29. 29. The apparatus according to claim 28, characterized in that the logical interconnection includes a programmable gate array in the field.
30. 30. An integrated circuit, characterized in that it comprises the apparatus according to claim 24.
31. 31. The integrated circuit according to claim 30, further characterized in that it comprises a coupled receiver.
32. 32. A system, characterized in that it comprises the integrated circuit according to claim 31 and a radio frequency label.
33. A method, characterized in that it comprises: directly synthesizing a digital signal that includes: the amplitude modulation of a channel in phase, with a first amplitude amplitude of four quadrants; and the amplitude modulation of a channel in quadrature phase with a second amplitude multiplier of four quadrants.
34. 34. The method of compliance with the claim 33, characterized in that it further comprises: the transformation of the channel in phase with a first compensator; and the transformation of the channel in the quadrature phase, with a second compensator.
35. 35. The method of compliance with the claim 34, characterized in that it further comprises: programming the first compensator; and the programming of the second compensator.
36. 36. The method according to claim 33, characterized in that it further comprises the transmission of the digital signal to a radio frequency label and the reception of a transformed version of the digital signal coming from the radio frequency label.
37. 37. An apparatus, characterized in that it comprises a direct digital synthesizer that includes a first multiplier of amplitude of four quadrants, and a second multiplier of amplitude of four quadrants, wherein the first multiplier of amplitude of four quadrants and the second multiplier of amplitude of four quadrants are coupled together in parallel.
38. 38. The apparatus according to claim 37, further characterized by comprising a first compensator coupled to the first four-quadrant amplitude multiplier, and a second compensator coupled to the second four-quadrant amplitude multiplier.
39. 39. The apparatus according to claim 38, characterized in that the first compensator includes a first finite impulse response filter, and a second compensator includes a second finite impulse response filter.
40. 40. The apparatus according to claim 39, characterized in that the first finite impulse response filter is programmable, and the second finite impulse response filter is programmable.
41. 41. The apparatus according to claim 37, further characterized in that it comprises a logic interface coupled to the integrated circuit.
42. 42. The apparatus according to claim 41, characterized in that the logical interconnection includes a programmable gate array in the field.
43. 43. An integrated circuit, characterized in that it comprises the apparatus according to claim 37.
44. 44. The integrated circuit according to claim 43, characterized in that it also comprises a coupled receiver.
45. 45. A system, characterized in that it comprises the integrated circuit according to claim 44, and a radio frequency label.
46. 46. A circuit board, characterized in that it comprises the integrated circuit according to claim 43.
47. 47. A transmitter, characterized in that it comprises the circuit board according to claim 46.
48. 48. A broadcasting network, characterized in that it comprises a transmitter in accordance with the claim 47.
49. 49. A method, characterized in that it comprises: modulating a signal by using a subset of bits from a pseudo-random code generator, to control a fast-jump frequency synthesizer; and the rapid jump of frequency of the signal with the fast jump frequency synthesizer, where the multiple frequency jumps occur within a single data bit time.
50. 50. The method according to claim 49, characterized in that the signal includes an extended spectrum signal.
51. 51. The method according to claim 50, characterized in that the extended spectrum signal includes a direct sequence spread spectrum signal.
52. 52. The method according to claim 49, characterized in that it also comprises the time jump of the signal.
53. 53. The method of compliance with the claim 49, characterized in that the frequency synthesizer of the fast hop provides a substantially constant envelope signal.
54. 54. The method according to claim 49, characterized in that the fast jump includes the frequency sweep.
55. 55. The method according to claim 49, characterized in that the modulation of the signal includes the oscillation in amplitude of the signal.
56. 56. The method of compliance with the claim 49, characterized in that it further comprises the modulation of a polarization of the signal.
57. 57. The method according to claim 56, characterized in that the modulation of the polarization of the signal includes the control of the power levels of power to the orthogonal polarization antennas.
58. 58. The method according to claim 49, characterized in that it further comprises the transmission of the signal to a radio frequency label, and the reception of a transformed version of the signal coming from the radio frequency label.
59. 59. A computer program, characterized in that it comprises the elements of the program readable in computer or machine translatable to increase the method according to claim 49.
60. The method according to claim 49, characterized in that it also comprises the transmission of the signal .
61. 61. An electronic means, characterized in that it comprises a program for carrying out the method according to claim 49.
62. 62. An apparatus, characterized in that it comprises: a pseudo-random code generator; and a fast hop frequency synthesizer, coupled to the pseudo-random code generator, wherein the multiple frequency jumps extend within a single data bit time.
63. 63. The apparatus according to claim 62, characterized in that it further comprises an amplitude controller coupled to the pseudo-random code generator.
64. 64. The apparatus according to claim 63, characterized in that it further comprises an amplification circuit "coupled to the amplitude controller.
65. 65. The apparatus according to claim 63, characterized in that it further comprises a signal attenuator circuit coupled to the amplitude controller.
66. 66. The apparatus according to claim 62, characterized in that the fast hop frequency synthesizer provides a substantially constant envelope signal.
67. 67. The apparatus according to claim 62, characterized in that it further comprises a matching gate coupled to the pseudo-random code generator, and a switch coupled between the matching gate and the fast-jump frequency synthesizer.