KR800000702B1 - Television signal processing apparatus incuding a transversal equalizer - Google Patents

Abstract

Appts. for improving the sharpness or crispness of the image generated by a TV receiver is disclosed. Signal delaying means, included in the luminance channel of a color TV receiver, are responsive to the TV video signal processed in the receiver. A portion of the signal delaying means serves to equalize the time delays of signals processed in the chrominance and the luminance channels of the receiver. A plurality of differently delayer video signals are developed at signal coupling means associated with the signal delaying means.

Description

TV signal processing unit

1 is a schematic configuration diagram of a color television receiver using an apparatus according to the present invention.

2 is a schematic configuration diagram of an embodiment according to the present invention

FIG. 3 shows frequency domain waveforms useful for understanding the circuit operation of FIG. 2. FIG.

4 shows a time domain waveform useful for understanding the second circuit operating rate.

5 is a schematic structural diagram of still another embodiment of the present invention

FIG. 6 shows frequency domain waveforms useful for understanding the operation of the present invention embodiment shown in FIG. 5. FIG.

FIG. 7 illustrates a time domain waveform useful for understanding the operation of the embodiment of the present invention, which is continually shown in FIG.

8 is a schematic diagram of the circuit performance of the embodiment of the present invention shown in FIG.

9 is a schematic structural diagram of still another embodiment of the present invention

10 is a schematic structural diagram of still another embodiment of the present invention

11 and 12 illustrate frequency domain waveforms useful for understanding the operation of the embodiment of FIG.

13 is a schematic structural diagram of still another embodiment of the present invention

14 and 15 show waveforms in the frequency domain useful for understanding the operation of the FIG. 13 embodiment.

The present invention relates to an apparatus for improving the instantaneous response of a television video signal processing apparatus, and more particularly to an apparatus for improving the sharpness of an image generated by a television.

The emergence of large kinescopes for television receivers has increased the importance of improving the instantaneous response of television video signal processing devices. This device is used to improve the detailed reproduction as well as the transition between voices in image quality.

It is known that the response of a video processing device can be improved primarily by increasing the amplitude transition sharpness in the video signal. The response will be approximated by generating a freshoot immediately before the transition and an overshoot immediately after the transition, ie the transition from white to black becomes prominent because the image just before the transition is normal. White is stronger than that of Esau, and the image immediately after the transition is stronger than that of the normal picture.

It is known that the sharpness of the amplitude transition is a preferential function of the high frequency response of the video signal processing apparatus. Therefore, it is necessary for the video processing apparatus to have a relatively wide bandwidth. However, as a receiver with a relatively wide bandwidth, the video processing device produces a sharper image than a narrower bandwidth device. This is because wide bandwidth devices exhibit phase nonlinearity or distortion as a function of frequency. That is, for example, since a wide bandwidth device generally has a more sensitive frequency upward transition characteristic (signal attenuation increases with increasing frequency) than a narrow bandwidth device, the high frequency video signal portion is delayed more than the low frequency video signal portion. Unnecessary asymmetric preshoots, overshoots and ringing phenomena appear in the processed video signal, resulting in phase distortion or nonlinearity first. Asymmetric preshoots, overshoots and ringing are unnecessary patents because they are not easily controlled. In particular, when processed on a receiver that improves the high-frequency response of the video signal but has inaccurate nonlinear phase characteristics, the image generated by the processed video signal is uncomfortable due to ringing, uncontrolled preshoot and overshoot. Do. As a result of this phase distortion, the instantaneous response of the wide bandwidth device is rougher than expected.

Various methods are known for improving the instantaneous response of a video processing device. In one of these devices, a batch parameter peak circuit is used, which is used to improve the high frequency response of the video processing device by increasing the amplitude of the high frequency component of the video signal compared to the amplitude of the low frequency component. Batch parameter peak circuits typically exhibit non-linear phase characteristics in the frequency function. These more general parametric peak circuits are formed for linear phase characteristics and peaking, and in general, a general parametric peak circuit that requires complex and expensive circuits is often required.

In another device that improves the transition response, the amplitude shift in the video signal is emphasized as the video signal passes through an overall parameter circuit arranged to produce sub-shoot overshoot. U.S. Patent No. 3,780,215, published by Shibata et al. On December 18, 1973, describes a circuit for generating preshoots and overshoots. In the Shibata circuit, the regional pass signal provided by the low pass filter in accordance with the video signal is extracted from the moderately delayed video signal to generate the preshoot and overshoot signal components. The appearance of voice transitions in the picture is emphasized more than anything else, but unless a lowpass filter is placed that has the same linear phase characteristics as the frequency function, the picture appears with unnecessary ringing and uncontrollable preshoots and overshoots. Gives offended

The color television video signal includes brightness, chroma and audio signal parts. The brightness signal portion covers a relatively wide bandwidth and extends to the low and high frequency ranges. The high frequency range has a relatively narrow bandwidth and includes chromatic and audio signal parts. The detail information of the image includes a high frequency component of the brightness signal portion.

The color television receiver for processing this signal includes a brightness channel for processing the brightness signal portion and a chroma channel for processing the chroma signal portion. In order to improve the sharpness and detail of the image, it is necessary to improve the instantaneous response of the brightness channel by increasing the high frequency response of the brightness channel. However, the appearance of chromatic and audio signal portions in the brightness channel causes unnecessary visible patterns in the image. Therefore, there is a need for an apparatus for removing chromatic and audio portions from the brightness channel. Conventionally, band elimination filters or trap circuits around the color carrier frequency have been lowered to remove chroma signal parts, and trap circuits have been centered around intermediate carrier acoustic frequencies to remove voice signal parts. In addition, a peaking circuit for relatively emphasizing the frequency components of the brightness signal portion is provided separately in the brightness channel. To reduce complexity and cost, a single circuit was needed to relatively emphasize the high frequency components of the brightness signal portion and to relatively attenuate the chromatic or audio signal portion or both.

In addition, it is necessary to adjust the peaking characteristics of the brightness channel. For example, depending on the quality of the received television signal, it is necessary to control the amplitude of a certain portion (relative high frequency component) of the brightness signal. Depending on the nature of the received signal, it may be necessary to adjust whether the high frequency components of the brightness signal are emphasized (i.e. peaked) or attenuated (having a minimum).

As such, if a received signal has been processed at the transmitter to provide a highlighted high frequency brightness signal component, such as a cable television (CATV) device, or if the transmitted signal contains a relatively high frequency noise component, the high frequency component of the brightness signal It is preferable to set the minimum to the peak rather than the peak. Regardless of whether the high frequency component is peaked or peaked, the adjustment of the amplitude of the high frequency component does not actually affect the trapping characteristics of the brightness channel. Since the brightness of the image is related to the direct current component of the video signal, it is necessary to ensure that such adjustment does not affect the direct current component of the video signal.

It is known how the amplitude or phase characteristic (all) required as a frequency function is formed in a device, in which a delay line or a delay signal generated at a signal coupling point (usually called a tap) along a device of that kind is known. Are combined in some way to obtain the required properties. Such a device is described in US Patent No. 2,263,376, published on November 18, 1941, entitled “Electrical Waveform Filters and Similar Devices,” published by A.D.Broomline, and published in July 1940, I.R.E. Pages 302 to 310 of Processing Unit 28, Vol. 7, published by H. E. Kellman, "The Transverse Filter," published in July 1955, Volume BTR-1 of the IRE Action of Radio and Television Receivers. Also found in the paper, “Selectivity and Immediate Response Analysis,” pages 1–8, R. V. Perry and D. Shurenian in Bell Systems Technical Journal, Vol. 39, No. 2, pages 405–422, March 1960. There is a statement entitled, “Side-Equalizer for Television Circuits”. These devices are sometimes called transverse equalizers or filters and are useful in many areas of signal processing. For example, it can be seen that such a device is useful for horizontal and vertical hole beam calibration. For example, see US Pat. No. 2,759,044 entitled "Bam Hole Correction in the Horizontal and Vertical Direction," published on August 14, 1956 by B. M. Oliver. U.S. Patent No. 2,922,965, "Above Equalizer and Correction for Televisions," published by Mr. Double U. Harrison on January 26, 1960, describes another form described in Oliver's patent. It is connected to a delay line with multiple taps to reduce the number of taps. In the delay line described in U.S. Patent No. 3,749,824, entitled "Suppression Filter for Carrier Chromaticity Signals Using Tapped Delay Lines," published on July 31, 1973 by T. Sagashima et al. It is selectively coupled to one end of the brightness channel delay line during color transmission to suppress the < RTI ID = 0.0 > The delay line is used to compensate for the delay time of the processed signal in the brightness and chroma channels.

A signal delay device according to an embodiment of the invention and responsive to a television video signal having at least a brightness and chroma signal portion is included in a brightness channel of the television video processing device. A plurality of signal combiners are connected to the signal delay device to develop a plurality of delayed video signals that are appropriately time-divided into predetermined sections. The two delayed video signals are combined to form a peak decision signal. The peak determination signal has an amplitude versus frequency characteristic with a relatively increased amplitude in the upper frequency region of the brightness signal portions. At least one of the delayed video signals is used to derive the bandwidth determination signal. The bandwidth determination signal is combined with the peak determination signal to form an output signal having an amplitude vs. frequency characteristic. Has amplitude.

According to another feature of the invention, part of the signal delay device serves to equalize the time delay of the processed signal in the chromaticity and brightness channels of the color television processing device.

According to another feature of the present invention, the two delay ratio signals control the form of the preshoot and overshoot of the amplitude transition of the output signal, and control the amplitude of the output signal within the upper frequency range of the brightness signal portion. An apparatus is provided.

According to another aspect of the present invention, the peaking determination signal and the bandwidth determination signal are combined so that the amplitude of the output signal at DC (zero frequency) and the range of the chromatic or audio signal portion is within the upper frequency range of the brightness signal portion. It is not affected by the amplitude adjustment of the output signal.

According to yet another aspect of the present invention, there is an apparatus for adjusting the amplitude of the output signal to the upper and lower values of the bandwidth determination signal within the upper frequency range of the brightness signal portion.

Hereinafter, the present invention will be described in more detail with reference to the accompanying drawings.

Referring to FIG. 1, a general apparatus of a color television receiver utilizing the present invention, which is composed of chromaticity, brightness, and voice and synchronization parts by an apparatus of an intermediate frequency (not shown) and a detection circuit (not shown). And a signal processor 12 responsive to the rf television signal received at the antenna to generate a video signal. The video output of the signal processor 12 is coupled to the chroma channel 14, which includes the chroma processor 16, and to the brightness channel 18, which is comprised of the first brightness processor 20 and the second brightness signal processor 22. The output signal of the chromaticity processing device 16 representing the B-Y, G-Y, and R-Y information is applied to the kinescope driver 34, in which these output signals are formed of the output Y of the brightness processing circuit 22 and the matrix. The signal processing device 20 is used to relatively attenuate unnecessary signal parts such as chroma or signal parts within the brightness channel 18, while increasing the amplitude of the high frequency components of the light signal part to improve the instantaneous response of the television receiver. The signal processing apparatus 20 is also used to equalize the delay times of the signals processed in the brightness channel 18 and the chroma channel 14. The output of the signal processor 20 is coupled to the brightness processor 22, which functions to amplify and process the video signal. The amplified and processed video signal is coupled to kinescope driver 34. Contrast control 32 is coupled to brightness processor 22 to control the amplitude of the video signal and to control the contrast of the image generated by kinescope 28. Contrast control devices are described in US Patent No. 3,804,981 entitled “Brightness Control” published by Jack Avins.

Another portion of the output signal from signal processor 12 is coupled to synchronous separator 24, which separates the horizontal and vertical sync pulses from the video signal. The sync pulse is coupled from the sync separator 24 to the deflection circuit 26. The deflection circuit 26 is coupled to the kinescope 28 and the high pressure device 30 to control the deflection or sweep of the electron beam in the kinescope 28 in a conventional manner. The deflection circuit 26 has a function of generating an erase signal from horizontal and vertical pulses. The erase signal is coupled to the brightness processor 22 to block the output of the brightness processor 22 during the vertical horizontal retrace period, thereby causing the kinescope 28 to shut off during each of these periods.

The circuit of FIG. 1 is the egg of Indianapolis, Indiana. Seed. a. Egg published by Copoishon. Seed. a. Color Television Service Data Suitable for use with color television receivers of the type shown in 1980.

The signal processing device 20 includes a signal delay device 36 shown as a delay line and a plurality of signal coupling devices or tabs 38 coupled to the delay line 36. The combination of signal delay device 36 and tab 38 is also referred to as a delay line with a tap. Although delay line 36 is shown as a delay line, delay device 36 may be another suitable device for delaying a video signal. For example, the signal delay device 36 may be formed in an arrangement of a charge coupling device (CCD) or a charge transfer device. Although taps 38 are shown connected directly to delay line 36 at consecutive points along the delay line, they may be coupled in other ways for signal coupling, such as capacitive coupling. The video signal is delayed between successive taps 38 in each constant time interval. Delay signals appearing at each of the taps 38 are fed to each of the devices 40. The device 40 has a function of weighting the signal and in this embodiment controls the signal amplitude. The device 40 modifies the amplitude of the video signal with its respective set gain value, thus generating multiple amplitude controlled or weighted signals. The weight device 40 is formed by a suitable gain control circuit including, for example, an amplifier, where the gain in the amplifier is set within a range above and below a constant value. Although the amplitude controller 40 is coupled to each tap 38 to show the general arrangement of the signal processing unit 20, this is not the case in all cases. If a predetermined gain such as 1 is required, the feature amplitude controller 40 is connected in series to the respective tab 38 and summing circuit 42.

The resulting amplitude control signal is geometrically coupled to summing circuit 42 to generate a video signal, which has an improved instantaneous response and is also independent of unnecessary signal portions such as chroma or audio signal portions. The summation circuit 42 is composed of circuit devices for adding or subtracting signals such as operational or differential amplifier devices. The amplitude control device 40 is included in the summation circuit 42. For example, Figure 9 illustrates such a device. As already explained, also in the form of a color television receiver as shown in FIG. 1, the delay device 36 needs to be arranged to equalize the time delay of the signal processed with the chromaticity and lightness channels.

FIG. 2 is a block diagram showing the configuration of the signal processing apparatus 20 of FIG. Taps 38a, 38b, and 38c of the world are coupled to delay line 36 in intervals corresponding to delay time intervals T _D , T _D + T ₁ and T _D + T ₁ + T ₂ , respectively. Delay line 36 includes a portion 37 having a time delay interval T _D before tap 38a selected for interval T ₁ to equalize the time delay of the processed signal in the brightness and chroma channels. That is, the sum of T _D and T ₁ is equal to the difference between the time delays associated with the signal processed by the chroma channel and the brightness channel. Also note that the signal resulting from the synthesis of the signals appearing in the taps symmetrically arranged around a certain point of the delay line has a time delay equal to the average of the time delays of the synthesized signals. Therefore, if tab 38b is placed in the middle between tabs 38a and 38c such that T ₁ and T ₂ are equal, the time delay of the output signal formed by synthesizing the signal from tabs 38a and 38c is processed within the chroma and brightness channels. It will be equal to the time delay required to equate to the time evolution of the given signals.

Respective tabs 38a, 38b and 38c are connected to respective weight devices 40a, 40b and 40c. The weight devices 40a, 40b and 40c are amplifiers (or attenuators) that modify the amplitude of the video signal shown in taps 38a, 38b and 38c with their respective constant values A, B and C, respectively. The adder 212 subtracts the amplitude that controls the output signals of the weight devices 40a and 40c from the amplitude that controls the output signal of the weight device 40b. Summer 212 consists of circuitry suitable for performing logarithmic summation, such as operational amplifiers and resistive metrics. The output signal of the summing circuit 212 is coupled to the peak control circuit 214, which modifies the output signal amplitude of the summing device 212. The peak control circuit 214 is composed of a conventionally known configurable gain device such as a variable gain amplifier. This peak control circuit is also arranged to calculate a gain range between a value smaller than a predetermined value and a value larger than a predetermined value. The output of the adder 212 and the weight device 40b is coupled to the adder 216, which is similar to the adder 212, where the two outputs are added logarithmically to form an output signal.

The operation of FIG. 2 is well understood by referring to FIG. 3 and FIG. The frequency domain and time domain waveforms are the waveforms at the respective circuit points of the apparatus of FIG.

Before explaining FIG. 3, the amplitude vs. frequency response characteristic of the tabbed delay line will be briefly described. It is well known in the art that the amplitude versus frequency transition characteristics of some delay lines that distribute the time delay T to the applied signal are ^expressed as constants, e ^-jwT and e being natural logarithms, varying as a function of e ^-jwT frequency, which is natural logarithm e. It is required by the ruler. The amplitude-to-frequency transition characteristic associated with the signal developed at the tap located at the reference point where T = 0 is flat since e ⁰ = 1. Note that the amplitude versus frequency characteristic associated with the two algebraically summed signals generated at each tap symmetrically located around the reference point changes as a cosine function.

Therefore, in FIG. 3, the time intervals T ₁ and T ₂ are selected identically to the time interval T, and the selection values A, B and C are equally selected to be the related values 1/2, 1 and 1/2 respectively. In Figure 2, the device of Figure 2 has an amplitude-to-frequency transition characteristic that changes with a cosine function with period 1 / T superimposed on level 312. The cosine portion of the transition characteristic is obtained by adding the signal appearing at taps 38a and 38c to the controlled signal. Level 312 is derived from the amplitude control signal shown at tap 38b. The apparatus of FIG. 2 has an overall amplitude-to-frequency transition characteristic 314a when the peak control circuit 214 is celled to 0.15 and an overall amplitude-to-frequency transition characteristic 314b when the peak control circuit 214 is celled to 0.5.

Observing FIG. 3 can understand the features of the signal processing device of FIG. The position along the frequency side of the maximum and minimum points of the amplitude versus frequency transition characteristic of FIG. 3 is set as necessary by selecting a time interval corresponding to the separation of the tabs 38a and 38c of FIG. It can be seen that the bandwidth of the output signal can be controlled by the cosine transition characteristic derived by combining the signals appearing at taps 38a and 38c and the coupling level 312 derived from the signal appearing at tap 38b. The maximum or peak amplitude of the characteristic is controlled by controlling the gain of the peak control circuit 214. However, the adjustment of the peak control circuit 214 does not affect the transition characteristics of the signal processor at direct current (i.e., zero frequency). This feature is particularly necessary because the screen brightness determined by the DC component of the brightness signal is not affected by the adjustment of the peaking control circuit. The adjustment of the peak control circuit 214 does not affect the amplitude of the minimum point. This is desirable because peak adjustment does not affect trapping or reduction of unnecessary signal amplitude. For example, if time T is chosen according to the inverse of the color subcarrier frequency (3.58 MHz), the frequency component between 0 and 3.58 MHz is amplitude adjusted to provide peaks without disturbing the minimum response to 3.58 MHz.

Therefore, while selecting the time interval between taps 38a and 38c, i.e., with respect to the second and third descriptions of the color television receiver state of FIG. 1, the signal processing apparatus of FIG. The same brightness is used to emphasize the high frequency content of the signal part.

Referring to FIG. 4, there are various time domain waveforms showing signals appearing at the circuit points of the signal processing device of FIG. 4A is a schematic representation of delayed video signals (indicated by a, b and c) which appear in taps 38a, 38b and 38c of the signal processing unit of FIG. 2, respectively. Various schematic combinations of delayed video signals a, b and c by the signal processing device of FIG. 2 are shown in FIG. 4C is a schematic representation of the output signal processing unit of FIG. For example, the transition time between amplitude 0 and amplitude 1 is 100 nS. For comparison, the transition time between amplitude 0 and amplitude 1 is obtained by measuring the sharpness of the waveform. Assuming that the rise time of delay line 36 in FIG. 2 can be ignored, the sharpness of the delayed video signals a, b and c is 100 nS. As an example, assuming that T ₁ and T ₂ are each equal to 100 nS, A, B, and C are 1/2, 1, 1/2, and the gain of the peak control circuit 214 is set to one. 4c, the output signal has a preshoot (amplitude less than or equal to 0) and an overshoot (amplitude greater than or equal to 1) having the same magnitude and the same time duration. The output signal has an amplitude transition sharper than that of the input video signal. The amplitudes of the preshoot and overshoot are controlled by selecting constant values A and C. On the other hand, the durations other than the preshoot and overshoot are adjusted by selecting the time interval between taps 38a, 38b and 38c. The amplitude transition sharpness and amplitude of the output signal are also controlled by selecting the gain of the peak control circuit 214.

The generation of preshoots and obishoots by the circuit relates to the phase line of the circuit depending on the frequency function. In addition, the occurrence of the same preshoot and overshoot corresponds to the linear phase algebraic characteristic. Corresponds to the phase-to-frequency characteristic of the peak circuit of FIG. The phase-to-frequency characteristic of the peak circuit of FIG. 2 is controlled by controlling the generation of the amplitude control signal associated with the taps 38a and 38c. For example, although the time intervals T ₁ and T ₂ are equally selected above, the preshoot and overshoot have a non-equalization time duration to compensate for phase-to-frequency nonlinearity within other parts of the video signal processor. It is necessary to choose differently to generate the chute.

It is necessary to select different selection values A and C to compensate for the phase-to-frequency nonlinearity in other parts of the video signal processor.

By selecting time intervals and constant intervals A and C corresponding to the separation of taps 38a, 38b and 38c, the amplitude transition of the brightness signal is emphasized by controlled preshoot and overshoot generation. The sharpness of the amplitude time lapse in the brightness signal may be controlled by the gain selection of the peak control circuit 214.

Referring to FIG. 3, if it is necessary to have a minimum amplitude at the color subcarrier frequency at 3.58 MHz to relatively attenuate the chroma signal part, T should be selected as 280 nS, the inverse of the color subcarrier frequency.

That is, by selecting T nearly 280nS seconds, the amplitude-to-frequency characteristic of the brightness signal is generated at about 1.79MHz. Brightness channel by causing the peak of the amplitude vs. frequency characteristic to occur in the high frequency component of the brightness signal, that is, the peak of the amplitude vs. frequency characteristic to occur in the frequency component closer to the color subcarrier frequency than half of the color subcarrier frequency (1.78 MHz). It is necessary to maximize the high frequency response of the signal processor while the signal processing device of FIG. 5 is better than the signal processing device of FIG.

The embodiment of the type shown schematically in FIG. 5 is used as the signal processing unit 20 of FIG. 1 so that it performs high frequency peaking consistent with effective tapping. The signal processing unit of FIG. 5 includes a delay line 36 '(or other device described above) in response to a video signal. Taps 38a ', 38b', 38c 'and 38d' are coupled to delay line 36 'at regular intervals corresponding to time intervals T ₁ ', T ₂ ', T ₃ ', respectively. Delay line 36 'includes a portion 37' having a time delay interval T _D 'before tap 38a' selected for the other part of the delay line to equalize the time delay of the processed signal in the brightness and chroma channels. Difference between the time delays of signals processed in chroma and brightness channels for the sum of T _D ′, T ₁ ′ and T ₂ ′ / 2 for the purpose of equalizing the time delay of the processed signals in the brightness and chroma channels. It is necessary to be equal to Thus, from the combination of signals appearing in the taps symmetrically arranged around the midpoint of the delay line, the signal has a time delay equal to the average signal delay of the combined signal. Because of this, if the time intervals T ₁ ′ and T ₃ ′ are equally selected, the output signal has the time delay necessary to equalize the time delay of the processed signal in the chromaticity and brightness channels.

The tabs 38a ', 38b', 38c 'and 38d', respectively, are coupled to the weight devices 40a ', 40b', 40c 'and 40d', respectively. The weight devices 40a ', 40b', 40c 'and 40d' act to modify the amplitude of the video signal by constant values A ', B', C 'and D', respectively. According to the tabs 38b 'and 38c', the amplitudes of the centrally controlled or weighted output signals of the two weight devices 40b 'and 40c' are coupled to the adder 412, where the adder 412 is logarithmically added. The output signal of the adder 412 is coupled to the peaking control circuit 416, and with this peaking control solution 416 serves to modify the amplitude of the output signal of the adder 414. The output signal of the peak control circuit 416 and the output signal of the summing device 412 are coupled to the summing device 418, where these output signals are combined algebraically to form an output video signal.

The peaking circuit operation of FIG. 5 is described as an example, where T ₁ ′, T ₂ ′, and T ₃ ′ are all selected as 140 nS, i.e., 1/2 of the inverse of the 3.58 MHz color subcarrier frequency. , Constant values A ', B', C 'and D' are each selected to be equal to 1/2.

Referring to FIG. 6, various frequency domain waveforms are shown combined with the peak circuit of FIG. This waveform is shown as an algebraic representation indicating a combination of delayed video signals which are represented by taps 38a ', 38b', 38c 'and 38d' as indicated by a ', b', c 'and d', respectively. This output shows the two gain settings of the peak circuit 416, i.e. gain settings at 50% and 75%. Examining FIG. 6, it can be seen that, as described above, the pairs of algebraic sums of the amplitude controlled signals result in the generation of amplitude versus frequency transition characteristics in accordance with the Cosine law. When the amplitude control signal according to the taps 38a 'and 38d' is added logarithmically to be divided into time intervals of about 3 x 140 nS, the cosine amplitude band frequency characteristic expressed by Equation 1/2 (a '+ d') is leaked.

This characteristic has a half-width ratio of 4/3 x 3.58 MHz. Similarly, when the amplitude control signals separated by time intervals equal to 140 nS and corresponding to taps 38b 'and 32c' are algebraically added, the cosine amplitude versus frequency characteristic expressed by Equation 1/2 (b '+ c') Spills. This characteristic has a repetition ratio of 4 x 3.58 MHz.

From the experiment of FIG. 6, the combination of the amplitude-controlled signal associated with taps 38a 'and 38d' is FIG. 6, which is represented by the equation 1/2 (a '+ d') and the amplitude band that controls the peak, or intensity, of the output signal. It can be seen that the outflow in the form of frequency transition characteristics. In addition, the combination of amplitude control signals associated with taps 38b 'and 38c' is represented by Equation 1/2 (b '+ c') in FIG. 6 and combined with the peak control characteristic, amplitude band frequency, which controls the bandwidth of the output signal. It can be seen that the outflow in the form of transition characteristics. The maximum value of the amplitude-to-frequency transition characteristic varies with the gain setting of the peak control circuit 416. Therefore, the adjustment of the peak control circuit 416 does not affect the characteristic amplitude at the DC zero frequency and the screen brightness is not affected by the adjustment of the peak control circuit 416. Adjustment of the peak control circuit 416 does not directly affect the trapping (minimum response) frequency.

The choice of delay periods T ₁ ′, T ₂ ′ and T ₃ ′ at 140 nS has an amplitude-to-frequency characteristic with peak amplitude at relatively high frequencies of about 2/3 × 3.58 MHz and an effective 3.58 MHz trapping. It is very advantageous because it can provide an output signal having However, it should be noted that the values given as examples can be modified as needed to suit a particular example. For example, selecting T ₂ ′ about 110 ns and T ₁ ′ and T ₃ ′ about 140 nS is also required. In this case, the amplitude versus frequency characteristic of the output signal will have a value of zero at about 4.5 MHz and a peak amplitude at about 2/3 x 3.58 MHz (ie, 2.4 MHz). Therefore, in the signal processing apparatus of FIG. 5, the frequency components in the chromaticity of the video signal and the region of the audio signal portion are relatively attenuated, and the relatively high frequency components of the brightness signal portion are increased in amplitude.

Therefore, by algebraically combining the amplitude control signals in the selected method, the peak device of FIG. 5 attenuates the chromatic or audio signal components or both components, and relatively emphasizes the amplitude of the relatively high frequency component of the brightness signal portion.

Referring to FIG. 7, various time domain waveforms associated with the peak circuit of FIG. FIG. 7A diagrammatically shows a delay video signal represented by taps 38a ', 38b', 38c 'and 38d' of the signal processing apparatus of FIG. 5 and denoted a ', b', c 'and d', respectively. FIG. 7B diagrammatically shows various combinations of delayed video signals a ', b', c 'and d' provided by the signal processing apparatus of FIG. FIG. 7C schematically shows the output signal of the signal processing device of FIG. Assuming that the input video signal has a transition time between 0 and 1 of 280 nS, if the rise time of delay line 36 'is assumed to be negligible, the delayed video signals a', b ', c' and d ′ Has a transition time of 280 nS. For simplicity, only the output signal with the gain setting of the peak control circuit 416 set to 1 is shown.

Referring to FIG. 7, it can be seen that the output signal includes an overshoot and a preshoot controlled by amplitude control signals associated with the taps 38a 'and 38d'. The amplitude of the preshoot and overshoot is controlled by selecting the selected values A 'and D', and the duration of the preshoot and overshoot is controlled by selecting the sections T ₁ 'and T ₃ '. The sharpness of the output signal is greater than the sharpness of the input video signal. Adjusting the gain setting of the peak control circuit 416 also adversely affects the sharpness and amplitude of the preshoot and overshoot of the output signal.

Thus, by algebraically combining the amplitude control signals in a predetermined manner, the peak circuit of FIG. 5 has controlled preshoots and overshoots that drive to emphasize amplitude transitions, and also provides relatively sharp amplitude transitions.

Note that the phase-to-frequency characteristic of the signal processing unit of FIG. 5 is directly controlled by controlling the amplitude control signal associated with taps 38a 'and 38d'. For example, the linear phase to frequency characteristic corresponds to the formation of the same preshoot and overshoot. Therefore, in the example described above, although the selected values A 'and D' are equally selected and the time delay intervals T ₁ 'and T ₃ ' are to provide a linear phase-to-frequency characteristic as evident by the same preshoot and overshoot, Although equally selected as described, the amplitude control signals associated with taps 38a 'and 38d' provide uneven preshoot and overshoot to compensate for the nonlinearity of phase-to-frequency in other parts of the video signal processor. To be controlled.

Fig. 8 shows the use of the schematic embodiment of the invention shown in Fig. 5, in which the actual operating part (indicated by dashed line 810) is suitable as an integrated circuit. The resistance value of FIG. 8 calculates the selected values of A '= 1/2, B' = 1/2, C '= 1/2, and D' = 1/2 in the embodiment in the FIG. 5 circuit operation description. It is selected according to. The circuit of FIG. 8 may be modified to provide different predetermined values and to suit a particular use case.

In FIG. 8, delay line 36 'is processed within chroma channel 14 and lightness channel 18 of FIG. 1 by delaying an input video signal between continuous taps 39a', 38b ', 38c' and 38d 'by a predetermined interval. It is chosen to equalize the time delay of the signal. A video signal source (not shown) typically has an output impedance approximately equal to the characteristic impedance of delay line 36 'to minimize signal reflections at the input of delay line 36'. The delay line 36 'is terminated with an impedance 812 selected so that its magnitude is approximately equal to the value of the characteristic impedance of the delay line 36' so as to minimize line termination reflection.

Taps 38a 'and 38d' are respectively connected to two inputs of differential amplifier 814 including NPN transistors 811 and 818, where delayed video signals appearing at taps 38a 'and 38d' respectively are weighted and mathematically added to 1/2 ( a signal at the junction of the resistors 820 and 822 of the differential amplifier 814, such as a '+ d'). The input impedance of the differential amplifier 814 becomes relatively large compared to the value of the characteristic impedance of the delay line 36 'by appropriately selecting the emitter resistor values of the transistors 811 and 818.

Tabs 38b 'and 38c' are coupled to the base of transistor 816 through resistors 824 and 826, respectively, and transistor 816 has a common emitter structure and also forms a summation circuit with resistors 824 and 826. Resistors 824 and 826 are chosen to be relatively large relative to the characteristic impedance of delay line 36 'so as not to load load line 36'. The signal at the emitter of transistor 816 is equal to 1/2 (b '+ c').

The 1/2 (b '+ c') signal is generated in the same way as the 1/2 (a '+ d') signal but in a summing circuit comprising transistors 816 and resistors 824 and 826 to hold the integrated circuit input stage. It should be noted that it is described as being.

The 1/2 (b '+ c') and 1/2 (a '+ d') signals are applied to the inputs of the differential amplifier 832 in common with the emitter follower stages, including the NPN transistors 828 and 830, respectively. . Differential amplifier 832 includes NPN transistors 836 and 834, where the 1/2 (a '+ d') signal is subtracted from the 1/2 (b '+ c') signal and G [1] is applied to the collector of transistor 834. / 2 · (b '+ c')-1/2 · (a '+ d')]. Where G is the gain of differential amplifier 832.

The gain G of the differential amplifier 832 is adjusted by adjusting the voltage at the peak control terminal of the peak control circuit including the NPN transistors 838,848 and 850. This adjustment is substituted for setting the gain of the peak control circuit 416 of FIG. The peak control circuit is coupled to the emitter and collector circuit of transistor 834 in such a way that the gain of differential amplifier 832 is adjusted without changing the DC voltage at the output of differential amplifier 832 in response to the peak control voltage. That is, the current applied from the collector of transistor 838 to the emitter circuit of transistor 834 and the current from the collector of transistor 848 to the collector circuit of transistor 834 are output from differential amplifier 832 with the same or opposite polarity in response to a change in the peak control voltage. Proportional to change the DC component in the signal.

The output of differential amplifier 832 is coupled to the base of NPN transistor 840, which includes a series connection with resistors 842, 844, and 846 and emitter follower circuit. The 1/2. (B '+ c') signal exiting the emitter of NPN transistor 828 is coupled to the junction of resistors 844 and 846, where this signal is G [1/2. (B '+ c')-. 1/2 · (a '+ d')] is added logarithmically to form an output signal.

In Fig. 9, an embodiment of the present invention similar to the embodiment of Fig. 5 is shown. Also shown is that when the delay line 36 " is properly terminated to generate a reflected signal, a simpler configuration can be made by utilizing the reflected signal (shown as an open circuit). Tap 912 is located at a position relative to the time delay interval [T ₁ ″ + T ₂ ″ + T ₃ ″ / 2), where T ₁ ″, T ₂ ″, T ₃ ″ are the time of FIG. 5 from the opening end of the delay line. Signals corresponding to the delay sections T ₁ ′, T ₂ ′, and T ₃ ′ and outgoing signals similar to the result of adding the signals shown in taps 38a ′ and 38d ′ in FIG. The tap position 914 is located at a position corresponding to the time delay T ₂ ″ / 2 from the opening end of the delay line 36 ″ and a signal similar to the result of the sum of the signals emitted from the taps 38b 'and 38c' of FIG. 5 appears. . Therefore, signal a ″, directly connected to reflection a ″ signal delayed from signal a ″ by a delay time interval such as delay interval T ₁ ″ + T ₂ ″ + T ₃ ″, taps to form the resulting signal indicated by a ″ + d ″. Leak at 912 Similarly, "by a delay interval equal to b" T ₂ delayed from the direct signal b "and signal reflected c" signal is flowing out of the tab 914 to form a resultant signal denoted by b "+ c".

The a ″ + d ″ signal is coupled to the weight device 916. The b ″ + c ″ signal is coupled to the weight device 918. The output signal of the weight device 916 is subtracted to the output of the weight device 918 in the summing device 920. The output signal of the adder 920 is applied to the adder 924 via the peak control circuit 922, where the output number is added to the output signal of the weight device 918 to form an output signal.

In FIG. 5, the amplitude of the amplitude control signal corresponding to the tap positions 38a ', 38b', 38c 'and 38d' is separately set by setting predetermined values of the weight devices 40a ', 40b' and 40c ', respectively. As a result of the control, the amplitudes of the amplitude control signal generated from the direct signal reflected from the taps 912 'and 914 and the reflected signal are controlled in pairs. That is, the weight device 916 controls the amplitudes of the amplitude control signals generated from the direct connection signal reflected from the tap 912 and the reflected signal, and the weight device 918 controls the amplitude control signal generated from the direct connection signal reflected from the tap 914 and the reflected signal. Control the amplitude of the

As in FIG. 8, the delay line forming the delay device 36 " includes a component that equalizes the time delay of the processed signals in the chromaticity and lightness channels. For this purpose, the overall length of the delay line must equalize the time delay difference between the signals processed in the chroma and brightness channels.

Referring to FIG. 10, a signal processing unit 1,000 suitable for use as the signal processing unit 20 of FIG. 1 is driven to relatively high frequency components of the brightness signal portions, while the signal processing unit shown in FIG. Similar forms are used to relatively attenuate unwanted portions of the video signal. The signal processing unit 1,000 also peaks or peaks the relatively high frequency components of the brightness signal.

The three taps 1,038a, 1,038b, and 1,038c are delay time intervals D, D + D ₁ and D + D ₁ + D ₂ to exit the delayed signals V _a , V _b , and V _c , respectively, for the incoming signal. Are connected to the delay line 1,036 in the corresponding intervals. These delay time intervals are the same as T _D , T _D + T ₁ , T _D + T ₁ + T ₂ of the signal processing unit of FIG. ₂ , respectively. Delay line 1,036 includes part 1,037 before tap 1,038a, which is similar to part 37 of the signal processing unit of FIG. 2 which equalizes the time delay of the processed signal in the brightness and chroma channels.

Delay line 1,036 is equal to the characteristic impedance of delay line 1,036 and terminates at an impedance of 1,036 shown as a resistor. A video signal source (not shown) must have an output impedance whose magnitude is approximately equal to the magnitude of the characteristic impedance of delay line 1,036.

Portion of the delayed signals V _a, V _b and V _c are through the resistors RA, RB, and RC each is connected to a common set of points 1042 to form a signal V _m.

Delayed signals V _a , V _m and V _c are applied respectively to the inputs “-”, “+”, “-” of the sum unit 1,012, equal to the sum unit 212 of FIG. 2 and the delayed signals V _a and V at V _m respectively. Subtract logarithmically _c to form V _P. In addition, summing device 1,012 drives to deform the amplitudes (ie the weight) of V _a and V _c before subtracting from V _m .

The output V _p of the adder 1,012 is applied to the amplitude control circuit 1,014 with a gain of K to form K _vP . For example, the amplitude control circuit 1,014 is a variable gain amplifier configured to provide a gain range from a value less than a certain reference value to a value greater than a certain reference value.

The output and junction 1,042 of the amplitude control unit 1,014 are connected to the "+" input of the sum unit 1,016, similar to the sum unit 216 of FIG. 2, where V _m and K _vP are algebraically added to form the output signal.

To describe the operation of the signal processing unit 1,000, the delay periods D ₁ and D ₂ are selected as t respectively and the summation device 1,012 is selected by V _a by means of the selected weights of lines 1/2, 1 and 1/2 respectively. , V _m , V _c are arranged to modify the amplitudes. For example, the value of RA is chosen to be equal to the value of RC. Assuming that the values of RA and RB are significantly larger than the values of the characteristic impedance of the delay line 1,036 by overlapping, the relationship of V _m to V _a , V _b , and V _c is expressed by the following equation.

The signals V _p and V _o are given by

Considering the tap 1,038b located at the reference point as already determined, the amplitude-to-frequency transition characteristics for V _m , V _o , and V _P are given by

의 피크 투 피크 진폭과, 레벨 RA/(RA+2RB)에 중첩된 1/t의 주기와(제4식 참조) DC(영 주파수)에서 최대진폭을 가지며 또한 1/t의 정수배수에서 최대진폭을 가지며 1/2t의 정수배수에서 최소진폭을 갖는 코싸인 함수이다.11 schematically shows the normal amplitude versus frequency transition characteristics associated with V _m and V _P. The transition property associated with V _m is

Peak-to-peak amplitude of, the period of 1 / t superimposed on level RA / (RA + 2RB) (see equation 4), and the maximum amplitude at DC (zero frequency), and also at an integer multiple of 1 / t. It is a cosine function that has a minimum amplitude at integer multiple of 1 / 2t.

Since the transition characteristics associated with V _m correspond to unnecessary phase inversion, it is necessary to select the values of the resistors RA, RB and RC in such a way that they do not fall below zero amplitude side (ie not to be negative). For example, this selects RA / (RA + 2RB) greater than or equal to 2RB / (RA + 2RB), or RA must be greater than or equal to 2RB. For this purpose, RA / (RA + 2RB) was chosen to be equal to 0.75 and 2RB / (RA + 2RB) was chosen to equal 0.25.

Regardless of the selection of RA, RB, RC, the transition characteristic associated with V _m at DC is always 1, so at DC, the delay signals at taps 1,038a, 1,038b, 1,038c all have the same amplitude. This is required because it allows for controllable peaking or peaking of relatively high frequency components of the brightness signal without adversely affecting the DC components of the brightness signal.

The transition characteristic associated with V _P (5) is a cosine function with minimum (0) amplitude at DC and integer multiples of 1 / t, period of 1 / t, and highest amplitude at integer multiples of 1 / 2t. .

The maximum amplitude points of the transition characteristic associated with V _P are opposite to the minimum amplitude points of the transition characteristic associated with V _m , and vice versa.

Since V _o is the algebraic sum of V _m and KV _P (see equation 3), the transition characteristics associated with V _o at integers of 1 / 2t do not adversely affect amplitude at DC or at integers of 1 / t. By controlling the value, it is either highlighted (peaked) or attenuated (de-peaked). That is, by varying the value of K, the transition characteristic associated with V _o at frequency 1 / 2t is greater (ie peaked) or smaller (depeaked) than the transition characteristic associated with V _b at this frequency. Note that the transition characteristic associated with V _b corresponds to the broadband transition characteristic associated with the signal exiting at tap 38b of the signal processing unit of FIG. 2 (level 312 of FIG. 3).

The variation of the transition characteristics associated with V _o as a function of K is shown in FIG. 12 which schematically shows the average amplitude versus frequency transition characteristics associated with V _m , V _P and V _o for various values of K. When the value of K is 2RB / RA, the transition characteristic associated with V _o is flat. That is, it is equal to the transition characteristic associated with V _b . When K is less than 2RB / RA, i.e., 1/2 (2RB / RA), the transition characteristic associated with V _o is peaked at 1 / 2t. When K is greater than 2RB / RA, ie 2 × (2RB / RA), the transition characteristic associated with V _o is peaked at 1 / 2t.

By selecting D ₁ and D ₂ of the signal processing unit 1,000, which is about 280 nS, the amplitude of the amplitude-to-frequency transition characteristic of the brightness channel 18 of FIG. 1 at about 1.78 MHz is peaked by changing the gain K of the amplitude control circuit 1,014. The state changes from the state to the peak state.

When the signal processing unit 1,000 is used in the brightness channel 18 of FIG. 1, the additional filter circuit before and after the signal processing unit 1,000 is arranged to attenuate the chromatic or audio signal part or both.

Referring to FIG. 13, another signal processing unit 1,300 suitable for use as the signal processing unit 20 of FIG. 1 is shown.

The four taps 1,338a, 1,338b, 1,338c and 1,338d are delay time intervals D, D '+ D ₁ ′, so as to leak delay signals e _a , e _b , e _c and e _d with respect to the input signal, respectively. D '+ D ₁ ′ + D ₂ ′ and D ′ + D ₁ ′ + D ₂ ′ + D ₃ ′ + respectively connected to delay line 1336 in the respective air gap sections.

These delay time intervals correspond to T ′ _D , T _D ′ + T ₁ ′, T _D ′ + T ₁ ′ T ₂ and ′ T _D ′ + T ₁ ′ + T ₂ ′ + T ₄ of the signal processing unit of FIG. similar. Delay line 1336 is similar to portion 37 'of the signal processing unit of FIG. 5 to equalize the temporal geometries of the processed signals in the lightness and chromaticity channels, and includes portion 1,337 immediately before tap 1,338a.

Delay line 1,33 is approximately equal to the characteristic impedance of delay line 1336 to terminate reflection at the end of the delay line and is terminated at impedance 1326 which is shown as a resistor. Equally, a video signal source (not shown) has an output impedance of approximately the same magnitude of the characteristic impedance of delay line 1336 to minimize signal reflection at the input.

Portions of delay signals e _a , e _b , e _c and e _d are applied to common junction 1,342 via resistors R ₁ , R ₂ , R ₃ and R ₄ , respectively, to form signal e _m .

The signals e _a , e _m and e _d are supplied to the “-”, “+” and “-” inputs of the summer 1312, respectively, which form the signal e _p . In addition, the adder 1312 drives to deform the amplitudes (weights) of e _a and e _d before subtracting from e _m .

The output of summing device 1312 is applied to amplitude control circuit 1316 to be used to modify the amplitude of e _p with gain p to form px _ep . The amplitude control circuit 1316 forms a gain range, for example, from a value greater than a certain value to a value greater than a certain value.

The output of the amplitude control device 1316 and the signal at junction 1342 are applied to the “+” input of the summation device 1318 which is similar to the summation device 216 of the signal processing unit of FIG. 2, where e _m and p · e _p are summed algebraically. Form the output signal e _o .

For this circuit, resistors are connected in series to the inputs of summing devices 1312 and 1318 to compensate for the time delay variation of the signals synthesized in the circuit due to the nonuniformity of parasitic capacitances associated with the input.

Signal processing operation of the oil nut 13 is selected as the delay interval D ₁ ', D _2' and D ₃ 'are each 2t' are summed by the device 1312 is selected for e weights of 1/2, 1 1/2, and _a will be described by way of example, which are respectively selected to modify the amplitudes of e _m and e _d . The value of R ¹ is selected to be equal to the value of R ₄ , and the value of R ₂ is selected to be equal to the value of R ₃ . If the values of R ₁ and R ₂ are larger than the characteristic impedance values of the delay lines 1,336 by overlapping, the relationship between e _m , e _a , e _b and e _d is given by the following equation.

The signals e _p and e _o are given by

Considering the midpoint between the taps 1,338b and 1,338c as reference points, the amplitude versus frequency shift characteristics associated with e _m , e _p , e _o , respectively, are given by

14 schematically shows the normal amplitude versus frequency transition characteristics associated with e _m and e _p . The amplitude versus frequency transition associated with e _m has the form of a cosine function (R ₂ / R ₁ + R ₂ ) coswt superimposed on another cosine function (R ₁ / R ₁ + R ₂ ) cos 3wt ′. Note that the transition characteristic associated with signal e _m has a maximum amplitude equal to 1 at DC (zero frequency) and after about (1/3) × (1 / 4t ′), a relatively sensitive upward transition (ie an increase in frequency). Amplitude decreases, and decreases from 1 / 4t 'to zero amplitude. The comparison between the broadband characteristics, indicated as 1/2 (b '+ c') in FIG. 6 and the transition characteristics associated with e _m in FIG. 14, corresponding to the signal leaked out at the output of the summation unit 412 of the signal processing unit of FIG. The latter transition shows a faster upshift than the former.

The transition characteristic associated with e _m corresponds to an undesirable phase inversion and thus selects values of resistors R ₁ , R ₂ , R ₃ and R ₄ so as not to fall below zero amplitude axis (ie not to be negative). It is necessary. For example, this selection results in 1/2 (R ₁ / R ₁ + R2) being greater than or equal to R ₂ / (R ₁ + R ₂ ) and R ₁ being greater than or equal to 2 (R ₂ ). For _{example, R 1 / (R 1 +} R2) was selected as _{0.75, R 2 / (R 1} + R 2) was selected as 0.25.

Regardless of the choice of R ₁ , R ₂ , R ₃ and R ₄ , the value of the amplitude versus frequency transfer characteristic associated with e _m at DC becomes 1, thus taps 1,338a, 1,338b, 1,338c and 1,338d at DC. The delay signals at all have the same amplitude. Depeaking and peaking of the high frequency components of the brightness signal can be controlled without affecting the DC component of the brightness signal.

The transition characteristic associated with e _p derived according to equation 11 has a minimum amplitude = 0 at DC and 1 / 4t 'and a maximum amplitude at (2/3) x 1 / 4t'. Therefore, the peak amplitude point of the transition characteristic associated with e _p is located in the relatively sensitive upward transition region between (1/3) x 1 / 4t and 1 / 4t 'of the transition characteristic associated with e _m .

Since the logarithmic fit of e _o -e _m and p x e _p (see equation 9), (2/3) the amplitude of the transition characteristic associated with e _o at 1/4 t 'adversely affects the amplitude at DC or 1/4 t' The value of P is emphasized (peak) or black is attenuated (de-peak) by controlling the value of P without giving. That is, by changing p, the amplitude of the amplitude versus frequency transition characteristic associated with e _o at frequency (2/3) 1 / 4t 'is equal to the signal (e _b + e) of FIG. 6 at frequency (2/3) 1 / 4t. _c ) is controlled to be greater (peak) or less (de-peak) than the amplitude of the transition characteristic associated with it. The transition characteristic associated with (e _b + e _c ) is the relatively wideband transition characteristic associated with the signal emanating from the output of the summation unit 412 of the signal processing unit of FIG. 5, that is, waveform 1/2 (b '+ c') of FIG. Corresponds to. The variation of the transition characteristic associated with V _o as a function of K is shown in FIG. 15. FIG. 15 is a plot of amplitude versus frequency transition characteristics associated with e _m , e _p and e _o for various values of P. FIG. When p is R ₂ / R ₁ , the transition characteristic associated with e _o is the same as the transition characteristic associated with e _b + e _c . When p is less than R ₂ / R ₁ , ie 1/2 R ₂ / R ₁ , the transition characteristic associated with e _o is depeaked at (2/3) 1 / 4t ′. When p is greater than R ₂ / R ₁ , that is, 3 · R ₂ / R ₁ , the transfer characteristic is peaked as long as it is associated with e _o .

By selecting D ₁ ′, D ₂ ′ and D ₃ ′ of the signal processing unit 1300 at about 140 nS, the amplitude of the amplitude-to-frequency transition characteristic of brightness channel 18 in FIG. 1 at 2.39 MHz does not change the amplitude at DC. By changing the gain P of the amplitude control circuit 1314 to change from the peak state to the peak state and effectively attenuate the signal components in the vicinity of 3.58 MHz, i.e., the chroma signal components.

In summary, the device relatively increases the amplitude of the high frequency components of the brightness signal portion and attenuates the amplitude of the chromatic or audio signal portion or both. The apparatus includes a delay line responsive to a television video signal. The multiple taps combine with the delay line to generate multiple delayed video signals. The two delayed video signals are synthesized to generate a peak control signal having an amplitude versus frequency characteristic that peaks in the frequency region between the frequencies f and DC in the region that attenuates the video signal. Note that it is necessary to space the two delayed video signals in the same time interval as NT / 2. Furthermore, N is an integer and T is the inverse of the frequency f. In the apparatus of FIG. 2, N was selected as 4. In the apparatus of FIG. 5, N was selected as three. Although suitable ranges of N include integers in the order of 2 to 5, other values of N may be used in certain examples. The band control signal or the broadband signal flows out of at least one third of the delayed video signal. The bandwidth control signal is combined with the peak control signal to form an output signal. For example, referring to the apparatus of FIG. 2, the wideband signal is derived from the delayed video signal leaked out of the center tap 38b. For example, referring to the apparatus of FIG. 5, the wideband signal is derived from the synthesis of the delay video signal developed at the two taps 38b 'and 38c' in the center.

It has been shown that the device provides an improved transition response that is consistent with the attenuation of the unwanted signal providing an unnecessary visible pattern without attenuation. It has also been shown that the apparatus provides a directly controllable preshoot and overshoot. It has also been shown that the apparatus provides a peak adjustment that does not substantially affect the amplitude of the frequency component or the amplitude of the DC component around the frequency f.

The device has also been described as providing control of the peak amplitude portion to allow peaking or depeaking depending on the quality of the transmitted signal. These devices include devices for reducing the amplitude of the output signal below the amplitude of the broadband signal at a frequency approximately equal to the frequency where the peak control signal has the maximum amplitude. For example, in the signal processing unit 100 of FIG. 10, the portions of two delayed signals separated by a delay time of 2T have the amplitude of the output signal greater than or equal to (peak) or less than the amplitude of the broadband signal at a frequency of 1 / 2T. Is added to the portion of the broadband signal emanating from the signal with the average delay middle of the two delayed signals. The portions of the two delayed signals separated by the delay time 3/2 in the signal processing unit 1,300 of FIG. It is added to the portion of the outgoing broadband signal by the algebraic addition of two delay signals each having a delay intermediate of the two delay signals separated by a delay of 2T to be adjusted to the following value (i.e., de-peak). A portion of the delay line can be used to equalize the time delay of the processed signals in the chroma and brightness channels.

Although the invention has been described with reference to specific embodiments, it is noted that various modifications are possible within the scope of the invention.

Claims (1)

Apparatus included in a television receiver comprising a video signal source for processing a television video signal, a brightness channel for processing a brightness signal, and a chroma channel for processing a chroma signal, as described herein and illustrated in the figures. A plurality of delayed video signals and a portion 37 connected to the video signal source, contained in the lightness channel, and for equalizing the time delay of signals processed by the lightness channel and the chromaticity channel. By combining from a plurality of signal connectors 38a, 38b, 38c connected to the delay device and at least two of the delayed video signals delayed from one another by a time interval such as NT / 2. And a first synthesis device 212 for developing the first synthesis signal, wherein N is configured to generate the first synthesis signal. A delay device 36 that is an integer greater than one so that T is the inverse of the frequency f such that the video signal is attenuated, and an apparatus B for outflowing a wideband signal from at least one other signal of the delayed video signals. And synthesize the broadband signal to provide a second synthesized signal, the first synthesized signal controlling the peak characteristic of the second synthesized signal and the first synthesized signal controlling the bandwidth characteristic of the second synthesized signal. And a utilization device connected to the second synthesis device in order to utilize the second synthesis signal.

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