KR20190133194A - Millimeter wave regeneration and retransmission for building penetration - Google Patents

Abstract

A system that enables signal penetration into a building is located outside the building and receives a signal of a first frequency that suffers loss upon penetration into the building and receives the received signal of the first frequency onto a wireless communication link within the building. And a first circuit for converting to a first format that overcomes the losses caused by penetration. The first circuit also implements a first transmission chipset for RF transmission in a first format corresponding to losses occurring upon penetration into a building, receiving a signal at a first frequency and receiving a received signal at a first frequency. The apparatus further includes a first transceiver for converting to a first format that overcomes the loss caused by penetration into the building. A second circuit located inside the building is communicatively linked with the first circuit and receives and transmits the converted received signal of the first format. The second circuit implements a first transmission chipset and further includes a second transceiver for receiving the converted signal of the first format from a first receiver outside the building and transmitting to the first receiver.

Description

Millimeter wave regeneration and retransmission for building penetration

The present invention relates to millimeter wave transmission, and more particularly, to a method for improving building permeability for millimeter wave transmission.

The millimeter wave transmission was developed as a bandwidth plan to create a spectrum of local multipoint distribution services (LMDS) available in the United States at 1300 MHz. Millimeter wave transmission meets the need for increased bandwidth availability due to the increasing bandwidth and application requirements for wireless mobile devices. However, if the bandwidth function is improved, the problem is that millimeter wave transmission has very poor building penetration capability. The signal drops dramatically when attempting to penetrate most building structures. This presents a serious problem because the majority of wireless signal traffic occurs in buildings, and when millimeter wave bandwidth becomes unavailable, the implementation of it in the modern market is severely limited. Therefore, there is a need for some method to improve the building penetration characteristics of millimeter wave transmission.

The present invention disclosed and described herein, in one aspect thereof, receives a signal of a first frequency located outside the building, which suffers loss upon penetration into the building, and wirelessly receives the received signal of the first frequency. And a system that enables signal penetration into the building, the first circuit converting to a first format that overcomes the loss caused by penetration into the building over the communication link. The first circuit implements a first transmit chipset for RF transmission in a first format corresponding to losses occurring upon penetration into a building, receives a signal at a first frequency, and builds a received signal at a first frequency. And a first transceiver for converting to a first format that overcomes the loss caused by infiltrating therein. A second circuit located inside the building is communicatively linked to the first circuit to receive and transmit the converted received signal of the first format. The second circuit further includes a second transceiver for implementing the first transmission chipset and receiving a converted signal of the first format from the first transmitter outside the building and transmitting to the first transmitter.

For a more complete understanding, reference is now made to the description provided below in connection with the accompanying drawings below.
1A is a block diagram of a building penetration system.
1B illustrates the bidirectional nature of a building penetration system for transmission from the outside.
1C shows the bidirectional nature of a building penetration system for transmission from the inside.
FIG. 1D shows the network layout of the building penetration system of FIG. 1A.
2 illustrates millimeter wave transmission between a receiver and a base station located both inside and outside the building structure.
3A shows a block diagram of an optical bridge for delivering millimeter wave transmission through a window.
3B shows a block diagram of an embodiment where the received signal is down converted to a level where it is more easily transmitted through a window or wall.
4 is a more detailed block diagram of millimeter wave regeneration and retransmission circuitry.
5 shows misalignment losses associated with millimeter wave regeneration and retransmission circuitry.
6 shows an RF transceiver circuit of millimeter wave regeneration and retransmission circuitry.
7 shows an optical focusing circuit of millimeter wave regeneration and retransmission circuitry.
8 illustrates various techniques for increasing spectral efficiency in a transmitted signal.
9 illustrates certain techniques for increasing spectral efficiency in a transmitted signal.
10 shows a general overview of a method of providing communication bandwidth between various communication protocol interfaces.
11 illustrates a method using multi-level overlay modulation over twisted pair / cable interface.
12 shows a general block diagram for processing multiple data streams in an optical communication system.
13 is a functional block diagram of a system for generating orbital angular momentum in a communication system.
14 is a functional block diagram of the orbital angular momentum signal processing block of FIG.
15 is a functional block diagram illustrating a method for removing orbital angular momentum from a received signal comprising multiple data streams.
FIG. 16 shows signal wavelengths with two quantum spin polarizations providing infinite number of signals with varying orbital angular momentum.
17A shows a plane wave with only a change in spin angular momentum.
17B shows a signal to which both spin and orbital angular momentum are applied.
18A-18C show various signals with different orbital angular momentum applied.
18D shows propagation of pointing vectors for various eigen modes.
18E shows a spiral phase plate.
19 illustrates a multilevel overlay modulation system.
20 shows a multilevel overlay demodulator.
21 illustrates a multilevel overlay transmitter system.
22 illustrates a multilevel overlay receiver system.
23A-23K illustrate exemplary multilevel overlay signals and their respective spectral power densities.
24 shows a comparison of multi-level overlay signals in the time and frequency domain.
25 illustrates the spectral alignment of a multilevel overlay signal for different signal bandwidths.
26 shows an alternative spectral alignment of the multi level overlay signal.
27 shows power spectral densities for various signal layers using a combined three layer multi level overlay technique.
FIG. 28 shows power spectral density on a logarithmic scale for layers using the combined three layer multi level overlay technique.
FIG. 29 shows a comparison of bandwidth efficiency for square root raised cosine versus multi-layer overlay for a symbol rate of 1/6.
30 shows a bandwidth efficiency comparison for square root raised cosine versus multi-layer overlay for a symbol rate of 1/4.
FIG. 31 shows a performance comparison between square root raised cosine and multi level overlay using ACLR.
32 shows a performance comparison between square root raised cosine and multi-level overlay using out-of-band power.
33 shows a performance comparison between square root raised cosine and multi-level overlay using band edge PSD.
34 is a block diagram of a transmitter subsystem for use with a multi level overlay.
35 is a block diagram of a receiver subsystem using a multilevel overlay.
36 shows an equivalent discrete time orthogonal channel of a modified multi-level overlay.
37 shows a PSD of a multilayer overlay, a modified multilayer overlay, and a square root raised cosine.
38 shows a bandwidth comparison based on -40 dBc out-of-band power between multilayer overlay and square root raised cosine.
39 shows an equivalent discrete time parallel orthogonal channel of a modified multilayer overlay.
40 shows the channel power gain of a parallel orthogonal channel of a modified multilayer overlay with three layers and T _sym = 3.
FIG. 41 shows a spectral efficiency comparison based on ACLR1 between a modified multilayer overlay and a square root raised cosine.
42 shows a spectral efficiency comparison between a modified multilayer overlay and square root raised cosine based on OBP.
FIG. 43 shows a spectral efficiency comparison based on ACLR1 between a modified multilayer overlay and a square root raised cosine.
FIG. 44 shows a spectral efficiency comparison based on OBP between a modified multilayer overlay and a square root raised cosine.
45 shows a block diagram of a baseband transmitter for a low pass equivalent modified multilayer overlay system.
46 shows a block diagram of a baseband receiver for a low pass equivalent modified multilayer overlay system.
47 illustrates a free space communication system.
48 shows a block diagram of a free space optical system using orbital angular momentum and multi-level overlay modulation.
49A-49C illustrate a manner of multiplexing a plurality of data channels into an optical link to achieve higher data capacity.
49D shows a group of concentric rings for a wavelength with multiple OAM valves.
50 illustrates a WDM channel comprising a plurality of orthogonal OAM beams.
51 illustrates one node of a free space optical system.
52 shows a network of nodes in a free space optical system.
53 illustrates a system for multiplexing between a free space signal and an RF signal.
54 illustrates an embodiment using a horn antenna for data transmission through a window or wall.
FIG. 55 illustrates downlink loss in the embodiment of FIG. 54.
FIG. 56 illustrates uplink signal strength in the embodiment of FIG. 54.
FIG. 57 illustrates the uplink signal strength when the power amplifier is located inside a building in the embodiment of FIG. 54.
FIG. 58 illustrates the gain and loss on the downlink of the embodiment of FIG. 59 when no power amplifier is included.
FIG. 59 illustrates signal strength at various points in the uplink when no power amplifier is provided in the embodiment of FIG. 54.
FIG. 60 illustrates shielding used with the embodiment of FIG. 56.
FIG. 61 illustrates an embodiment of using a patch antenna to transmit data through a window or wall.
FIG. 62 illustrates a patch antenna array used in the embodiment of FIG. 61.
63 shows a patch antenna of a patch antenna array.
64 shows antenna gain simulation for a patch antenna.
65 illustrates generation of a high directivity, high gain beam using a patch antenna array.
66 illustrates another embodiment of a microstrip patch antenna array.
67 shows a patch antenna element.
68 shows the electron radiation field of the patch antenna.
69 shows a top view of a multilayer patch antenna array.
70 shows a side view of a multilayer patch antenna array.
71 shows a first layer of a multilayer patch antenna array.
72 shows a second layer of a multilayer patch antenna array.
73 illustrates a transmitter for use with a multilayer patch antenna array.
74 illustrates multiplexed OAM signals transmitted from a multilayer patch antenna array.
75 illustrates a receiver for use with a multilayer patch antenna array.
76 shows a 3D model of a single rectangular patch antenna.
FIG. 77 illustrates a radiation pattern of the patch antenna of FIG. 10.
78A shows the radiation pattern of the circular array for OAM mode order l = 0.
78B shows the radiation pattern for OAM mode order l = 0 near the array axis.
78C shows the radiation pattern for OAM mode order l = 1 near the array axis.
78D shows the radiation pattern for OAM mode order l = 2 near the array axis.
79 is a flowchart showing a design and layout process of a patch antenna.
80 is a flow chart showing a process of patterning a copper layer on a laminate for a patch antenna.
81 is a flowchart showing a test process of the manufactured patch antenna.
82A illustrates an embodiment for transmitting an RF signal through a window or wall using an RF transceiver chipset.
82B illustrates one embodiment of a system for transmitting wireless signals through a window or wall using a Ferraso chipset.
83 illustrates an implementation of a repeater using a Ferraso chipset.
84A is a top-level block diagram of a Peraso transceiver.
84B and 84C show detailed application diagrams of the Peraso chipset.
85 illustrates serial transmission between Peraso transceivers.
86 illustrates parallel transmission between a Peraso transceiver.
87 is a functional block diagram of a Peraso transceiver located in an external window.
88 shows a method of using a laser to power an external Peraso transmitter.
89 illustrates alignment holes in the VCSEL.
90 illustrates the use of alignment holes to align the optical circuit of the VCSEL.
91 illustrates optical power coupling between VCSELs.
92 illustrates a method for powering external system components using a solar panel.
93 illustrates a method of powering external system components using a laser.
94 illustrates a method of powering external components from an internal power source using inductive coupling.
95 shows a pair of circular loops connected by mutual inductance.
96 shows a table providing information related to coil efficiency.
97 is a schematic diagram of a coil coupled via inductive coupling.
98 is a schematic diagram of coils coupled through magnetic resonance.
99 shows a functional block diagram of a self-resonant wireless power transfer system.
100 is a schematic diagram of a magnetic coupling resonator.
101 is a schematic diagram of a simple power generation circuit.
102 schematically illustrates the use of impedance matching to overcome eddy current losses.
FIG. 103 illustrates a perspective view of a Peraso transceiver circuitry located outside and inside the structure.
104 illustrates a side view of a Peraso transceiver circuitry located outside and inside the structure.
105 shows a table of various parameters related to the transmission of signals through the window glass.
106 shows another table of various parameters related to signal transmission through the window glass.
107 shows an additional table of various parameters related to signal transmission through the window glass.
108 shows how a millimeter wave system can be combined with a home IP network to provide broadband data transmission.
109 shows a functional block diagram of a combined IP home network system.
110 is a functional block diagram of a home IP network system.
111 shows how a millimeter wave system can be used to send information to a home IP network system.
112 shows a first embodiment for wireless transmission of broadband data to a home IP network system.
113 shows a second embodiment for the wireless transmission of broadband data to a home IP network system.
114 shows a third embodiment for wireless transmission of broadband data to a domestic IP network system.
115 illustrates a combined optical data transmission system and RF data transmission system for providing broadband data to a home gateway.
116 illustrates how a load balancing technique may be used to control the network configuration between the optical network data transmission system and the RF network data transmission system.
117 illustrates various optical connections between the central office and the customer premises.
118 illustrates a GPON architecture.
119 shows upstream and downstream GTC frames.
120 illustrates a downstream GTC frame format.
121 illustrates an upstream GTC frame format.
FIG. 122 illustrates a Virtual Optical Line Termination Hardware Abstraction (vOLTHA) layer.
123 shows an implementation of vOLTHA on OLT and ONU links.
124 illustrates a broadband link between an OLT and a home gateway.
125 illustrates an interface between an ONU and a plurality of home gateways.
126 shows a first embodiment of a broadband data communication link between an OLT and a home gateway.
127 shows a second embodiment of a broadband data communication link between an OLT and a virtual reality goggles.
128 is a functional block diagram of a 60 GHz transceiver dongle.
129 shows a six byte MAC address for an Ethernet interface in one of the broadband communication links.
130 illustrates a switch in the aforementioned PON network.

Referring now to the drawings, various drawings and embodiments of millimeter wave regeneration and retransmission for building penetration and various embodiments related thereto are shown and described, and other possible embodiments are also described, and like elements throughout Similar abbreviations have been used to represent. The drawings are not necessarily drawn to scale, and in some cases the drawings have been exaggerated and / or simplified in various places for illustrative purposes only. Those skilled in the art will understand various possible applications and modifications based on the examples of possible embodiments below.

One problem with wireless communications is that high frequency RF waves cannot penetrate the windows and walls of homes and offices. If the window includes infrared (IR) shielding to save energy in a home or office building, the loss of signal transmitted through the shield is typically up to 40 or 50 dB. Thus, the millimeter wave systems described herein provide the ability to provide tunneling of such optical and high frequency radio waves without the need to drill holes in glass, windows, or buildings, thereby providing a physical portal through them, thereby greatly enhancing wireless communication technology. This can provide an advantage. This can be done at any frequency with the problem of penetrating glass or buildings. Glass is one of the most popular and versatile because of its continuous improvement in solar and thermal performance. One way to achieve this performance is to use passive and solar controlled low emissivity coatings. Such low emissivity glass materials cause significant losses in millimeter wave spectral transmission and cause significant problems in millimeter wave transmission through such glass. The system described below provides the ability to handle frequencies that are having trouble passing through glass or buildings so that signals can be transmitted into and out of the home or building.

Millimeter wave signaling was developed when the FCC devised a band plan to create a local multipoint distribution service (LMDS) spectrum of 1300 MHz that could be used within each basic trading area across the United States. The plan allocated two LMDS licenses per "BTA", "A Block" and "B Block", respectively. The A block license includes a total of 1150 MHz of bandwidth and the B block license consists of a total of 150 MHz of bandwidth. Licensee Telegent has developed a system for fixed wireless point to multipoint technology that can transfer high-speed broadband from the rooftop to nearby small and medium-sized businesses. However, this system as well as other systems provided by Winstar and NextLink have not been successful, and many LMDS licenses have been passed back to the FCC hands. These licenses and related spectrum are likely to be useful for 5G testing and services.

Referring now to FIG. 1A, a general block diagram of a building penetration transmission system 102 is shown. The building penetration transmission system 102 uses a 5G fixed millimeter wave arrangement to overcome high building penetration losses due to RF and optical obstacles such as windows, bricks and concrete walls. Building penetration transmission system 102 greatly increases the number of enterprise and home buildings in which 5G millimeter wave signals can be used to provide Gigabit Ethernet services. The system provides an optical or RF tunnel through the window or wall 106 without the need for drilling of any hole in the window or wall or the creation of any type of signal transmissive portal. Generating directional propagation using the described system allows the generation of directional beams to tunnel through low glass or walls. This system can meet the link cost between the internal transceiver and the external transceiver. The system significantly increases the number of buildings that can use millimeter-wave signals to deliver Gigabit Ethernet using consumer installations.

Building penetration transmission system 102 generally includes an external repeater transmitter 104 located outside the window or wall 106. Repeater transmitter 104 transmits multiple frequencies including 2.5 GHz band, 3.5 GHz band, 5 GHz band, 24 GHz band, 28 GHz band (A1, A2, B1 and B2), 39 GHz band, 60 GHz band, 71 GHz band and 81 GHz band And receive. The 3.5 GHz band is CBRS (Citizens Band Radio Service), the 60 GHz band is the V band, and the 71 GHz and 81 GHz are the E bands. The repeater transmitter 104 is powered using magnetic resonance or inductive coupling such that the external unit does not require an external power source. Repeater 104 transmits the signal received through window or wall 106 to transceiver 108 located inside the building. The transceiver 108 includes an antenna 110 for providing an Ethernet and / or power connection. Building penetration transmission system 102 may provide 1 gigabit per second throughput traffic tunneling through building structures such as windows or walls. The transceiver 108 may include a port 112 that provides femtocell connectivity, but generally uses the antenna 110 to transmit Wi-Fi indoors. Alternatively, an Ethernet or power connection may be hard wired to the transceiver 108. The building penetration transmission system 102 may be located at any point in the wall or window of the structure. The building penetration transmission system 102 is designed to work with different types of walls and windows such that millimeter wave signals can penetrate different types of structures. Repeater 104 and transceiver 108 are constructed of metal / plastic designs to withstand the harshest environments, including precipitation, hot / cold weather, and high / low humidity.

The transceiver 108 includes a gigabit Ethernet port, a power output, at least one USB 2.0 port, and dual flash image support. The building penetration transmission system 102 provides a range of up to 200 feet (60 meters). The system requires 24V / M passive Gigabyte PoE, and in one embodiment has a power consumption of up to 20W that can be powered using self resonant wireless charging. The system provides a channel bandwidth of 60 GHz with 2 GHz.

1B and 1C show bidirectional communication between a transceiver 104 located on the outside side of the window or wall 106 and a transceiver 108 located on the inside side of the window or wall 106. The remote base station transmitter 109 transmits the radio signal to the external transceiver 104. Communication transmissions from the external transceiver 104 to the internal transceiver 108 occur over the communication link 114. The signals transmitted therein may be sent from the internal router 115 to the consumer premises equipment (CPE) 111 using beam shaping or WiFi 113. As shown in FIG. 1C, an internal device 117 (eg, mobile device or Internet of Things device) transmits a signal to an internal router 115. The internal router 115 provides a signal to the internal transceiver 108. The transmission from the inside of the window or wall 106 to the outside is on the communication link 116 from the transceiver 108 to the transceiver 104. The external transceiver 104 then sends a signal to the external base station 109. Thus, the system allows for bi-directional communication that can utilize RF, optical or other types of communication techniques as described in more detail below.

Referring now to FIG. 1D, a network arrangement of the building penetration system described in connection with FIGS. 1A-1C is shown. The provider network 130 interfaces with the local network through an optical fiber point of presence cabinet 132. Cabinet 132 has an optical fiber link 134 to access point 136. Each access point 136 is located on a light pole of a local area via a wireless communication link using any number of communication frequencies, for example, as described herein. Communicate wirelessly with your network. The access point 136 communicates with a transceiver system 138 comprising a building penetration system as described herein, wherein the signal is wirelessly transmitted to an external transceiver and then transmitted to the interior of a business or home so that the information is stored. It may be transmitted in both directions from / to the provider network 130 to devices located within the various structures. In this way, data may be provided between the network provider 130 and any type of devices located within the structure, using wireless communications that would typically not be able to penetrate the structure due to the loss caused by the signal penetrating into the structure. Can be.

Referring now to FIG. 2, a millimeter wave transmission system 102 for communication is shown. Base station 104 generates millimeter wave transmissions 206 and 208 for transmission to various receivers 210 and 202. The millimeter wave transmission 206 that is moved directly from the base station 204 to the receiver 210 can be easily received without much ambient interference. The millimeter wave transmission 208 from the base station 204 to the receiver 212 located inside the building 214 will have a significant interference problem. The millimeter wave transmission 208 does not easily penetrate the building 214. When passing through transparent windows or buildings, you experience significant signal loss. Frequencies above 28 GHz do not penetrate building walls and windows, and 85% of communications traffic comes from inside buildings.

Given that millimeter wave spectral transmissions do not propagate very far and lack the ability to penetrate the room, these frequencies will be used in very short range applications of approximately one mile. As a prospect, low-power Wi-Fi at 2.4 GHz can cover most homes less than 3000 square feet (sq. Ft.), But 5 GHz Wi-Fi signals can't travel farther in the higher frequency range. It will cover only about 60% of storey houses. For 5G applications, the power is higher, but the higher the frequency, the greater the loss when propagating through space and other media.

The losses incurred by millimeter-wave signals penetrating a building lower the data rate to near zero. For example, when transmitting on a downlink from a base station through transparent glass onto a home or building interior, the maximum data rate is 9.93 Gb per second. When transmitting over color, the data rate is 2.2 Mb per second. When sending over bricks, the data rate is 14Mb per second, and when sending over concrete, the data rate drops completely to 0.018bps. Likewise, when transmitting on the uplink from inside a building to the base station, the maximum data rate through the transparent glass is 1.57 Gb per second and 0.37 Mb per second through the colored glass. The signal transmitted on the uplink has a data rate of 5.5 Mb per second when transmitted through bricks and 0.0075 bits per second when transmitted through concrete. In addition, there are differences for downlink and uplink in transmissions to / from old buildings or new buildings. Old buildings are defined as buildings that use a composite model that includes 30% standard glass and 70% concrete walls. The new building is defined as a composite model that includes 70% infrared reflective glass (IRR glass) and 30% concrete walls. Base station transmission on the downlink into the building is 32 Mb per second for the old building and 0.32 Mb per second for the new building. Similarly, the uplink transmission from the home / building to the base station is 2.56 Mb per second for the old building and 25.6 kb per second for the new building.

Despite these drawbacks, in order to meet the increasing demand for bandwidth, RF service providers will gradually move to higher frequency carrier frequencies. In particular, 28 GHz is a promising frequency band for providing a local multipoint distribution service (LMDS). The 28 GHz and 39 GHz frequency bands are being considered by the FCC for small cell deployment to support 5G networks up to subscriber premises using beamforming and beam steering. These high frequency bands have many advantages, but also have the disadvantages caused by large penetration losses when passing through building materials or windows. These advantages include higher frequency rates, more sophisticated beam shaping capabilities, and more effective beam steering at smaller footprints of components that provide millimeter wave frequencies.

3A illustrates one way to transmit millimeter wave signals into a building using an optical bridge 302 mounted to window 304. Optical bridge 302 includes a first portion 306 included outside of window 304 and a second portion 308 included inside of window 304. The first portion 306 includes a 28 GHz transceiver 310 mounted outside the window 304. The 28 GHz transceiver 310 receives, for example, millimeter wave transmissions being transmitted from the base station 104 as described in connection with FIG. 1. The received / transmitted signal is transmitted to and from the transceiver 310 using a receiver optical subassembly (ROSA) / transmitter optical subassembly (TOSA) 312. The receiver optical assembly is a component used to receive an optical signal in a fiber optic system. Similarly, transmitter optical subassemblies are components used to transmit optical signals in optical fiber systems. ROSA / TOSA component 312 transmits or receives an optical signal through window 304 to ROSA / TOSA component 314 located within window 304. These signals are sent from the ROSA / TOSA 314 to the Wi-Fi transceiver 316 for in-building transmission.

3B illustrates another embodiment where a received frequency that does not readily pass through the colored window or wall 330 down converts the received signal to facilitate communication between the window or wall 330. Outside the building, signals are received at the antenna 332 of the transceiver 334 at a frequency that does not readily pass through a window or wall. The transceiver 334 sends the signal to a down / up converter 336 to down convert the signal into a frequency band that more easily passes through the window / wall 330. The other transceiver 338 takes the frequency down converted signal from the converter 336 and transmits it through the window / wall 330. The transmitted signal is received by a transceiver 340 located inside the building at a down converted frequency. The received signal is sent to up / down converter 342 to convert the signal to a level for transmission inside the building. In many cases, this can be a Wi-Fi band. The up-converted signal is sent to the router 344 for transmission inside the building. Outgoing signals received from devices located inside the building are processed and transmitted in the reverse manner to transmit signals from the transceiver 334 to the outside of the building.

Referring now to FIG. 4, a more detailed view of a component for sending millimeter wave transmissions through a window or wall of a building is provided. The transceiver 210 includes an optional antenna gain element 402 for receiving millimeter wave transmissions transmitted from the base station 204 onto the down / up link 404. The down / up link 404 includes 28 GHz beam transmission. However, other frequency transmissions may also be used. The RF receiver 406 is used to receive information from the base station 204 via the down / up link 404. Similarly, RF transmitter 408 is used to send information to base station 204 over down / up link 404. The received signal is provided to a demodulator 410 for demodulation of any received signal. The demodulated signal is provided to a groomer 412 which places the signal in a suitable configuration for transmission by the optical transmission component. When converting other modulations (in other words, from higher-order QAM to on-off keying), a signaling conversion that requires some grooming (or signal conditioning) to ensure that all bits are properly converted and still provide low BER. This exists. The system converts from RF at high QAM rate to the native bit rate of the OOK to allow transmission using the VCSEL to pass through the window. Since VCSEL only works with OOK, conversion with groomer 412 is required. If the received signal is down-converted from 28 GHz directly to 5.8 GHz (because 5.8 GHz passes through walls and glass), then there is no need to worry about the complexity of converting to lower order modulation. The problem is that down-converting signals from 28 GHz to 5.8 GHz require expensive components. The groomer 412 completes the conversion of the received 28 GHz signal to frequency for transmission over glass or wall without the need for more expensive components.

The signals to be transmitted pass through an amplifier 414 which amplifies the signal for transmission. VCSEL 416 is a vertical cavity surface emitting laser, which is a type of semiconductor laser diode with a laser beam missing vertically from the top surface. In a preferred embodiment, the VCSEL 416 includes a Finisa VCSEL having a wavelength of approximately 780 nm, a modulation rate of 4 Gb per second, and an optical output power of 2.2 mW (3.4 dBm). In alternative embodiments, components for optical signal transmission through window 404 may include LEDs (light emitting diodes) or ELLs (edge emitting lasers). Different lasers allow for different light retransmissions at different frequencies based on different characteristics of the window, for example colored. The VCSEL 416 includes a transmission optical subassembly (TOSA) for generating an optical signal for transmission from the VCSEL 416 to the VCSEL 418 located outside the window 204.

VCSELs 416 and 418 include laser sources for generating optical signals for communication across window 404. In one embodiment, the VCSEL includes a Finisa VCSEL that provides a 780 nm optical signal having a maximum modulation rate of 4 Gb per second and an optical output power of 3 mW (5 dBm) when operating at 1 Gb per second. The TOSA includes a laser device or LED device for converting an electrical signal from the amplifier 414 into an optical signal transmission. Transmissions from the outer VCSEL 416 are directed to the inner VCSEL 418 and associated receiver optical assembly (ROSA).

The optical signal is transmitted through window 404 using optical focusing circuit 417. Optical focusing circuit 417 will be described in more detail with respect to the transmitter and receiver side in conjunction with FIG. The optical link 428 between VCSEL 416 and VCSEL 418 still defines an acceptable loss while transferring information between the VCSELs 416 and 418. Has The VCSEL has an output power of approximately 5dBm. The detector of the receiver in the VCSEL can detect a signal of approximately -12 dBm. The glass loss associated with the optical signal passing through the glass at 780 nm wavelength is 7.21 dB. The coupling loss and lens gain associated with the transmission is approximately 0.1 dB. The maximum displacement loss caused by a 3.5mm lens displacement is 6.8dB. Therefore, the total link margin is equal to 2.88 dB based on the subtraction of detector sensitivity, glass loss, coupling loss and lens gain, and maximum displacement loss from the VCSEL output power. 2.88dB link margin is provided for additional unexpected losses such as lens loss and unexpected power fluctuations.

Lens displacement or misalignment can account for a significant portion of link loss in the system. As shown in FIG. 5, the acceptable range of misalignment 402 ranges from approximately -6.5 mm to 6.5 mm from the center of the power spectrum received by the detector. Misalignment loss 404 is in the range of 0.6 dB to 6.8 dB when the misalignment moves between ± 6.5 mm. As shown at 406, the maximum allowable misalignment loss is 9.4 dB.

The VCSEL 418 inside the window 204 uses TOSA to transmit optical signals through the window 204 to the ROSA in the VCSEL 416 located outside the window at a data rate of 0.5 Gbps. The received optical signal is provided to a de-groomer component 32 for processing from the raw bit rate of the OOK to high QAM rate RF to enable RF transmission after signal reception by the VCSEL. The degroomed signal is modulated in modulator 422. The modulated signal is transmitted over the uplink 404 using the RF transmitter 408. Transceiver 310 is powered by power input 424, and components inside the window are similarly powered by power input 426. The signal is provided inside the building using a Wi-Fi transmitter 428 connected to receive the optical signal received by the VCSEL 418 and provide the signal to the VCSEL 418 for transmission through the window 304. . Wi-Fi transmitters use the 802.11 transport protocol.

Referring now to FIG. 6, a more detailed block diagram of the transceiver 310 is shown. Receiver portion 602 includes an RF receiver 604 for receiving an RF signal transmitted from a base station on the downlink 606. Receiver 604 generates an output signal with real part BBI 608 and imaginary part BBQ 610. The RF receiver 604 generates a real signal 608 and an imaginary signal 610 in response to the received signal and the input from the phase locked loop / voltage controlled oscillator 605. Phase locked loop / voltage controlled oscillator 605 provides an input to RF receiver 604 in response to a reference oscillator signal provided from reference oscillator 607 and a voltage controlled oscillator signal provided from oscillator 609. Real signal 608 and imaginary signal 610 are provided to analog-to-digital converter 612 for conversion to a digital signal. Analog-to-digital converter 612 is clocked by an associated clock input 614 provided from clock generation circuit 616. Clock generation circuit 616 also receives an input from reference oscillator 607. Real and imaginary digital signals 618 and 520 are input to digital down converter 622. These digital signals are down converted to low frequencies and output as a beam stream 624 to the optical transmission circuit VCSEL for transmission across the glass panes.

Transmitter portion 624 receives the digital bitstream 626 from the optical circuit and provides the bitstream to the real and imaginary portions of the digital up-converter 628 to convert the digital data to higher frequencies for transmission. . The real and imaginary parts of the up-converted digital signal are provided to a crest factor reduction processor (CFR) 630. Some signals (particularly OFDM-based systems) have a high peak-to-average power ratio (PAR) that negatively affects the efficiency of the power amplifier (PA). The CFR scheme implemented by the processor helps to reduce the PAR and is used for many networks (CDMA and OFDM). However, CFR schemes developed primarily for CDMA signals have poor performance when used in OFDM (given a tight error vector magnitude (EVM) requirement). With a well-designed CFR algorithm on the FPGA, high performance can be achieved by lowering latency and significantly reducing the PAR of the output signal, thereby improving PA efficiency and reducing cost.

Real and imaginary signals are provided from the CFR processor 630 to the digital to analog converter 632. Digital to analog converter 632 converts this real and imaginary digital signal into real and imaginary analog signals BBI 634 and BBQ 636. Real and imaginary analog signals are input to the RF transmitter 638. RF transmitter 638 processes real signal 634 and imaginary signal 636 in response to input from phase locked loop / voltage controlled oscillator 604 to generate and transmit millimeter waves and uplink 640. Generates an RF signal for transmission over it.

Referring now to FIG. 7, an optical focusing circuit 317 is shown coupled to an optical transmission interface across window 304. The optical focusing circuit 417 is included with the VCSELs located on each side of the window 304 and includes a transmitter 602 and a receiver 604. Transmitter 602 and receiver 604 will be included on each side of window 304 when the system provides two-way communication across the window. Transmitter 602 includes a VCSEL 606 provided by Finisa, which in one embodiment transmits a 780 nm optical signal at 4 Gb per second and has a power output of 3.42 dBm. The optical signal generated by the VCSEL 606 is provided to an acromatic doublet 608 having a focal length of 7.5 mm that collimates the optical signal generated by the VCSEL 606 into a small opening. Collimated beam 610 is transmitted across window 304. The collimated beam 610 exits the window 304 and passes through a biconvex lens 612 having a focal length of 25 mm on the receiver portion 604. The biconvex lens 612 focuses the beam column 610 onto the half ball lens 614 that focuses the optical signal onto the semiconductor aperture of the photo detector 616. In one embodiment, photo detector 616 has an aperture diameter of 10 mm and a detector sensitivity of 12 dBm.

The transmission between the VCSEL 606 and the RF transceiver 10, in one particular embodiment, is filed on Nov. 21, 2016, which is hereby incorporated by reference in its entirety in "SYSTEM AND METHOD FOR COMMUNICATION USING ORBITAL ANGULAR MOMENTUM WITH MULTIPLE. Orthogonal function signal transmission techniques such as those described in US Application Serial No. 15 / 357,808 entitled "LAYER OVERLAY MODULATION". However, it should be understood that various other data transfer techniques may be used.

7 shows two ways to increase the spectral efficiency of a communication system. In general, there are two basic ways to increase the spectral efficiency 702 of a communication system. This improvement can be achieved by the signal processing technique 704 in the modulation scheme, or by using multiple access techniques. In addition, spectral efficiency can be increased by creating a new eigen channel 706 in the electromagnetic wave. These two technologies are completely independent of each other, and one class of innovation can be added to the second class of innovation. Therefore, the combination of these technologies introduces new innovations.

Spectral efficiency 702 is a major driver of the business model of a communication system. Spectral efficiency is defined in bits / second / hz and the higher the spectral efficiency, the better the business model. This is because spectral efficiency can be interpreted as more users, higher throughput, higher quality or some of them in the communication system.

Regarding a technique using a signal processing technique or a plurality of access techniques. These technologies include innovations such as TDMA, FDMA, CDMA, EVDO, GSM, WCDMA, HSPA and the latest OFDM used in 4G WIMAX and LTE. Almost all of these techniques use decades-old modulation techniques based on sinusoidal eigenfunctions called QAM modulation. In a second class of technologies involving the creation of new eigen channels 706, the innovations include eigen channels and electromagnetic waves that are independent of the uncorrelated radio path, as well as diversity techniques including spatial and polarization diversity. It includes multiple inputs / multiple outputs (MIMO) that generate radio waves.

Referring now to FIG. 8, the communication system configuration introduces two techniques, one from signal processing technique 804 and one from the creation of a new eigen channel 806 category, which are completely independent of each other. These combinations provide a unique way to disrupt the access portions of end-to-end communications systems, from twisted pairs and cables to RF used in fiber optics, free space, cellular, backhaul, and satellites. The first technique involves the use of a new signal processing technique that uses a new orthogonal signal to upgrade QAM modulation using a non-sine function. This particular embodiment is called quantum level overlay (QLO) 902 shown in FIG. The second embodiment involves the application of a new electromagnetic wavefront that utilizes the characteristics of an electromagnetic wave or photon called orbital angular momentum (QAM) 904. Each application of quantum level overlay technology 902 and orbital angular momentum application 904 uniquely provides a further higher spectral efficiency 906 within the communications system with which they are combined.

With respect to quantum level overlay technology 902, a new eigen function is introduced that greatly increases the spectral efficiency of the system when overlapped (on top of each other within one symbol). Quantum level overlay technology 902 borrows from quantum mechanics a special orthogonal signal that reduces the time bandwidth product to increase the spectral efficiency of the channel. Each orthogonal signal overlaps within a symbol and serves as an independent channel. This independent channel differentiates the present technology from existing modulation techniques.

In connection with the orbital angular momentum application 904, this embodiment introduces a twisted electromagnetic wave or light beam, with a helical wavefront carrying the orbital angular momentum (QAM). Different OAM carrying waves / beams can be orthogonal to one another in the spatial domain, which allows the waves / beams to multiplex and demultiplex efficiently within the communication link. OAM beams are noted in communications because of their potential for special multiplexing multiple independent data carrying channels.

With regard to the combination of quantum level overlay technology 902 and orbital angular momentum application 904, this combination is unique because the OAM multiplexing technology is compatible with other electromagnetic wave technologies such as wavelength and polarization split multiplexing. This means the possibility of further increasing system performance. The combination of these technologies in high-capacity data transmissions disrupts the access portion of end-to-end communication systems, from twisted pairs and cables to RF in fiber optics, free-space optics, cellular / backhaul, and satellites.

Each of these techniques can be applied independently of each other, but this combination not only increases spectral efficiency, but also provides a unique opportunity to increase spectral efficiency without sacrificing distance or signal to noise ratio.

Using "Shannon Capacity Equation", a determination can be made whether the spectral efficiency is increased. This can be mathematically translated into more bandwidth. Because bandwidth is valuable, spectral efficiency gains can be easily converted into economic gains for business impacts using higher spectral efficiency. In addition, when sophisticated forward error correction (FEC) techniques are used, the net impact is higher quality, but some bandwidth is sacrificed. However, if higher spectral efficiency (or more virtual bandwidth) can be achieved, some of the gain bandwidth for FEC can be sacrificed, and thus higher spectral efficiency can also be converted to higher quality.

Operators and suppliers are interested in increasing spectrum efficiency. However, the problem with it is cost. Each technology in a different layer of the protocol has a different price tag associated with it. Techniques implemented at the physical layer have the greatest impact since other techniques can be superimposed on top of the underlying layer techniques, thus further increasing spectral efficiency. The price list for some technologies may be drastic when considering other associated costs. For example, multiple input multiple output (MIMO) technology utilizes additional antennas to create additional paths where each RF path can be treated as an independent channel, increasing the total spectral efficiency. In the MIMO scenario, the operator has other associated soft costs that deal with structural issues such as antenna installation. Not only are these technologies expensive, they have enormous time problems because structural activities take time, and achieving higher spectral efficiencies can lead to significant financial losses.

Quantum level overlay technology 902 has the advantage that independent channels are created in the symbol without requiring a new antenna. This will have enormous cost and time advantages over other technologies. In addition, the quantum level overlay technology 902 is a physical layer technology, which means that there is another technology at the higher layer of the protocol that can all ride on top of the QLO technology 902, thus making the spectral efficiency even more. Increase. QLO technology 902 utilizes standard QAM modulation used in OFDM-based multiple access techniques such as WIMAX or LTE. QLO technique 902 basically intensifies QAM modulation at the transmitter by injecting new signals into the baseband I & Q components and superimposing them prior to QAM modulation, which will be described in more detail below. At the receiver, the reverse process is used to separate the superimposed signals, and the net effect is pulse shaping which allows for better localization of the spectrum compared to standard QAM or even root raised cosine. The impact of this technique is much higher spectral efficiency.

Referring now to FIG. 10 in more detail, in order to increase the number of communication channels, a combination of applications of multi-level overlay modulation 1004 and orbital angular momentum 1006 have been enhanced at various communication protocol interfaces 1002. A general overview of how to provide communication bandwidth is shown. The discussion below about orbital angular momentum processing and multi-level overlay modulation illustrates two techniques, which may or may not be implemented with RF transmissions in the systems and embodiments described below. The RF transmission may be configured to implement one or both of the techniques in the described embodiments, or both.

Various communication protocol interfaces 1002 include various communication links, such as RF communications, wired communications such as cable or twisted pair connections, or optical communications utilizing optical wavelengths, such as fiber optic communications or free space optical communications. Various types of RF communications include RF microwave or RF satellite communications, as well as a combination of real time multiplexing between RF and free space optics.

By combining the multi-level overlay modulation technique 1004 with the orbital angular momentum (OAM) technique 1006, high throughput on various types of communication links 1002 can be achieved. Using multilevel overlay modulation alone without OAM increases the spectral efficiency of the communication link 1002, which is wired, optical, or wireless. However, with OAM, the spectral efficiency increase is even greater.

Multilevel overlay modulation technique 1004 provides new degrees of freedom beyond conventional two degrees of freedom, where time T and frequency F are independent variables in a two-dimensional notational space defining an orthogonal axis in the information diagram. . This involves a more general approach than modeling a fixed signal in either the frequency or time domain. Previous modeling methods using fixed time or fixed frequency are considered a more restrictive case of the general approach using multi-level overlay modulation 1004. In multi-level overlay modulation technique 1004, signals may be differentiated in two-dimensional space rather than along a single axis. Therefore, the information transfer capacity of the communication channel can be determined by a number of signals occupying different time and frequency coordinates, and can be differentiated in the notation two-dimensional space.

In the notational two-dimensional space, the minimization of the time-bandwidth product, ie, the area of the signal occupied in that space, allows for more dense packing, thus enabling more signal usage in the allocated channel and more resulting information delivery capacity. Let's do it. Given a frequency channel difference Δf, a given signal delivered through it at minimum time Δt will have an envelope described by a particular time-bandwidth minimization signal. The time bandwidth product for these signals takes the form:

Where n is an integer ranging from 0 to infinity and represents the order of the signals.

These signals form an orthogonal set of infinite elements, where each has a finite amount of energy. They are finite in both the time domain and the frequency domain and can be detected from a mixture of other signals and noise through correlation, for example by match filtering. Unlike other wavelets, these orthogonal signals have similar signal and frequency forms.

Orbital angular momentum process 1006 provides a twist for the wavefront of an electromagnetic field carrying a data stream that may enable the transmission of multiple data streams on the same frequency, wavelength, or other signal support mechanism. This will increase the bandwidth on the communication link by allowing a single frequency or wavelength to support multiple eigen channels, with each individual channel having a different orthogonal and independent orbital angular momentum associated with it.

Referring now to FIG. 11, there is shown an additional communication implementation technique utilizing the techniques described above as a twisted pair or cable that carries electrons (not photons). Rather than using multilevel overlay modulation 1004 and orbital angular momentum technology 1006, respectively, only multilevel overlay modulation 1004 is associated with a single wired interface, more specifically a twisted pair communication link or cable communication link 1002. Can be used together. The operation of the multi-level overlay modulation 1004 is similar to that described above in connection with FIG. 10, but is used by itself without using the orbital angular momentum technique 1006, and with a twisted pair communication link or cable interface communication link 1002. Used together.

Referring now to FIG. 12, shown is a general block diagram of processing multiple data streams 1202 for transmission in an optical communication system. Multiple data streams 1202 are provided to one multilayer overlay modulation circuit 1204 where the signals are modulated using a multilayer overlay modulation technique. The modulated signal is provided to an orbital angular momentum processing circuit 1206 that applies torsion to each wavefront being transmitted over the wavelength of the optical communication channel. Torsional waves are transmitted over the optical interface 1208 over an optical communication link, such as an optical fiber or a free space optical communication system. 12 may illustrate an RF mechanism in which the interface 1208 includes an RF interface rather than an optical interface.

Referring now in more detail to FIG. 13, an orbital angular momentum in a communication system as described in connection with FIG. 10 to provide a data stream that can be combined with a plurality of other data streams for transmission over the same wavelength or frequency. A functional block diagram of a system for creating a "torsion" is shown. Multiple data streams 1302 are provided to transmission processing circuitry 1300. Each data stream 1302 includes, for example, an end-to-end link connection carrying a voice call or a packet connection carrying non-circuit switch packed data on the data connection. Multiple data streams 1302 are processed by modulator / demodulator circuit 1304. The modulator / demodulator circuit 1304 modulates the received data stream 1302 on a wavelength or frequency channel using a multilevel overlay modulation technique, which will be described in more detail below. The communication link may include an optical fiber link, a free space optical link, an RF microwave link, an RF satellite link, a wired link (no twist), and the like.

The modulated data stream is provided to an orbital angular momentum (OAM) signal processing block 1306. Each modulated data stream from the modulator / demodulator circuit 1304 is provided with a different orbital angular momentum by the orbital angular momentum electromagnetic block 1306 such that each modulated data stream has its own different orbital angular momentum associated therewith. . Each modulated signal having this associated orbital angular momentum is provided to an optical transmitter 1308 that transmits each modulated data stream having a unique orbital angular momentum on the same wavelength. Each wavelength has a selected number of bandwidth slots B and can increase its data transfer capacity by a factor of the number of degrees of the orbital angular momentum l provided from the OAM electromagnetic block 1306. An optical transmitter 1308 that transmits signals at a single wavelength may transmit B groups of information. The optical transmitter 1308 and the OAM electromagnetic block 1306 may transmit l × B groups of information in accordance with the configurations described herein.

In the receive mode, the optical transmitter 1308 will have a wavelength that includes a plurality of signals transmitted therein with different orbital angular momentum signals embedded therein. The optical transmitter 1308 sends these signals to an OAM signal processing block 1306 that separates each of the signals having different orbital angular momentum and provides the separated signal to a demodulator circuit 1304. The demodulator process extracts the data stream 1302 from the modulated signal and provides it to the receiving end using a multilayer overlay demodulation technique.

Referring now to FIG. 14, a more detailed functional description of OAM signal processing block 1406 is provided. Each input data stream is provided to an OAM circuit 1402. Each OAM circuit 1402 provides a different orbital angular momentum to the received data stream. Different orbital angular momentum is achieved by applying different currents for the generation of the signal being transmitted to produce the specific orbital angular momentum associated with it. The orbital angular momentum provided by each OAM circuit 1402 is unique to the data stream provided to it. Infinite orbital angular momentum can be applied to different input data streams using a number of different currents. Each individual generated data stream is provided to a signal combiner 1404 that combines the signals on the wavelength for transmission from the RF receiver 1406.

Referring now to FIG. 15, a manner in which the OAM processing circuit 1306 can separate a received signal into a plurality of data streams is shown. Receiver 1502 receives the combined OAM signal onto a single wavelength and provides the information to signal separator 1504. Signal separator 1504 separates each of the signals having different orbital angular momentum from the received wavelength and provides the separated signal to OAM torsion canceling circuit 1506. OAM torsion canceling circuit 1506 removes the associated OAM torsion from each associated signal and provides the removed modulated data stream for further processing. The signal separator 1504 separates each received signal from which the orbital angular momentum has been removed into individual received signals. Individually received signals are provided to the receiver 1502 for demodulation using, for example, multilevel overlay demodulation described in more detail below.

FIG. 16 illustrates how a single wavelength or frequency with two quantum spin polarizations can provide infinite twist with varying orbital angular momentum associated therewith. The l axes represent various quantized orbital angular momentum states that can be applied to a particular signal at a selected frequency or wavelength. The symbol omega (ω) represents various frequencies that can be applied to signals of different orbital angular momentum. Upper grid 1602 represents a potentially usable signal for left hand signal polarization, and lower grid 1604 is for a potentially usable signal with right hand polarization.

By applying different orbital angular momentum states to a signal of a particular frequency or wavelength, potentially infinite states can be provided at that frequency or wavelength. The state at frequency Δω or wavelength 1606 in both left hand polarization plane 1602 and right hand polarization plane 1604 can provide infinite signals in different orbital angular momentum states Δl. Blocks 1608 and 1510 represent a specific signal having orbital angular momentum Δl at a frequency Δω or wavelength in both the right hand polarization plane 1604 and the left hand polarization plane 1610, respectively. By changing different orbital angular momentum within the same frequency Δω or wavelength 1606, different signals may also be transmitted. Each angular momentum state corresponds to a different determined current level for transmission from the optical transmitter. By estimating an equivalent current for generating a specific orbital angular momentum in the optical domain and applying that current to signal transmission, signal transmission in the desired orbital angular momentum state can be achieved.

Therefore, the example of FIG. 16 shows two possible angular momentum, spin angular momentum and orbital angular momentum. The spin version appears in the macroscopic electromagnetic polarization and has only left and right hand polarization due to the up and down spin direction. However, the orbital angular momentum represents an infinite number of states quantized. There are two or more paths and the theoretically quantized orbital angular momentum level can be infinite.

Using the orbital angular momentum state of the transmitted energy signal, physical information can be embedded in the radiation transmitted by the signal. The Maxwell-Heaviside equation can be expressed as

는 델 연산자이고, E는 전기장 강도이고, B는 자속 밀도이다. 이러한 식들을 이용하면, 맥스웰의 오리지널 식으로부터 23개의 대칭/보존량을 유도할 수 있다. 그러나, 10개의 잘 알려진 보존량이 존재하고, 그 중 몇 개만이 상업적으로 사용된다. 역사적으로, 맥스웰 방정식이 그 원래의 쿼터니언 형태를 유지한다면, 대칭/보존량을 쉽게 볼 수 있을 것이지만, 헤비사이드에 의해 현재의 벡터 형태로 변환된 때에는, 맥스웰 방정식에서 이러한 내재적 대칭성을 보는 것은 더 어렵게 되었다.here,

Is the Dell operator, E is the electric field strength, and B is the magnetic flux density. Using these equations, 23 symmetry / preservation quantities can be derived from Maxwell's original equation. However, there are ten well known reserves, only a few of which are used commercially. Historically, if the Maxwell's equations retained their original quaternion shape, it would be easy to see the symmetry / conservation, but when converted to the current vector form by the heavyside, it became more difficult to see this inherent symmetry in the Maxwell's equation. .

Maxwell's linear theory is U (1) symmetry with Abelian commutation relations. They can be extended to form a higher symmetric group SU (2) with non-Abel commutation relationships that deal with global (non-local in space) characteristics. The interpretation of Max-Theory's Wu-Yang and Harmuth suggests the presence of magnetic monopoles and magnetic charges. As far as the classical field is concerned, this theoretical structure is pseudo-particle or instanton. The interpretation of Maxwell's work actually begins in quite a different way from Maxwell's original intentions. In Maxwell's original formula, Faraday's electrophoretic state (Aμ field) became central and made them compatible with the Yang-Mills theory (before heavyside). A mathematical dynamic entity called soliton can be classical or quantum, linear or nonlinear and describes an EM wave. However, soliton is in the form of SU (2) symmetry. In order for the classical Maxwell's U (1) symmetry theory of traditional interpretations to describe such an entity, it must be extended to the SU (2) form.

In addition to six physical phenomena (which cannot be explained by the traditional Maxwell theory), the recently formulated Harmuth Ansatz also deals with the incompleteness of Maxwell's theory. The Maxwell's Maxwell's equation can be used to calculate the EM signal rate if a magnetic current density and magnetic charge are added that match the equation filled by the sheep-mills. Therefore, with accurate geometry and topology, the Aμ potential always has a physical meaning.

Conservation and electromagnetic fields can be expressed according to the conservation of system energy and the conservation of system linear momentum. Temporal symmetry, that is, conservation of system energy, can be expressed using the theorem of pointing according to the following equation.

The conservation of the system linear momentum representing spatial symmetry, i.e., electromagnetic Doppler shift, can be expressed by the following equation.

Conservation of the system energy center can be expressed by the following equation.

Similarly, the conservation of the system angular momentum causing the azimuth Doppler shift is expressed by the following equation.

For the radiation beam in free space, the EM field angular momentum J ^em can be separated into two parts.

For each single Fourier mode of real value representation:

The first part is EM spin angular momentum S ^em , the classical expression of which is wave polarity. And, the second part is the EM orbital angular momentum L ^em , and its classical expression is par helicity. In general, both EM linear momentum P ^em and EM angular momentum J ^em = L ^em + S ^em are radiated in all directions up to the distance.

By using the pointing theory, the light vorticity of the signal can be determined according to the luminous flux equation below.

Where S is the pointing vector

And U is the energy density.

E and H each contain an electric field and a magnetic field, and ε and μ ₀ are the permittivity and permeability of the medium, respectively. Then, even with light, V can be determined by the curl of the light beam according to the following equation.

Referring now to FIGS. 17A and 17B, the scheme of a signal and its associated pointing vector in a planar wave situation is shown. In the plane wave situation shown generally at 1702, the transmitted signal may take one of three configurations. When the electric field vectors are in the same direction, a linear signal is provided, as generally shown at 1704. In circularly polarized light 1706, the electric field vectors rotate by the same magnitude. In elliptical polarization 1708, the electric field vector rotates but has a different magnitude. The pointing vector remains constant with respect to the signal configuration for FIG. 17A and is always perpendicular to the electric and magnetic fields. Referring now to FIG. 17B, when the intrinsic orbital angular momentum is applied to a signal as previously described herein, the pointing vector, S 1710, will be spiral about the direction of propagation of the signal. This helix may be changed to enable signals to be transmitted on the same frequency as described herein.

18A-18C show differences in signals with different helicity (ie orbital angular momentum). Each helical pointing vector associated with signals 1802, 1804, and 1806 provides a different shaped signal. Signal 1802 has an orbital angular momentum of +1, signal 1804 has an orbital angular momentum of +3, and signal 1806 has an orbital angular momentum of -4. Each signal has a distinct angular momentum and associated pointing vector, which allows the signal to be distinguished from other signals within the same frequency. Since these signals are detectable individually and do not interfere with each other (Eigen channel), it is possible for different types of information to be transmitted at the same frequency.

18D shows propagation of pointing vectors for various eigen modes. Each ring 1820 exhibits a torsion representing different orbital angular momentum within different eigen modes or at the same frequency. Each ring 1820 represents a different orthogonal channel. Each eigen mode has a pointing vector 1822 associated with it.

Topology charges can be multiplexed at frequencies for linear or circular polarization. In the case of linear polarization, the topological charge will be multiplexed on the vertical and horizontal polarizations. In the case of circular polarization, the topological charge will be multiplexed on the left hand and right hand circular polarization. Topology charge is a helix index "I", or another name for the magnitude of the torsion or OAM applied to the signal. The helicity index may be positive or negative. In RF, different topological charges can be generated and multiplexed together and de-multiplexed to separate the topological charges.

Topological charge (l) reduces distortion of the RF wave (opposite of the light beam) by adjusting the appropriate material and machine capability or phase mask with a certain refractive index, or the voltage on the device causing the torsion of the RF wave with the specific topology charge. It can be generated using a spiral phase plate (SPP) as shown in FIG. 18E using a hologram generated with a new technique or a new material for generating an RF version of the spiral optical modulator (SLM). The spiral phase plate can convert the RF plane wave (l = 0) into a twisted RF wave of a certain helix (ie, l = +1).

Crosstalk and multipath interference can be corrected using RF multiple input multiple output (MIMO). Most channel damage can be detected using control channels or pilot channels and corrected using algorithmic techniques (closed loop control systems).

As described above in connection with FIG. 13, each multiple data stream applied within the processing circuit has a multilayer overlay modulation scheme applied to it.

Referring now to FIG. 19, reference numeral 1900 generally represents one embodiment of a multi-level overlay (MLO) modulation system, but it should be understood that the term MLO and illustrated system 1900 are exemplary embodiments. The MLO system may include such as disclosed in US Pat. No. 8,503,546 entitled Multilayer Overlay Modulation, incorporated herein by reference. In one example, modulation system 1900 will be implemented within multi-level overlay modulation box 504 of FIG. System 1900 takes as input the input data stream 1901 from digital source 1902, which is separated by input demultiplexer (DEMUX) 1004 into three parallel discrete data streams of logic ones and zeros. Data stream 1001 may represent a data file to be transmitted, or an audio or video data stream. It should be understood that more or fewer separate data streams may be used. In some embodiments, each separate data stream 1901A-1903C has a data rate of 1 / N of the original rate, where N is the number of parallel data streams. In the embodiment shown in FIG. 19, N is three.

Each separate data stream 1901A-1903C is connected to an M-QAM constellation, eg, orthogonal amplitude modulation (QAM) symbol within 16 QAM or 64 QAM by one of the QAM symbol mappers 1905A-C. Mapped. The QAM symbol mapper 1905A-C is connected to the respective output of the DEMUX 1904 and has phase I (1906A, 1908A, and 1910A) and quadrature phase (Q) 1906B, 1908B, and 1910B at discrete levels. Generated in parallel as data streams. For example, at 64 QAM, each I and Q channel uses eight separate levels to transmit 3 bits per symbol. Each of the three I and Q pairs 1906A-1906B, 1908A-1908B, and 1910A-1910B are used to weight the output of the corresponding pairs of function generators 1907A-1907B, 1909A-1909B, and 1911A-1911B. The generator generates signals such as the modified Hermit polynomial described above in some embodiments and weights them based on the amplitude values of the input symbols. This provides 2N weighted or modulated signals, each of which carries a portion of the original data from the incoming data stream 1901, which is a raised cosine filter, as is done for prior art QAM systems. Instead of modulating each symbol within the I and Q pairs 1906A-1906B, 1908A-1908B, and 1910A-1910B. In the illustrated embodiment, three signals SH0, SH1, and SH2 corresponding to variations of H0, H1, and H2, respectively, are used, although it should be understood that other signals may be used in other embodiments.

The weighted signals are not subcarriers, but rather the lower layers of the modulated carrier, and are combined and superimposed at both frequency and time using summers 1912 and 1916 without mutual interference in each of the I and Q dimensions due to signal orthogonality. Summers 1912 and 1916 serve as signal combiners for generating synthesized signals 1913 and 1917. The weighted orthogonal signal is used for both the I and Q channels that were equivalently processed by the system 1900 and summed before the QAM signal is transmitted. Therefore, some embodiments additionally use QAM for transmission even if a new orthogonal function is used. Due to the tapering of the signal in the time domain as shown in FIGS. 16A-16K, the time domain waveform of the weighted signal will be limited to the duration of the symbol. In addition, due to the tapering of the special signal and frequency domain, the signal is also confined to the frequency domain, which minimizes the interface of the signal and adjacent channels.

The synthesized signals 1913 and 1917 are converted to analog signals 1915 and 1919 using digital to analog converters 1914 and 1919 and then modulated 1921 to the local oscillator (LO) 1920. Used to modulate the carrier signal at a frequency. Modulator 1921 includes mixers 1922 and 1924 connected to DACs 1914 and 1919, respectively. The 90 degree phase shifter 1923 converts the signal from the LO 1920 into the Q component of the carrier signal. The outputs of mixers 1922 and 1924 are summed in summer 1925 to produce output signal 1926.

MLO can be used with a variety of transmission media such as wired, optical, and wireless, and can be used in combination with QAM. This is because MOL uses spectral overlay of various signals rather than spectral overlap. Due to the expansion of the available spectrum resources to multiple layers, bandwidth utilization efficiency can be further increased. The number of orthogonal signals is increased from two (cosine and sine of the prior art) to a number limited by the precision and jitter limits of the generator used to generate the orthogonal polynomial. In this way, the MLO uses each of the I and Q dimensions of QAM to define GSM, code division multiple access (CDMA), wideband CDMA (WCDMA), high speed downlink packet access (HSPDA), evolution-data optimization (EV-DO), OFDM It extends to any multiple access technology such as orthogonal frequency division multiplexing, worldwide interoperability for microwave access (WIMAX), and long term evolution (LTE) systems. MLO can also be used with other multiple access (MA) schemes such as frequency division duplexing (FDD), time division duplexing (TDD), frequency division multiple access (FDMA), and time division multiple access (TDMA). Overlapping individual orthogonal signals in the same frequency band creates a virtual bandwidth that is wider than the physical bandwidth, adding a new dimension to signal processing. This modulation can be applied to twisted pairs, cables, fiber optics, satellites, broadcast, free space optics and all types of wireless access. This method and system includes current and future multiples including EV-DO, UMB, WIMAX, WCDMA (w / w), Multimedia Broadcast Multicast Service (MBMS) / Multiple Input Multiple Output (MIMO), HSPA Evolution, and LTE Compatible with the access system.

Referring now to FIG. 20, although an MLO demodulator 2000 is shown, it is to be understood that the terms MLO and illustrated system 2000 are exemplary embodiments. Modulator 2000 takes MLO signal 1126 as input, which may be similar to output signal 1826 from system 1800. The synchronizer 2027 extracts phase information input to the local oscillator 2020, allowing the modulator 2021 to generate basebands for the analog I signal 2015 and the Q signal 2020. Modulator 2021 includes mixers 2022 and 2024 coupled to LO 2020 through a 90 degree phase shifter 2023. I signal 2015 is input to each of signal filters 2007A, 2009A, and 2011A, and Q signal 2020 is input to each of signal filters 2007B, 2009B, and 2011B. Since orthogonal functions are known, they can be separated using correlation or other techniques to recover the modulated data. Information of each of the I and Q signals 2015 and 2020 can be extracted from the superimposed function summed in each symbol because the functions are orthogonal in a correlational sense.

In some embodiments, signal filters 2007A-2007B, 2009A-2009B, and 2011A-2011B use a locally generated copy of the polynomial as the known signal in the match filter. The output of the match filter is the recovered data bits, eg, the equivalents of the QAM symbols 2006A-2006B, 2008A-2008B, 2010A-2010B of the system 2000. Signal filters 2007A-2007B, 2009A-2009B, and 2011A-2011B produce 2n streams of n, I, and Q signal pairs, which are input to demodulators 2028-2033. Demodulators 2028-2033 incorporate energy into their respective input signals to determine the value of the QAM symbol, and thus the logical 1 and 0 data bit stream segments represented by the determined symbol. The outputs of modulators 2028-2033 are then input to multiplexer (MUX) 2005A-2005C to produce data streams 2003A-2003C. When system 2000 demodulates a signal from system 1800, data streams 2003A-2003C correspond to data streams 1803A-1803C. Data streams 2003A-2003C are multiplexed by MUX 2004 to produce data output stream 2001. In summary, the MLO signals are superimposed on top of each other at the transmitter and separated at the receiver.

MLO can be distinguished from CDMA or OFDM by the way that orthogonality between signals is achieved. MLO signals are orthogonal to each other in both time and frequency domain and may overlap in the same symbol time bandwidth product. Orthogonality can be obtained with the correlation characteristics of the superimposed signals, eg, the sum of least squares. In comparison, CDMA uses orthogonal interleaving or displacement of signals in the time domain, while OFDM uses orthogonal displacement of signals in the frequency domain.

By allocating the same channel to multiple users, the bandwidth efficiency of the channel can be increased. It is also possible for individual user information to be mapped to special orthogonal functions. The CDMA system superimposes a large number of user information, observes orthogonal code sequences between time symbols to distinguish individual users, and assigns a unique signal to each user who is not overlapping but orthogonal in the frequency domain. Neither CDMA nor OFDM increases bandwidth efficiency. CDMA uses more bandwidth than necessary to transmit data when the signal has a low signal-to-noise ratio (SNR). OFDM spreads data over multiple subcarriers to achieve good performance in a multipath radio frequency environment. OFDM uses cyclic prefix OFDM to reduce multipath effects, guard time to minimize inter-symbol interference (ISI), and each channel is mechanically adapted to behave as if the transmitted waveform is orthogonal. Is made. (Sync function for each subcarrier in frequency domain)

In contrast, MLO uses a set of functions that effectively form an alphabet that provides more available channels at the same bandwidth, thereby enabling high bandwidth efficiency. Some embodiments of the MLO do not require the use of cyclic prefix or guard time, resulting in better performance than OFDM in terms of spectral efficiency, peak-to-average power ratio, and power consumption, and require fewer operations per bit. In addition, embodiments of MLO are more tolerant to amplifier nonlinearity than CDMA and OFDM systems.

21 illustrates one embodiment of an MLO transmitter system 2100 receiving an input data stream 1901. System 2100 includes DEMUX 1904, QAM symbol mapper 1905A-C, function generators 1907A-1907B, 1909A-1909B, and 1911A-1911B, and summer (1906) of system 1900 shown in FIG. A modulator / controller 2001 incorporating functionality equivalent to 1919. However, it should be understood that the modulator / controller 2001 may use a larger or smaller amount of signal than shown in the system 1900. Modulator / controller 2001 may include other components, whether application specific integrated circuit (ASIC), field programmable gate array (FPGA), and / or discrete circuitry or integrated into a single integrated circuit (IC) chip. have. The modulator / controller 2101 may include an application specific integrated circuit (ASIC), a field programmable gate array (FPGA) and / or discrete components or other components integrated into a single integrated circuit (IC) chip.

Modulator / controller 2101 is coupled to DACs 2104 and 2107 which communicate 10-bit I signals 2102 and 10-bit Q signals 2105, respectively. In some embodiments, I signal 2102 and Q signal 2105 correspond to composite signals 1913 and 1917 of system 1900. However, it should be understood that the 10 bit capacities of I signal 2102 and Q signal 2105 represent only one embodiment. As shown, modulator / controller 2101 also controls DACs 2104 and 2107 using control signals 2103 and 2106, respectively. In some embodiments, DACs 2104 and 2107 include AD5433, a complementary metal oxide semiconductor (CMOS) 10 bit current output DAC, respectively. In some embodiments, multiple control signals are sent to respective DACs 2104 and 2107.

DACs 2104 and 2107 output analog signals 2115 and 2119 to quadrature modulator 1921 connected to LO 1920. Although the output of modulator 1921 is shown coupled to transmitter 2108 for wirelessly transmitting data, in some embodiments modulator 1921 is coupled to a fiber optic modem, twisted pair, coaxial cable, or other suitable transmission medium. May be

22 shows an embodiment of an MLO receiver system 2200 capable of receiving and demodulating signals from system 2100. System 2200 receives input signals from receiver 2208, which may include input media such as RF, wired, or optical. A modulator 2021 driven by LO 2020 converts the input to baseband I signal 2015 and Q signal 2019. I signal 2015 and Q signal 2019 are input to an analog-to-digital converter (ADC) 2209.

ADC 2209 outputs 10-bit signal 2210 to demodulator / controller 2201 and receives control signal 2212 from demodulator / controller 2201. The demodulator / controller 2201 may include an application specific integrated circuit (ASIC), a field programmable gate array (FPGA) and / or other components integrated into discrete circuit elements or a single integrated circuit (IC) chip. Demodulator / controller 2201 correlates the received signal with a locally generated replica of the signal set used to perform demodulation and identify the transmitted symbol. Demodulator / controller 2201 also recovers the data clock used to estimate frequency error and read data from ADC 2209. The clock timing is sent back to the ADC 2209 using the control signal 2212 so that the ADC 2209 can segment the digital I and Q signals 2015 and 2019. In some embodiments, a number of control signals are sent by the demodulator / controller 2201 to the ADC 2209. Demodulator / controller 2201 also outputs data signal 2001.

Hermit polynomials are classical orthogonal polynomial sequences and are the eigen states of quantum harmonic oscillators. Signals based on the Hermit polynomial have the minimum time-bandwidth product characteristics described above and can be used in embodiments of the MLO system. However, it should be understood that other signals can also be used, such as orthogonal polynomials such as, for example, Jacoby polynomial, Gezen Bauer polynomial, Legendr polynomial, Chebyshev polynomial, and Laguerre polynomial. Q-functions are another class of function that can be used as the basis for MLO signals.

In quantum mechanics, the coherent state is the state of the quantum harmonic oscillator that is most closely related to the vibrational behavior of the classical harmonic oscillator system. The compact coherent state is any state of quantum mechanical Hilbert space that causes the uncertainty principle to saturate. That is, the products of those two operators take their minimum value. In an embodiment of the MLO system, the operator corresponds to the time and frequency domain where the time-bandwidth product of the signal is minimized. The compression characteristics of the signals allow scaling simultaneously in the time and frequency domains without losing mutual orthogonality between signals in each layer. This feature enables flexible implementation of the MLO system in various communication systems.

Since signals with different orders are orthogonal to each other, the signals may overlap to increase the spectral efficiency of the communication channel. For example, when n = 0, the optimal baseband signal will have a half-time-bandwidth product that is a Nyquist intersymbol interference (ISI) criterion to avoid ISI. However, signals with 3/2, 5/2, 7/2 and more time-bandwidth products can be superimposed to increase spectral efficiency.

An embodiment of the MLO system uses a function based on the modified Hermit polynomial, 4n, which is defined as follows.

Where t is time and ξ is the bandwidth utilization parameter. Along with their Fourier transform (amplitude squared), a plot of Ψ _{n over} n in the 0-9 range is shown in FIGS. 5A-5K. The orthogonality of the different orders of the function can be verified by the following integration.

The Hermit polynomial is defined by the contour integral:

Here, the contour is moved counterclockwise around the origin. Hermit polynomials are described by George Arken in a mathematical method for physicists, for example, on page 416, the contents of which are incorporated by reference.

23A-23K show representative MLO signals and their respective spectral power densities based on the modified Hermit polynomial Ψn for n in the range 0-9. 23A shows plots 2301 and 2304. Plot 2301 includes curve 2327 representing Ψ 0 plotted about time axis 2302 and amplitude axis 2303. As can be seen in plot 2301, curve 2327 approximates a Gaussian curve. Plot 2304 includes curve 2337 representing the power spectrum of Ψ 0 plotted about frequency axis 2305 and power axis 2306. As can be seen in plot 2304, curve 2337 also approximates a Gaussian curve. Frequency domain curve 2307 is generated using Fourier transform of time domain curve 2327. Since the time axis and frequency units are related by Fourier transform, it should be understood that the desired time or frequency span in one domain represents the units of the corresponding curve in the other domain, but the time and frequency units on axes 2302 and 2305 are Normalized for baseband analysis. For example, various embodiments of the MLO system may include a non-zero duration of a symbol represented by a curve rate and a curve rate in the megahertz (MHz) or gigahertz (GHz) range, that is, the curve 2327. Time periods above zero will be compressed to the appropriate length calculated using the inverse of the desired symbol rate. For the bandwidth available in the megahertz range, the nonzero duration of the time domain signal is in the microsecond range.

23B-23J show plots 2307-2324 with time domain curves 2328-2336 and their corresponding frequency domain curves 2338-2446 representing Ψ 1 Ψ 9, respectively. As can be seen in FIGS. 23A-23J, the number of peaks in the time domain plot corresponds to the number of peaks in the frequency domain plot, whether positive or negative. For example, in plot 2323 of FIG. 23J, time domain curve 2336 has five positive peaks and five negative peaks. Thus, in the corresponding plot 2324, the frequency domain curve 2346 has ten peaks.

23K shows overlay plots 2325 and 2326 overlaying curves 2327-2336 and 2337-2346, respectively. As shown in plot 2325, the various time domain curves have different durations. However, in some embodiments, the nonzero duration of the time domain curve is of similar length. For MLO systems, the number of signals used represents an improvement in the number of overlays and the spectral efficiency. Although 10 signals are disclosed in FIGS. 23A-23K, it should be understood that more or less amounts of signals may be used, and that a different set of signals may be used rather than the plotted Ψ n signals.

The MLO signal used in the modulation layer can improve spectral efficiency and has a minimum time-bandwidth product that can be integrally quadratic. This is achieved by overlaying multiple inverse multiple parallel data streams and transmitting them simultaneously within the same bandwidth. The key to successfully separating the overlaid data streams at the receiver is that the signals used within each symbol period are orthogonal to each other. MLO overlays an orthogonal signal within a single symbol period. This orthogonality prevents ISI and intercarrier interference (ICI).

Since the MLO operates at the baseband layer of signal processing and some embodiments use a QAM architecture, conventional radio technologies will also work with the MLO to optimize the air interface or radio segment to other layers in the protocol stack. Techniques such as channel diversity, equalization, error correction coding, spread spectrum, interleaving and space-time encoding are applicable to MLO. For example, time diversity using a multipath mitigating rake receiver may also be used with the MLO. MLO, like fading channels, provides an alternative to higher-dimensional QAMs when channel conditions are only suitable for lower-order QAMs. MLO can also be used with CDMA to extend the number of orthogonal channels by overcoming CDMA Walsh code limitations. MLO may also be applied to each tone of the OFDM signal to increase the spectral efficiency of the OFDM system.

Embodiments of the MLO system amplitude modulate a symbol envelope to produce a sub-envelope rather than a sub-carrier. For data encoding, each subenvelope is modulated independently in accordance with N-QAM, resulting in each subenvelope carrying information independently, unlike OFDM. Instead of spreading information over many subcarriers, as in OFDM, each sub-envelope of the carrier carries separate information in the case of MLO. This information may be recovered due to the orthogonality of the sub-envelopes defined for the sum of the squares for the duration and / or spectrum. Since the MLO is transparent above the symbol level, pulse train synchronization or time code synchronization is not a problem if needed for CDMA. MLO deals with symbol variations, while CDMA and TDMA are spreading techniques for multiple symbol sequences over time. MLO can be used with CDMA and TDMA.

24 shows a comparison of MLO signal widths in the time and frequency domain. The time domain envelope representations 2401-2403 of the signals SH0-SH3 are all shown to have a duration, TS. SH0-SH3 may represent PSI0-PSI2 or may be another signal. The corresponding frequency domain envelope representations are 2405-2407, respectively. SH0 has a bandwidth BW, SH1 has three times the bandwidth of BW, and SH2 has a bandwidth of 5BW, which is five times the bandwidth of SH0. The bandwidth used in the MLO system will be determined, at least in part, by the widest bandwidth of any used signal. If each layer uses only a single signal type within the same time window, the spectrum will not be fully utilized because lower order signals use less available bandwidth than higher order signals.

25 illustrates the spectral alignment of MLO signals, taking into account different bandwidths of signals and making the spectrum use more uniform, using SH0-SH3. Blocks 2501 through 2504 are frequency domain blocks of an OFDM signal with multiple subcarriers. Block 2503 has been enlarged for further details. Block 2503 includes a first layer 2503x composed of multiple SH0 envelopes 2503a-2503o. The second layer 2503y of the SH1 envelope 2503p-2503t has 1/3 envelope as the first layer. In the example shown, the first layer 2503x has 15 SH0 envelopes, and the second layer 2503y has 5 SH1 envelopes. This is because the SH1 bandwidth envelope is three times SH0 and the 15 SH0 envelope occupies the same spectral width as the five SH1 envelopes. The third layer 2503z of block 2503 includes three SH2 envelopes 2503u-2503w because the SH2 envelope is five times the SH0 envelope.

The total bandwidth required for this implementation is a multiple of the least common multiple of the bandwidths of the MLO signal. In the example shown, the minimum common multiple of the bandwidth required for SH0, SH1 and SH2 is 15 BW, which is a block in the frequency domain. An OFDM-MLO signal can have multiple blocks, and the spectral efficiency of the illustrated implementation is proportional to (15 + 5 + 3) / 15.

FIG. 26 illustrates another spectral alignment of MLO signals that may alternatively be used for the alignment scheme shown in FIG. 25. In the embodiment shown in FIG. 26, the OFDM-MLO implementation stacks the spectra of SH0, SH1 and SH2 in such a way that the spectrum of each layer is used uniformly. Layer 2600A includes envelopes 2601A-2601D that include both SH0 and SH2 envelopes. Similarly, layer 2600C comprising envelopes 2603A-2603D includes both SH0 and SH2 envelopes. However, layer 2600B including envelopes 2602A-2602D only includes an SH1 envelope. Using the ratio of envelope sizes described above, it is easy to see that BW + 5BW = 3BW + 3BW. Thus, each SH0 envelope in layer 2600A, one SH2 envelope in layer 2600C, and two SH1 envelopes in layer 2600B.

3 Scenario Comparison

1) MLO with three layers is defined as follows.

(Current FPGA implementations use truncation intervals of [-6, 6].)

2) Conventional scheme using rectangular pulse

3) Conventional scheme using square-raised raised cosine (SRRC) pulses with a rolloff coefficient of 0.5

In the case of the MLO pulse and the SRRC pulse, the cutoff interval is indicated as [-t1, t1] in the following figure. For simplicity, the MLO pulses defined above can be easily adjusted to achieve the desired time interval (eg, microseconds or nanoseconds). In case of SRRC pulse, the cutting interval of [-3T, 3T] is determined. Where T is the symbol duration for all results herein.

Bandwidth efficiency

X-dB boundary power spectral density Bandwidth is defined as the smallest frequency interval where the power spectral density (PSD) is XdB below the maximum value of the PSD. X-dB can be considered as out-of-band attenuation.

Bandwidth efficiency is expressed in symbols per second per Hertz. The bits per second per hertz can be obtained by multiplying the symbols per second by the symbols per second and the number of bits per symbol (ie multiply log2M for M-ary QAM).

Truncation of MLO pulses introduces inter-layer interferences (ILI). However, the truncation interval of [-6, 6] yields negligible ILI, while [-4,4] results in some acceptable ILI.

The bandwidth efficiency of MLO can be improved by allowing intersymbol interference (ISI). To achieve this improvement, we can design the sender parameters and develop the receiver-side detection algorithm and error performance evaluation.

Referring now to FIG. 27, the power spectral density of each layer (SH0-SH2) in ML0 and the power spectral density of the combined three layer MLO are shown. 2702 shows the power spectral density of the SH0 layer, 2704 shows the power spectral density of the SH1 layer, 2706 shows the power spectral density of the SH2 layer, and 2708 the combined power of each layer. Spectral density.

Referring now to FIG. 28, the power spectral densities of the three layers combined as well as the power spectral densities of each layer are shown on a logarithmic scale. (2702) represents the SH0 layer. 2704 represents an SH1 layer. 2706 represents an SH2 layer. 2708 represents the combined layer.

Referring now to FIG. 29, there is a band efficiency comparison versus out-of-band attenuation (X-dB) with quantum level overlay pulse truncation interval [-6,6] and symbol rate 1/6. Referring to FIG. 30, a band efficiency comparison versus out-of-band attenuation (X-dB) with quantum level overlay pulse truncation interval [-6,6] and symbol rate 1/4 is shown.

The QLO signal is generated from the following physicist's special Hermit function.

를 이용하고,

는 자신의 부분과 일관성을 유지하기 위해M 스펙트럼 효율과 관련된 모든 수치에서 사용됨을 이해해야 한다.Early hardware implementation

Using

It should be understood that in order to be consistent with its part, it is used at all figures related to M spectral efficiency.

The low pass equivalent power spectral density (PSD) of the combined QLO signal is X (f) and its bandwidth is B. Where bandwidth is defined by one of the following criteria:

ACLR1 (the first adjacent channel leak rate) in dBc is

ACLR2 (second adjacent channel leak rate) in dBc is

The out-of-band power to total power ratio is

The band-edge PSD of dBc / 100 kHz is as follows.

Referring now to FIG. 31, a performance comparison is shown using ACLR1 and ACLR2 for both square root raised cosine schemes and multilayer overlay schemes. Line 3102 shows the performance of MLO 3104 using ACLR1 versus square root raise cosine 3102 using ACLR1. Additionally, a comparison of MLO 3108 using ACLR2 and square root raised cosine 3106 using ACLR2 is also shown. Table A shows the performance comparison using ACLR.

Referring now to FIG. 32, a performance comparison is shown between square root raised cosine 3202 and MLO 3204 using out-of-band power. Referring now to Table B, a more detailed comparison of the performance using out of band power is shown.

Referring now to FIG. 33, a performance comparison is further provided between square root raised cosine 3302 and MLO 3304 using band edge PSD. Details of the performance comparison are provided in Table C.

Referring now to FIGS. 34 and 35, the transmission subsystem (FIG. 34) and the receiver subsystem (FIG. 35) are more specifically shown. The transceiver is realized using basic building blocks that are provided as products that are not sold commercially. Modulation, demodulation, and special Hermit correlation and de-correlation are implemented on the FPGA board. The FPGA board 3402 of the receiver 3400 recovers the data clock (and data) used to estimate frequency error and read data from the analog-digital (ADC) board 3406. The FGBA board 3400 also segments digital I and Q channels.

On the transmitter side 3400, the FPGA board 3402 controls the digital-to-analog (DAC) board 3404 to generate analog I and Q baseband channels for subsequent up conversion within the direct conversion quad modulator 3406. In addition to the control signals needed to achieve this, a special Hermit Correlated QAM signal is realized. The direct conversion quad modulator 3406 receives an oscillator signal from the oscillator 3408.

ADC 3506 receives I and Q signals from quad demodulator 3508 which receives oscillator signals from 3510.

Since the communication takes place over a short distance, neither the transmitter's power amplifier nor the receiver's LNA are used. Although a frequency band of 2.4-2.5 GHz (ISM band) is selected, any frequency band of interest may be used.

MIMO uses diversity to achieve some spectral efficiency increase. Each signal from the antenna serves as an independent orthogonal channel. With QLO, the gain in spectral efficiency occurs within the symbol, and each QLO signal acts as an independent channel because it is orthogonal to each other in every permutation. However, since QLO is implemented at the bottom of the protocol stack (physical layer), any technique of higher level protocol (i.e., transport) works with QLO. Therefore, all existing technologies can be used with QLO. This includes a rake receiver and equalizer to prevent fading, cyclic prefix insertion to prevent time dispersion, all other techniques using beam shaping, and MIMO to further increase spectral efficiency.

Given the spectral efficiency of a real wireless communication system, the following approach may be more appropriate because of the substantially different bandwidth definitions (and also not the strict band limiting characteristics of the actual transmission signal).

Referring now to FIG. 36, consider an equivalent discrete time system and obtain a Shannon capacity for that system (denoted Cd). For example, with respect to discrete time systems such as AWGN's existing QAM system, the system is as follows.

Where a is a scalar representing channel gain and amplitude scaling, x [n] is an input signal (QAM symbol) with unit average energy (scaling is contained in a), and y [n] is a demodulator (matching filter) Output symbol, index n is a discrete time index.

The corresponding Shannon capacity is as follows.

Where σ2 is the noise variance (complex dimension) and | a | 2 / σ2 is the SNR of the discrete time system.

Second, calculate the bandwidth, W, according to the adopted bandwidth definition (e.g., a bandwidth defined as -40 dBc out-of-band power). If the symbol duration corresponding to a sample of the discrete time (or time required to transmit the C _d bits) is T, the spectral efficiency can be obtained as follows.

In discrete time systems of AWGN channels, the use of turbo or similar codes will provide performance very close to the Shannon limit C _d . Performance in the discrete time domain is the same regardless of the pulse shape used. For example, the same C _d (or C _d / T) can be obtained by using an SRRC (square root raised cosine) pulse or a rectangular pulse. However, considering the continuous time real system, the bandwidths of the SRRC and rectangular pulses will be different. For a typical real bandwidth definition, the SRRC pulse's bandwidth is less than that of a rectangular pulse, so SRRC provides better spectral efficiency. In other words, there is little room for improvement in the discrete time system of the AWGN channel. However, in continuous practical systems, there may be considerable room for improvement of spectral efficiency.

Referring now to FIG. 37, a PSD plot (BLANK) of MLO, modified MLO (MMLO), and square root raised cosine (SRRC) is shown. The figure of FIG. 37 shows better localization characteristics of the MLO. The advantage of MLO is bandwidth. 37 also shows that interferences for adjacent channels will be much smaller for MLO. This provides additional benefits in managing, allocating, or packaging the spectral resources of multiple channels and systems, further improving overall spectral efficiency. In-band PSD of MLO and SRRC when the bandwidth is defined as -40 dBc out-of-band power is shown in FIG. 37. The bandwidth ratio is about 1.536. Thus, there is considerable room for improvement of spectral efficiency.

The modified MLO system is based on block processing in which each block contains N MLO symbols and each MLO symbol has L layers. The MMLO may be converted to parallel (virtual) orthogonal channels with different channel SNRs as shown in FIG. 39. The output provides an equivalent discrete time parallel orthogonal channel of the MMLO.

It should be understood that the pulse overlap of the MLO due to intersymbol interference was solved by parallel orthogonal channel conversion. As an example, the power gain of a parallel orthogonal virtual channel of an MMLO with three layers and 40 symbols per block is shown in FIG. 40. 39 shows the channel power gain of a parallel orthogonal channel of MMLO with three layers and T _sim = 3. By applying a water filled solution, an optimal power distribution over the orthogonal channel for a fixed transmit power can be obtained. The transmit power on the kth orthogonal channel is denoted by P _k . Then the discrete time capacity of MMLO can be given as

It is to be understood that K depends on the number of MLO layers, the number of MLO symbols per block, and the MLO symbol duration.

During the MLO pulse period defined by [-t1, t1], and symbol duration T _mlo , the MMLO block length is as follows.

Assuming that the bandwidth of the MMLO signal based on the adopted bandwidth definition (ACLR, OBP or others) is W _mmlo , the actual spectral efficiency of the MMLO is given by

41-42 show _{MMLO with} N = 40 symbols per block, L = 3 layers, T _mlo = 3, t ₁ = 8, and SNR of duration [-8T, 8T], T = 1, and 5 dB Shows the spectral efficiency comparison of SRRC with roll-off factor β = 0.22. Two bandwidth definitions are used based on ACLR1 (first adjacent channel leakage power ratio) and OBP (out-of-band power).

43-44 show spectral efficiency comparisons of MMLO with L = 4 layers. The spectral efficiency and gain of the MMLO for a specific bandwidth definition are shown in the following table.

Referring now to FIGS. 45 and 46, a basic block diagram of a low pass equivalent MMLO transmitter (FIG. 45) and a receiver (FIG. 46) is provided. The low pass equivalent MMLO transmitter 4500 receives a number of input signals 4502 in block-based transmitter processing 4504. Transmitter processing outputs a signal to an SH (L-1) block 4506 that produces an I & Q output. These signals are then all combined together in a combining circuit 4508 for transmission.

Within the baseband receiver (FIG. 46) 4600, the received signal is separated and applied to a series of matched filters 4602. The output of the matched filter is then provided to block-based receiver processing block 4604 to generate various output streams.

Consider an N MLO symbol block in which each MLO symbol carries L symbols from L layers. Then, there is an NL symbol in the block. Let c (m, n) = the symbol transmitted by the mth MLO layer in the nth MLO symbol. All NL block symbols are written as column vectors as follows. c = (c (0,0), c (1,0), ..., c (L-1, 0), c (0,1), c (1,1), ..., c ( L-1, 1), ..., c (L-1, N-1)] T. The output of the receiver match filter for a block transmitted in the AWGN channel defined by column vector y of length NL can then be represented by y = Hc + n, where H is an NL × NL matrix representing the equivalent MLO channel, n Is the correlated Gaussian noise vector.

Applying SVD to H results in H = U D VH, where D is a diagonal matrix containing a single value. Transmitter side processing and receiver side processing (UH) using V provide an equivalent system with NL parallel orthogonal channels (ie, y = H Vc + n and UH y = Dc + UH n). This parallel channel gain is given by the diagonal element of D. The channel SNR of this parallel channel can be calculated. It is to be understood that the transmit and receive block based processing solves the ISI problem by obtaining a parallel orthogonal channel.

Since the channel SNRs of these parallel channels are not the same, an optimal water charging solution can be applied to calculate the transmit power of each channel given a fixed total transmit power. This transmit power and corresponding channel SNR can be used to calculate the capacity of the same system as provided in the previous report.

Fading, Multipath, and Multicell Interference Issues

Techniques used to neutralize channel fading in conventional systems (eg, diversity techniques) may also be applied to the MMLO. In the case of a slowly changing multipath distributed channel, if the channel impulse response can be fed back, it can be integrated into the equivalent system mentioned above, and can handle the channel induced ISI and the intentionally introduced MMLO ISI together. If fast time varying channels or channel feedback is not possible, channel equalization should be performed at the receiver. Block-based frequency domain equalization can be applied and oversampling will be required.

Considering the same adjacent channel power leakage for the MMLO and the existing system, the interference power of adjacent cells is nearly the same in both systems. If interference cancellation techniques are required, they may be developed for the MMLO.

Scope and system description

This report shows the symbol error probability (or symbol error rate) performance of the MLO signal in an additional white Gaussian noise channel with various intersymbol interference levels. For reference, the performance of conventional QAM without ISI is included. The same QAM size is considered for all layers of MLO and conventional QAM.

The MLO signal is generated from the physicist's special Hermit function.

에 대응하고, 일관성을 위해

이 이 보고서에서 사용된다는 것에 유의해야 한다.Where Hn (? T) is the n-order Hermit polynomial. The function used for lab setup is

In order to ensure consistency

Note that this is used in this report.

In the above function, an MLO signal with 3, 4 or 10 layers corresponding to n = 0 to 2, 0 to 3 or 0 to 9 is used and the pulse duration (range of t) is [-8, 8].

Fully synchronized AWGN channels are considered.

The receiver consists of a matched filter and a conventional detector without any interference cancellation, i.e. without QAM slicing at the matched filter output.

Where T _p is the pulse duration (16 in the set considered) and Tsym is the inverse of the symbol rate of each MLO layer. The considered cases are listed in the following table.

Signal induction used for modulation

For this purpose, it would be convenient to express the signal amplitude s (t) in a complex form close to the quantum mechanics form. Therefore, the complex signal can be expressed as follows.

Where s (t) and sigma (t) are Hilbert transforms of each other, and s (t) is a quadrature of s (t), and thus have similar spectral components. That is, if they are the amplitudes of sound waves, the ear cannot distinguish one form from another.

Fourier transform pairs are defined as follows:

Also, all moments are normalized to M0.

Then, the moment is as follows.

In general, one may consider representing the signal, s (t), as a polynomial of order N in order to fit close to s (t) and use the coefficients of the polynomial as data representations. This is equivalent to specifying a polynomial in such a way that the first N "moments" M _j represent data. In other words, the moment can be used instead of the coefficient of the polynomial. Another way is to extend the signal, s (t), with a set of N orthogonal functions, φ _k (t), instead of the power of time. Here, data can be regarded as an orthogonal expansion coefficient. One class of orthogonal functions is the sine and cosine functions (same as Fourier series).

Therefore, the above moment can be expressed by using the orthogonal function ψ with the following moment.

Similarly,

If you don't use complex signals,

To represent the average value from the time domain to the frequency domain, replace as follows:

 This is the same as the somewhat mysterious rule in quantum mechanics where classical momentum becomes an operator.

Therefore, using the above assignment, the following equation is obtained.

And

now. The effective duration and the effective bandwidth can be defined as follows.

However, we know that:

This can be simplified by assigning

Also, we know the following:

therefore,

대신에, 등식

을 강제하고 어떤 신호가 이 등식을 충족하는지 확인하는데 관심이 있다. 고정 대역폭 Δf가 주어지면, 가장 효율적인 전송은 시간-대역폭 프로덕트

를 최소화하는 것이다. 주어진 대역폭 Δf에 대하여, 최소 시간에서 송신을 최소화하는 신호는 가우시안 엔벨로프가 될 것이다. 그러나 유효 대역폭은 종종 주어지지 않지만, 전체 대역폭 f₂-f₁은 항상 주어진다. 이제, 최단 유효 시간에 이 채널을 통해 전송될 수 있는 신호 형상, 및 그 유효 기간은 무엇인가? now,

Instead, the equation

We are interested in forcing and determining which signal satisfies this equation. Given a fixed bandwidth Δf, the most efficient transmission is a time-bandwidth product

To minimize. For a given bandwidth Δf, the signal that minimizes transmission at the minimum time will be a Gaussian envelope. However, the effective bandwidth is often not given, but the total bandwidth f ₂ -f ₁ is always given. Now, what is the signal shape that can be transmitted through this channel in the shortest valid time, and its validity period?

Here, φ (f) is zero outside the range f ₂ -f ₁ .

To perform the minimization, a univariate method (Lagrange's Multiplier technique) is used. Since the denominator is a constant, it is necessary to minimize the numerator as shown below.

This is only possible if

The solution is as follows.

Now, if a wave disappears at infinity but still needs to meet the following minimum time-bandwidth product:

The wave equation of the harmonic oscillator is then obtained.

It disappears at infinity only if:

Where H _n (τ) is the Hermit function,

Therefore, the Hermit function H _n (τ) occupies information blocks of 1/2, 3/2, 5/2, ..., and the minimum information quantum is 1/2.

Squeezed

Here, Dirac Algebra's quantum mechanics approach will be used to derive the most common form of the complete eigenfunction. First define the following operators:

Now we are ready to define Δx and Δp as follows:

Now, we will vary the parameterization and use only one variable ξ as follows instead of the two variables λ and μ.

Now, the eigen state in the case of being compressed is as follows.

Now, consider the compressed operator

For the distribution P (n), we will have the formula

Thus, the final result is as follows.

Free space communication

Additional configurations are used in conjunction with free space optical communication that can show that the optical angular momentum processing and multilayer overlay modulation techniques described above are useful within an optical network framework. Free space optical systems offer a number of advantages over traditional UHF RF-based systems by improving inter-system isolation, receiver / transmitter size and cost, the absence of RF licensing laws, and combining space, lighting, and communication into one system. . Referring now to FIG. 47, an operation example of a free space communication system is shown. The free space communication system utilizes a free space optical transmitter 4702 that transmits a light beam 4706 to the free space optical receiver 4704. The main difference between a fiber optic network and a free space optical network is that the information beam is transmitted through the void, not the fiber optic cable. This causes a number of link difficulties, which will be discussed more fully below. Free space optics use a line of sight to provide an optical bandwidth connection that can transmit and receive data, voice, and video communications of up to 2.5 Gbps between transmitter 4702 and receiver 4704 using invisible light. Technology. Free space optics use the same concept as optical fibers, except that they do not use fiber optic cables. The free space optical system provides a light beam 4706 in the infrared (IR) spectrum at the bottom of the light spectrum. Specifically, the optical signal is within the range of 300 GHz to 1 terahertz in terms of wavelength.

Existing free space optical systems can provide data rates of up to 10 gigabits per second at distances of up to 2.5 kilometers. Although the communication range of free space optical communications in outer space is currently thousands of kilometers, there is the potential to connect millions of kilometers of interplanetary distances using optical telescopes with beam expanders. NASA used lasers in January 2013 to transmit images of Mona Lisa to the Moon's reconnaissance satellite, approximately 240,000 miles away. An error correction code algorithm similar to that used in compact discs has been implemented to compensate for atmospheric interference.

Distance recording for optical communications involves the detection and emission of laser beams by space probes. A two-way distance record for communications was established by a waterborne laser altimeter device mounted on the MESSENGER spacecraft. Designed as a laser altimeter for aqueous exploration missions, these infrared diode neodymium lasers could communicate at approximately 15,000,000 miles (24,000,000 kilometers) when the spacecraft flew near Earth in May 2005. Previous records were set for unidirectional detection of laser light from the Earth by Galileo probes, two ground-based lasers seen in 1992 at 6 million kilometers by outbound probes. The researchers used a white LED-based spatial lighting system for indoor local network communications.

Referring now to FIG. 48, shown is a block diagram of a free space optical system using orbital angular momentum and multi-level overlay modulation in accordance with the present disclosure. While the present disclosure is directed to systems using OAM and MLO modulation, it will be understood that the system may implement only one of these techniques, or none of them. OAM twisted signals may be transmitted using free optics in addition to being transmitted over optical fibers. In this case, the transmission signal is generated in the transmission circuit 4802 of each of the FSO transceivers 4804. Free space optical technology is based on connectivity between FSO-based optical radio units, each consisting of an optical transceiver 4804 with a transmitter 4802 and a receiver 4806 providing full duplex open pair and bidirectional closed binary functionality. Each optical radio transceiver unit 4804 further includes an optical source 4808 plus a lens or telescope 4810 for transmitting light to other lenses 4810 that receive information over the atmosphere. At this point, the receiving lens or telescope 4810 is connected to the high sensitivity receiver 4806 via an optical fiber 4812. The transmission transceiver 4804a and the reception transceiver 4804b must have eyes on each other. Trees, buildings, animals, and atmospheric conditions can all obstruct the field of view needed for these communications media. Because line of sight is so important, some systems use a beam divergence or diffuse beam approach that includes a large field of view that can withstand significant line of sight interference without significantly affecting the overall signal quality. The system may also include an automatic tracking mechanism 4814 that maintains a tightly focused beam on the receiving transceiver 3404b even when the transceiver is mounted on a shaking tall building or other structure.

The modulated light source used with light source 4808 is typically a laser or light emitting diode (LED) that provides a transmitted light signal that determines all transmitter performance of the system. Only detector sensitivity in receiver 4806 plays an equally important role in overall system performance. For communication purposes, only lasers capable of modulating from 20 megabits per second to 2.5 gigabits per second can meet current market demands. In addition, how the device is modulated and how much modulated power is generated are both important for device selection. Lasers in the 780-850 nm and 1520-1600 nm spectral bands meet the frequency requirements.

Commercially available FSO systems operate in the near IR wavelength range of 750-1600 nm developed to allow one or two systems to operate at an IR wavelength of 10,000 nm. The physical and transmission properties of light energy as they travel through the atmosphere are similar throughout the visible and near infrared wavelength range, but there are many factors that influence which wavelength is selected for a particular system.

The atmosphere is considered to be very transparent at visible and near infrared wavelengths. However, certain wavelengths or wavelength bands may experience severe absorption. At near-infrared wavelengths, absorption occurs mainly in response to water particles (ie, moisture), which are inherent parts of the atmosphere, even under clear weather conditions. There are several transmission windows that are almost transparent (ie have less than 0.2 dB attenuation per kilometer) within the 700 to 10,000 nm wavelength range. These wavelengths are around a certain center wavelength with most free space optical systems designed to operate in windows of 780-850 nm and 1520-1600 nm.

Wavelengths in the 780-850 nm range are suitable for free space optical operation, and higher laser sources can operate in this range. Inexpensive CD lasers can be used at 780 nm, but the average lifetime of these lasers can be problematic. This problem can be greatly extended by driving the laser at a fraction of the maximum rated output power. At about 850 nm, optical source 4808 may include inexpensive, high performance transmitter and detector components that are readily available and commonly used in network transmission equipment. Highly sensitive silicon (SI) avalanche photodiode (APD) detector technology and advanced vertical cavity emitting lasers can be used within the optical source 4808.

VCSEL technology can be used to operate in the 780-850nm range. A possible drawback of this technique is the use of night vision scopes for beam detection, which still makes it impossible to demodulate the detected rays.

Wavelengths in the 1520-1600 nm range are well suited for free space transmission, and high quality transmitter and detector components are readily available for use within the optical source block 4808. The combination of low attenuation and high component availability within this wavelength range enables the development of wavelength division multiplexing (WDM) free space optical systems. However, components are generally more expensive and detectors are generally less sensitive and have a smaller reception surface area compared to silicon avalanche photodiode detectors operating at 850 nm wavelength. These wavelengths are compatible with erbium-doped fiber amplifier technology, which is critical for high power (greater than 500 milliwatts) and high data rates (greater than 2.5 gigabytes per second). 50-65 times more power can be transmitted at wavelengths 1520-1600 nm than can transmit at wavelengths 780-850 nm for the same eye safety rating. Disadvantages of this wavelength include the inability to detect beams with night vision scopes. Night vision scopes are one technique that can be used to align beams through alignment circuitry 4414. Class 1 lasers are safe under reasonably foreseeable operating conditions, including the use of optics for intrabeam viewing. Class 1 systems can be installed in any location without limitation.

Another potential light source 4808 includes a class 1M laser. Class 1M laser systems operate in the wavelength range of 302.5 to 4000 nm, which is safe under reasonably foreseeable conditions, but can be dangerous if the user uses optics within a portion of the beam path. Therefore, Class 1M systems should only be installed in locations where unsafe use of optical aids can be prevented. Examples of various characteristics of Class 1 and Class 1M lasers that may be used for the light source 4808 are illustrated in Table G below.

The 10,000 nm wavelength is relatively new to commercial free space optics and is being developed due to better fog transmission capabilities. There is a great deal of debate on these characteristics at present, because it depends heavily on the type and duration of the fog. Few components are available at the 10,000 nm wavelength because they are not used in ordinary communications equipment. In addition, the arrangement behind the window is not suitable since the energy of 10,000 nm does not penetrate the glass.

In this wavelength window, the FSO system should have the following characteristics: The system must be able to operate at high power levels, which is important for long distance FSO system transmission. The system must be able to provide the high speed modulation capabilities that are important for high speed FSO systems. The system must provide a small footprint and low power consumption that are important for overall system design and maintenance. The system must be able to operate over a wide temperature range without compromising performance before the system can be useful for outdoor systems. Also the average failure interval should exceed 10 years. Existing FSO systems typically use VCSELS to operate in the short IR wavelength range, and use Fabry-Perot or distributed feedback lasers to operate in the longer IR wavelength range. Several other types of lasers are suitable for high performance FSO systems.

Free space optical systems using orbital angular momentum processing and multilayer overlay modulation will provide several advantages. This system will be very convenient. Free space optics provide a wireless solution for last-mile connections or connections between two buildings. There is no need to dig or bury fiber optic cables. Free space optics also do not require an RF license. The system is upgradable and its open interface supports equipment from various vendors. The system can be installed behind a window, eliminating the need for expensive rooftops. It is also not affected by radio frequency interference or saturation. The system is also very quick. The system delivers 2.5 Gigabit data throughput per second. This provides enough bandwidth to transfer files between the two locations. As file sizes grow, free space optics provide the bandwidth needed to efficiently transfer these files.

Free space optics also provide a secure wireless solution. Laser beams cannot be detected by spectrum analyzers or RF meters. It is difficult to find because the light is not visible. The laser beam used to transmit and receive data is very narrow. This means that it is almost impossible to intercept the transmitted data. To intercept data, it must be within the field of view between the receiver and the transmitter. In this case, you will be alerted that the connection is lost at the receiving site. Therefore, only minimal security upgrades will be required for free space optical systems.

However, there are some drawbacks to free space optical systems. The distance of the free space optical system is very limited. The current operating distance is about 2 km. This is a powerful system with excellent throughput, but distance limitations are a major obstacle to large scale implementations. In addition, all systems must maintain visibility at all times during transmission. Obstacles such as the environment or animals can interfere with transmission. Free space optics technology must be designed to cope with changes in the atmosphere that can affect free space optical system performance capacity.

It is fog that can affect free space optical systems. Dense fog is a major challenge for the operation of free space optical systems. Rain and snow have little effect on free space optics, but fog is different. Fog is steam consisting of droplets only a few hundred micrometers in diameter, but a combination of absorption, scattering, and reflection can modify the light's properties or completely obstruct its path. The main answer to fog when deploying free-space optical-based wireless products is through network design, which reduces FSO linked distances and adds network redundancy.

Absorption is another problem. Absorption occurs when water molecules suspended in the Earth's atmosphere digest photons. This reduces the power density (attenuation) of the free space optical beam and directly affects the availability of the system. Absorption occurs more easily at some wavelengths than at other wavelengths. However, proper power usage and space diversity (multi-beam in FSO-based devices) based on atmospheric conditions help maintain the required level of network availability.

Solar interference is also a problem. Free space optical systems use high sensitivity receivers with larger aperture lenses. As a result, natural background light can interfere with free space optical signal reception. This is especially true for the high levels of background radiation associated with intense sunlight. In some cases, direct sunlight can cause link failure for several minutes when the sun is in the field of view of the receiver. However, it is easy to predict when the receiver is most sensitive to the effects of direct sunlight. If direct exposure to the equipment is unavoidable, narrowing the field of view or using a narrow bandwidth optical filter can improve system performance. Interference due to sunlight reflected from the glass surface is also possible.

Scattering problems can also affect connection availability. Scattering occurs when wavelengths collide with scatterers. The physical size of the spawner determines the type of spawning. If the scatterer is smaller than the wavelength, it is called Rayleigh scattering. If the scatterer is of a size comparable to the wavelength, this is known as Mie scattering. When scattering is much larger than the wavelength, it is known as nonselective scattering. In scattering, unlike absorption, there is no energy loss and there is only a directional redistribution of energy that can significantly reduce beam intensity over long distances.

I find that physical obstacles, such as birds or construction cranes, can temporarily block a single beam free space optical system, but this tends to cause only short interruptions. If the obstacle moves, the transmission automatically resumes easily. Optical wireless products use multi-beam (spatial diversity) to improve availability by handling transient disturbances as well as other atmospheric conditions.

Building movement can upset the receiver and transmitter alignment. Free-space optical-based optical radios use divergent rays to maintain connectivity. Combined with a tracking mechanism, a multi-beam FSO-based system offers much greater performance and simplicity of installation.

Scintillation is caused by artificial devices such as heating ducts that create temperature changes between hot air or other air pockets coming out of the earth. This can cause variations in signal amplitude, which results in "image dancing" at the free space optical based receiver. This glare effect is called "refractive turbulence". This mainly affects two things. Beam wander is caused by a violent vortex that is not larger than the beam. Beam diffusion is the diffusion of light beams that propagate through the atmosphere.

Referring now to FIGS. 49A-49D, to achieve higher data capacity within the optical link, additional degrees of freedom must be used from multiplexing multiple data channels. In addition, using two different orthogonal multiplexing techniques together can significantly improve system performance and bandwidth.

One multiplexing technique that can exploit this possibility is mode division multiplexing (MDM) using orbital angular momentum (OBM). OAM mode refers to a laser beam in a free space optical system or a fiber optic system, ^having a phase term of e ^{ilφ at} the wavefront, φ is the azimuth and l determines the OAM value (topological charge). OAM mode generally has a strength distribution in the form of a "donut" ring. Multi-spatial array laser beams with different OAM values are orthogonal to each other and can be used to transmit multiple independent data channels at the same wavelength. As a result, system capacity and spectral efficiency can be greatly improved in terms of bits / S / Hz. Free space communication links using OAM can support 100 Tbits / capacity. Various techniques for implementing this, as shown in FIGS. 49A-49B, include a combination of multiple beams 4902 having a number of different OAM values 4904 at each wavelength. Thus, beam 4902 includes OAM values, ie OAM1 and OAM4. Beam 4906 includes OAM value 2 and OAM value 5. Finally, beam 4908 includes an OAM3 value and an OAM6 value. Referring now to FIG. 49B, a single beam wavelength 4910 is shown using a first group of OAM values 4912 having both a positive OAM value 4912 and a negative OAM value 4914. Similarly, the OAM2 value may have a positive value 4916 and a negative value 4918 at the same wavelength 4910.

49C illustrates the use of wavelength 4920 with polarization multiplexing of OAM values. Wavelength 4920 may have multiple multiplexed OAM values 4922. Left hand or right hand polarization can be applied to the OAM value to further increase the number of available channels. Finally, FIG. 49D shows two groups of concentric rings 4960 and 4962 for wavelengths having multiple OAM values.

Wavelength division multiplexing (WDM) has been widely used to improve optical communication performance in both fiber optic systems and free space communication systems. OAM mode multiplexing and WDM are orthogonal to each other so that they can be combined to achieve a dramatic increase in system capacity. Referring now to FIG. 50, a scenario is illustrated where each WDM channel 5002 includes multiple orthogonal OAM beams 5004. Thus, using a combination of orbital angular momentum and wavelength division multiplexing, a significant improvement in the communication link to capacity can be achieved.

Current optical communication architectures present significant routing problems. Routing protocols for use with free space optical systems must take into account the optical communication requirements within the free space optical system. Thus, the free space optical network must be modeled as a directed hierarchical random sector geometry graph in which sensors route data through multi-hop paths through the cluster head to the base station. This is a new efficient routing algorithm for local neighborhood discovery and base station uplink and downlink discovery algorithms. The routing protocol must order Olog (n) storage on each node compared to the O (n) commands used within current technology and architecture.

Current routing protocols are based on link state, distance vector, route vector, or source routing and are very different from the new routing technology. First, current technology assumes that some of the links are bidirectional. This is not true for free space optical networks where all links are unidirectional. Second, many current protocols are designed for special networks where routing protocols are designed to support multi-hop communication between all node pairs. The goal of the sensor network is to route sensor readings to the base station. Therefore, the main traffic pattern is different from that of the ad hoc network. In sensor networks, node-to-base stations, base station-to-node and local neighbor communications are mainly used.

Recent studies have reported that 5 to 10 percent of link and wireless ad hoc networks are unidirectional due to various factors, taking into account the effects of unidirectional links. Routing protocols such as DSDV and AODV implicitly ignore these unidirectional links and therefore are not appropriate in this scenario because they use reverse path technology. Other protocols, such as DSR, ZRP, or ZRL, are designed or modified to accommodate unidirectional by detecting unidirectional links and then providing bidirectional abstraction over those links. Referring now to FIG. 51, the simplest and most efficient solution for dealing with unidirectionality is tunneling, which establishes a tunnel by bidirectional emulation for the unidirectional link using a bidirectional link on the reverse channel. Tunneling also prevents the bursting of acknowledgment packets and looping by simply pressing the link layer acknowledgment for tunneled packets received on the unidirectional link. However, tunneling works well for most bidirectional networks with few unidirectional links.

Within a network using only unidirectional links, such as free space optical networks, systems such as those shown in 51 and 52 would be more applicable. Nodes in the unidirectional network utilize directional transmission 5102 that transmits from node 5100 in a defined single direction. Each node 5100 also includes an omni-directional receiver 5104 capable of receiving signals coming to the node in any direction. In addition, as noted above, node 5100 will also include 0log (n) storage 5106. Thus, each node 5100 provides only a unidirectional communication link. Thus, the series of nodes 5200 shown in FIG. 52 can unidirectionally communicate with any other node 5200 and can pass from one desk location to another through a sequence of interconnected nodes.

Topological charge can be multiplexed on the wavelength for linear or circular polarization. In the case of linear polarization, the topological charges on the vertical and horizontal polarizations will be multiplexed. In the case of circular polarization, the retardation charge is multiplexed into the left hand and right hand circular polarizations.

Topological charge produces an appropriately varying refractive index, adjusting the voltage on the SLM causing distortion of the beam with a particular topological charge, such as a spiral phase plate (SPP), phase mask hologram, or spatial distortion modulator (as shown in FIG. 17E). It can be generated using SLM: Spatial Light Modulator. Different topological charges can be generated and mixed together to separate into separate charges.

Spiral phase plates can convert plane waves (l = 0) into torsional waves of a certain heli- city (i.e. l = +1), so the QWP (Quarter Wave Plate) converts linearly polarized light (s = 0) into circularly polarized light (i. = +1).

Crosstalk and multipath interference can be reduced by using multiple-input-multiple-output (MIMO).

Most channel corruptions can be detected using control or pilot channels and corrected using algorithmic techniques (closed loop control systems).

In addition to free space optics in real time, multiplexing of topology charges to RF provides redundancy and better capacity. When atmospheric disturbances or channel damage from glare affect the information signal, the free space optics can be toggled to RF and returned in real time. This approach still uses torsional waves in free space optics as well as RF signals. Most channel damage can be detected using control or pilot channels and corrected using algorithmic techniques (closed loop control systems) or by switching RF and free space optics.

In another embodiment shown in FIG. 53, both the RF signal and free space optics may be implemented within dual RF and free space optics 5302. The dual RF and free-space optics 5302 provide orbital angular momentum and multi-layer overlay for the RF signal 5310 and the free-space optical projection unit 5304 that transmits the light waves to which the orbital angular momentum is applied by multi-level overlay modulation. And an RF unit 5308 including circuitry necessary for transmitting information having the same. The dual RF and free space optics 5302 may be multiplexed in real time between the free space optical signal 5308 and the RF signal 5310 according to operating conditions. In some situations, free space optical signal 5308 will be best suited for transmitting data. In other situations, free space optical signal 5308 is not available and RF signal 5310 will be most suitable for transmitting data. Dual RF and free space optics 5302 can multiplex in real time between these two signals based on available operating conditions.

Multiplexing the topological charge in RF as well as free space optics in real time provides redundancy and better capacity. When atmospheric disturbances or channel damage from glare affect the information signal, the free space optics can be toggled to RF and returned in real time. This approach still uses torsional waves in free space optics as well as RF signals. Most channel damage can be detected using control or pilot channels and corrected using algorithmic techniques (closed loop control systems) or by switching RF and free space optics.

Referring now to FIG. 54, an alternative embodiment is shown where a horn or conical antenna is used for the transmission of a signal through the window or wall, instead of using the VCSEL for the transmission of the signal through the window or wall. The signal transmitted through the horn antenna is amplified for transmission to overcome the losses due to the transmission of the signal through the window / wall. The device provides an optical or RF tunnel through a window or wall without drilling a hole. Millimeter wave transmission system 5402 includes an outer portion 5404 located outside the window or wall 5406 and an inner portion 5408 located inside the wall or window. The outer portion 5404 includes an antenna 5410 for transmitting and receiving signals to an external source. In a preferred embodiment, the antenna comprises a 28 GHz antenna. However, those skilled in the art will appreciate that other antenna operating bandwidths may be used.

The transmitted and received signals are processed at 28 GHz circulator 5412. Circulator 5412 includes an RF switch for switching between three ports in outer portion 5404 and has good isolation. The signal input at port 2 in circulator 5412 is output at port 3 and the signal input at port 1 is output to port 2. Thus, the signal received by antenna 5410 is provided to port 2 of circulator 5412 and output to port 3. Port 3 signal is provided to an input of a power amplifier 5414. Similarly, the output of power amplifier 5416 is connected to input port 1 such that a transmission signal is provided to port 2 of circulator 5412 for transmission by antenna 5410.

Power amplifier 5412 amplifies signal strength for transmission through a window or wall. The signal output from the power amplifier 5414 is provided to the horn antenna 5418. Horn antenna 5418 transmits via a window or wall 5406 to an RF signal provided to horn antenna 5520 received from power amplifier 5414. The horn antenna can transmit and receive in a wide frequency band from 24 GHz to the electronic band. Within this range a specific operating band for the horn antenna is used. This band includes the 24 GHz band; 8 GHz A1 band; 28 GHz B1, A3 and B2 bands; Including but not limited to the 31 GHz band and the 39 GHz band. Horn antennas may also be of different sizes, for example to provide a gain of 10 dB or 20 dB.

The received signal is output from horn antenna 5520 to demodulation circuit 5542 for demodulation. The demodulator 5542 receives a signal output from the phase locked loop / local oscillator 5424 in addition to receiving the received signal from the antenna 5520. Phase locked loop / local oscillator 5624 is controlled in response to clock generation circuit 5526. The demodulated signal is provided from demodulator 5542 to analog-to-digital converter 5428 to produce a digital output. The digital signal is routed through router 5432 to the appropriate receiving side in the structure.

The signal to be transmitted is received from inside the building at the router 5430. Router 5430 provides a digital signal to a digital-to-analog converter 5432 that converts the digital signal into an analog format. The analog signal is next modulated by modulator 5434. The modulator 5342 modulates the signal in response to an input from the phase locked loop / local oscillator 5424 under the control of the clock generation circuit 5526. The modulated signal from modulator 5342 is transmitted through window / wall 5406 using horn antenna 5434. The signal transmitted by the horn antenna 5436 is received by an externally located receive horn antenna 5438. The output of horn antenna 5438 is provided at the input of power amplifier 5416, which amplifies the signal for transmission from antenna 5410 after passing through circulator 5412. Although the above description has been made in connection with the use of a horn antenna for transmission through a window / wall, conical antennas can also be used for transmission through a window or wall.

Referring now to FIG. 55, downlink loss between the transmit antenna 5410 and the receive circuitry in the inner portion 5408 is shown. The signal is received at -110 dBm. The gain of the receive antenna is 45dB and the loss is 2dB. Thus, the signal output from the receive antenna 5410 has an intensity of -67 dBm. Circulator 5412 has a loss of 2 dB and the signal from circulator 5412 has an intensity of -69 dBm. Power amplifier 5414 provides 27 dB to boost the signal to -42 dBm for transmission across the window / wall. Horn antenna 5418 provides a gain of 10 dBi for transmitting signals at 32 dBm. The window / wall provides about 40dB of loss. Receive horn antenna 5520 provides a gain of 10 dBi for receiving a signal at -72 dBm and outputting a -62 dBm received signal to internal circuit components.

Referring now to FIG. 56, the uplink signal strength is shown when the power amplifier is located outside the window / wall 5406. The transmitted signal has an intensity of 18 dBm before reaching the input of the horn antenna 5436. Antenna 5436 provides a gain of 10 dBi for transmitting signals at 28 dBm. Window / wall 5406 causes an overall loss of approximately 40 dB, dropping the signal strength to -12 dB. Horn antenna 5438 provides a 10 dBi gain in the signal and outputs the signal at -2 dBm. The power amplifier 5416 provides a gain of 26 dB for outputting a signal of 24 dBm to the Port 1 input of the circulator 5412. Power circulator 5412 provides an additional loss of 2 dB to output the signal to antenna 5410 at 22 dBm. The signal is transmitted from antenna 5410 with a gain of 45 dB and a loss of 2 dB to provide a transmitted signal strength of 65 dBm.

Referring now to FIG. 57, the uplink signal strength is shown when the power amplifier 5702 is located inside the building. Internal power amplifier 5702 is used when more power needs to be transmitted from an internal terminal. Before being input to the power amplifier 5702, the signal has an intensity of 18 dBm in the building. Power amplifier 5702 provides a 26 dB gain to transmit a signal at 44 dBm to the input of horn antenna 5436. Horn antenna 5436 provides a 10 dBi gain and the transmit RF signal is 54 dBm. The transmitted signal suffers approximately 40 dB loss through window / wall 5404, which drops signal strength to 14 dBm outside of window / wall 5404. Receive horn antenna 5438 provides a gain of 10 dBi to increase the signal strength to 24 dBm at the output of horn antenna 5438 provided to port 1 of circulator 5412. Circulator 5412 has a signal strength drop of 22 dBm due to a loss of 2 dB. Transmit antenna 5410 provides an additional gain of 45 dB and a loss of 2 dB to provide a transmitted output signal strength of 65 dBm.

Referring now to FIG. 58, the gain and loss in the downlink when no power amplifier is included is shown. A signal with a -103 dBm strength is received by the antenna 5410. Antenna 5410 provides a gain of 45 dB and a loss of 2 dB. This provides a 58 DBM signal at the output of antenna 5410 input to port 2 of circulator 5412. Circulator 5412 provides an additional 2 dB loss to the signal providing a -62 dBm signal from port 3 provided at the input of horn antenna 5518 providing a gain of 20 dBi. A signal having a value of -42 dBm is transmitted from the horn antenna 5418 through the window / wall 5406. Window / wall 5406 provides a 40 dB loss to the transmit signal providing a -82 dBm signal at the received horn antenna 5520. Horn antenna 5520 provides an additional 20 dBi gain for the signal output at -62 dBm to the remaining circuitry of the inner portion 5408 of the device.

Referring now to FIG. 59, the signal strength at various points in the uplink when the power amplifier is not provided is shown. The transmitted signal is provided to the input of the horn antenna 5432 at an intensity of 18 dBm. Horn antenna 5432 provides a gain of 20 dBi for outputting a signal of 38 dBm through window / wall 5406. Window 5406 causes a 40 dB loss in signal so that receive horn antenna 5438 receives the signal at -2 dB. Receive horn antenna 5438 amplifies the signal to 18 dBm with a gain of 20 dBi. The 18 dBm signal is input to port 1 of circulator 5412. Circulator 5412 causes a 2 dB loss in the signal output through port 2 at 58 dBm. The transmit antenna has a gain of 45dB and a loss of 2dB, making the signal transmitted from the antenna to 59dBm.

The dB losses described above for windows / walls and various system components are all approximate. Systems including other dB loss values and gains may also be used in connection with the embodiments described herein. It will be known to those skilled in the art how to determine the dB loss to be associated with a particular wall or window and related system components. One example of how the dB value can be determined is August 1, 2016, entitled "REGENERATION, RETRANSMISSION OF MILLIMETER WAVES FOR INDOOR PENETRATION," incorporated herein by reference, respectively. On November 22, 2016, filed in U.S. Provisional Application No. 62 / 369,393 and entitled "REGENERATION, RETRANSMISSION OF MILLIMETER WAVES FOR BUILDING PENETRATION USING HORN ANTENNAS." US Provisional Application No. 62 / 425,432, filed.

(Horn antenna)

Referring now to FIG. 60, another alternative embodiment is shown in which a horn antenna is used for the transmission of a signal through a window or wall. As mentioned above, the millimeter wave transmission system 5402 includes an outer portion 5404 located outside of the window or wall 5406 and an inner portion 5408 located inside the wall or window. The outer portion 5404 includes an antenna 5410 for transmitting and receiving signals to an external source.

The transmitted and received signals are processed at 28 GHz circulator 5412. The port 3 signal is provided to the input of the power amplifier 5414. Similarly, the output of power amplifier 5416 is connected to input port 1 such that a transmission signal is provided to port 2 of circulator 5412 for transmission by antenna 5410. The signal output from the power amplifier 5414 is provided to a 28 GHz horn antenna 5418. Horn antenna 5418 is transmitted from power amplifier 5414 via a window or wall 5406 to an RF signal provided to receive horn antenna 5520. The received signal is output from horn antenna 5520 to modulator circuit 5242 for demodulation. The demodulator 5542 receives a signal output from the phase locked loop / local oscillator 5424 in addition to receiving the received signal from the antenna 5520. Phase locked loop / local oscillator 5624 is controlled in response to clock generation circuit 5526. The demodulated signal is provided from the demodulator 5542 to the analog-to-digital converter 5428. The digital signal is routed through router 5432 to the appropriate receiver.

The signal to be transmitted is received from inside the building at the router 5430. In one embodiment, this will include a Wi-Fi router. Router 5430 provides a digital signal to digital analog converter 5432 to convert the signal into an analog format. The analog signal is modulated by modulator 5434. The modulator 5342 modulates the signal in response to an input from the phase locked loop / local oscillator 5424 under the control of the clock generation circuit 5526. The modulated signal from modulator 5342 is output through window / wall 5406 via horn antenna 5434. This signal is transmitted by the horn antenna 5438 or by an externally located receive horn antenna 5438. The output of horn antenna 5438 is provided to input power amplifier 5416 which amplifies the signal for transmission from antenna 5410 after passing through circulator 5412.

Horn antennas 5418, 5420, 5436 and 5438 may have high gains of up to 20 dB. The antenna pattern of this antenna has side lobes and front lobes. The front protrusion is projected toward the receiving antenna. In order to shield the surrounding environment from radiation from the side lobes of the horn antennas 5518, 5420, 5436 and 5438, a shield 6002 can be added above the horn antenna to provide adequate protection for the environment near the device. Shield 6002 acts as an absorber to block signals from the surrounding environment and may include any material necessary to include and absorb the emission of the horn antenna in the local area contained within shield enclosure 6002. .

(Patch antenna)

Referring now to FIG. 61, there is shown an alternative embodiment of using a patch antenna 6102 for the transmission of a signal through a window or wall 6104. The signal transmitted via patch antenna 6102 is processed in one of the manners described above for transmitting the signal through window or wall 6104. Patch antenna 6102 generates a directional radio wave beam to tunnel through the low-emissivity glass or wall. The device provides a light or RF tunnel through the window or wall 6104 without drilling any hole or requiring the creation of some type of signal transmissive portal. Millimeter wave transmission system 6100 includes an outer portion 6106 located outside of window or wall 6104 and an inner portion 6108 located inside of window or wall 6104. The outer portion 6106 includes an antenna 6110 for transmitting and receiving signals to an external source, such as a base station. The link budget between the base station and the antenna must be satisfied. In a preferred embodiment, the antenna comprises a 28 Hz antenna. However, it will be understood by those skilled in the art that other antenna operating bandwidths such as 24 GHz, 39 GHz, 60 GHz and other bandwidths may be used.

The transmit and receive signals received at the antenna 6110 are provided from the interior 6108 and processed by the transceiver processing circuit 6112. The transceiver processing circuit 6112 places the received signals in the antenna 6110 or places the signals received from the interior 6108 of the building described above to enable their transmission through the window or wall 6104 or the antenna Any circuit described above may be included to convert from a format that may pass through the window or wall 6104 for external transmission through 6110. The transceiver processing circuit 6112 can downconvert the frequency to a lower frequency EM wave that can penetrate glass and walls and can also be amplified using an antenna array. Components in the transceiver processing circuit 6112 may include, but are not limited to, RF circulators, power amplifiers, up-down converters, RF transmission circuits, optical transmission circuits, and the like.

The transceiver processing circuit 6112 places the signal in a format for transmission through the window or wall 6104. The signal output from the transceiver processing circuit 6112 on the line 6114 is provided to a patch antenna 6102a. Patch antenna 6102a transmits an RF or optical signal provided from transceiver processing circuit 6112 to receive patch antenna 6102b through window or wall 6104. The patch antenna 6102 may transmit and receive over a wide frequency band from 24 GHz to the e-band. Within this range, a specific operating band for patch antenna 6102 is used. These bands are the 24 GHz band; 28 GHz A1 band; 28 GHz B-1, A3 and B2 bands; 31 GHz band; 39 GHz band; And the 60 GHz band. Patch antenna 6102 can be of various configurations to provide various levels of gain therefrom. In one embodiment, the antenna may be configured to provide a gain of 18 dB.

The received signal is output from the patch antenna 6102b to the transceiver processing circuit 6118 on line 6161 for demodulation and further processing. The transceiver processing circuit 6118 may include any of the various configurations described above in connection with the internal transceiver circuitry. The demodulated and processed signal is provided from the transceiver processing circuit 6118 to the Wi-Fi router 6120 and transmitted to a receiving device in the structure.

The signal to be transmitted to the external receiver is received from the inside of the building in the Wi-Fi router 6120. Wi-Fi router 6120 provides the signals to transceiver processing circuit 6118 which converts the Wi-Fi data signal into RF format to be transmitted via wall or window 6104 as discussed above. The RF signal is output from the transceiver processing circuit onto the line 6120 to the patch antenna 6102c. The modulated signal from patch antenna 6102c is transmitted through window / wall 6104. The signal transmitted by patch antenna 6102c is received by receive patch antenna 6102d located outside of the building. The output of patch antenna 6102d is provided to transceiver processing circuit 6112 on line 6224. The signal is converted into a format necessary to enable transmission of the signal from the antenna 6110. This format includes 24 GHz, 28 GHz, 39 GHz, 60 GHz; Current cellular LTE frequencies; 3.5 GHz CBRS; 5 GHz; 24, 28, 39, 60, 70, 80 GHz mm-band or any other frequency band that suffers from signal loss problems when transmitted through windows or walls.

Referring now to FIG. 62, an example of a patch antenna array 6202 is provided. Patch antenna array 6202 includes a first layer 6202 and a second layer 6204 located over first layer 6202. First layer 6202 is directly connected to window or wall 6104. Each layer 6202/6204 includes a number of patch antennas 6206. Each of the first and second layers 6202/6204 transmits a signal across a window or wall 6104. Patch antenna array 6202 may transmit in all millimeter wave bands, such as 24 GHz, 28 GHz, 39 GHz, 60 GHz, and the like. The plurality of patch antennas 6206 may be configured in a rectangular, circular or elliptical configuration to generate a directional beam for carrying traffic payloads.

Referring now to FIG. 63, one of patch antennas 6206 of patch antenna array 6202 of FIG. 62 is shown. Patch antenna 6206 has an overall width along first edge 6302 of 1.23 mm and is 1.56 mm in length at second edge 6204. Patch antenna 6206 forms a slot 6308 to which transmission line 6308 is connected to patch antenna 6206. Slot 6308 has a length of .36 mm along first edge 6310 and a width of 0.1 mm at each side 6312 of transmission line 6308. Patch antenna 6206 is created on substrate 6314 made of FR408. Patch antenna 6206 has a relative dielectric constant of 3.75, a loss tangent of 0.018, and a thickness of 0.127 mm.

Referring now to FIG. 64, a transmission beam simulation for the antenna of FIG. 63 using a high-frequency structure simulator (HFSS) is shown. The single patch antenna produces a transmit beam 6402 with a peak gain of 3.8 dB and a 3 dB beam width of 80 ° as shown in FIG. The design and simulation of the patch antenna is carried out using the 'ANSYS HFSS' of the microstrip feed structure in preparation for manufacturing. The side lobe radiation can be absorbed by the absorbing material and the main lobe is directed to the receiving unit.

By using a patch antenna array as shown in FIG. 62, a high directional high gain beam can be generated as shown generally in FIG. 65. A plurality of patch antennas 6502 in the patch antenna array 6504 may each transmit a separate beam 6506. Each of the individual beams 6506 has an associated direction and gain. The output of patch antenna array 6504 will combine each of the individual patch antenna beams 6506 to produce a combined array transmission beam 6508. The combined transmit beam 6508 will have better directivity and greater gain than each of the individual beams 6506 generated by the individual patch antennas 6502. Thus, by using a patch antenna array 6504 to generate the transmit beam, a beam 6508 with sufficient gain and direction to pass through the window or wall to the receiver and overcome the associated signal loss would be possible.

Referring now to FIG. 66, another embodiment of a microstrip patch antenna array 6602 using a microstrip antenna array for 60 GHz band applications is shown. Microstrip patch antenna array 6602 includes a 2x8 microstrip patch antenna array 6604 using a conductor-backed coplanar waveguide (CB-CPW) loop feed 6605. Patch antenna array 6604 is comprised of an upper substrate layer 6604 and a lower substrate layer 6660. Conductor backed coplanar waveguide 6605 is located on lower substrate layer 6660 including a 32 mm by 28 mm plane made of quartz having a dielectric constant of 3.9, loss tangent of 0.0002, and thickness of 0.525 mm. Plane 6660 defines an input 6610 that connects to a transmission line providing an input to a patch antenna 6612 of a 2 × 8 patch antenna array formed on a plane 6660 forming a CPW-feed loop. Top substrate layer 6604 forms a plurality of patch antennas 6614 on a 'Rogers RO3003' substrate having a thickness of approximately 0.254 mm, a dielectric constant of 3.00 and a loss tangent of 0.001. This type of antenna provides 18dB of board-side gain at 61GHz and has a bandwidth of about 57GHz to 64GHz.

Referring now more specifically to FIG. 67, a patch antenna element 6700 is shown. Most of these patch antenna elements 6700 are located on the multilayer patch antenna array as described above. Antenna element 6700 includes a patch 6702 having a length L and a width W. Patch 6702 is supplied from input transmission line 6704 connected to the supply network and overlying substrate 6706 with height h. The micro strip patch antenna includes a first radiation slot 6710 along the first edge of the patch 6702 and a second radiation slot 6710 along the second edge of the patch 6702. The electric field at the opening of each slot can be broken down into X and Y components as shown in FIG. The Y component is out of phase and canceled out by the half wavelength transmission line 6704. The radiation field can be determined by treating the antenna as an opening 6800 as shown in FIG. 68 having a width W 6802 and a height h 6804.

 The transmission line model can be further analyzed in the following manner. Gr is the slot conductance and Br is the slot susceptance. These can be determined according to the following formula.

The input admittance of the patch antenna 6700 can be approximated as follows.

Is the end effect of the microstrip.

Rectangular patch antenna 6700 will resonate when the imaginary part of the input admittance becomes zero.

The terminal effect can be calculated according to the following formula.

The resonance frequency of the patch antenna 6700 is determined by the following equation.

In general, the width W of the aperture is determined by the following equation.

In addition to using patch antennas to generate directional, high-gain beams for signal transmission through windows or walls, patch antennas can be used to increase the bandwidth on the communication link between patch antennas through windows or walls. The application of orbital angular momentum (OAM) to the transmitted signal can be used. This is explained in more detail in the description below beginning with FIG. 69.

69-76 are filed on Jan. 4, 2017, entitled “MODULATION AND MULTIPLE ACCESS TECHNIQUE USING ORBITIAL ANGULAR MOMENTUM,” which is hereby incorporated by reference in its entirety. Laguerre-Gaussian (LG), Hermite-Gaussian (HG), Ince-Gaussian (IG) or orbital angular momentum described in patent application No. 15 / 398,5611 A multilayer patch antenna array 6802 can be used to transmit signals such as (OAM) signals. The multi-layer patch antenna array 6802 can transmit a first antenna layer 6904 to transmit a first aligned beam, a second antenna layer 6906 to transmit a second aligned beam, and a third aligned beam. A third layer 6908 for. Respective layers 6904, 6906 and 6908 are stacked in the same center. Although this embodiment is illustrated with respect to a multi-layer patch antenna array 6802 including only three layers, it should be understood that more or fewer layers can be implemented in a manner similar to that described herein. Patch antenna 6910 is disposed on the surface of each layer 6904, 6906 and 6908. Each patch antenna is arranged not to be obscured by the layers. Layers 6904, 6906 and 6908 are separated from each other by layer separating members 6912 providing a gap between each layer 6904, 6906 and 6908. The configuration of the layers of the patch antennas may be rectangular, circular or elliptical configurations to produce Hermit-Gaussian, Langer-Gaussian or In-Gaussian beams.

The patch antenna 6910 used in the multilayer patch antenna array 6902 is manufactured by Flame Retardant 408 (FR408) laminate manufactured by Isola Global of Arizona Chandler and having a relative dielectric constant of about 3.75. The total height of this antenna is 125 μm. The metal of the antenna is copper with a thickness of approximately 12 μm. The patch antenna is designed to have an operating frequency of 73GHz and 4.1mm of free space wavelength. The input 50 ohm line of the antenna is 280μm and the input size of the 100 ohm line is 66μm.

Each patch antenna 6910 is configured to transmit a signal in a predetermined phase that is different from the phase of each of the other patch antennas 6910 in the same layer. Thus, as further shown in FIG. 71, four patch antenna elements 6910 are included on layer 6904. Each antenna element 7504 has a separate phase associated with it, as shown in FIG. These phases include π / 2, 2 (π / 2), 3 (π / 2) and 4 (π / 2). Similarly, as shown in FIG. 72, layer 6906 has phase π / 2, 2 (π / 2), 3 (π / 2), 4 (π / 2), 5 (π, as indicated. / 2), eight different patch antenna elements 6910 including 6 (π / 2), 7 (π / 2) and 8 (π / 2). Finally, referring back to FIG. 69, twelve patch antenna elements 6910 are included on layer 6908. Each of these patch antenna elements 6910 is assigned a phase in the manner shown in FIG. This phase is π / 2, 2 (π / 2), 3 (π / 2), 4 (π / 2), 5 (π / 2), 6 (π / 2), 7 (π / 2), 8 (π / 2), 9 (π / 2), 10 (π / 2), 11 (π / 2) and 12 (π / 2).

Each antenna layer 6904, 6906, and 6908 is connected to a coaxial end-run connector 6916 to feed each layer of the multilayer patch antenna array 6902. Each connector 6916 is connected to receive a separate signal allowing transmission of separate aligned antenna beams in a manner similar to that shown in FIG. The emitted beams are multiplexed together by the multilayer patch antenna array 6902. Orthogonal wavefronts are provided that are spatially transmitted from each layer of the multilayer patch antenna array 6902 to increase capacity when each wavefront acts as an independent unique channel. The signal is multiplexed on a single frequency and propagated without interference or crosstalk between the multiplexed signals. The example for FIG. 70 illustrates the transmission of an OAM beam at OAM 1, OAM 2, and OAM 3 order levels.

It should be understood that other types of Hermit Gaussian and Langer Gaussian beams may be transmitted using the multilayer patch antenna array 6902 shown. The Hermit-Gaussian polynomials and the Langer Gaussian polynomials are examples of classic orthogonal polynomial sequences that are the eigen states of quantum harmonic oscillators. However, it should be understood that other signals may be used, such as orthogonal or Jacobi polynomials, Gegenbauer polynomials, legend polynomials, and Chebyshev polynomials. Legendre, Bessel, spherical, and in-Gaussian functions may be used. Q-functions are another class of functions that can be used as the basis for orthogonal functions.

The feed network 6718 illustrated in each layer 6904, 6906, 6908 uses delay lines of different lengths to determine the phase of each patch antenna element 6910. By constructing the phase as shown in Figs. 69-72, OAM beams of different orders are generated and multiplexed together.

Referring now to FIG. 73, a transmitter 7302 is shown for generating a multiplexed beam for transmission. As mentioned above, the multilayer patch antenna array 6802 includes a connector 6916 associated with each layer 6904, 6906, 6908 of the multilayer patch antenna array 6902. Each of these connectors 6916 is connected to a signal generating circuit 7304. Signal generation circuit 7304 includes a 60 GHz local oscillator 7308 for generating a 60 GHz carrier signal in one embodiment. The signal generation circuit 7304 may also operate with other frequencies, such as 70/80 GHz. The 60 GHz signal is output from the local oscillator 7306 to a power divider circuit 7308 that separates the 60 GHz signal into three separate transmission signals. Each of these separate transmission signals is provided to an IQ mixer 7310, each connected to one of the layer input connectors 6916. IQ mixer circuit 7310 is coupled to an associated additive Gaussian noise circuit 7312 to insert noise components into the generated transmission signal. The AWG circuit 7312 may also generate a 'SuperQAM' signal for insertion into the transmission signal. IQ mixer 7310 is entitled, "SYSTEM AND METHOD FOR COMMUNICATION USING ORBITAL ANGULAR MOMENTUM WITH MULTIPLE LAYER OVERLAY MODULATION", incorporated herein by reference in its entirety. HG, LG, IG, OAM signals in the same manner as described in US patent application Ser. No. 14 / 323,082, filed July 3, 2014 (US Pat. No. 9,331,875, registered May 3, 2016). Create

Usage of the transmitter 7302 is shown in FIG. 73. Multiplexed beams (Hermit Gaussian, Langer Gaussian, Ins Gaussian, etc.) as shown in FIG. 74 may be generated at a specific frequency for fast tunneling. This type of mode division multiplexing (MDM) achieves higher throughput with one frequency and several LG, HG or IG beams. As shown, the multilayer patch antenna array 6802 will generate a multiplexed beam 7402 for transmission. In this example, multiple OAM beams are shown with twists for various order OAM signals in a manner similar to that disclosed in US patent application Ser. No. 14 / 323,082. The associated receiver detector will detect various OAM rings 7404 as shown in each ring associated with a separate OAM processed signal.

When a signal is transmitted in free space (vacuum), the signal is transmitted in a plane wave. These can be expressed as described below. Free space includes a non-conductive medium (σ = 0), so J = σE = 0.

From the experimental results, Ampere's law and Faraday's law are expressed as follows.

Propagates in the z direction and therefore E and H are in the xy plane.

E can be oriented in the x-direction and H can be oriented in the y-direction without loss of generality, thus providing propagation in the z-direction. Given Amper's Maxwell's equation,

Next, the vector wave equation can be expressed as follows.

So in general:

therefore:

In free space

now:

이므로, 근축 가정

Since, parliamentary home

if,

The cylindrical coordinates can be expressed as

This gives the paraxial wave equation at the cylindrical coordinates

then:

In general, Eo can rotate in the xy plane and the wave still propagates in the z direction.

q to curvature of the phase front near the optical axis.

Where q ₂ is the output plane and q ₁ is the input plane.

은 z 축과 교차하는 파면의 곡률이다. here,

Is the curvature of the wavefront intersecting the z axis.

Therefore, for a complete plane wave R = ∞, the equation becomes

Where W ₀ is the beam waist.

Ray length is shown below.

Where n is the refractive index.

The multiple phase shift is expressed as follows.

The real part of P (z) represents the phase shift between Gaussian beam and ideal plane wave. Therefore, the basic mode is provided as follows.

here,

Higher order modes may provide other solutions. The solution to the rectangular equation is

In the rectangular coordinates can be determined as follows.

The cylindrical coordinate solution of the equation is

In cylindrical coordinates can be determined as follows.

은 또한

로 표현될 수도 있다.expression

Is also

It may be represented by.

The lowest mode is the most important mode and in fact this landscape mode is the same in both rectangular and cylindrical coordinates.

then

Referring now to FIG. 75, shown is a receiver 7502 for demultiplexing a signal received from a multiplexed signal generated using the transmitter 7302 of FIG. 73. Receiver 7502 includes a multi-layer patch antenna array 7502 as described above. The multilayer patch antenna array 7502 receives the received multiplexed signal 7504 and each layer 7504, 7506, 7508 of the antenna array 7502 is a multiplexed signal received from each connector output 7516 of a particular layer. Will extract the specific order of. The signal from each connector 7516 uses a 60 GHz local oscillator signal from oscillator 7508 to demultiplex the received signal in a manner similar to that discussed in connection with US patent application Ser. No. 14 / 323,082. 7506). The demultiplexed signal can be read, for example, by a real time oscilloscope 7510 or other signal reading device. Thus, each of the three transmitted signals is decoded at the receiver 7502 transmitted with each aligned OAM signal received from the transmitter 602. In another embodiment, a demultiplexing approach using a spiral phase plate (SPP) may be applied to detect OAM channels.

The signal transmitted by the transmitter 7302 or receiver 7502 can be used for information transfer between two locations in a variety of problems. This includes use for both front and back haul communications within a communications or data network.

The multilayer patch antenna array 7402 may transmit Hermit Gaussian beams using the processing discussed in connection with US Patent Application No. 14 / 323,082 or the Langer Gaussian beam. When transmitting a Langer Gaussian beam, information may be transmitted in various ways. A spiral phase plate and beam splitter approach may be used, a dual OAM mode antenna approach may be used, or the patch antenna described herein may be used. Such an implementation would be beneficial for both front hall and back hole applications.

In order to transmit several OAM modes of order l and amplitude alOAM, the antenna element must be supplied by the input signal according to the following equation.

It should be noted that the number of elements in the multilayer patch antenna array 7502 limits the number of possible OAM modes due to sampling. Due to aliasing, order modes greater than N / 2 are actually negative order modes.

Single Mode Link Budget

Asymptotic Formulation

The goal is to determine the asymptotic formula of the link budget at a distance, i.e., D → + (∞). We want to find the leading term for each value of l link budget -l. The link budget is given as follows:

로부터, 링크 버짓은 점근적으로 1/D^2|l|+2에서의 감쇠와 일치하는, 디케이드(decade)당 기울기 -20(|l|+1)dB의 직선으로 나타나는 경향이 있다. Fraunhof Street

From the link budget tends to appear as a straight line with a slope of -20 (| l | +1) dB per decade, which is asymptotically attenuated at 1 / D ^{2 | l | +2} .

(Asymptotic Representation with Gain and Free Space Loss)

The gain and free space loss can be determined by the equation

If | l | is a fixed value, each equivalent gain is R ^{2 | l |} And thus the link budget is R ^{4 | l |} It is improved by 2 times. Conversely, if R is a fixed value, | l | increases, the link budget decreases because asymptotically the effect of D is greater than that of Rt and Rr.

Referring now to FIG. 76, a 3-D model of a single rectangular patch antenna designed for 2.42 GHz and only one linear polarization is shown. The radiation pattern of this antenna is shown in FIG.

78A shows the radiation pattern of the circular array for OAM mode order l = 0 due to the higher lattice lobe. 78B, 78C and 78D show the radiation pattern for OAM mode orders at l = 0 (FIG. 78B), l = 1 (FIG. 78C), and l = 2 (FIG. 78D) near the array axis.

The asymptotic OAM path loss is

Given the e-band frequency, distance of 1000 m, and appropriate patch antenna element gain, other parameters can be calculated including the diameter of the transmitter and receiver array rings, the number of antennas, and the like.

Fabrication of patch antenna 7510 is performed through a design and layout process generally shown in FIG. 79, a clean room and lithography process for the manufacture of the antenna generally shown in FIG. 80, and a final test process shown in FIG. . Referring now to FIG. 79, the design and layout process is described in greater detail. Initially, the patch antenna is designed and simulated in step 7902 using ANSYS HFSS with a micro strip feed structure. ANSYS HFSS consists of a high frequency structure simulator. The software in the device stimulates the 3D radio wave electromagnetic field. ANSYS HFSS generates a GDSII file (graphic database system file used to control integrated circuit photo mask plotting) from the HFSS simulation in step 7904 and places the GDSII file in the AWR (Applied Wave Research Corporation) microwave office (MWO) layout. Export to To measure the antenna with ground-signal ground probe feeding, a previously designed conductor-supported coplanar waveguide for the microstrip transition design made using Agilent Momentum is an AWR MWO as a GDSII Agilent momentum file at step 7806. Imported into the layout. These two designs are brought together in step 7908 to add 12 μm weight and etch compensation to the lateral dimensions to account for the isotropic wet etch used in the manufacturing process. The final GDSII file for the layout is exported in step 7910 and provided to the clean room in step 1912.

Referring now to FIG. 80, a clean room process for patterning a copper layer on an FR408 laminate is shown. Initially, the double-sided Cu FR408 laminate is cut to the appropriate size (typically 1.5 "x 1.5") using scissors in step 8002. The FR408 laminate is washed by rinsing the laminate with acetone, isopropanol (IPA) and nitrogen (N2) in step 8004 and dried using program 2 of CPK solvent spinner in a solvent hood or with an appropriate chuck. In step 8006 the laminate is dehydrated bake at 130 ° C. for 2 minutes on a hot plate (eg, a Cole Farmer digital hot plate). Next, in step 8008, hexa methyldisilyan (HMDS) is deposited onto the laminate in a lane method using a yield engineering YES-310 vacuum hood oven. Laminate samples are placed in an HMDS oven for 20 minutes to improve resist adhesion. Next, in step 8010, the mask is cleaned using Program 2 of the CPK solvent spinner with an appropriate chuck. The mask is further cleaned using an automated mask cleaner (Ultratech Mask Cleaner) using program 0 DIW only in step 8012.

The lithographic process is performed in steps 8014-8034. First, the Shipley S1813 photoresist is spun on the backside of the laminate in step 8014 to protect the ground layer, for example using a Blower Science Cee Spin Coater System. In one embodiment, the spin coater system will operate at 3000 rpm at 3000 rpm / s for 60 seconds. The samples are soft baked in step 8016 for 90 seconds at 115 ° C. on a hot plate and hard baked at 130 ° C. for 60 seconds in a hot plate at step 8018. In step 8202, the S1813 resist is spun on the top pattern copper layer. In one embodiment, this is performed for 60 seconds at 3000 rpm at 3000 rpm / s. In step 8202, soft bake the sample for 90 seconds at 115 ° C. on a hot plate. In step 8024 the top of the sample is exposed at 110 mJ / cm 2 using Karl Suss MA6 BA6 contact aligner / printer. Next, the circuit is developed in step 8026 with Microposit MF-319 for 60 seconds in a beaker. Samples are rinsed in deionized water (DIW) and N2 in the base hood. A reactive ion etch process is performed at step 8032 to remove excess photoresist using a Techniques Series 85 RIE. This is achieved by applying 02 only at 180 mTorr of 50 W for 15 seconds. In step 8034 the samples are hard baked on a hot plate at 130 ° C. for 60 seconds. Lithography is examined under a Leica Inm optical microscope at step 8036 to ensure that the lithography is accurate and that the gaps are defined and not excessively developed.

In steps 8038-8046 a 12 μm copper layer is etched. Copper is etched at 1 minute intervals in step 8038 by stirring the sample in a Cu etchant. Inquiry step 8040 determines whether the Cu etching process is complete; otherwise, rotates the sample 90 ° in step 8082 and returns to stirring the sample in the Cu etchant in step 8038. If inquiry step 8040 determines that the Cu etching process is complete, control proceeds to step 8044 where the sample is cleaned with DIW and N2 and dried in the base hood. In the inquiry step 8046, the sample is inspected using a microscope to determine if Cu has been completely removed. If not, control returns to step 8038 for further agitation in the Cu etchant. Once all Cu is removed, control passes to stripping of the photoresist process.

Stripping of the photoresist occurs first by rinsing the sample with acetone, IPA, DIW and N2 and drying in a solvent hood or using Program 2 of CPK Solvent Spinner as an appropriate chuck. In step 8050 the samples are dehydrated bake at 130 ° C. for 5 minutes on a hot plate. The etched laminate sample is examined under a microscope at step 8302 to see if the gap is etched without undue etching of the area within the sample.

The generated patch antenna may be tested as shown in FIG. 81 to confirm the operation of the antenna. Initially, in step 8102, a DC test is performed on the antenna so that the G-S-G feed is not shorted. An RF test is performed in step 8104 to measure S11-return loss in the frequency band using an Agilent VNA at the Cascade M150 probe station. The radiation pattern of the antenna may then be measured at the appropriate frequency using an NSI spherical near field scanner at step 8106.

In a further configuration, the patch antenna can be used with the horn antenna to overcome the 40 dB loss that occurs through the window or wall. The foregoing embodiment will also be configured to meet FCC and OSHA requirements. In addition to the techniques described herein, other near-field techniques may be used to transmit information through a window or wall.

(Transceiver Chipset)

Referring now to FIG. 82A, through a window or wall 8202 without significant signal loss using an RF transceiver chipset that transmits a frequency to receive a signal from base station 8204 at a frequency that will not penetrate window / wall 8202. An embodiment of transmitting an RF signal is shown. Base station 8204 transmits a radio signal 8206 to building transmission penetration system 8230. Building transmission penetration system 8230 includes a first transceiver 8232 that implements a transmission chipset for receiving signal 8206 from base station 8204. The first transceiver 8232 is a second transceiver that implements a transmission chipset via a bidirectional transmission link 3236 for a signal transmitted to the structure and a transmission link 8238 for a signal transmitted to the base station 8204 out of the structure. And connected to 8212.

A second transceiver 8234 located inside the structure communicates with the Wi-Fi router 8220 via the transmission line 8222 and the receiving line 8224. Wi-Fi router 8220 communicates with a wireless device located within the structure. Transmission lines 8222 and 8224 are bidirectional communications between Wi-Fi router 8220 and second transceiver 8218 in a manner similar to allowing lines 8214 and 8216 to allow bidirectional communication between second transceivers 8234. Allow. The chipset implemented in the first transceiver 8232 and the second transceiver 8342 is not limited and may receive any number of frequencies including 3.5 GHz, 24 GHz, 28 GHz, 39 GHz, 60 GHz, 71 GHz and 81 GHz from the base station. And convert the windows / walls 8202 into a format that will penetrate into and out of the building. The signal may use any protocol, including but not limited to 2G, 3G, 4G-LTE, 5G, 5G New Radio (NR) and WiGi.

Referring now to FIG. 82B, a more specific embodiment of the system of FIG. 82A is shown for system 8200 for transmitting 60 GHz or other bandwidth wireless signal through window or wall 8202. In this embodiment, the Peraso chipset is used to enable transmission within the system 8200. Base station 8204 transmits a 60 GHz radio signal 8206 to millimeter wave system 8208. Millimeter wave system 8208 includes a first 60 GHz transceiver 8210 that implements a Ferraso chipset for receiving 60 GHz signal 8206 from base station 8204. The first Ferraso transceiver 8210 implements a second 60 GHz chipset that implements the Ferraso chipset through a transmission connector 8214 for signals transmitted to the structure and a transmission line 8216 for signals transmitted to the base station 8204 outside the structure. Is connected to the transceiver 8212.

The second Ferraso transceiver 8212 is located outside of the window or wall 8202 and transmits wireless signals to a third 60 GHz transceiver 8218 that implements the Ferraso chipset inside the window or wall 8202. A third Peraso transceiver 8218 located inside the structure communicates with the Wi-Fi router 8220 via a transmission line 8222 and a receiving line 8224. Wi-Fi router 8220 communicates with a wireless device located within the structure. Transmission lines 8222 and 8224 provide a Wi-Fi router 8220 in a manner similar to allowing lines 8214 and 8216 to allow bidirectional communication between the second Ferraso transceiver 8212 and the first Ferraso transceiver 8210. And two-way communication between the third Ferraso transceiver 8218. In the case of TDD, since three time slots are generally allocated to TX and one time slot is allocated to RX, slots are separated according to time and thus do not collide. Thus, in the case of bidirectional communication, there is no problem in terms of interference at the same frequency. When TX is performed with +1 helicity and RX with -1 helicity in the same frequency and time is used, full-duplex isolation using OAM twisted beams may be used.

Referring now to FIG. 83, an additional Ferraso chipset implementation is shown. 83 shows a base station 8302 communicating with a 60 GHz transceiver 8304 that implements a Ferraso chipset over a 60 GHz wireless communication link, and the signal is between a 2 and 60 GHz transceiver that implements the Ferraso chipset between the Ferraso transceivers 8304. 8230 shows a repeater configuration being transmitted in both directions. The second Ferraso transceiver 2908 generally has a wireless 60 GHz communication link 8310 with a third 60 GHz transceiver 8312 implementing the Ferraso chipset over the distance indicated as 8314. Repeater 8316, consisting of the Ferraso transceiver 8304 and the Ferraso transceiver 8308, allows signals from the base station 8302 to be boosted and transmitted over a longer distance to the Ferraso transceiver 3812. The Ferraso transceiver 8312 communicates bidirectionally with the router 8318 via communication lines 8222 and 8324. The repeater configuration as described above may be used to extend the transmission range of the 60 GHz signal transmitted from the base station 8308.

84A shows a top block diagram of a Peraso transceiver 8260 that may be used for transmission as described above. A pair of antennas 8846 are used to transmit (8460B) and receive (8462A) 60 GHz signals. The received signal according to one of the above embodiments is passed from antenna 8846A to demodulator 8264, and the signal received from antenna 8846A is responsive to the signal provided from phase locked loop / local oscillator block 8466. Is demodulated. The demodulated signal is passed to an analog-to-digital converter 8468 to convert the analog signal into a digital format in response to the demodulated signal and the clock signal provided by the clock generator 8270. The digital signal is provided at output 8852.

The signal to be transmitted is provided to the input 8424 in digital format and converted from digital to analog format in a digital-to-analog converter 8476 in response to a clock signal from the clock generator 8270. The analog signal is modulated in modulator 8478 in response to the analog and control signals from phase locked loop / local oscillator block 8466. The modulated signal is transmitted from the antenna 8846B in one of the configurations described above from the ferraso transceiver 8260. The Ferraso chipset is described in more detail in the Ferraso W110 WiGig chipset product overview of December 18, 2015, which is incorporated herein by reference.

Referring now to FIG. 84B, a more detailed application diagram of the Peraso chipset is shown. While the Peraso chipset in the 60 GHz band has been described, those skilled in the art will appreciate that the chipset may use any frequency that allows the repeater to expand its signal transmission capability. Examples include, but are not limited to, millimeter band, 28 GHz band, 39 GHz band, 2.5 GHz band, CBRS band (3.5 GHz), and Wi-Fi band (5 GHz). The Ferraso chipset consists of the W110 chipset used with Wiggig applications. The Ferraso chipset uses the PRS1125 integrated circuit 8402 and the PRS4001 integrated circuit 8404 to implement IEEE 802.11ad functionality. The Ferraso chipset delivers a complete ultra-fast USB 3.0 to WiGig solution. The PRS4001 low-power Wig baseband integrated circuit 8402 integrates an analog front end 8406 that includes a digital-to-analog converter 8080, an analog-to-digital converter 8410, and a phase locked loop 8412. The PRS 4001 circuit 8402 further includes a baseband physical layer 8414, a Mac layer 8416 and two RISC CPU cores. The PRS4001 circuit 8402 is compatible with IEEE 802.11ad. USB 2.0 and 3.0 interfaces 8424 enable USB communication. The PRS4001 circuit 8402 supports seamless connectivity with all Ferraso 60 GHz radios.

The PRS1125 integrated circuit 8404 is a single chip direct conversion RF transceiver that provides a 60 GHz single-ended receiver and transmission interface. The PRS1125 circuit 8404 provides up to 14dBm transmit output power, better than -21dB transmit EVM (16-QAM), less than 5dB receiver noise, and more than 70dB receiver conversion gain. The integrated single-ended 60 GHz antenna interface includes a transmit data path 8418 and a receive data path 8420. Phase locked loop 8822 is tuned to all channels of IEEE 802.11ad using an integrated controller. The Ferraso chipset delivers wireless storage, wireless display and multi-gigabyte mobile wireless applications. Antenna 8826 includes an NA grade patch antenna with 8.5 dBi gain in the entire 60 GHz band.

Communication between the Ferraso chipset transceivers can be performed in many ways to control the throughput between them. As shown in FIG. 85, communication between the first Ferraso transceiver 8502 and the second Ferraso transceiver 8504 may be performed in series over a single communication channel 8506. In this case, data is sequentially transmitted one item at a time through a single communication channel 8506. 86 shows a parallel transmission configuration. In this configuration, the transmission between transceiver 8502 and transceiver 8504 occurs over a number of channels 8608 operating in parallel. In this configuration, to increase data throughput, different data streams may be transmitted simultaneously over parallel communication channel 8908. In a parallel configuration, the data stream is petitioned into two multiple sub-streams and transmitted over a separate parallel channel 8908. The results can be combined together at the receiver 8504.

87 shows a functional block diagram of a Peraso transceiver 8702 located on the outer side of window or wall 8006. Since the Ferraso transceiver 8702 is located outside of the window or wall 8704, some manner is needed to provide power to the Ferraso transceiver 8702. The power unit 8706 located on the outer side of the wall 8704 can power the Ferraso transceiver 8702 in a number of ways. In one implementation, the power unit may include a solar cell and solar generation circuit for generating power. In one embodiment, the maximum power consumption for a Peraso transmitter located on an exterior wall or window is 15W. In order for the transceiver to provide a transmission power of 14 dBm or approximately 25 mW, 15 W power consumption is generated. If 15W of power is needed for 20 hours of the day, about 300Whr of energy is needed to support the transceiver each day. A power unit with an efficiency of 1.25 operating for 24 hours can provide approximately 375 Whr of energy. The 375Whr is divided by 3.5 (approximate number of solar hours in winter) to provide the solar capacity required for a 100W transceiver.

Another method of powering an external Peraso transmitter is described in FIG. 88. An internally located laser 8802 is used to transfer energy within the laser beam 88042 to a photodiode 8906 located on an external Ferraso transmitter. Since the wall blocks the beam, the laser beam 8904 will be transmitted through the window. The power required for the transmitted laser beam is defined as follows.

The efficiency of the optics, Eff _Optics , depends on the type of glass being transmitted. The pane can be commercial or household. For home window glass such as ClimaGuard 70/36, the optical efficiency is 0.68 at a transmission wavelength of 445 nm. For commercial window glass such as SunGuard SN 68, the optical efficiency is 0.64 at a transmission wavelength of 445 nm.

The efficiency of the silicon photodiode, Eff _PVCell, is defined as

Thus, the optical power that must be transmitted at 450 nm can be determined using optical and photodiode efficiency in the following manner.

Thus, the number of laser diodes required to deliver 34W to a 450nm, 4.5W blue diode is 45W / 4.5W or about 8 diodes.

(VCSEL alignment and power)

Referring now to FIG. 89, a VCSEL 8902 is shown. Since one VCSEL is located outside the window and the second VCSEL is located inside the window, there must be a way to align the optical transmission links provided from one VCSEL to the other. One way in which this alignment can be achieved is to have alignment holes 8904 located in multiple locations on the VCSEL 8902. In the embodiment shown in FIG. 89, alignment holes 8904 are located at each corner of VCSEL 8902. This alignment hole 8904 is used in the manner shown in FIG. 90 to align the first VCSEL 8902a with the second VCSEL 8902b. Thus, by visually aligning each alignment hole 8904 located at each corner of the VCSEL 8902a and VCSEL 8902b, the light transmission circuit in the VCSEL can be aligned through the window 9002.

Instead of using an external power input, the VCSEL 8902 located in the window can be powered in another way, as shown in FIG. 91, and FIG. 91 shows the VCSEL 9102 and the window or wall inside the window or wall 9002. The VCSEL 9106 is located outside of it. The power supply 9108 is provided directly to the internal VCSEL 9102 through some type of input connection. Power combiner 9210 in inner VCSEL 9102 is coupled with similar power combiner 9112 in external VCSEL 9106. If the VCSELs 9102 and 9106 are located in transparent windows, photo inductors or other types of optical power couplers may be used in the power coupling devices 9110 and 9112. VCSELs 9102 and 9106 are located opposite the opaque walls, or optical signal inductive coupling devices such as coils and doctors may be used in power coupling devices 9110 and 9112. In this manner, power combiner 9210 provides power to power combiner 9112 to power external VCSEL 9106.

(System power)

Referring now to FIG. 92, illustrated is the manner in which power may be provided to external system components 9202 located outside of the system and internal system components 9304 located within interior 5608. Internal system component 9304 includes the antennas, modulators, demodulators, and other components described above to generate signals for transmission and determine received signals. External system component 9202 is comprised of the circulator, power amplifier, and horn antenna described above. Internal system component 9304 is connected to an internal power system 9206 that can be plugged into a power system located within a building. Since internal system component 9304 and external system component 9202 are separated by window / wall 9002, there must be some way to transmit or provide power to external system components. One way of doing so includes the use of a power system 9208 powered by a number of solar cell panels 9210 located outside of the building to which external system components 9202 are connected.

The power required from power system 9290 to external system component 9202 is approximately .76W. One way to provide this .76W power is to use solar panel 9210. A solar cell plate providing .76 W or 1 W to the solar cell plate 9210 can be used. For a 24 hour power supply system, 0.76 W would require 18.24 W hours of power. If 18.24WW hours are provided with an efficiency of 1.25%, 22.8W hours are required. Dividing the efficiency of 22.8W hours by 3.5 hours (#daylight hours in winter) gives a total of 6.52W. Likewise for a 1W system, 24W hours are required to provide 1W per day. At 1.25% efficiency, 24W hours requires 30W hours. Dividing 30 hours by 3.5 hours of daylight available in winter, you get 8.57W hours. The solar cell plate 9210 used for powering may be similar to the solar cell plate used to charge smart phones and tablets. This type of panel includes both 7W and 9W charging panels that meet .76W and 1W energy level requirements.

7W portable solar chargers with high efficiency solar charging panels typically weigh 0.8 pounds. The typical size of this device is 12.8 x 7.5 x 1.4 inches (32.5 x 19 x 3.5 cm). The other 7W amorphous solar battery charger panel measures 15.8 × 12.5 × 0.8 inches (40 × 31.75 × 2 cm) and weighs 3 pounds. An alternative 9W charging panel with a single crystal cell measures 8.7 × 10 × 0.2 inches (22 × 25.5 × 0.5 cm) and the flexible solar panel measures 12 × 40 inches (30.5 × 100 cm). Another 9W high efficiency solar panel is 8.8 x 12.2 x 0.2 inches (22.35 x 31 x 0.5 cm).

Referring now to FIG. 93, instead of using a solar panel, external system component 9202 may use the transmitted laser power to power external system components rather than using a solar drive system. Internal system component 9304 has a power system 9302 that provides power to all components inside the window or wall 9304. Power system 9302 has internal power connections 9308 to a power outlet, for example, located within a building. The power system 9302 provides system power to the internal system components 9304 in a known manner. The power system 9302 also provides power to the laser transmitter 908. The laser transmitter may comprise a laser diode in one embodiment. The laser transmitter 9908 generates a laser beam 9910 that is transmitted through a window 9304 to a photovoltaic receiver (PV receiver) 9312 located outside of the window 9304. The laser transmitter 9308 includes a set of optical elements for forming the beam size to be transmitted to the PV receiver 9312. The generated laser power can be defined according to the following equation.

The optical power required for a PV receiver to sense energy at 445 nm can be defined in the following way.

This is the wavelength of the receiver laser.

(하마마쓰 Si-포토다이오드)

(Hamamatsu Si-photodiode)

Thus, a 2W laser diode is needed to supply power at 445 nm. The PV receiver 9312 converts the received laser light energy back into electricity. Power generated by the PV receiver 9312 in response to the received laser beam 9310 is provided to the power system 9314. Power system 9314 powers external system components 9202 to enable their operation.

Referring now to FIG. 94, there is shown another way of powering external components from an internal power source using inductive coupling, rather than using a solar panel or laser source, wherein external system components 9202 are external systems. The power provided by magnetic induction or magnetic resonance coupling to an internal power source through window / wall 9504 can be used to power the components. Internal system component 9304 has a power system 9402 that provides power to all components on the interior portion of window or wall 9504. The power system 9402 has internal power connections 9906 to a power outlet, for example, located within a building. Power system 9402 provides system power to internal system components 9304 in a known manner. The power system 9402 also provides power to the induction coil or magnetic resonator 9908. Induction coil or magnetic resonator 9908 enables magnetic coupling with a second induction coil or magnetic resonator 9412 located outside window / wall 904. Induction coils or magnetic resonators 9408 and 9412 enable induction or resonant coupling of power from internal power system 9402 to external power system 9414. Power received from the induction coil or the magnetic resonator 9412 is provided to the power system 9414 in response to the received electromagnetic energy 9410. The power system 9414 provides power to external system components 9202 so that their operation transmits signals through the window / wall 904.

In addition, in addition to the actively powered devices shown in FIGS. 92, 93 and 94, passive power devices that do not power external components but provide shorter distances or higher power from internal components in a building may be used. Can be.

Induction coils 9408/9412 provide inductive coupling of power between internal and external circuits, while magnetic resonators 9408/9412 use magnetic resonant coupling to transfer power between circuits. With regard to the induction coil, the coupling coefficient between the coils can be calculated in the following manner. Referring now to FIG. 95, the mutual inductance between two circular loops separated by a distance d (9502) with radii of a (9504) and b (9506), respectively, can be calculated using Neumann's equation: Can

Where ds and ds' are the incremental sections of the circular filament, r is the distance between the two sections, and is defined as

Substituting the above contents in Neumann's equation, it becomes as follows.

The integral of the above equation can be rewritten using elliptic integral and is calculated as

Where K (m) and E (m) are elliptical integrals of the first and second types, respectively, and m is defined as

Assume a value between 0 and 1.

The solution of elliptical integration of the first and second kind can be approximated by

For low m values, the representation of the power series shows reasonable accuracy. However, as m increases, both ellipses deviate from this numerical integral. For the first kind of lip-to-inward goal, there is a tendency to be progressively infinity than the solution computed by numerical integration, as when approaching unity.

Substituting the equations for K (m) and E (m) into the equation for M (m), we get

Next, substituting the equation for m into the equation above gives the equation for mutual inductance, which is a function of the distance between two circular coaxial loops:

For two coils with n1,2 turns, the equation can be adjusted as follows.

This represents the mutual inductance of the two coils with n1,2 as a function of distance d, permeability μ of the material surrounding the coils and the inner radius of the two coils.

Figure-of-Merit (U) can be described in terms of the Q factor that describes the ratio between the energy stored by the inductor's loop and the power consumed in a given cycle. The figure of merit is based on the following equation: other coil parameters such as wire radius Ra, loop radius a, permeability of free space μ _{0 (} since the core of the loop is air), the conductivity of the core material and the distance between the primary and secondary Depends on

In one embodiment, the transmitting coil will have the characteristics of a loop radius of 6.25 cm, a wire radius of 10.25x10-3, four coil turns, a distance between the primary and secondary loops of 46 mm and an operating frequency of 6.78 MHz.

Referring now to FIG. 96, a table is provided that provides information regarding the efficiency of coils over various coil radii, the number of turns in a coil, and different coil heights.

Referring now to FIGS. 97 and 98, the manner in which coils 9702 are inductively coupled to each other and resonator circuit 9802 inductively resonates with each other is shown. 97 shows how alternating current is provided from oscillator 9704 to L1 coil 9702 in response to input voltage 9706. An alternating current in the L1 coil 9702 creates an alternating magnetic field 9908, which in turn induces an alternating current in the secondary coil L2. This allows current to be provided to the rectifier 9710 provided to the load 9712. The magnetic field generated by the primary coil 9702 radiates equally in almost all directions. The flux produced by the magnetic field falls off quickly with distance according to the inverse square law. Therefore, the secondary coil L2 9702 should be disposed as close as possible to the primary coil L1 9702 to block the maximum magnetic flux.

Referring now to FIG. 98, magnetic resonance wireless charging may be used that requires precise alignment and proximity between coils in order to overcome the major drawback of inductive wireless charging due to the requirement to closely couple the coils. Magnetic resonance can be used to charge any active component from inside a building to outside the building using coils of different sizes. The basic concept of self-resonant power transfer is to tunnel energy from one coil to another in a specified way across a window or wall, instead of spreading the energy forward in the main coil. The self-resonant wireless charging circuit receives an input voltage VS across an input portion 9804 that applies a voltage VS to the oscillator 9906. The output of the oscillator 9906 passes through the drive coil 9808. Drive coil 9808 generates a current that includes coil 9810 with capacitor 9812 coupled across coil 9810 in primary resonator circuit 9802. Resonator circuit 9802a is coupled with resonator circuit 9802b to provide self resonant wireless charging. Resonator circuit 9802b includes coil 9814 with capacitor 9816 connected across the coil. Resonator circuit 9802b is coupled with drive coil 9818 coupled to rectifier 9820, which is used to drive load 9822. The basic concept of magnetic resonator power transfer is that energy from resonator circuit 9802a tunnels into resonator circuit 9802b instead of diffusing omnidirectionally from primary coil 9802a.

In order to use inductive coupling and magnetic resonance coupling to provide wireless power transfer from inside a building to outside of the building through a window or wall using the above-described millimeter wave transmission system, it depends on whether inductive coupling or magnetic resonance coupling is used Design considerations should be addressed. In order to provide wireless power transfer using inductive coupling, high magnetic coupling is required where the distance between the transmit power unit and the receive power unit must be very small. Standards available for inductively coupled wireless power transfer include Qi and PMA. Using standards between 5W and 15W can be transmitted over small distances of 5-10mm.

Wireless power transfer using magnetic resonant coupling, also known as HR-WPA (High Resonant Wireless Power Transfer), uses loosely coupled magnetic resonance for power transfer. High-quality coefficient magnetic resonators enable efficient energy transfer at lower coupling speeds, which can deliver power over longer distances between transmit and receive power supplies while providing more positional freedom. Existing standards include Rezence (WiTricity) and WiPower (Qualcom).

Referring now to FIG. 99, shown is a functional block diagram of a self-resonant wireless power transfer system that can be used to power the millimeter wave system of the present invention. The AC voltage signal is provided to an AC input 9902. The AC voltage signal is applied to an AC / DC converter 9904 that converts the AC signal into a DC signal. The direct current signal from the AC-DC converter 9904 is applied to the DC / RF amplifier 9906. The DC / RF amplifier 9906 is a high efficiency switching amplifier that converts the DC voltage into an RF voltage waveform used to drive the source resonator. The RF voltage waveform from the DC / RF amplifier 9906 is applied to add the impedance matching network 9908. Impedance matching network 9908 provides impedance matching and improves system efficiency. The signal from impedance matching network 9908 is provided to a transmitting side source resonator 9910 that links the signal to receiver side device resonator 9912. Source resonator 9910 and device resonator 9912 are high quality factors that enable efficient energy transfer at low coupling speeds (larger distance and / or positional freedom) between the transmitting side and the receiver side located opposite the window or wall. It is a resonator. This energy coupling is called high resonant wireless power transfer (HR-WPT). Power delivered to the device resonator 9912 goes to the second impedance matching network 9914 and the RF / DC rectifier 9916. Rectifier 9916 is used for load 9918 requiring a DC voltage and converts the received AC power back to a DC signal.

The source resonator 9910 and the device resonator 9912 have characteristics that can be described by two basic parameters, the resonance frequency ω 0 and the intrinsic loss factor Γ. The ratio of these two parameters defines the quality factor (Q) of the resonator, (Q = ω 0 / 2Γ), a measure of how well the resonator stores energy. The resonator energy vibrates at the resonant frequency between the inductor (energy stored in the magnetic field) and the capacitor (energy stored in the electric field) and dissipates in the resistance. The resonant frequency and quality factor of the resonator are defined as follows.

The equation for Q shows that reducing circuit losses, that is, reducing R, increases the quality factor of the system. Q electromagnetic resonators are typically made of low absorption conductors and components, resulting in a relatively narrow resonant frequency width.

인 상호 인덕턴스(M)를 통해 결합되는 인덕터(LS)(10006) 및(LD)(10008)로 표현된다. 각 코일은 공진기(CS(10010) 및 CD(10012))를 형성하기 위한 커패시터를 갖는다. 저항(RS(10014) 및 RD(10016))은 코일(10006, 10008) 및 각각의 공진기의 공진 커패시터(10010 및 10012)의 옴 및 방사 손실을 포함하는 기생 저항이다. 부하는 RL(10018)로 표시된다.By placing source resonator 9910 close to device resonator 9912, coupling between devices that allow the resonators to exchange energy can be achieved. A schematic diagram of a coupled resonator is shown generally in FIG. 100. The source voltage is a sinusoidal voltage source 10002 having an amplitude Vg at a frequency ω having an equivalent generator resistance Rg 10004. Source and device resonator coils

It is represented by inductors LS 10006 and LD 10008 that are coupled through phosphorus mutual inductance M. Each coil has a capacitor for forming resonators CS 10010 and CD 10012. Resistors RS 10014 and RD 10016 are parasitic resistors that include the ohmic and radiation losses of coils 10006 and 10008 and the resonant capacitors 10010 and 10012 of each resonator. The load is represented by RL 10018.

The analysis of the circuit of FIG. 100 provides the power delivered to the load resistor 10018 divided by the maximum power available from the source, resonating at ω according to the equation below in both the source and the device.

Where U is the figure of merit for the system.

Generator resistors 10014 and 10016 and load resistors 10018 are selected to provide the best system performance (performed by an impedance matching network) according to the following equation.

The efficiency of power transfer defined above is maximized according to the following equation.

 The best efficiency of the wireless power transfer system depends on the system performance index, which can be written in terms of the magnetic coupling coefficient k of the resonant period and the resonator quality coefficients QS and QD at no load.

The magnetic coupling coefficient k is a function of the relative size of the resonator, the resonance period distance and the relative direction of the resonator. The above equation shows that using a high-quality factor resonator allows efficient operation even at low coupling speeds. This does not require accurate positioning between the source resonator and the device resonator, and the distance between the coils is wider, providing more positional freedom and movement freedom. Since accurate positioning is not necessary, it is possible for a consumer to install internal and external transceivers inside and outside the window or wall.

The figure of merit U depends on other coil parameters such as the wire radius Ra, the loop radius a, the permeability of free space μo, the distance d between the primary and secondary loops and the conductivity of the core material. The figure of merit U can be expressed as a Q factor that describes the ratio between the energy stored by the loop and the power consumed in a given cycle.

Here, sigma represents the kind of commit of the material, and c is the speed of light.

Referring now to FIG. 101, a circuit diagram of a generator for converting 50 Hz grid AC to kHz, such as the DC / RF amplifier 9906 of FIG. 99, is provided. This shows the potential power source of a wireless energy transfer system that uses a rectifying and switching network to convert the power grid AC to the operating frequency of the energy transfer system. FIG. 101 shows a simple example of a power supply comprising a rectifier 10102 comprising four diodes 10104 and a switching network 10104 comprising four power MOSFET transistors 10108. Capacitor 10110 is connected between rectifier 10102 and switching network 10106. The resistance of the power supply is in the range of 250 m to 400 m. Input to the power source is provided across rectifier 10102 across terminal 10112. The output v1 10114 from the switching network 10106 is approximately square voltage. The Fourier component of the normalized square signal, f (t), is

Referring now to FIG. 102, illustrated is how impedance matching can be used to overcome the losses between resonators 10202 and 10204. Schematic representations of the resonators 10202 and 10204 are provided in the manner described above for transmission through the window 10206. Two resistors (Rthin) 10208 are inserted into the inductors (LS and LD) respectively to mimic and model the eddy current losses due to the thin silver layer of the 'Low-e' class. With proper impedance matching through resistance and / or matching control using the impedance matching network, coils and resistors described above, the loss is overcome by modifying the coil's turn, coil area, coil's permeability (material type) and frequency of the resonant frequency can do.

103 and 104 show perspective and side views of an external transmission circuit 10302 and an internal transmission circuit 10304 using inductive or resonant coupling for transferring power from a Ferraso chipset and an internal transmission circuit to an external transmission circuit. The external transmission circuit 10302 consists of an antenna 10306 that receives millimeter wave transmission from a base station or other external transmission source. In alternative embodiments, antenna 10306 may also include a Peraso transceiver that enables direct reception of transmissions from other Peraso transceivers. The Ferraso transceiver 10308 is used to transmit a signal through a window or wall that separates the external transmission circuit 10302 from the internal transmission circuit 10304. The coil 10310 is used for inductive power transfer or self resonant power transfer from inside a building in the manner described above. Circuit board 10312 is used to interconnect the electronic components of the antenna 10306, the Ferraso transceiver 10308, and the coil 10310.

Internal transmission circuit 10304 includes a Ferraso transceiver 10312 for transmitting and receiving signals with the Ferraso transceiver 10308 in the external transmission circuit 10302. Internal coil 10314 enables inductive or self-resonant power coupling with external transmission circuit 10302. In addition, the circuit board 10316 enables the interconnection between the Ferraso transceiver 10312, the coil 10314, and any other internal electronic circuit.

With regard to the panes through which signals or power must pass, the relative dielectric constant, power transfer rate, phase and reflectivity can be calculated according to the following Drude model.

ω _p : bulk plasma frequency

γ: in-band damping period

For silver: ω _p = 9.6ev, γ = 0.0228ev

k 손실 loss due to absorption

Absorbed power:

Phase:

The values of ε _r , n, k, absorption power and absorption loss are illustrated in FIG. 105.

로 정의될 수 있고, 두 층에 대한 반사 손실은

로 정의될 수 있다. 흡수 손실은

로 정의된다. 이들의 값은 도 106 및 107에 더 구체적으로 예시되어 있다. 이 값들은 반사율(R)을 기초로 하여 아래와 같이 결정될 수 있다.Return loss for one layer

Where the return loss for both layers is

It can be defined as. Absorption loss

Is defined as Their values are illustrated more specifically in FIGS. 106 and 107. These values can be determined as follows based on the reflectance R.

The absorption coefficient is as follows.

(Application using a home IP network system)

Current broadband systems use wired broadband with fiber optic connections to transfer information from network providers to consumers. For example, the AT & T U-verse has fiber to the node and copper to the premises, or in some cases provides fiber to all the paths to the premises. Fiber optics to premises systems are expensive and require a lot of time to deploy. Other solutions include DirectTV, DLS modems from Frontier and cable boxes from Charter or Comcast. Another solution is to implement broadband wireless delivery. However, when providing broadband using wireless high frequency RF waves, problems arise with signals that cannot penetrate the windows and walls of homes and buildings.

In traditional cable TV or satellite networks using broadcast RF video technology, all content flows continuously downstream to each customer, and the customer switches the content on the set-top box. Customers can choose from a number of options offered by cable / satellite providers, which are provided through pipes that flow into homes / businesses. Broadcast networks are just one way of transferring data from a provider to a consumer. Up to now, antennas were placed on the roof to receive signals from the hub, then drilled through different floors to allow signals to penetrate the building. This approach from the building roof to individual units in the building can be very costly and time consuming for the operator. Another approach is to send beams from the hub to individual devices, but this can cause the signal to reach the windows or walls of the building. Losses are caused by walls or windows when a wireless beam tries to penetrate a building. This loss is enormous for millimeter wave wireless signals, so a method of providing broadband transmission using the techniques described above would be very advantageous.

One way to overcome the above mentioned problem with regard to wireless broadband transmission is shown in FIG. 108. An improved combined home IP network system 10806 may be provided by combining an existing home IP network system 10802 with a millimeter wave transmission system 10804. Millimeter wave transmission system 10804 has the advantages of higher bit rates, more accurate beam shaping and steering, and smaller footprint components. Residential IP network system 10802 includes a combination service consisting of the Internet, TV, and VoIP telephony services. These services can be ordered in bundles or separately, and not all service combinations are available. TV services are based on IPTV (Internet Protocol Television), which is used to provide TV services. Network system 10802 also uses IP technology (Internet Protocol technology) to operate TV, computer, home telephone, and wireless devices together using Internet technology. This provides many useful features and gives you better control of the device in a service delivery manner. The use of IP technology also provides more personalization so that services can be tailored to the exact needs of consumers. Examples of this type of service include AT & T U-verse, DirectTV, Fronttier's DSL modem, Charter or Comcast cable box. The residential IP network system 10802 and video backbones provide high quality video, advanced features, and other applications. The residential IP network system 10802 is a two-way IP network provided to a customer's home via fiber optics or fiber optic node technology.

The millimeter wave system 10804 enables the transmission of signals through windows or walls as described above. By combining the millimeter wave system 10804 with the home IP network system 10802, wireless broadband transmission is a loss caused by the transmission of signals through windows or walls that degrade system performance from network providers to user devices located inside buildings. Can be provided without. Within the combined home IP system 10806, the content is maintained in the network and will be provided to the customer only upon request. Within the combined home IP system 10806, the IP network is bidirectional. Switchable video transmission is not limited by the size of the "pipe" to the home / business. These networks can provide more content and functionality. The network creates the potential to give customers more choices, including niche programming and high definition programming that interests a wide variety of viewers.

Compared to "traditional" cable or satellite TV, the combined system 10806 that provides IPTV is another improved configuration that enables more flexibility and creativity within the network. Combined system 10806 using IPTV enables two-way interaction over existing unidirectional cable or satellite broadcast networks. Two-way home IP networks give viewers more options to interact, personalize, and control the viewing experience. IP technology also provides more flexibility within the home network. With a combined system home IP network, all system receivers in the home or enterprise are connected to the same high-speed network. This allows game consoles, laptops and other devices to connect to the on-premises home IP network.

Watching IPTV in the coupled system 10806 is different from video streaming over the public Internet. IPTV allows network providers to run programs over their home IP networks, allowing them to control video quality and reliability of services. Highest quality internet video can be delayed due to low bandwidth, heavy traffic, or poor connection quality. IPTV allows TVs to communicate with other services so integrated high-speed Internet-based content and features can appear on TV screens. For example, online photos uploaded to a public or private cloud can be viewed directly on the TV.

Referring now to FIG. 109, a functional block diagram of the home IP coupled system 10806 of FIG. 108 is shown in more detail. Network content 10902 is provided from a service provider to a millimeter wave transmission system 10904. Network content 10902 may include video, audio, Internet web pages, or any other network based material. The millimeter wave system 10904 may operate at multiple wavelengths in accordance with the system described above in connection with the transmission of signals from the exterior of the building to the interior of the building and from the interior of the building to the exterior of the building. The millimeter wave system 10904 will include all of the various systems for transmitting in both directions between the interior and exterior of the building described above. The millimeter wave system 10904 transmits broadband data to the home IP system 10906 located inside the building. The millimeter wave system 10904 may be on either side of the glass or wall, which makes it possible to tunnel the radio wave through light or RF. The millimeter wave system 10904 is directly connected to the home gateway 10906 via electronic integration in the window unit. In an alternative embodiment, the millimeter wave system 10904 is wirelessly connected to the home gateway 10906 on a licensed band or unlicensed Wi-Fi with beamforming. The home IP system 1106 provides broadband content to a number of home devices 10908 located inside the building via wired connection 10110 and wireless connection 10112.

110 shows a functional block diagram of a home IP network system 11002. Input 11004 from the millimeter wave transmission system, which enables the transmission of millimeter waves from the external transmission unit into the structure, provides a wideband signal to the home IP network gateway 11006. The home IP network gateway 11006 determines where the signal from the input 11004 needs to be routed and provides one of the plurality of possible outputs to the appropriate destination IP address associated with the device requesting the broadband information. . The output line may include coaxial cable 11008, Ethernet cable 11010 or existing telephone line 11012. Coaxial cable 11008 may provide input to set top box 11014 and then provide output to living room TV 11016 via, for example, an HDMI connection 11018. The first Ethernet connection 11010 may be connected to the set top box / DVR 11020. Additional Ethernet connection 11022 provides data to second television 11024. Ethernet connections 11010 may also provide data to PC 11026 or network drive 11028. Existing telephone line connection 11012 will be provided at telephone outlet 11030 for telephone connection. Finally, Wi-Fi antenna 11032 provides the ability for home IP network gateway 11006 to provide a Wi-Fi network connection within the structure. Wi-Fi network connectivity allows devices such as PC 11034, laptops 11036, iPad 11038, or iPhone 11040 to wirelessly connect to home IP network gateway 11006 to receive broadband data. .

111 shows how a millimeter wave system can be used to send information to a home IP network system. An access unit 11102 located outside of the structure wirelessly transmits broadband IP to a CPE (customer premises equipment) unit 11104 located within one or more structures associated with the home IP network system. The access unit 11102 can receive broadband data for transmission to the CPE unit 11104 via wireless transmission or wired connection. The wireless access provided between the access unit 11102 and the CPE unit 11104 is arbitrary including millimeter bands 24 GHz, 28 GHz, 39 GHz, 60 GHz and 2.5 GHz, CBRS band 3.5 GHz, Wi-Fi bands 2.4 and 5 GHz. It may be provided in a plurality of frequency bands of but is not limited thereto. Such a signal is transmitted from outside the structure to the inside of the structure using any of the foregoing transmission techniques for transmitting the signal through a wall or window. Within the structure, the CPE unit 11104 includes an Internet of Things (IOT) device 11106, a PC 11108, an IP TV 11110, a closed circuit television 11112, an IP phone 11114 and a Wi-Fi extender 11116. Wi-Fi or other unlicensed bands are used on site to transmit signals. These are only some examples of IP-based devices and any type of Wi-Fi connectable device in the structure may be used for communication with the CPE 1104. The manner in which the broadband data can be transmitted from the outside of the structure to the inside of the structure may be configured by using the above-described millimeter wave transmission system in various ways.

112 shows a first embodiment in which the access unit 11102 wirelessly transmits broadband data to a millimeter wave system transceiver 11202 located outside the window or wall 11204. This system is installed in the consumer with a repeater (transceiver 11202) outside the building and a transceiver 11206 inside the building. This configuration can use an optical signal or an RF signal to tunnel the radio using a millimeter wave transmitter on either the glass or the wall. The broadband signal is directly connected to the CPE device 11104 via electronic integration in the unitary window unit to provide access to the home IP network 11208. Wireless transmission to the millimeter wave transceiver 11202 includes, but is not limited to, millimeter wave bands such as 24 GHz, 28 GHz, 39 GHz, 60 GHz, and 2.5 GHz; CBRS band, such as 3.5 GHz; And Wi-Fi bands such as 2.4 and 5 GHz. Millimeter wave transceiver 11202 transmits a signal through window or wall 11204 to a second millimeter wave transceiver 11206 located inside the structure. The configuration of the millimeter wave transceivers 11202 and 11206 may be any of those discussed herein in connection with a system for transmitting signals through the window or wall 11204. Internal millimeter wave transceiver 11206 outputs the received data to or receives data from customer premises equipment 11104 associated with the home network IP 11208. The millimeter wave transceiver 11206 and the CPE 11104 are integrated within the same box or device to receive signals from the millimeter wave transceiver 11202 located outside of the structure and provide data to the home IP network 11208 and related devices. It may include equipment.

FIG. 113 illustrates an alternative embodiment where the access unit 11102 wirelessly transmits a wideband data signal to an external millimeter wave transceiver 11202 as described above with respect to FIG. 112. In this embodiment, the millimeter wave transceiver is provided on the side of the window or wall 11204 that enables tunneling of the radio wave using an optical signal or an RF signal. The signal transmitted through window or wall 11204 is then wirelessly connected to CPE 11104 using beam shaping using unlicensed band or unlicensed Wi-Fi. External millimeter wave transceiver 11202 transmits data to internal millimeter wave transceiver 11206 through a window or wall 11204 as described herein. The internal millimeter wave transceiver 11206 includes a beam shaping device or Wi-Fi router that transmits the received signal to the integrated millimeter wave transceiver 12002 and the CPE 11104 using a beam shaping permit or Wi-Fi. The CPE 11104 sends data to the home IP network 11208 and related devices.

Referring now to FIG. 114, there is shown another embodiment of a system for transmitting broadband IP signals to a home IP network 11208, where the access unit 11102 sends signals to the outside side of a window 11404 of a building or structure. Wirelessly transmits to the millimeter wave transceiver 11402 located in the. The millimeter wave transceiver 11402 is located outside of the window glass and at a distance away from the window 11404 by radio frequency tunneling using licensed band or unlicensed Wi-Fi using a high power phased array and beam shaping circuit 11403. Enable wireless connection to the located CPE 11408. Millimeter wave transceiver 11402 wirelessly transmits signal through window 11404 to a millimeter wave transceiver 11406 positioned within the structure but not directly opposite the window 11404. A high power phased array 11403 providing a Wi-Fi router function. The millimeter wave transceiver 11406 is integrated with the CPE 11408, which transmits broadband data to the home IP network 11208 and related devices.

The described system provides an optical or RF tunnel that allows signals to be transmitted from outside the building to devices in the building. If broadband access is delivered to a building (home or commercial), other unlicensed bands may be used within the building. Optical or RF tunnels may be used to allow signals from IoT devices in a building to travel from interior to exterior. In addition to the techniques described herein, other near-field techniques may be used to transmit information through a window or wall.

(Millimeter wave using optical network)

One of the challenges faced by next-generation broadband access at gigabyte speeds is the need to install fiber in the home or business. Fixed millimeter-wave 5G wireless access technology allows the existing optical network unit (ONU), a passive optical network (PON) endpoint, to be used in a set of self-installed fixed wireless access points. 115 illustrates a combination of an optical data transmission system 11504 and a millimeter wave system 11150 such as GPON / NG PON2 / vOLTHA. This combined system enables control of bandwidth allocation from OLT to millimeter wave RU as will be described in more detail below. Each of these optical data transmission systems 11504 provides a way to transfer data from a central network location to a millimeter wave system 11150, which allows users, such as homes or businesses, to transfer data in RF format via the final drop (last 100 m). Allow transmission to the premises. Millimeter wave system 11150 may use millimeter wave beam shaping and beam steering techniques to ensure quality of service for home applications in response to dynamically changing network conditions. The millimeter wave system 11150 provides a connection to the home gateway 11506 (eg, the IP network gateway described above) to provide services to users located in homes or businesses. Millimeter wave system 11150 greatly increases the number of corporate and residential buildings where 5G millimeter waves can be used to provide Gigabit Ethernet. Accordingly, millimeter wave system 11150 uses TDMA and SDN controlled beam steering for wireless final drop access to the millimeter wave system in a structure sent to residential gateway 11506.

The residential gateway 11506 does not have the ability to dynamically trigger or coordinate data flow behavior between the millimeter wave system 11150 and the optical data transmission system 11504 based on network conditions, so that the hybrid as described below ONU and millimeter wave remote units can implement innovative SDN-enabled beam steering mechanisms to achieve a high quality experience with dynamic network slicing mechanisms and an optimized OLT-ONU signaling framework. Thus, as shown in more detail in FIG. 116, the optical network data flow 11602 in the GPON / NG PON2 / vOLTHA network 11504 and the data flow 11604 of the RF network 11150 are transmitted to the control system 11606. Balanced by, the RF network 11150 can provide sufficient resources to support the required optical network data flow 11602 and the optical network 11504 can provide sufficient resources to support the RF network data flow 11604. have. This configuration includes configuring network devices in optical network 11504 and RF network 1152. Accordingly, optical network 11504 and RF network 1152 are composed of components that can be remotely reconfigured by a central controller to enable balancing of loads transmitted by the network. If there are not enough resources in either of the networks, the network configuration 11608 changes to balance network data flow between the optical network 11504 and the RF network 11150 to avoid bottlenecks at the interface between the two networks. Can be. Such network reconfiguration is described in U.S. Patent Application "ULTRA-BROADBAND VIRTUALIZED TELECOM AND INTERNET," filed July 31, 2017, incorporated herein by reference (No. 15-664,764). Network configuration control techniques can be used.

Optical data transmission system 11504 (FIG. 15) includes a GPON, NG PON2, vOLTHA or similar type of system. Referring now to FIG. 117, two major active transmission equipment used for gigabyte passive optical network (GPON), optical line terminal 11702 (OLT) and ONU (optical network unit 11704 or ONT) (There is an optical network terminal 11706. The optical line terminal 11702 is in the central office 11708 and controls the information going in both directions through the optical distribution network.) The OLT 11702 is a CSM (control and switch module). ), The ELM (EPON Link Module, PON Card), and the redundancy protection function The optical network unit 11704 transmits the optical signal transmitted through the optical fiber 11710 from the OLT 11702 to the individual subscriber at the customer premises 1712. The ONU 11704 may transmit data coming from the subscriber to the OLT 11702. The optical network terminal 11706 is essentially the same as the ONU 11704. The ONT 11706 is ITU-T (ITU Telecommuni cation Standardization Sector and ONU 11704 are IEEE terms, both of which refer to user-side equipment in a GPON system, which supports high bandwidth, triple play services and distances up to 20 km.

Within the optical fiber for business configuration 1714, the OLT 11702 is connected to the ONU 11704 via the optical fiber 11710. The ONU 11704 is connected to the PON endpoint 1716 via copper wire 1718. Within the optical fiber for cabinet configuration 1720, OLT 11702 is connected to ONU 11704 via optical fiber 1710. The ONU 11704 is connected to the PON endpoint 1716 via copper wire 1718. Finally, with a groove connection 1722 in the optical fiber, the OLT 11702 is connected with the ONT 11706 through the optical fiber 1710.

Referring now to FIG. 118, a single optical fiber 11710 from the OLT 11702 is coupled to the passive optical splitter 11802. Splitter 11802 splits power into a separate path 11804 to user 11806. There may be from 2 to 128 user paths 11804 at any location. GPON has two multiplexing mechanisms. In the downstream direction (OLT to user), data packets are encrypted and broadcast to the user. In the upstream direction (user to OLT), data packets are sent using TDMA.

The ONU-ID is an 8-bit identifier that the OLO 11702 assigns to the ONU 11704 via a PLOAM message during ONU activation. The ONU-ID is unique at the PON and is maintained until the ONU 11704 is powered off or deactivated by the OLT 11702. OLT 11702 also assigns 12-bit allocation identifier (ALLOC_ID) to ONU 11704 to identify traffic bearing entities that are recipients of upstream bandwidth within that ONU.

The transport container (T-CONT) is a logical connection group that appears as a single entity for upstream bandwidth allocation for the ONU 11704. The number of T-CONTs supported for the designated ONU 11704 is fixed. ONU 11704 autonomously creates all T-CONT instances supported during ONU activation and OLT 11702 detects the number of T-CONT instances supported by the designated ONU. There are five types of T-CONT. Type 1 is fixed bandwidth and is used for delay and priority sensitive services. Types 2 and 3 are guaranteed bandwidth types and are mainly used for video services and high priority data services. Type 4 is best and is primarily used for data services such as the Internet and low priority. Type 5 is a mixed type that includes all bandwidth types.

The ONU 11704 may be located at various distances from the OLT 11702, which means that the transmission delay of each ONU is unique. Ranging is performed by the OLT 11702 to measure the delay and set a register in each ONU 11704 to equalize the delay. OLT 11702 sends an acknowledgment to each ONU 11704 to set a predetermined time interval for transmission. Grant maps are dynamically recalculated every few milliseconds and are used to allocate bandwidth to all ONUs for such requests.

Dynamic bandwidth allocation (DBA) allows you to quickly adopt your bandwidth allocation based on your current traffic requirements. GPON uses TDMA to manage upstream access by ONU 11704, which provides an unshared time slot for each ONU for upstream transmission. Using a DBA allows upstream time slots to shrink and grow depending on the distribution of upstream traffic loads. If the OLT 11702 does not support DBA, upstream bandwidth is statically allocated to the T-CONT and cannot be shared and can only be changed through the management system.

There are two types of DBAs: status reporting DBAs (SR-DBA) and stateless reporting DBAs (NSR-DBA). In SR-DBA, OLT 11702 requests the T-CONT buffer status and ONU 11704 responds with a separate report for each T-CONT. After receiving the report, the OLT 11702 recalculates the bandwidth allocation and sends a new map to the ONU 11704. ONU 11704 sends data in the designated time slot. The ONU 11704 sends idle cell upstream to the OLT 11702 to inform the ONU that there is no information to send. The OLT 11702 may then assign a grant to another T-CONT.

In the NSR-DBA, the OLT 11702 continuously allocates a small amount of additional bandwidth to each ONU 11704. OLT 11702 increases bandwidth allocation for the ONU when it observes that ONU 11704 does not transmit idle frames. If ONU 11704 transmits an idle frame, the OLT reduces the allocation accordingly. The NSR-DBA has the advantage that the ONU 11704 does not need to know the DBA, but has the disadvantage that the OLT 11702 does not know how to allocate bandwidth to several ONUs 11704 in the most efficient manner.

Referring now to FIG. 119, an upstream GTS frame 11902 and a downstream GTS frame 11904 are shown. 120 shows a more detailed view of the downstream GTC frame 11904. The downstream GTC frame 11904 is 38880 bytes long with a duration 11906 of 125 us, which corresponds to the downstream data rate of 2.48832 Gbps. OLT 11702 broadcasts PCBd (GTC header) 11908 to all ONUs 11704 and the ONUs operate according to the relevant information. PCBd includes a 'Psync' field 12002 that indicates the start of a frame for ONU 11704. The 'Ident' field 12004 includes an 8-KHz super frame counter field. The 'PLOAMd' field 12006 handles functions such as OAM related alarms or threshold crossing alerts. The 'BIP' field 12008 is bit interleaved parity used to estimate the bit error rate. Downstream payload length indicator (Plend) 12010 provides the length of the upstream bandwidth (US BW) map. Each entry in the upstream bandwidth (US BW) map field 12012 represents a single bandwidth allocation for a particular T-CONT.

The allocation ID field 12014 indicates the receiver of the bandwidth allocation and directly processes the ONU 11704 using the lowest 254 allocation ID value. The 'Flag' field 12016 allows upstream transmission of the physical layer overhead block for the designated ONU 11704. 'Slot Start' field 12018 and 'Slot Stop' field 12020 indicate the start and end of an upstream transmission window. CRC field 125022 provides error detection and correction for the bandwidth allocation field.

The GTC payload field 12024 includes a series of GPON Encapsulation Method (GEM) frames 12026. The downstream GEM frame stream is filtered at ONU 11704. Each ONU 11704 is configured to recognize which port ID belongs to, and the port ID uniquely identifies the GEM frame 12026.

Reference is again made to FIG. 119. The upstream GTS frame period 11914 is 125us and is 19440 bytes long, which provides an upstream data rate of 1.24416 Gbps. Each frame 1119 includes a number of ONU transmit bursts 1912 from ONU 11704 and each burst includes a physical layer overhead (PLOu) section 1914 and one or more bandwidth allocation intervals 1192. do. The BW map indicates the arrangement of bursts in the frame and the allocation interval in each burst.

Referring now to FIG. 121, an upstream GTS frame 11910 is shown in more detail. The 'PLOu' burst 12102 includes a preamble that ensures proper physical layer operation. The 'PLOu' field 12102 also includes an ONU-ID field 12104 indicating the unique ONU-ID of the ONU 11704. The upstream physical layer OAM (PLOAMu) field 12106 is responsible for management functions such as scope, activation of ONT 11706 and alert notification. Upstream power leveling sequence (PLSu) field 12108 includes information about the laser power level in ONU 11704 as seen by OLT 11702.

GEM frame 12026 is sent from OLT 11702 to ONU 11704 using GTC frame payload section 12112. OLT 11702 may allocate up to all downstream frames to meet its downstream needs. ONU filters incoming frames based on Port-ID. Frames are sent from ONU 11704 to OLT 11702 using the configured GEM allocation time. ONU 11704 buffers and sends frames in bursts when time is allocated by OLT 11702. OLT 11702 multiplexes the frames received from ONU 11704.

Another system that may be used for optical data transmission system 11504 (FIG. 115) is a Next-Generation Passive Optical Network (NG-PON2) system. NG-PON2 is a 40Gbps capable multi-wavelength PON system that can scale up to 80Gbps. In the NG-PON2 system, there are three types of channel speeds: 10 / 2.5Gbps base rate or optionally 10 / 10Gbps and 2.5 / 2.5Gbps.

NG-PON2's main target requirements include increased total capacity per Optical Line Terminal (OLT) PON port, sustainable bandwidth of all Optical Network Units (ONU) at 1 Gbps downstream and 0.5-1 Gbps downstream, support for more than 64 ONUs per port, Compatibility with legacy PON infrastructure, 40 km differential distance and smooth movement, support for multiple applications on the same optical distribution network (ODN), built-in test and diagnostics, and PON resiliency.

There are many applications that drive demand for the next generation of PONs, including FTTB, enterprise, mobile backhaul, fronthaul, and cloud-RAN. Content is currently a major driver of high access bit rate requirements. Content service providers must prepare for future access networks, and we can conclude that future access networks are truly multi-service solutions.

Currently, software packages and personal data are downloaded and stored in the data center. This requires very high upload and download speeds as well as symmetry and low latency. This means that the "cloud opportunity" from NG-PON2 is also a very important reason for evolving into a new network.

NG-PON2 is compatible with legacy loss budget classes. The NG-PON2 requires at least 14dB optical path loss and allows for 40km differential reach. There are three classes defined by NG-PON2 of Tx / Rx wavelength channel tuning time. Class 1 components may include a switch on laser array, class 2 components may be based on an electronically tuned laser (DBR), and class 3 components may be thermally tuned DFBs.

The NG-PON2 transport convergence layer provides new features supported by the protocol, such as multiple wavelengths, TWDM and point-to-point channels. Communication starts with a single channel and later adds more channels and distributed OLT channel terminations (CT), leading to a single fiber. The new protocol feature allows for multiple wavelengths, so the protocol handles ONUs with new identities for tuning, identifying systems and wavelength channels, new management protocols for PtP, WDM and TWDM activation, and uncorrected lasers that should not be sent to the wrong wavelength channel. It also supports new rogue scenarios that enable cross-channel messaging, detection, and mitigation for some procedures through distributed OLT channel termination.

NG-PON2 has an interchannel termination protocol. OLT CTs (Channel Termination) are decentralized, so some procedures are OLT CT quiet window synchronization, ONU tuning, ONU activation, isolated ONU parking, ONU connected to wrong ODN, guided hand-off of ONU between OLT CTs, and bad Message passing between OLT CTs, such as ONU isolation, is required.

The NG-PON2 also supports ONUs such as when the ONU transmitter hops to the wrong upstream channel, when the ONU transmitter begins to transmit on the wrong upstream wavelength, or when the OLT CT transmits on the wrong downstream wavelength channel, and when interfering from a coexisting device. It covers various protection scenarios and bad behaviors.

Currently, NG-PON2 OLT optics are based on Bidirectional Optical Subassemblies (BOSA) integrated into the XFP form factor. These optics are suitable for TWDM PON, 10Gbps downstream, 2.5Gbps or 10Gbps upstream. XFP integrates an electro-absorption integrated laser diode and a semiconductor optical amplifier (SOA) to reach Type A N1 class NG-PON2 optical requirements. The receiver components, the high-sensitivity burst mode avalanche photodiode (APD), preamplifier and limiting amplifier, are packaged in a single-mode fiber-stub integrated package with sensitivity equivalent to -28.5 dBm at 10 Gbps and -32 dBm at 2.5 Gbps. Is mounted.

NG-PON2 ONU optical element is based on the Bi-directional Optical Sub Assembly (BOSA). BOSA emits high optical power on a +1 to 9dBm N1 type A link capable of performing four upstream channels with a burst mode adjustable distributed feedback laser (DFB) of 10 Gbps or 2.5 Gbps. On the receiver side, high-sensitivity four-channel tunable APDs, preamps and limiting amplifiers can operate with a sensitivity of -28dBm at 10Gbps.

Another implementation that can be used for optical data transmission system 11504 (FIG. 115) is a virtual optical line termination hardware abstraction (vOLTHA) that can be used within one of the systems described above. As shown in FIG. 122, vOLTHA is an abstraction layer on legacy and next generation network equipment. This is the first time for PON (G-PON, E-PON, XGS-PON) and ultimately for Fast NG-PON2 DOCSIS and Ethernet. vOLTHA is an access network that acts as an abstract programmable switch and works with legacy and virtualized devices. vOLTHA can be run on a device or a general purpose server.

Each access technology brings its own protocols and concepts, which means that control and management of legacy access devices can be problematic. vOLTHA confines differences in access technologies to access regions and hides them from the upper layers of the OSS stack. Referring now to FIG. 123, an implementation of vOLTHA with an OLT 12302 and ONU 12304 links is shown. OLT 12302 communicates with multiple ONUs 12304 via splitter 12305. vOLTHA containers communicate via gRPC. The base container publishes events to Kafka and maintains data in Consul for service discovery. Southbound OLT adapter 12306 and ONU adapter 12308 will also be their containers. OLT adapter 12306 and ONU adapter 12308 enable OLT-ONU interoperability through vOLTHA core 12310. The ONU adapter 12308 transmits the ONT Management Control Interface (OMCI) to the OLT 12302 via the OLT adapter 12306.

By using vOLTHA to create a hardware abstraction layer for wave agility, you can integrate it into your home network IP gateway via millimeter-wave fixed wireless access (gigabit access with dynamic QoS-application and network slicing support). . One of the challenges facing next-generation broadband access at gigabit speeds is the need to install fiber in the home or business. Referring to FIG. 124, a fixed millimeter wave 5G radio access technology may be used to integrate self-installed fixed wireless access points using ONU 12402 (PON endpoint).

Almost all recent Fiber to the home (FTTH) deployments, as well as what is currently planned, use passive optical networking. The concept of passive optical network (PON) 12212 involves the use of a passive fiber splitter that allows multiple customers (typically 32-128 people) to share a single fiber pair. GPON has also seen trials and initial deployments of several large telecommunications companies, but is primarily used as the basis for transmitting Ethernet via encapsulation in the GEM frame (GPON encapsulation method) 12026 (FIG. 120). GPON is designed with very strict timing requirements. Thus, both EPON and GPON use TDMA (Time Division Multiple Access), which is also informally referred to as "time sharing". The time is divided into slots that are fixed or variable in length or long enough to contain one or more data frames (typically about 100-1000 msec). During a given slot, only one ONU 12402 is allowed to transmit, and everything else must turn off their laser. The OLT 12410 is responsible for determining the transmission schedule and sending it to the ONU 12402 (which is considered a form of batch polling by the OLT), which is responsible for sending the correct clock synchronized with the OLT to transmit at the correct time. It must be maintained.

The number of time slots assigned to each ONU 12402 need not be fixed. Both EPON and GPON provide a flexible mechanism for the OLT 12410 to dynamically allocate bandwidth to the ONU based on demand and network operator policy. This mechanism is particularly nonspecific for the algorithm used in the case of EPON, where a very simple request-based protocol is outside the scope for interesting dynamic bandwidth allocation algorithms. It is desirable to extend the bandwidth allocation with millimeter wave technology using PON technology, the channel is broadcast to all ONU 12402, and each frame is labeled with the address of the target ONU. The ONU 12402 passes the frame through the home gateway 12406 to the end user's LAN and all other ONUs drop the frame. This is a form of TDMA where the OLT 12510 determines its transmission schedule and each time slot sustains the duration of the frame.

Millimeter wave system 12004 may also use millimeter wave beamforming and beam steering techniques to ensure QoS for home applications accessed through home gateway 12406 in dynamically changing network conditions. Hybrid ONU 12404 and millimeter wave remote unit (RU), although given current home gateway (RGW) devices 12406 do not have the ability to trigger and adjust service flow operations directly and dynamically based on network conditions. Remote Unit (12408) is designed with an innovative SDN-enabled beam steering mechanism to achieve a high quality user experience through a dynamic network slicing mechanism and an optimized OLT-ONU (gPON) signal framework. The millimeter wave frequencies implemented by millimeter wave system 12404 are defined in bands of approximately 24, 28, 39, and 60 GHz. However, this approach can also be applied to 3.5GHz CBRS. Millimeter wave system 12404 offers a lot of potential for use as a wireless broadband service with beam steering under the control of the SDN towards a self-installed millimeter wave home modem. As mentioned above, the SDN beam steering mechanism and dynamic network slicing mechanism may use the techniques described in US patent application Ser. No. 15 / 664,764, which is incorporated herein by reference.

In the vOLTHA scenario, home gateway 12406 uses ONU (millimeter wave technology) within millimeter wave system 12404 at the last drop (hundreds of meters) where a wireless access point is directly connected to ONU 12402 via millimeter wave RU 12408. 12402). Hybrid virtual OLT (vOLTHA) 12410 and millimeter wave fixed broadband wireless technology via millimeter wave system 12404 can provide home and business self-installation access. In addition, the synchronization characteristics of gOLON-based vOLTHA can be extended to beam steering control technology itself for mapping / distributing ONU traffic between multiple millimeter wave modems 12408 supporting slicing control in home networks. In this scenario, a single PON 12212 is considered a collection of point-to-point links, one per hybrid ONU 12402 + millimeter wave wireless device 12408 in an Ethernet switch. PON 12212 typically connects up to 128 ONUs 12402 to each OLT 12210 and hybrid ONU-RUs to multiple millimeter wave modems using beam steering control schemes. The millimeter wave modem 12408 is self-installing and not only reduces the need for fiber connections to the home / apartment, but also provides additional statistical gains and aggregation points at the ONU + RU in the Ethernet layer, providing customers with this PON 12212 It will be on a single large Ethernet. In addition, if latency and cost are not critical factors, ONU + RU can be treated as an integrated, IP router with load balancing and slicing, providing statistical gain and aggregation points.

Thus, from an operator's point of view, by bridging all headquarters' PONs 12212 and providing ONU + RUs 12402/12408 at the Ethernet layer, customers served by these PONs 12212 are single large. It will be on Ethernet. In addition, if latency and cost are not critical factors, it can threaten ONU + RU 12402/12408 as an IP router with load balancing and additional slicing. The system can also be designed where transmission is performed on an externally high internal 60 GHz band channel and internally externally on a low 60 GHz band channel.

vOLTHA's current ONU 12402 is complemented by a millimeter wave RU 12408 to perform beam steering via a modem installed in each home. In a practical scenario, a small cell placed with each ONU 12402 in an urban outdoor environment is regularly affected by trees and passing objects. In millimeter wave beam forming systems, environmental problems such as wind movement and tree blockage can be solved by beam steering techniques under the control of SDN using beam patterns with very narrow wavelengths. The practical obstacles to lamp post placement scenarios must be integrated into the beam forming system and system design.

Nearly all modern PONs 12512 are either used as the base protocol in EPON or encapsulated in GPON as GEM, and the physical and logical topology of a simple Ethernet PON deployment runs on Ethernet at some level: Ethernet is currently used as the basis for the data link layer and Internet Protocol (IP) is used as the ubiquitous network layer protocol. Some networks still use separate fibers to transmit in each direction (1310nm and 1490nm for bidirectional use). The Ethernet PHY provides serialized bit stream functionality (dedicated) at the Medium Access Control (MAC) layer. MAC divides the bit stream into frames. The frame is labeled with a header containing the source and destination MAC addresses. This allows statistical multiplexing of frames from multiple hosts on a single link.

125 illustrates an interface between ONU 12402 and a plurality of home gateways 12406. Single fiber pair 12502 provides data to and receives data from ONU 12402. The ONU 12402 interfaces with the millimeter wave remote unit 12408 and has the ability to generate an RF beam 12504 that can be directed to one or more millimeter wave radio units 12408B associated with a home or business. The interface between the millimeter wave remote units 12408A and 12408B may include one or more building penetration techniques described herein. The millimeter wave radio unit 12408 provides beam steering techniques and slice control techniques that enable bidirectional data transmission control between the ONU 12402 and the home gateway 12406. Millimeter wave remote unit 12408B associated with a home or business interface with home gateway 12406 provides a broadband data connection to the associated home or business structure.

Referring now to FIGS. 126 and 127, there is a more specifically illustrated embodiment for broadband data communication between OLT 12410 and a device located within a structure. In connection with FIG. 126, the OLT 12410 is located in the central office / MEC 12602, which may be part of a Virtual Base Band Unit (VBBU). OLT 12410 reserves transmission to ONU 12402 over optical fiber 12604. OLT 12410 is connected to multiple ONUs 12402 through fiber pair 12604. ONU 12402 maintains an accurate clock for synchronization with OLT 12410. Remote unit 12408 is coupled to ONU 12402. The remote unit 12408 is part of the millimeter wave system 12404 described above. The combined ONU / RV is treated as an IP router that provides load-balancing and slicing and provides statistical gain for signal transmission and serves as an aggregation point for received data. Combined ONV / RV also provides wireless communication with remote devices associated with the structure. The remote unit 12408 is located in a light pole or tower located near the structure and provides a wireless final drop connection that replaces fiber optics, DSL and cables in a home or business.

The remote unit 12408 uses a controlled beamforming and slice control technique to generate a wireless beam 12606 that is transmitted to an external millimeter wave transceiver 12608 located outside of the structure. External millimeter wave transceiver 12608 repeats the signal received from the external hub and allows the signal to pass through glass or a building. External millimeter wave transceiver 12608 transmits a wideband data signal through window or wall 12610 and internal millimeter wave transceiver 12612 using one of the techniques described above for transmitting through a wall or window. Internal millimeter wave transceiver 12612 also employs beam forming and slicing techniques as described herein to transmit wireless beams 12614 to residential gateway 12616 within the structure. The home gateway 12616 is a self-installed home modem that provides interconnection between devices located within the structure that communicates with the home gateway 12616 over a wired or wireless connection with broadband data received from an internal millimeter wave transceiver 12612. It includes. OLT 12410, ONU 12402, RU 12408, millimeter-wave transceivers 12608/12612, and home gateway 12616 all control SDN access to the last drop connection by controlling the entire end-to-end configuration of the component. It includes the hardware abstraction layer from vOLTHA described above that enables it.

FIG. 127 shows the same structure described with respect to FIG. 126 for wideband data transmission between OLT 12410 and internal millimeter wave transceiver 12126. Rather than show a connection to the home gateway 12616 that the system can still perform, a 60 GHz wireless connection to a pair of virtual reality (VR) goggles 12702 is shown. The 60 GHz transceiver dongle 12704, which will be described in more detail below, is inserted into the USB port of the internal millimeter wave transceiver 12612. This provides the ability for the internal millimeter wave transceiver 12612 to bidirectionally communicate with the VR goggles 12702 located inside the structure via the 60 GHz transceiver dongle 12704. The VR goggles 12702 can be used wirelessly with any internal computer or central office without a local computer. 127 shows a 60 GHz wireless link to the VR goggles 12702, but other types of devices may also be wirelessly connected to the 60 GHz transceiver dongle 12704 to enable wideband data transmission. 127 shows a 60 GHz wireless link to the VR goggles 12702, but other types of devices may also be wirelessly connected to the 60 GHz transceiver dongle 12704 to enable wideband data transmission. As described above, data transmission control between the optical data transmission unit and the RF data transmission unit using the SDN slicing is applicable to the embodiments of FIGS. 126 and 127. OLT 12410, ONU 12402, RU 12408, millimeter-wave transceivers 12608/12612, and VR goggles 12702 all allow the SDN to control the entire end-to-end configuration of the component to access the last drop connection. It includes the hardware abstraction layer of vOLTHA, as described earlier.

Referring now to FIG. 128, a functional block diagram of a 60 GHz transceiver dongle 12704 is shown. The 60 GHz transceiver dongle 12704 implements a 60 GHz chipset using, for example, the Peraso transceiver described above with respect to FIG. 84B. The chipset brings low-cost, low-power, high-performance SuperSpeed USB 3.0 to WiGig devices. The chipset includes a USB 2.0 and 3.0 device / host system interface 11280. The interface 1802 supports link speeds of up to 2.0 Gbps at 10 m and 1 Gbps at 20 m, and the control interface 1804 can configure the chipset with multiple gigabit Wig USB adapters or 60 GHz wireless connections of peripheral devices.

The 60 GHz transceiver dongle 12704 integrates two processors 12806 to provide maximum flexibility to support 801.11ad MAC functionality. CPU code boot loading is supported at USB interface 12802 or external serial flash 12808. The MAC includes internal memory 12810 sufficient to receive / transmit packets to the host interface as well as buffer data transmissions with the PHY. No additional RAM is needed.

The physical layer can modulate / demodulate all control and single carrier π / 2 □ BPSK, π / 2 □ QPSK, and 16 □ QAM WiGig coding schemes up to 4.62Gbps. Low Density Parity Check (LDPC) forward error correction maximizes the performance of unreliable or noisy communication channels. The baseband includes a highly configurable programmable IO subsystem consisting of GPIO, UART, SPI, TWI, PWM and JTAG. The firmware integrates multiple layers of debug capabilities such as logging and extensive statistics and event counters.

The 60 GHz transceiver dongle (12704) provides mobile multi-gigabyte wireless connectivity, high quality, low latency wireless UHD 4k displays, wireless docking stations, I / oh and mobile "sync and move", small cell backhaul and Wi-Fi infrastructure, and other multi-gigabit It can be used for many different applications, including byte links. The system can be deployed to transmit on the higher band channel from all center frequencies (3.5, 24, 28, 39, 60 GHz) from the outside to the inside and from the lower band channel to the outside.

As shown in FIG. 129, every Ethernet interface is assigned a unique 6 byte MAC address 12902 at the time of manufacture to represent a locally managed address. This MAC address 12902 contains three bytes 12904 that identify the manufacturer of the device using an Organizationally Unique Identifier (OUI) assigned by the IEEE Institute, with the remainder being assigned by the manufacturer. Some local schemes may override MAC addresses assigned by the manufacturer. One bit 12906 of the first byte serves as a flag indicating such a locally managed address. This bit 12906 is set to zero in all manufacturer assigned addresses. This allows the ONU 12410 to map to millimeter wave radio beams and maintain a table that reproduces the glue logic between the fixed radio and the OLT / ONU allocation slots.

Referring now to FIG. 130, switch 13002 in the optical network (which may be OLT, ONV, or ONT) uses a MAC address to bridge multiple point-to-point or shared-middle Ethernet segments 1308 together. do. When the frame passes through switch 13002, the switch knows the sender's location. The source address of the frame is stored in the forwarding database 13004 of the switch memory along with the interface on which the frame arrived. This is used to indicate subsequent frames. The switch 13002 finds the destination address of the frame in the database 13004 and determines the interface to which the frame is to be delivered. If the switch 13002 does not have a record of the location of the address, the frame can be flooded to all interfaces. This wastes link capacity very much, and the intention is to prevent it.

The MAC address may represent different host groups using different flag bits 13008. Currently, Ethernet does not provide built-in multicast routing by default, which broadcasts to all group addresses, but some switches (13002) connect to IP multicast using a technology called Internet Group Management Protocol (IGMP) snooping, and Ethernet Infer multicast groups.

In summary, the goal is to use 5G fixed wireless millimeter wave and 5G cores with slicing, transmit over vOLTHA and provide home-like modems with speeds similar to Gig power fiber service (eg 1Gbps). This enables balancing of data flow between the optical and RF networks associated with PON 12212 using millimeter wave system 12404 in one example. Assume that the home street column is equipped with ONU 12402 and millimeter wave remote unit 12408.

One skilled in the art having the benefit of the present disclosure will understand that the regeneration and retransmission of millimeter waves for building penetration provides a way to provide millimeter wave signals inside buildings where signals do not penetrate effectively. It is to be understood that the drawings and detailed description herein are to be considered as illustrative and not restrictive, and are not intended to be limited to the particular forms and examples disclosed. On the contrary, as defined by the following claims, any additional modifications, changes, rearrangements, substitutions, alternatives, design choices, and embodiments apparent to those skilled in the art without departing from the spirit and scope of the present invention are included. Accordingly, the following claims are intended to be construed to include all such further modifications, changes, rearrangements, substitutions, alternatives, design choices and examples.

Claims (30)

A system that allows signal penetration into a building,
Located outside the building and receiving a signal of a first frequency that suffers a loss upon penetration into the building and overcoming the loss caused by penetration of the received signal of the first frequency into the building through a wireless communication link. A first circuit for converting into one format, the first circuit comprising:
Implementing a first transmission chipset for RF transmission of the first format corresponding to losses incurred upon penetration into a building, receiving the signal of the first frequency and receiving the received signal of the first frequency into the building The first circuit further comprising a first transceiver for converting to the first format that overcomes the loss caused by infiltration; And
A second circuit located inside a building and communicatively linked with the first circuit, the second circuit for receiving and transmitting the converted received signal of the first format, the second circuit comprising:
And a second transceiver for implementing the first transmission chipset and receiving a converted signal of the first format from the first transceiver outside the building and transmitting to the first transceiver. A system for enabling signal penetration into a building, comprising. The Wi-Fi communication device of claim 1, further comprising: a Wi-Fi coupled to the second transceiver for conversion between the converted signal of the first format and the Wi-Fi version signal and for transmitting and receiving the Wi-Fi version signal inside a building. And further comprising a Fi router. 2. The signal penetration of a building according to claim 1, wherein the first frequency is selected from the group consisting of 3.5 GHz, 24 GHz, 28 GHz, 39 GHz, 60 GHz, 71 GHz, and 81 GHz. Letting system. 10. The system of claim 1, wherein the received signal of the first frequency is transmitted using a first protocol. 5. The system of claim 4, wherein the first protocol is selected from the group consisting of 2G, 3G, 4G-LTE, 5G, 5G New Radio (NR) and WiGi. . 2. The apparatus of claim 1, wherein the first transceiver implements the first transmission chipset, transmits the converted signal of the first format to the first transceiver and receives from the first transceiver, and And a third transceiver for transmitting the converted signal to the second transceiver in the building and receiving from the second transceiver. 2. The system of claim 1, wherein said first transmit chipset comprises a Ferraso chipset. 2. The building of claim 1, further comprising at least one solar cell plate connected to the first circuit, wherein the at least one solar cell plate provides power for operating the first circuit. A system that enables signal penetration of the signal. 10. The system of claim 1, further comprising a laser power system for providing power to the first circuit, the laser power system comprising:
A laser for generating a beam for transmitting light energy from inside the building to outside the building; And
And a photovoltaic receiver for generating electrical energy for receiving light energy from a laser beam and for powering said first circuit. 2. The system of claim 1, further comprising an inductive coupling system for providing power to the first circuit, wherein the inductive coupling system:
A first induction coil located inside the building and connected to the power system located inside the building; And
And a second induction coil located outside the building and inductively coupled with the first induction coil, wherein the second induction coil is coupled to provide electrical energy to the first circuit. System to enable this. 2. The magnetic resonance coupling system of claim 1, further comprising a magnetic resonance coupling system for providing power to the first circuit.
A first magnetic resonance coil located inside the building and connected to the power system located inside the building; And
A second magnetic resonance coil located outside the building and magnetically coupled with the first magnetic resonance coil, wherein the second magnetic resonance coil is connected to provide electrical energy to the first circuit. A system that enables signal penetration of the signal. A system that allows signal penetration into a building,
Located outside the building, receiving millimeter wave signals that suffer from intrusion into the building, and converting the received millimeter wave signals into a first format that overcomes the losses caused by infiltrating the inside of the building onto the wireless communication link. As a first circuit, the first circuit is:
Implement the Ferraso chipset for RF transmission of the first format corresponding to the loss incurred during penetration into the building, receive the millimeter wave signal, and penetrate the received millimeter wave signal into the building. The first circuit further comprising a first transceiver for converting to the first format to overcome; And
Located inside a building and communicatively linked to the first circuit, receiving a converted millimeter wave signal of the first format from the first transceiver and transmitting to the first transceiver, and transmitting the converted millimeter wave signal. A second circuit for converting to a second format for transmission to a wireless device in a building, wherein the second circuit is:
And the second device implementing the Ferraso chipset and including a second transceiver for transmitting the converted millimeter wave signal of the first format to the first transceiver outside the building and receiving from the first transceiver. A system that enables signal penetration into a building, characterized in that. 13. The Wi-Fi device of claim 12, further comprising: a Wi-Fi coupled to a third transceiver for conversion between the converted signal of the first frequency band and the Wi-Fi version of the signal and for transmitting and receiving the Wi-Fi version of the signal inside a building. And a router further comprising a router. 13. The system of claim 12, wherein the received signal at the first frequency is transmitted using a first protocol. 15. The system of claim 14, wherein the first protocol is selected from the group consisting of 2G, 3G, 4G-LTE, 5G, 5G New Radio (NR) and WiGi. . 13. The apparatus of claim 12, wherein the first transceiver implements a first transmit chipset, transmits the converted signal of the first format to the first transceiver, receives from the first transceiver, and converts the first format. And a third transceiver for transmitting the received signal into the building and receiving from the inside of the building. 13. The building of claim 12, further comprising at least one solar cell plate coupled to the first circuit, wherein the at least one solar cell plate provides power for operating the first circuit. A system that enables signal penetration of the signal. 13. The system of claim 12, further comprising a laser power system for providing power to the first circuit, the laser power system comprising:
A laser for generating beams for transmitting light energy from inside the building to the outside of the building; And
And a photovoltaic receiver for receiving optical energy from a laser beam and generating electrical energy for powering the first circuit. 13. The system of claim 12, further comprising an inductive coupling system for providing power to the first circuit, the inductive coupling system:
A first induction coil located inside the building and connected to the power system located inside the building; And
And a second induction coil located outside the building and inductively coupled with the first induction coil, the second induction coil being coupled to provide electrical energy to the first circuit. System to enable this. 13. The magnetic resonance coupling system of claim 12, further comprising a magnetic resonance coupling system for providing power to the first circuit.
A first magnetic resonance coil located inside the building and connected to the power system located inside the building; And
And a second magnetic resonance coil located outside the building and magnetically coupled with the first magnetic resonance coil, wherein the second magnetic resonance coil is connected to provide electrical energy to the first circuit. System to enable signal penetration into the furnace. A system that allows signal penetration into a building,
A first location located outside the building and receiving a millimeter wave signal that suffers a loss upon penetration into the building and converting the received millimeter wave signal into a first format that overcomes the losses caused by penetrating the inside of the building onto the wireless communication link. Wherein the first circuit is:
Implementing the Ferraso chipset for RF transmission of the first format corresponding to losses incurred during penetration into a building, and receiving losses caused by receiving millimeter wave signals and penetrating the received millimeter wave signals into the building. A first transceiver for converting to the first format to overcome; And
Implements the first transmission chipset, transmits the converted signal of the first format to the first transceiver and receives from the first transceiver, transmits the converted signal of the first format to a building interior and from inside the building A second transceiver for receiving;
The first circuit comprising a; And
Located inside a building, communicatively linked with the first circuit, receiving a converted millimeter wave signal of the first format and converting the converted millimeter wave signal to a WiFi format for transmission to a wireless device in the building A second circuit, wherein the second circuit is:
A third transceiver that implements the Ferraso chipset and receives the converted millimeter wave signal of the first format from the second transceiver outside the building and transmits to the second transceiver; And
For conversion between the converted millimeter wave signal of the first format received from the third transceiver and transmitted to the third transceiver and the WiFi format for reception by a wireless device inside a building, and the Wi- A Wi-Fi router coupled to the third transceiver for transmitting and receiving Fi format signals;
And said second circuit comprising: a system for enabling signal penetration into a building. 22. The system of claim 21, wherein the received signal at the first frequency is transmitted using a first protocol. 23. The system of claim 22, wherein the first protocol is selected from the group consisting of 2G, 3G, 4G-LTE, 5G, 5G New Radio (NR) and WiGi. . 22. The building of claim 21, further comprising at least one solar cell plate connected to the first circuit, the at least one solar cell plate providing power for operating the first circuit. A system that enables signal penetration of the signal. 22. The system of claim 21, further comprising a laser power system providing power to the first circuit, the laser power system:
A laser for generating a beam for transmitting light energy from inside the building to outside the building; And
And a photovoltaic receiver for generating electrical energy to receive optical energy from a laser beam and to power the first circuit. 22. The system of claim 21, further comprising an inductive coupling system for providing power to the first circuit, the inductive coupling system:
A first induction coil located inside the building and connected to the power system located inside the building; And
And a second induction coil located outside the building and inductively coupled with the first induction coil, the second induction coil being coupled to provide electrical energy to the first circuit. System to enable this. 22. The magnetic resonance coupling system of claim 21, further comprising a magnetic resonance coupling system for providing power to the first circuit.
A first magnetic resonance coil located inside the building and connected to the power system located inside the building; And
And a second magnetic resonance coil located outside the building and magnetically coupled with the first magnetic resonance coil, wherein the second magnetic resonance coil is connected to provide electrical energy to the first circuit. A system that enables signal penetration of the signal. A system that allows signal penetration into a building,
Implement a first transmit chipset for RF transmission in a first format that is located outside of a building and that copes with losses incurred during penetration into a building, receives signals of a first frequency that suffers loss upon penetration into a building A first transceiver for converting the received signal of the first frequency into a first format that overcomes the loss caused by penetration of the received signal into a building over a wireless communication link;
Located outside the building, implementing the first transmission chipset, transmitting the converted signal of the first format to the first transceiver and receiving from the first transceiver, and receiving the converted signal of the first format inside the building A second transceiver for transmitting to and receiving from inside a building;
A third transceiver located inside a building, implementing the first transmission chipset, and receiving the converted signal of the first format from the second transceiver outside the building and transmitting to the second transceiver; And
A Wi-Fi router coupled to the third transceiver for conversion between the converted signal of the first format and the Wi-Fi version of the signal and for transmitting and transmitting the Wi-Fi version of the signal inside a building. A system that allows for signal penetration into a building. 29. The system of claim 28, wherein said first transmit chipset comprises a Peraso chipset. 2. The method of claim 1, wherein the first frequency is selected from the group consisting of 3.5 GHz, 24 GHz, 28 GHz, 39 GHz, 60 GHz, 71 GHz, and 81 GHz. System.

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