KR20100135272A - Micro-miniature monolithic electromagnetic resonators - Google Patents

Abstract

The filter includes a substrate and one or more resonator structures formed on the planar side of the substrate. Each of the one or more resonator structures has a resonant frequency and includes a plurality of adjacent line segments and a folded transmission line patterned to form a plurality of gaps disposed between the adjacent segments. The ratio of the sum of the average width of the adjacent lines and the average width of the gaps to the substrate thickness is no more than 0.50. The filter further includes an input terminal coupled to one end of the one or more resonator structures and an output terminal coupled to the other end of the one or more resonator structures.

Description

Miniature Monolithic Electromagnetic Resonator {MICRO-MINIATURE MONOLITHIC ELECTROMAGNETIC RESONATORS}

Statement on federally sponsored research and development

The U.S. Government has licensed the payment for the present invention under reasonable terms as defined by the terms of Contract No. H94003-05-C-0508 provided by the Defense MicroElectronics Activity (DMEA) established by the US Department of Defense. May have the right to require a patent owner to license others to the invention in limited circumstances.

Related application

This application claims priority from US Patent Provisional Application No. 61 / 070,634, filed March 25, 2008 and US Patent Provisional Application No. 61 / 163,167, filed March 25, 2009, which provisions are incorporated herein by reference. Included clearly.

Field of invention

FIELD OF THE INVENTION The present invention relates generally to microwave filters, and more particularly to microwave filters designed for narrowband applications.

Electrical filters have long been used for the processing of electrical signals. In particular, electrical filters are used to select desired electrical signal frequencies from the input signal by passing the desired signal frequencies while blocking or attenuating other undesirable electrical signal frequencies. Filters can be classified into several general categories including low pass filters, high pass filters, band pass filters and band cut filters indicating the type of frequencies selectively passed by the filter. In addition, the filters can be used by types such as Butterworth, Chebyshev, Inverse Chebyshev and Ellipse, indicating the type of band-shaped frequency response (frequency cutoff characteristics) the filter provides in relation to the ideal frequency response. Can be classified.

The type of filter used often depends on the intended use. Typically, in communication applications, band pass filters are used to filter or block RF signals of all bands except one or more predefined bands in cellular base stations and other communication equipment. For example, such filters are typically used at the front end of a receiver to filter out noise and other unwanted signals that are harmful to components of the receiver in a base station or communication equipment. Placing a sharply defined band pass filter directly at the receiver antenna input will often eliminate various adverse effects resulting from strong interfering signals having frequencies near the desired signal frequency. Due to the position of the filter at the receiver antenna input, the insertion loss must be very low so as not to degrade the sensitivity of the receiver as measured by the noise value of the receiver. In most filter techniques, achieving low insertion loss requires a compromise of the corresponding filter stiffness or selectivity.

Often, in commercial communications applications, the minimum possible pass using narrowband filters to allow the fixed frequency spectrum to be divided into the maximum possible number of frequency bands to increase the actual number of users that may fit into the fixed spectrum. It is desirable to filter the band. In addition to the enormous increase in wireless communications, such filtering must provide both high selectivity (the ability to distinguish signals separated by small frequency differences) and sensitivity (the ability to receive weak signals) in an increasingly unfavorable frequency spectrum. . Of particular importance is the frequency range of about 800-2200 MHz. In the United States, the 800-900 MHz range is used for analog cellular communications. Personal Communication Service (PCS) is used in the range of 1,800-2.200 MHz.

In general, microwave filters couple two electromagnetic blocks, i.e., a plurality of resonators that store energy very efficiently at one frequency f ₀ , and combine electromagnetic energy between the resonators to form multiple stages or poles. It is constructed using combinations. For example, a four-pole filter may include four resonators and five couplings between signal inputs, resonators, and signal outputs. The strength of a given bond is determined by its reactance (ie inductance and / or capacitance). The relative intensities of the bonds determine the filter shape, and the topology of the bonds determines whether the filter will perform a band pass or band cut function. The resonant frequency f ₀ is mainly determined by the inductance and capacitance of each resonator. In typical filter designs, the frequency at which the filter operates is determined by the resonant frequencies of the resonators constituting the filter. Each resonator must have a very low internal resistance so that the response of the filter can be sharp and highly selective for the reasons described above. This low resistance requirement tends to dictate the size and cost of the resonators for a given technology. Typically, microwave filters have multiple resonant frequencies, which enable microwave filters to operate in different modes. These resonant frequencies may be multiples of the fundamental frequency f ₀ and the fundamental frequency f ₀ (eg, 2f ₀ , 3f _0, etc.) or multiples of the divisor of the fundamental frequency f ₀ (eg, 2f ₀ / n, 3f ₀ / n and the like).

Traditionally, filters have been manufactured using conventional conductors, i.e. non-superconductors. These conductors have inherent losses, and as a result, circuits formed therefrom have varying degrees of loss. Loss is particularly important in the case of resonant circuits. The quality factor Q of a device is a measure of its waste or loss of power. For example, resonators with higher Q have lower losses. Typically, resonant circuits made from conventional metals in microstrip or stripline structures have a Q of at most about 400. With the discovery of high temperature superconductivity in 1986, attempts have been made to fabricate electrical devices from high temperature superconductor (HTS) materials. The microwave characteristics of HTSs have improved greatly since their discovery. Epitaxial superconductor thin films are now routinely formed and commercialized.

Currently, there are various applications where microstrip narrowband filters are as small as possible. This is especially true for wireless applications where HTS technology is used to obtain small size filters with very high resonator Q. The necessary filters are often very complex, perhaps with more than 12 resonators and some cross couplings. However, the available size of available substrates is generally limited. For example, wafers available for HTS filters generally have a maximum size of only 2 or 3 inches. Thus, there is a great need for means to obtain filters as small as possible while maintaining high quality performance. For narrowband microstrip filters (eg, bandwidths of about 2 percent, but especially less than 1 percent), this size problem can be very serious.

The factors that determine the size of these types of filters vary. In general, the filter size is determined by the decrease in the center frequency of the filter, the decrease in insertion loss target, the increase in the number of resonators required, the increase in power processing requirements (compression, intermodulation), or the closest. It will increase if the stray coupling between neighboring resonators is too large to be ignored. Either of these can render the filter impractical due to the limitations imposed by the limited small substrate size.

In order to maintain the high quality performance of the filter, it is desirable to minimize the peak current densities in the filter's structure as large as possible. As described in US Pat. No. 6,026,311, the peak current densities in the filter structure can be reduced by increasing the widths of the microstrip lines and the gaps between the lines with respect to the substrate thickness. That is, wider microstrip lines can be used in areas of the filter structure where high current is expected to minimize the current density in those areas, thereby improving the power handling capability of the resulting filter. However, the relatively high current flowing through the microstrips creates a relatively large electromagnetic field that interferes with the surrounding structures. Thus, in the case where the filter has a plurality of resonators, box-shaped structures can be arranged around the respective resonators to prevent the electric fields generated in each of the resonators from interfering with each other. However, these box shaped structures add to the size and cost of the filter.

In addition to size and loss considerations, a particular concern of the present invention is the minimization of intermodulation distortion (IMD), which is becoming increasingly important in microwave and RF amplifier designs. IMD is an undesirable phenomenon that occurs when two or more signals of different frequencies are present at the input of a nonlinear device to produce pseudo radiations having frequencies different from the desired harmonic frequencies of the filter. The frequencies of the intermodulation products are mathematically related to the frequencies of the original input signals and can be calculated by the equation mf ₁ ± nf ₂ , where f ₁ is the frequency of the first signal and f ₂ is the frequency of the second signal. M, n = 0, 1, 2, 3, ... The intermodulation products are generated in various orders, and the order of the distortion products is given by the sum of m + n. Conventional filter design techniques operate the filter in higher order modes (i.e., the mode corresponding to the second resonant frequency from the fundamental frequency f ₀ or higher frequency), resulting in higher order intermodulation modes of congestion. This indicates that it is impractical.

Thus, there is a need to provide a filter that has a smaller size, has the least unwanted mode activity, and achieves a very high unloaded Q.

In accordance with the present invention, a monolithic filter comprises a substrate (eg, a substrate composed of dielectric material), and one or more resonator structures (which may be virtually planar) formed on the planar side of the substrate. In one embodiment, the filter has the form of a microstrip filter and thus includes a continuous ground plane disposed on the other planar side of the substrate. Each of the resonator structure (s) has a resonant frequency, for example, in the microwave range (eg, in the range of 800-2,200 MHz). Each resonator structure includes a plurality of adjacent line segments and a folded transmission line (eg, spiral-in, spiral-out) patterned to form a plurality of gaps disposed between adjacent line segments. out) structure). In one embodiment, the folded transmission line is comprised of a high temperature superconductor (HTS) material. The filter further includes an input terminal coupled to one end of the one or more resonator structures, and an output terminal coupled to the other end of the one or more resonator structures. The input terminal and output terminal can be coupled to the resonator structure (s), so that the filter can operate as a narrowband filter.

The ratio of the sum of the average width of the adjacent lines and the average width of the gaps to the thickness of the substrate is 0.50 or less. In one embodiment, the ratio is 0.30 or less. In another embodiment, the ratio is 0.20 or less. In yet another embodiment, the ratio is 0.10 or less. Each of the resonator structures may have any shape, for example rectangular or circular. In another embodiment, each resonator structure has a nominal linear electrical length of full wavelength at the resonant frequency of each resonator structure. If the filters include multiple resonator structures, they may be coupled in series with each other. In this case, each of the resonator structures can have a nominal linear electrical length of the full wavelength at the resonant frequency of each resonator structure, and the input terminal and output terminal can be coupled to the resonator structures, so that the filter is of higher order. Can operate in the mode of.

Other and further aspects and features of the invention will become apparent upon reading the following detailed description of the preferred embodiments which are intended to illustrate rather than limit the invention.

The figures illustrate the design and use of preferred embodiments of the invention, wherein like elements are indicated by common reference numerals. To better understand how the above and other benefits and objects of the present invention are achieved, a more detailed description of the invention briefly described above will be provided in connection with specific embodiments thereof shown in the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS The present invention will be described and described in more detail using the accompanying drawings, with the understanding that these drawings only illustrate exemplary embodiments of the invention and therefore should not be considered as limiting its scope.
1 is a plan view of a spiral-in, spiral-out resonator filter of the prior art.
2 is a cross-sectional view of the prior art spiral-in, spiral-out resonator of FIG. 1 taken along line 2-2.
3 is a plan view of a basic spiral-in, spiral-out resonator structure constructed in accordance with the present invention.
4 is a cross-sectional view of the spiral-in, spiral-out resonator structure of FIG. 3 taken along lines 4-4.
5 is an enlarged view of the spiral-in, spiral-out resonator structure of FIG. 4 taken along line 5-5.
6 is a plan view of a resonator constructed in accordance with the present invention using two spiral-in, spiral-out resonator structures shown in FIG.
FIG. 7 is a plan view of the two-wavelength resonator of FIG. 6 compared to a two-wavelength spiral-in, spiral-out resonator of the prior art.
8 is a plan view of another resonator constructed in accordance with the present invention using two circular spiral-in, spiral-out resonator structures.
9 is a plan view of a single resonator filter constructed in accordance with the present invention using the eight resonators shown in FIG. 6 coupled to each other to form a single higher order resonator.
10 is a graph of the calculated frequency response of the filter of FIG. 9.
11 is a normalized input power plotted for two types of resonators constructed in accordance with the present invention, one type of resonator with contiguous / rounded corners and a second type of resonator with uncontiguous / rounded corners. Graph showing intermodulation distortion.
12 is a plan view of a multiple resonator filter constructed in accordance with the present invention using the four resonators shown in FIG.
13 is a graph of the calculated frequency response of the filter of FIG. 12.
14 is a plan view of a multiple resonator filter constructed in accordance with the present invention using the two resonators shown in FIG.
15 is a graph of the calculated frequency response of the filter of FIG. 14.
16 is a plan view of a multiple resonator filter constructed in accordance with the present invention using the ten resonators shown in FIG.
Figure 17 is a plan view of a multiple resonator filter using eight two wavelength resonators constructed in accordance with the present invention.
18 is a graph of the calculated frequency response of the filter of FIG. 17.

Unlike conventional approaches to maximize the widths of the resonator lines to reduce the peak current density in the resonator, reducing the resonator lines and gaps with respect to the filter substrate results in a relatively small filter exhibiting high quality factor Q and inherent power handling capabilities. It was found to provide. In addition, contrary to conventional thinking, higher order filters operating in higher order even modes do not easily stimulate neighboring modes to provide a very clear wideband response with a much wider band without reentrant mode, and high temperature superconductor (HTS) materials The use of has been found to further reduce nonlinear effects.

In the illustrated embodiments of the radio frequency (RF) filters described below, full wavelength (λ) spiral-in, spiral-out resonators are used because they have the ability to reduce peak current near the edges of the resonator lines. Because. The filters are used as band pass filters with a pass band in the desired frequency range, such as 800-900 MHz or 1,800-2,220 MHz. In a typical scenario, RF filters are placed in the front end of a receiver (not shown) behind a broadband pass filter that removes energy outside the desired frequency range.

The conventional filter 10 as shown in FIGS. 1 and 2 is first described. The conventional filter structure 10 includes a substrate 12 and a spiral-in, spiral-out (SISO) resonator structure 14 patterned on one planar side (top) of the substrate 12.

For ease of manufacture, the resonator structure 14 may be formed in a monolithic manner on the substrate 12 using conventional techniques such as photolithography. In the illustrated embodiment, the resonator structure 14 may be comprised of an HTS material such as epitaxial thin film thallium barium calcium cuprate (TBCCO) or yttrium barium cuprate (YBCO). Alternatively, the resonator structure 14 may be composed of superconductors, such as magnesium diboride (MgB 2), niobium, or other superconductors having a transition temperature lower than 77K, which may lead to substrates where the designer does not fit HTS materials. This is because it is allowed to use. As an alternative, the resonator structure 14 may be comprised of conventional metals such as aluminum, silver or copper, but an increase in resistive losses in these materials may limit the applicability of the present invention.

Substrate 12 may be composed of a dielectric material such as LaAlO ₃ , magnesium oxide (MgO), sapphire or polyimide. In the illustrated embodiment, the conventional filter 10 has a microstrip structure and thus further comprises a continuous ground plane 16 disposed on the other planar side (bottom) of the substrate 12 opposite the resonator structure 14. Include. As an alternative, the conventional filter 10 has a stripline structure, in which case the filter 10 may instead comprise another dielectric substrate (not shown), the resonator structure 14 having a respective It can be inserted between the dielectric substrates. The filter 10 further includes an input terminal (pad) 18 and an output terminal (pad) 20 coupled to the resonator structure 14 in a manner that configures the filter 10 to have narrowband characteristics.

The resonator structure 14 includes folded transmission lines 22 patterned to form a SISO structure. In general, the SISO structure is a conductor folded over itself to form two parallel lines 24 connected to each other by a single 180 ° bend 26. The two lines 24 are then spirally wound together in the same direction around the bend 26, with the ends of one line 24 coming out of the structure in one direction and coupled to the input terminal 18, the other line. The end of 24 comes out of the structure in the opposite direction and is coupled to the output terminal 20. That is, one end of the transmission line 22 has a plurality of left-turned turns that rotate through at least 360 ° when engaged, and the other end of the transmission line 22 has at least 360 ° when engaged. It has a plurality of turns to the right that rotate through. At least one turn wound to the left is disposed between at least two turns wound to the right and at least one turn wound to the right is disposed between at least two turns wound to the left. Further details describing various types of such SISO resonator structures are disclosed in US Pat. No. 6,026,311.

As shown in FIG. 2, the transmission line 22 forms a plurality of line segments 32 and a plurality of gaps or spaces 34 between the line segments 32. Transmission line 22 generates an electromagnetic field with an induction field 36 which tends to be the same as the widths of line segments 32 and the gaps 34 between line segments 32. In particular, as a result of the SISO structure, the currents in adjacent line segments 32 are unidirectional, which tends to reduce the peak magnitude of the current near the edges of the transmission line 22 in the resonator structure 14.

In the embodiment shown in FIG. 2, the average width of the line segments 32 (0.250 mm in this example) and the average of the gaps 34 relative to the thickness 38 (0.500 mm in this example) of the substrate 12. The ratio of the sum of the widths (0.250 mm in this example) is relatively large (1 in this example), which creates an electromagnetic field that extends far beyond the resonator structure 14 itself, so that it is below the resonator structure 14 and the substrate 12. A relatively large induction field 36 is provided between the disposed ground plane 16 and any metal elements including electrically grounded lids over the substrate 12.

Now, a filter 50 constructed in accordance with one embodiment of the present invention as shown in FIGS. 3 and 4 is described. Like the conventional filter structure 10, the filter 50 has a substrate 52, a spiral-in, spiral-out (SISO) resonator structure 54 patterned on one planar side (top) of the substrate 52. The resonator structure 54 in such a way that the filter 50 is configured to have narrow band characteristics, and a continuous ground plane 56 disposed on the other planar side (bottom) of the substrate 52 opposite the resonator structure 54. And an input terminal (pad) 58 and an output terminal (pad) 60 coupled to it. Like the resonator structure 14, the resonator structure 54 is patterned to form a SISO structure to form a plurality of line segments 72 and intervening gaps 74 between the line segments 72. ).

Importantly, unlike the conventional resonator structure 14, the average width of the line segments 72 (0.050 mm in this example) and the line segments 72 relative to the thickness of the substrate 52 (0.500 mm in this example). The ratio of the sum of the average widths (0.025 mm in this example) of the gaps 74) is relatively small. In the illustrated embodiment this ratio is 0.15, but this ratio may be 0.50 or less, preferably 0.30 or less, more preferably 0.20 or less. The widths and intervening gaps 74 of the line segments 72 are uniform, but the widths of the line segments 72 as well as the gaps 74 as long as the ratio of the widths and gaps to the substrate thickness remains relatively small. Note that the widths may also be non-uniform.

Thus, having such a low ratio creates an electromagnetic field that does not extend far beyond the resonator structure 14, and thus above the substrate 52 and the ground plane 56 disposed below the resonator structure 14 and the substrate 52. A relatively small induction field 76 is created between any metal elements including the electrically grounded leads. This allows the physically planar resonator structure 54 not to be coupled to the multiloss metal elements surrounding the resonator structure 54 but to exhibit a three-dimensional aspect (ie, a donut or toroidal electromagnetic field), resulting in more efficient energy storage. cause. That is, the resonator structure 54 is more energy efficient due to the minimal interaction between the resonator structure 54 and the outside world.

Direct capacitive coupling of the resonator structure 54 through the input terminal 58 and the output terminal 60 may be achieved at the high voltage ends of the resonator structure 54. As described in US Pat. No. 6,026,311, the lengths of the high voltage ends of the resonator structure 54 can be adjusted according to external loading, so that current nodes occur at the geometric center of the resonator structure 54, thereby transmitting the transmission line 62. Edge current reduction occurs at the edges of the < RTI ID = 0.0 >

The current density of the resonator structure 54 was calculated using a full-wave planar program Sonnet with the same cell sizes as the width of the line segments and the gaps therebetween. Sonnet uses red for the strongest current densities, and as the current weakens, the colors change downward along the rainbow, turning blue for the weakest current densities. As shown in gradation, the corresponding current densities range from very dark gray for the strongest current densities to very light gray or white for the mid range current densities and almost black for very low current densities.

As shown in FIG. 3, a relatively low current density region 80 is located in the center of the resonator structure 54 and at the ends of the transmission line 62, which reflects three current nodes for the propagation length structure. (I.e. for a full wavelength transmission line, the first zero current node is located at the beginning of the transmission line, the second zero current node is located in the middle of the transmission line, and the third zero current node is located at the end of the transmission line) , A relatively high current density region 82 is located around the resonator structure 54, which reflects two current peaks for the propagation length structure (i.e., for the full wavelength transmission line, the first current peak is the transmission line). At the midpoint of the spiral-in portion of the second current peak will be at the midpoint of the spiral-out portion of the transmission line).

Single resonator structures constructed in accordance with the present invention can be used as building blocks for the design of higher order resonators that can operate at much smaller and / or lower frequencies than similar conventional monolithic resonators. Such resonators may be used in filters designed to operate the resonator in higher resonant modes. These higher order modes do not readily stimulate neighboring modes, providing a very clear broadband response with no entry mode and no sign of pseudo mode out of up to three times the desired resonance. Such resonators may operate in any propagation mode of operation (nλ, where n is any integer) (eg, second (λ), fourth (2λ), sixth (3λ), etc.). These higher order resonators also have higher power processing capability. The resonators can be designed around the desired higher order mode. That is, the resonator can be tailored to the selected higher order mode, so very little energy will be coupled to other modes.

For example, as shown in FIG. 6, two basic resonator structures 54 are connected in series to form a two wavelength (2λ) resonator 100. As a result of the relatively low ratio of the widths of the transmission lines to the substrate thickness and the gaps of the basic resonator structures 54, more electromagnetic fields are defined closer to the substrate surface, thus the far field effect of the basic resonator structures 54. Are greatly reduced, permitting closer access of the resonator structures 54 and enabling smaller higher order filters.

Direct capacitive coupling to resonator structures 54 can be achieved at or near the central current node, so that other modes of resonator structures 54 are not easily stimulated, because the resonator can This is because the local voltage at the central capacitive coupling node is nearly zero when resonating in nλ / 2 mode. Modeling suggests that approximation (n ± 1) λ modes can be stimulated, even though the filter is often misaligned at such frequencies and the energy combined can actually be very small. Compared to the conventional two wavelength (2λ) resonator 40 shown in FIG. 8 designed to operate at the same frequency, the footprint of the two wavelength (2λ) resonator 100 is much smaller.

As shown in FIG. 6, a relatively low current density region 80 is formed in the center of each resonator structure 54, at the ends of the transmission line 62 and between the resonator structures 54. ), Which reflects five current nodes for a two-wave length structure, with two relatively high current density regions 82 located at the peripheries of the resonator structures 54 This reflects four current peaks for the two wave length structure.

The reduced size of the higher order resonator 100 reduces costs due to the reduced substrate area and smaller microwave packaging, and allows conventional metal (non-HTS) filters to be made small enough to be used in cellular handset type applications. It can be seen that it offers the possibility. For HTS applications, the smaller size filter can also significantly reduce the overall cold head load, allowing the use of smaller, less power consuming coolers. The increased length of the resonator (above quaternary mode) helps to reduce some of the nonlinear effects in materials such as HTS by causing multiple peaks along the length of the transmission line to reduce the peak current in the resonator. In addition, these higher order modes radiate much less than lower modes, allowing for further reduction of filter size. This is mainly due to the low ratio of the widths of the lines and gaps to the substrate thickness, because the electromagnetic fields do not extend very far from the resonators and preferentially with other parts of the same resonators, not the ground planes of neighboring resonators Because they interact.

Although the basic resonator structures have been described as being rectangular, it should be appreciated that the basic resonator structures may have other shapes. For example, referring to FIG. 8, two circular resonator structures 154 are connected in series to each other to form a two wavelength (2λ) resonator 150. Like the basic resonator structures 54, each of the circular resonator structures 154 is patterned to form a SISO structure to form a plurality of line segments 172 and intervening gaps 174 between the line segments 172. A folded transmission line 162. The ratio of the sum of the average width of the line segments 172 to the substrate thickness and the average width of the gaps 174 between the line segments 172 is relatively small. The current density of the resonator structures 154 was calculated using a propagation plane program sonnet with cell widths equal to the width of the line segments and the gaps therebetween.

As shown in FIG. 8, a relatively low current density region 180 is located in the center of each resonator structure 154, at the ends of the transmission line 162 and between the resonator structures 154. Located at the center of), which reflects five current nodes for the two-wave length structure, with two relatively high current density regions 182 located at the peripheries of the resonator structures 154, which are two wave lengths. Reflect the four current peaks for the structure.

It should be appreciated that the resonator structures disclosed herein can be combined to provide resonators that can operate at wavelengths greater than 2λ to improve their power processing capability when used for signal transmission purposes. For example, referring to FIG. 9, a single resonator 190 is constructed with eight basic resonator structures connected together in series between the input port 194 and the output port 196 to provide an improvement of 9 dB of power processing. 192 (ie, three consecutive doublings of the resonators 2 ³ ). Each of the resonator structures 192 is similar to the resonator structure 54 described above in that the widths of the line segments and intervening gaps are relatively small relative to the substrate thickness. The resulting resonator 190 has an effective wavelength of 8λ that can operate in many nλ modes. The calculated frequency response S ₁₁ , S ₂₁ of the resonator 190 is shown in FIG. 10. Further details describing techniques for improving power processing of filters using multiple basic resonator structures are disclosed in US Patent Publication No. 2008-0278262 A1.

The corners of the resonators described herein may have a shape to achieve a desired IMD slope. For example, referring to FIG. 11, the IMD for a resonator with rounded / continuous or rounded corners and the IMD for a resonator with square corners were measured for normalized input power. As shown, the resonator with rounded corners showed an IMD slope of three, while the resonator with square corners showed an IMD slope of four. Thus, it may be advantageous for the corners of the resonators to have a shape that depends on whether the filter should operate at a relatively low power level or at a relatively high power level.

The aforementioned resonators can be combined together to form multiple resonator filters. For example, referring to FIG. 12, the band pass filter 200 is an input coupled to the first resonator 100 (1) through four two wavelength (2λ) resonators 100, a capacitive coupling 212. Output terminal 210 coupled to fourth resonator 100 (4) via terminal 208 and capacitive coupling 214. The second resonator 100 (2) and the third resonator 100 (3) are connectors which are means for improving the inherent coupling between the second resonator 100 (2) and the third resonator 100 (3). Through 216 are coupled to each other at their tops and bottoms. The current density of the resonators 100 was calculated using a propagation plane program sonnet with cell widths equal to the width of the line segments and the gaps therebetween. As shown in FIG. 12, relatively low current density regions 220 are located in the center of each resonator 100 and at the ends of the transmission line, and relatively high current density regions 222 are formed in the second and second regions. 3 is located around the resonators 100 (2) and 100 (3). The calculated frequency response S ₁₁ , S ₂₁ of the resonator filter 200 is shown in FIG. 13.

As another example, referring to FIG. 14, the band pass filter 250 includes two 16 wavelength (16λ) resonators 190, an input terminal 252 and a second resonator coupled to the first resonator 190 (1). An output terminal 254 coupled to 190 (2). The calculated frequency response S ₁₁ , S ₂₁ of the resonator filter 250 is shown in FIG. 15. As another example, referring to FIG. 16, the band pass filter 300 includes ten sixteen wavelength (16λ) resonators 190, an input terminal 302 coupled to a first resonator 190 (1), and a tenth. And an output terminal 304 coupled to the resonator 190 (10). The second resonator 190 (2) and the fifth resonator 190 (5) are coupled to each other at their tops and bottoms through cross couplings 306, and the sixth resonator 190 (6) and the fifth The nine resonators 190 (9) are coupled to each other at their tops and bottoms via cross couplings 306 to produce transmission zeros in the approximate cutoff band, moving from a Chebysef-like response to a pseudo elliptic response. do. This is done to improve the approximate band selectivity (the slope of the cutoff) of the filter at the cost of blocking further away from the passband.

As another example, referring to FIG. 17, a band pass filter 350 is input coupled to a first resonator 352 (1) through eight two wavelength (2λ) resonators 352, a capacitive coupling 362. An output terminal 360 coupled to an eighth resonator 352 (8) via a terminal 358 and a capacitive coupling 364. Each of the resonators 352 has a line width of 0.01 mm for the line segments. And the width of the gaps between the line segments is 0.005 mm, which is the same as the resonator 100 shown in Fig. 6. Thus, the ratio of the sum of the average width of the line segments and the average width of the gaps to the substrate thickness is 0.03. The calculated frequency responses S ₁₁ and S ₂₁ of the resonator filter 350 are shown in FIG. 18.

While specific embodiments of the invention have been shown and described, the foregoing description is not intended to limit the invention to those embodiments. It will be apparent to those skilled in the art that various changes and modifications can be made without departing from the spirit and scope of the invention. Accordingly, the present invention is intended to cover alternatives, modifications and equivalents that may fall within the spirit and scope of the invention as defined by the claims.

Claims (17)

Board;
One or more resonator structures formed on the planar side of the substrate—each of the one or more resonator structures having a resonant frequency and patterned to form a plurality of adjacent line segments and a plurality of gaps disposed between the adjacent line segments A transmission line, wherein the ratio of the sum of the average width of the adjacent lines and the average width of the gaps to the thickness of the substrate is less than 0.50;
An input terminal coupled to one end of the one or more resonator structures; And
An output terminal coupled to the other end of the at least one resonator structure
Monolithic filter comprising a. The monolithic filter of claim 1, wherein the input terminal and output terminal are coupled to the one or more resonator structures such that the filter can operate as a narrowband filter. The monolithic filter of claim 1, wherein the folded transmission line has a spiral-in, spiral-out structure. The monolithic filter of claim 1, wherein the ratio is 0.30 or less. The monolithic filter of claim 1, wherein the ratio is 0.20 or less. The monolithic filter of claim 1, wherein the ratio is 0.10 or less. The monolithic filter of claim 1, wherein the substrate is comprised of a dielectric material. 8. The monolithic filter of claim 7, further comprising an electrically conductive ground plane disposed on the other planar side of the substrate. The monolithic filter of claim 1, wherein each of the one or more resonator structures is rectangular. The monolithic filter of claim 1, wherein each of the one or more resonator structures is circular. The monolithic filter of claim 1, wherein each of the one or more resonator structures is a planar structure. The monolithic filter of claim 1, wherein the folded transmission line is comprised of a high temperature superconductor (HTS) material. The monolithic filter of claim 1, wherein each of the one or more resonator structures has a nominal linear electrical length of full wavelength at the resonant frequency of the respective resonator structure. The monolithic filter of claim 1, wherein the one or more resonator structures comprise a plurality of resonator structures coupled in series with each other. 15. The apparatus of claim 14, wherein each of the resonator structures has a nominal linear electrical length of full wavelength at the resonant frequency of each resonator structure, wherein the input terminal and output terminal are configured to allow the filter to operate in a higher order mode. Monolithic filter coupled to the resonator structures. The monolithic filter of claim 1, wherein the resonant frequency is within a microwave range. 17. The monolithic filter of claim 16, wherein the resonant frequency is in the range of 800-2,200 MHz.

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