KR19990007352A - NTS interference signal detection / blocking device and method in digital television receiver - Google Patents

Abstract

In a digital receiver such as a digital TV receiver, in order to suppress an artifact component of NTSC co-channel interference, a code decoding result is obtained by using blocking filtering, and then the selected code decoding result is compared by comparing the code decoding result to a final code decoding. Get the result.

In order to suppress the artifact component of the NTSC cochannel interference, the code decoding result obtained using the blocking filtering is selected as the final code decoding result when the NTSC cochannel interference occurs. The digital TV signal is detected to determine whether an NTSC intercarrier of 4.5 MHz has been obtained to ascertain whether actual NTSC co-channel interference has occurred.

Description

NTS interference signal detection / blocking device and method in digital television receiver

The present invention relates to a digital TV device, such as a high definition digital television (HDTV), used for terrestrial broadcasting in the United States in accordance with the TV Development Subcommittee (ATSC) standard. And a circuit for detecting co-channel interference occurring in an analog TV signal.

The TV Development Subcommittee (ATSC) is a digital television standard (DTV) in the 6 MHz bandwidth TV channel, which is used by the National Television Subcommittee (NTSC) in the United States for analog television over-the-air signals, in the Digital TV Standards released September 16, 1995. ) Specifies the residual sideband (VSB) signal for signal transmission. The spectrum of the residual sideband (VSB) DTV signal is designed as if the spectrum of the same channel interferes with the NTSC analog TV signal. This is 1 of the NTSC analog TV signal, which is lowered between the even times that the luminance and chrominance component energy of the NTSC co-channel interfering analog TV signal is lowered, and the even times of the 1/4 horizontal scan line ratio of the NTSC analog TV signal. The odd amplitude modulation sideband frequencies of the pilot carrier and the DTV signal are located when they are an odd multiple of the / 4 horizontal scan line ratio. The video carrier of the NTSC analog TV signal is 1.25 MHz offset from the low end frequency of the TV channel. The carrier of the DTV signal is an offset value obtained by multiplying 59.75 times the horizontal scanning line ratio of the NTSC analog TV signal in order to arrange the DTV carrier signal of about 309,877.6 KHz obtained from the low limit frequency of the TV channel. Thus, the carrier of the DTV signal is about 2,690122.4 Hz obtained from the intermediate frequency of the TV channel.

The exact code ratio shown in the digital TV standard is multiplied by (684/286) times, which gives a speech carrier offset value of 4.5 MHz from the video carrier of the NTSC analog TV signal. Here, the horizontal scan line / code number of the NTSC analog TV signal is 684, and multiplied by 286, which is a horizontal scan line ratio factor of the NTSC analog TV signal, to obtain a voice carrier offset value of 4.5 MHz from the video carrier included in the NTSC analog TV signal. The code rate is 10.762238 mega code / sec, which can be included in the VSB signal extended by 5.381119 MHz from the DTV carrier signal. In other words, the VSB signal may be limited from the low limit frequency of the TV channel to the 5.690997 MHz extended band.

In general, the ATSC standard for terrestrial broadcasts of digital HDTV signals in the United States allows transmission of both high-definition TV formats with a 16: 9 ratio. The first HDTV display format uses 1920 sample / scan lines and 1080 active horizontal scan lines / 30 Hz frame format with a 2: 1 interlaced area. Another HDTV display format uses a 1280 luminance sample / scan line and a sequentially scanned TV image / 60 Hz frame format. The ATSC standard also uses a DTV display format rather than an HDTV display format that parallelly transmits four TV signals with normal resolution when compared to NTSC analog TV signals.

DTV, transmitted by residual sideband amplitude modulation in terrestrial broadcasts in the United States, consists of a continuous data area containing 313 consecutive data segments each. The data area is regarded as a continuous counting of modulo-2, in which data areas counted in radix and data areas counted in even numbers constituting a data frame are successive. The frame rate is 20.66 frames / second. Each data segment is active for 77.3 microseconds. Thus, if the code rate is 10.78 MHz, it becomes 832 code / data segment. Each segment of data will have four sets of sync codes with + S, -S, -S, and + S consecutively on one line. The value of + S is a level below the maximum positive data deviation, and the value of -S is a level above the maximum negative data deviation. The initial line of each data area includes a sync code set area that encodes an experimental signal for equalization of channels and prevention of multipath processing. The experimental signal is followed by three 63-PN sequential samples followed by 511 pseudo noise sequential samples (PN sequential). The median value of the 63-PN sequential sample in the area synchronization code is according to the initial logic rule of the first line of the data area counted in radix and the second logic rule of the first line of the data area counted in even. And the first and second logic rules are complementary to each other.

Data in the data line is trellis coded using twelve inserted trellis codes, each of which is a 2/3 ratio trellis code without one bit encoded. The inserted trellis codes are Reed-Solomon coding, which is prepared as an error correction countermeasure for correcting an unexpected situation occurring in a noise source such as an ignition device of an exposed vehicle. The Reed-Solomon coding result is coded and transmitted in 8-level (3 bits / sign) one-dimensional alignment code for airwave transmission, and the transmission is performed without the trellis coding procedure and precoding of the identified code. The Reed-Solomon coding result is coded by one-dimensional alignment codes of 16 levels (4 bits / sign) for transmission of the cable TV broadcast, and the transmission is also performed without precoding. Residual sideband (VSB) signals have their own transmission wave, whose amplitude depends on the modulation rate.

The unique carrier is replaced with a pilot carrier of fixed amplitude according to the modulation ratio described above. This fixed amplitude pilot carrier is generated by shifting the DC component when modulating the voltage applied to the balanced modulator that generates an amplitude modulation sideband that is supplied to the filter and supplies the VSB signal in response. If eight levels of the 4-bit code coding have normal values of -7, -5, -3, -1, +1, +3, +5, and +7 as carrier modulated signals, the pilot carrier is normal. Has the value 1.25. The normal value of + S is +5 and the normal value of -S is -5.

When transmitting in the prior art of DTV, it was required to determine whether to use a code precoder in DTV broadcasting, which performs a code generation circuit and precode filters the code. This decision of the broadcaster depends on co-channel interference regardless of the needs of the NTSC broadcaster. The code precoder complements the post-coding of the sign accidentally caught on each DTV receiver with a cutoff filter, used prior to the use of a data slicer inside the code decoder circuit, to prevent interfering co-channel interfering NTSC signals. do. Sign precoding is not used on the sign sync set of the data line or on the data line to which the sync data of the data area is transmitted.

Co-channel interference is eliminated far from NTSC stations, which is likely to occur when a certain ionospheric condition is established or when a hot summer day is likely to cause co-channel interference. However, such interference does not occur without the same channel occurring at NTSC stations. If NTSC interference is confined to its broadcast area only, HDTV broadcasting may use a code precoder to operate HDTV signals more easily than NTSC interference. This would also allow the cutoff filter to be used as a code post coder for fully matched filtering at the DTV receiver. If the exclusion or possibility of NTSC signal interference is sparse, the flat spectral noise that causes code value errors in the trellis decoder is reduced, and DTV broadcasts stop using the code precoder, so that each DTV receiver has a code post. The coder will be unnecessary. Without the broadcaster's knowledge of this situation, co-channel interference of NTSC signals becomes a disturbance requirement for some broadcast receiving areas, causes short circuits in cable broadcasting, causes intermediate frequency video disturbances unsuitable for NTSC receivers, and analog TV recording. Magnetic tape may need to be used for digital TV recording, which may result in, or other inherent problems of abnormal conditions may arise.

Referring to the co-channel interference detection included in the digital TV signal of Nielsen et alii with the patent application No. 5,594,496 on Jan. 14, 1997, the NTSC co-channel interference detector in the data segment containing the area sync code is based on the received noise. In order to obtain a data region blocking filter response having a period independent of the code coding in which cochannel interference is measured, data region blocking filtering is required. The data area cutoff filter response in these periods is compared to the response aimed at better cutoff filtering, which blocks NTSC co-channel interference results. If at the signal level there is a better cut-off filtering result with a significant reduction, then the signal is the actual result of NTSC co-channel interference, and cut-off filtering is used to block the NTSC co-channel interference result prior to code decoding. Consider. If there is no significant reduction in noise detected using better cut filtering, the noise is Johnson noise and it is assumed that no cut filtering is used to block NTSC co-channel interference results. The reason is that blocking filtering in differentially delayed code decoding combines linearly with respect to Johnson noise increased by 3 dB or more.

The current ATSC DTV standard does not recognize a transmission scheme using code precoding. The blocking of the same channel that interferes with the analog TV signal is determined to be performed in the trellis decoding process after the data slicing process related to the code decoding. This processing omits whether or not to determine whether precoding should be made at the time of transmission. Unfortunately, the same channel that interferes with the analog TV signal causes errors in the data slicing process, which places more strain on the error correction decoding procedure, trellis decoding, and Reed Solomon decoding. These errors limit the range of the broadcast area, resulting in lower revenues for commercial DTV broadcasts. Thus, although the current ATSC DTV standard does not allow sign precoding in DTV transmission, it is desirable to provide co-channel interfering analog TV signal protection prior to data slicing.

In general, the linear combination has a meaning added or subtracted depending on whether it is applied by classical or modular equations. Modular combinations applied to and performed on linear combinations are treated with modular calculations. A type of coding that records a series of digital codes through differential delay, a linear combination of differentially delayed intervals, and an example by post coding of a code used in the prior art of an HDTV receiver, is described herein. Defined as signed recording. Also described herein is a type of coding that records a series of digital codes through the modular combination itself with delayed results of the modular combination and pre-coding of the symbols used in the prior art of HDTV transmitters. Defined as recording.

The problem of co-channel interference derived from an analog TV signal can be solved by installing an appropriate filter circuit inside the receiver in the past problem of interference of the receiver. If the active area of the system channel is not exceeded, co-channel interference can be prevented by blocking signal transmission during DTV modulation, and the performance of the system can be regarded as a signal overlap problem. The filter circuit inside the receiver is applied to select a digital signal derived from co-channel interference by the analog TV signal, and in order to sufficiently reduce the energy of the aforementioned system channel, it is inversely correlated with the correlation of the analog TV signal. Take advantage of characteristics.

Looking at the co-channel interference derived from the analog TV signal, it enters the system channel between the DTV transmitter and the DTV receiver. The use or no use of sign precoding in a DTV transmitter has no effect on co-channel interference derived from analog TV signals. In a DTV receiver, if co-channel interference is not wide enough to span the end of the receiver to capture the system channel, this is preferable to a data slicing circuit with a cutoff filter to reduce the energy of the upper energy spectral component of the co-channel interference. Thus, errors occurring during data slicing can be reduced. The DTV station must match the carrier frequency exactly, which is 310 KHz, which is close to the TV channel assigned low threshold frequency. Therefore, this carrier frequency is the optimum offset value of the frequency obtained from the video carrier of the co-channel NTSC analog TV signal close to the interference. The optimum offset value of this carrier frequency is exactly 59.75 times the horizontal scanning line frequency f _H of the NTSC analog TV signal. The result of co-channel interference included in the demodulated DTV signal includes bits when 59.75 times the horizontal scan line frequency (f _H ) of the NTSC analog TV signal, and the same channel video carrier interferes the digital HDTV carrier and the analog TV signal. Bits of 287.75 times f _H generated by heterodyne between heterogeneous chrominance subcarriers of the same channel interfering with the digital HDTV carrier and the analog TV signal, these bits being 59.75 times f _H. This bit is close to the fifth harmonic. This result encompasses bits close to 345.75 times f _H generated by heterodyne between the digital HDTV carrier and the audio channel of the same channel that interferes with the analog TV signal, which are the sixth at 59.75 times f _H. It is a bit close to harmonics. The harmonic relationship of these bits is a precisely designed single-block filter that incorporates a few codes with differential delays. The use of the NTSC cancellation comb filter prior to data slicing inside the DTV receiver is intended to additionally perform the first type of sign recording and to correct the sign by data slicing.

The data slicing operation following the first type of sign recording inside the DTV receiver performs the non-destructive quantification processing of the code resulting from the first type of sign recording, so that as far as data transmission is concerned, the data quantification level should match the code level. Because it is designed. Quantification is identified with the co-channel interfering analog TV signal remaining after the filtering associated with the first type of sign recording, but the extent is less than the step between code sign levels.

This is a kind of phenomenon capture that consumes a small signal in the quantification process to obtain a good signal gain.

As far as data transmission is concerned, a series of digital data codes is made over the entire length of the system channel. When the second type of sign recording is processed with sign precoding at the DTV transmitter, an additional combination of differentially delayed series of data codes does not boost the transmit power or more interfere with the propagation of the analog TV signal. It is based on the modular principle of increasing the average inner code distance to suppress. Instead, the basic mechanism for suppressing the jamming of the analog TV signal is to be provided by blocking filtering on the DTV receiver side by facing the DTV signal with its attenuator and suppressed by the quantification effect inside the data slicer. As a result, the analog TV signal results included in the cutoff filter response are immediately sent along the cutoff filter.

The order of progression of the first and second types of code recording processes has little effect on signal transmission through the system channel in such a situation, since the coding combination of a series of codes does not reduce the transmission of the signal. . The order of progression of the first and second types of code recording processes is that the co-channel interfering analog TV signal is recorded while the second type of code is not superposed between the first type of code recording and the subsequent data slicing. It does not affect the receiving power of digital receiver very much. This view is a general basis upon which the present invention is based, citing a digital TV receiver having an NTSC co-channel interference blocking adaptive filter with US Patent Application No. 08 / 839,691 April 15, 1997. The adaptive filter circuit receives to code decode the series of 2N-level codes that are susceptible to the results of co-channel interfering analog TV signals. N at this time is a positive integer. When the NTSC co-channel interference is detected, the NTSC co-channel interference is detected in the data slicing process used to directly perform code decoding on the baseband signal recovered by detecting the VSB-AM DTV signal simultaneously It is determined whether or not this energy has enough energy. If it is determined that the NTSC co-channel interference does not have enough energy to make error correction impossible, the baseband signal is code decoded using a first data slicer to produce a sign decoding result. If it is determined that the NTSC co-channel interference has enough energy so that error correction is not possible, the baseband signal is first decoded to reduce the energy of the co-channel interference before being decoded using a second data slicer. It is filtered using a comb filter. Incidentally, the first comb filter is to perform a first type of sign recording procedure that causes an error in the sign decoding result value generated in the second data slicer. The first type of sign recording procedure performed prior to data slicing in the second data slicer is considered a precoding procedure for filtering suitable for NTSC co-channel interference blocking. The second comb filter performs a second type of sign recording procedure after data slicing in the second data slicer to compensate for the first type of sign recording procedure and to perform a post coding procedure to generate a sign decoding result. do.

The first type of sign recording procedure records a series of input codes through a differential delay and a first linear combination of the differentially delayed sections. The second type of sign recording procedure recovers the partially decoded sign decoding result recovered by the second data slicer. The second type of sign recording procedure uses a second linear combination of partially filtered sign decoding results obtained by feeding back previous sign decoding results having a delay similar to the differential delay shown in the series of input codes. . While generating data area synchronization information and data segment synchronization information, by forcing the code decoding result to confirm an ideal code decoding result drawn from a memory inside the DTV receiver, The driving error included in the coded decoding result is shortened. One of the first and second linear combinations performs a subtraction process and the other is an addition process.

The main point of the present invention is that the final code decoding result is NTSC co-channel rather than the measurement value selected from the intermediate code decoding result obtained by data slicing the base code code code without using a comb filter to suppress NTSC co-channel interference. In order to suppress the interference, it is to be corrected based on the measurement value selected from the post coded code decoding result obtained by performing the comb filtering. The decision is obtained by comparing each post coded code decoding result corresponding to the intermediate code decoding result over each data segment. From the corresponding value of the intermediate code decoding result, a substantial deviation of the post coded code decoding result is considered to occur, due to the appearance of an NTSC artifact component in the baseband, whereby the post coded code decoding result is intermediate. It is preferentially selected over the decoding result value and included in the final code decoding result value, otherwise it indicates that the selection is wrong.

Accordingly, the present invention has been made to solve the above-mentioned problems of the prior art, and includes a digital TV signal receiver including a digital TV signal detection apparatus that provides a series of 2N-level codes each having a code interval of a predetermined time length. As related, the series of 2N-level codes are likely to be accompanied by artifact components of the co-channel interfering analog TV signal, where N represents a positive integer. The codes are grouped into contiguous data segments with headers of each data segment sync code, and the data segments are grouped into contiguous data regions with initial data segments of each data region containing data region sync codes. The data area is converted from the data area to the data area. The digital TV signal receiver includes circuitry to provide M single comb filters, responsive to a series of 2N-level codes, wherein each single comb filter response is co-channel interference than the series of 2N-level codes. Less is accompanied by the artifact component of the analog TV signal. A plurality of code decoders are included in the digital TV signal receiver to generate respective code decoding measurement results. A first code decoder of the plurality of code decoders directly responds to the series of 2N-level codes measured first code decoding result. Each of these plurality of code decoders generates respective signed decoding result values respectively and responds to one of the M single comb filters, each of these code decoding measurement results is fully post coded to each matched filtering, The response of each of the single comb filters of the M is obtained from the respective sign decoding measurement results. Compared to the first code decoder, the plurality of code decoders include at least a second code decoder to produce a measured second code decoding result. According to the present invention, the digital TV signal receiver includes a circuit for detecting whether there is a deviation between the measured first and second code decoding result values, and an optimum value from each code decoding measurement result value. A final code decoding result is generated between times when the sync code occurs, including an optimal selection circuit for selecting (best estimates). The selection of the optimum value by the optimum selection circuit depends on the current deviating from the first measurement code decoding result value of the respective code decoding measurement result values.

In an embodiment of the present invention, M is 2, and the plurality of code decoders generate a second code decoding result value including only the second code decoder except the first code decoder. The optimal selection circuit takes the following form in an embodiment of the present invention. A multiplexer is connected to provide a selectability between the measured first and second code decoding result values, thereby generating a final code decoding result value in a section in which a sync code occurs. A square result is used as an absolute measurement value of the deviations using a squarer for the deviation between the measured first and second code decoding result values, and an average value of the square result is generated using an averager. Threshold detectors form a condition in which the multiplexer selects the measured second code decoding result value in response to an average value of the squared result value and exceeds a predetermined threshold value so that a sync code occurs. Generate a final code decoding result value, otherwise, the multiplexer selects the measured first code decoding result value to form a condition for generating the final code decoding result value between intervals in which a sync code occurs.

1 is a code decoding result obtained by using an NTSC cancellation comb filter before decoding a code, post-coding the comb filter after decoding a code, and taking measurements to block NTSC co-channel interference, Block diagram of a digital TV signal receiver using a co-channel interference detector, which compares the sign decoding results obtained without taking measurements to block NTSC co-channel interference.

2 illustrates in detail the NTSC co-channel interference detector used in the digital TV signal receiver of FIG. Block diagram showing an example of an optimal selection circuit for selecting a most suitable value among code decoding results.

Fig. 3 is a block diagram showing in detail a part of the digital TV signal receiver of Fig. 1 when the NTSC cancellation comb filter takes a 12 code delay;

Fig. 4 is a block diagram showing details of a part of the digital TV signal receiver of Fig. 1 when the NTSC cancellation comb filter takes 6 code delays.

Fig. 5 is a block diagram showing details of a part of the digital TV signal receiver of Fig. 1 when the NTSC cancellation comb filter takes a 2-video line delay.

Fig. 6 is a block diagram showing details of a part of the digital TV signal receiver of Fig. 1 when the NTSC cancellation comb filter takes a 262-video line delay.

Fig. 7 is a block diagram showing details of a part of the digital TV signal receiver of Fig. 1 when the NTSC cancellation comb filter takes a 2-video frame delay.

FIG. 8 is a block diagram showing in detail a part of the digital TV signal receiver of FIG. 1 for generating a predetermined code decoding result value in a data synchronization period; FIG.

9 is a block diagram of a digital TV receiver using multiple NTSC cancellation comb filters to perform parallel code decoding.

Fig. 10 is a combined configuration diagram of Figs. 10A and 10B detailing a suitable code selection circuit used in the digital TV signal receiver of the type shown in Fig. 9;

Fig. 10A is a block diagram showing in detail the circuit arrangement of the digital TV signal receiver of Fig. 9 for calculating the code decoding result in the data synchronization section described above.

10B includes an example of an optimal selection circuit for selecting the most suitable one of a plurality of code decoding results in order to generate a final code decoding result in a period in which a sync code occurs. Block diagram depicting a TV signal receiver in detail.

Hereinafter, an embodiment of the present invention will be described in detail with reference to the accompanying drawings. In various parts of the circuit, shimming delays must be inserted in the order in which the results of the operation are correct, which is well known in the field of electronic circuit design. Otherwise, an abnormal condition occurs with respect to a particular shimming delay request, which is omitted herein.

Fig. 1 shows that a digital TV signal receiver is used for recovering corrected error data, which data is displayed on a TV set as suitable for recording by a digital video cassette recorder or MPEG-2 decoding. The DTV signal receiver of FIG. 1 receives a TV broadcast signal from the reception antenna 8, but may instead receive a wired broadcast signal. The TV broadcast signal is delivered as an input signal to the device termination 10. In general, the device termination 10 includes a radio-frequency amplifier and a first decoder for converting a radio-frequency TV signal into an intermediate frequency TV signal, thereby providing Passed as an input signal. The DTV receiver preferentially prioritizes a plurality of conversion types of the IF amplifier chain 12, an IF amplifier for amplifying a DTV signal and converting it into a UHF band by a first detector, and a second amplifier for converting the amplified DTV signal into a VHF band. A second detector and an IF amplifier for amplifying and converting the DTV signal into the VHF band. If demodulation to baseband is performed in the digital sector, the IF amplifier chain 12 further includes a third detector to convert the amplified DTV signal to the final intermediate frequency band close to the baseband.

First, surface acoustic wave (SAW) filters are used in the IF amplifier for the UHF band, shaping the channel selection response and removing adjacent channels. The SAW filter quickly removes and cuts frequencies above 5.38 MHz from the carrier frequency and pilot carrier of the suppressed residual sideband (VSB) DTV signal, which becomes frequency and fixed amplitude. The SAW filter removes according to the amount of frequency-modulated acoustic carrier of any co-channel interfering analog TV signal. Eliminating the FM acoustic carrier of any co-channel interfering analog TV signal in the IF amplifier chain 12 prevents the artifact component of the carrier from being generated when the final IF signal is detected, thereby recovering the baseband sign. In addition, it is possible to prevent artifact component interference due to data slicing of the baseband code during code decoding. In the prevention of artifact component interference due to data slicing of the baseband code while code decoding is performed, it is more preferable to perform comb filtering before data slicing.

The final IF output signal from the IF amplifier chain 12 is passed to a complex demodulator 14, which demodulates the residual sideband amplitude modulated DTV signal in the final intermediate frequency band to recover the real and imaginary parts of the baseband signal. . Demodulation in the digital sector occurs after analog-to-digital conversion of the final intermediate frequency band in a few megacycle ranges, which is included in an HDTV receiver with US Patent No. 5,479,449, registered December 26, 1995 in the United States. Reference was made to a digital VSB detector with a tracker. Alternatively, if demodulation is done in the analogue section, the result is generally analog-to-digital conversion to facilitate future procedures. The complex demodulation is first performed by in-phase (I) synchronous demodulation and quadrature phase (Q) synchronous demodulation. In general, the in-phase (I) synchronous demodulation procedure represents a 2N-level code that has 8-bit or more precision and encodes N bits of data. In general, 2N is 8 when the DTV signal receiver of FIG. 1 receives the airwaves through the antenna 12, and 2N is 16 when the DTV signal receiver of FIG. 1 receives the wired broadcast wave. An aspect of the present invention is to receive broadcast waves from the ground to the air, and in FIG. 1, a portion of the DTV receiver that provides code decoding and error correction decoding for the received wired broadcast transmission is omitted.

A code synchronizer and equalizer circuit 16 receives at least the real part sample of the digitized in-phase (I channel) baseband signal from the complex demodulator 14, and in FIG. Indicates receiving an imaginary part sample of a phase (Q channel) baseband signal. The circuit 16 has a digital filter having an effective weighting effect, which compensates for the dual phase (ghost) and tilt angle (tilt) included in the received signal. The sign synchronization and equalization circuit 16 performs sign synchronization or rotation, similar to amplitude equalization and double phase cancellation. Sign synchronization and equalization circuits used for sign synchronization are performed prior to amplitude equalization, which is referred to US application number 5,479,449. In this design, the demodulator 14 delivers an overextracted demodulator response that includes real and imaginary numbers of baseband signals to the code synchronization and equalization circuits 16. After sign synchronization, the overextracted data is removed by about one-tenth, to extract baseband I-channel signals at normal code rates, and to reduce the sampling rate through digital filtering used for amplitude equalization and double phase cancellation. to be. Rotation or phase tracking, in code synchronization and equalization circuits where amplitude equalization proceeds with code synchronization, is also a well known technique in the field of digital signal receiver design.

Each sample of the output signal of circuit 16 is divided into 10 or more bits, effectively one analog code digitally representing one of the (2N-8) levels. The output signal of the circuit 16 is carefully gain controlled by one of several known methods, and the level of the ideal step for the sign is known. Since the response speed of such gain control is extremely fast, one method of gain control selected adjusts the DC component of the baseband signal real part delivered from the complex demodulator 14 to the normal level of +1.25. In general, such a gain control method is well described in U.S. Patent Application No. 5,479,449, and is shown in more detail in the automatic gain control of a radio receiver for the reception of digital HDTV signals of U.S. Patent No. 5,573,454, registered on June 3, 1997, the present invention See this.

The output signal from the circuit 16 is transmitted to the data synchronization circuit 18 as an input signal, which is the synchronization information F of the data area derived from the signal of the equalized baseband I channel and the data segment synchronization information S. ). Alternatively, the input signal delivered to the sync detection circuit 18 may be obtained prior to equalization.

At a normal code rate, signal samples of the equalized I-channel, delivered as an output signal at circuit 16, are passed as input signals to NTSC cancellation comb filter 20. The comb filter 20 comprises a first delay device 201 for generating a pair of differentially delayed series of 2N-level codes, and a differential delay for generating a response of the comb filter 20. A first linear combiner 202 is provided for linearly combining a series of symbols. Referring to what is described in US Pat. No. 5,260,793, the first delay device 201 provides a delay equal to 12 cycles of a 2N-level code, and the first linear combiner 202 becomes an adder. Each sample of the comb filter 20 output signal is divided into 10 or more bits, effectively one analog code digitally representing one of (4N-1) = 15 levels.

The code synchronization and equalization circuit 16 is considered to be designed to limit the direct current bias component of its input signal, which has a normalized level of +1.25 and is a complex demodulator 14 due to pilot carrier detection. Appears in the real part of the baseband signal transmitted from Thus, each sample of the output signal of the circuit 16 is applied to the input signal of the comb filter 20, and effectively one analog code has the following normal levels: -7, -5, -3, -1 It is a digital representation of one of +1, +3, +5, and +7.

These code levels are referred to as odd code levels and are detected by the odd level data slicer 22 to produce the temporary code decoding results of 000, 001, 010, 011, 100, 101, 110, and 111, respectively. A sample of each output signal of the comb filter 20 effectively shows that one analog code has the following normal levels: -14, -12, -10, -8, -6, -4, -2, 0, One of +2, +4, +6, +8, +10, +12, and +14 is represented digitally. These code levels are called even code levels and are excellent to generate the temporary code decoding results of 001, 010, 011, 100, 101, 110, 111, 000, 001, 010, 011, 100, 101, 110, and 111, respectively. Level data slicer 24 is detected.

The data slicers 22, 24 in this sense are referred to as difficult solutions or as simple solutions used to perform the Viterbi decoding scheme. Using the multiplexer connection to shift its position in the circuit and applying a bias to modify its slicing range, the odd-level data slicer 22 and the even-level data slicer 24 are connected to a single data slicer. An array can be replaced with However, these arrangements are not suitable because of their complex operation.

Next, the code synchronization and equalization circuit 16 will be described a method for suppressing the DC bias component included in its input signal. The DC bias component at this time has a normal level of +1.25, and appears in the real part of the baseband signal transmitted from the complex demodulator 14 due to pilot carrier detection. Alternatively, the sign synchronization and equalization circuit 16 is designed to retain the direct current bias component contained in its input signal, which, in a sense, simplifies the design of the equalization filter in the circuit 16. In this case, the data slicing levels in the odd-level data slicer count the DC bias component accompanying the data processes included in its input signal and take it as an offset value. Whether the circuit 16 is designed to block or maintain a DC bias component contained in an input signal with discontinuity, which is regarded as the data slicing level in the even-level data slicer 24, the first linear combiner 202 is provided with an adder. However, if the differential delay delivered by the first delay device 201 is selected, the first linear combiner 202 becomes an adder and the data slicing levels in the even-level data slicer 24 are themselves. The superimposed DC bias component accompanying the data processes included in the input signal of is counted and taken as an offset value.

The comb filter 26 is used after the data slicers 22, 24 to produce a post coding filter response as a precoding filter response of the comb filter 20. The comb filter 26 includes a multiplexer 261 having three inputs, a second linear combiner 262, and a second delay device 263 having the same delay as the first delay device 201 in the comb filter 20. It is composed of The second linear combiner 262 is a modulo-8 adder if the first linear combiner 202 is a subtractor, and a modulo-8 subtractor if the first linear combiner 202 is an adder. The first linear combiner 202 and the second linear combiner 262 may be configured with respective ROMs to sufficiently speed up the linear combining operation to support the included sample rate. The output signal from the multiplexer 261 carries the response obtained from the post coding comb filter 26 and is delayed by the second delay device 263. The second linear combiner 262 combines the precoded coded decoding result obtained from the even-level data slicer 24 with the output signal obtained from the second delay device 263.

The output signal of the multiplexer 261 is applied to the multiplexer 261 when it is selected in response to the first, second, and third states of the multiplexer control signal applied from the controller 28 to the multiplexer 261. Play one of the three input signals. First input of the multiplexer 261 while data area synchronization information F and data segment synchronization information S obtained from the equalized baseband I-channel signal are recovered by the data synchronization detection circuit 18. The port receives the ideal sign decoding result from the memory in the controller 28. While forming a condition for providing the final coding result of the output signal and the ideal code decoding result from the memory in the controller 28 in the multiplexer 261, the controller 28 is configured to provide a first control signal of the multiplexer control signal. The state is passed to the multiplexer 261. The odd level data slicer 22 transfers the intermediate code decoding result as an output signal to the second input port of the multiplexer 261. The multiplexer 261 is conditioned to the second state of the multiplexer control signal to reproduce the intermediate code decoding result as the final coding result which is its output signal. The second linear combiner 262 transmits the code decoded result post-coded as its output signal to the third input port of the multiplexer 261. The multiplexer 261 is conditioned to the third state of the multiplexer control signal to reproduce the post coded code decoding result as its final coding result which is its output signal.

While the data synchronization detection circuit 18 recovers the data area synchronization information F and the data segment synchronization information S, the post coding comb is fed back by feeding back an ideal code decoding result value transferred from the memory in the controller 28. Execution errors occurring in the post-coded sign decoding results from the filter are reduced. This part is the main part of the present invention and will be described in more detail later.

The output signal from the multiplexer 261 in the post-coded comb filter 26, which contains the final code decoding results contained in the three parallel bit groups, is a data assembler for application to the data interleaver 32. It is collected as (30). The data interleaver 32 rectifies the collected data into a series of parallel data and sends it to the trellis decoder circuit 34. In general, the trellis decoder circuit 34 uses a 12 trellis decoder. The trellis decoding result applied from the trellis decoder circuit 34 is transferred to the data de-interleaver circuit 36 for rectification. The byte analysis circuit 38 performs Reed Solomon decoding on the output signal of the data de-interleaver 36 to generate a modified series of error bytes that are passed to the data randomizer 42. The data is transferred to the performing Reed Solomon decoder circuit 40, and converted into a Reed Solomon error correction coding byte. The data randomizer 42 transfers the reproduced data to the other receiver (not shown). The other of the complete DTV receivers includes a packet classifier, an audio decoder, an MPEG-2 decoder and others. Another of the DTV receivers integrated into a digital tape recorder / player would have circuitry for converting the data into the format required for recording.

The co-channel interfering NTSC signal detector 44 provides a controller 28 capable of knowing that the co-channel interfering NTSC signal is in a state sufficient to cause an uncorrectable error included in the data slicing performed by the data slicer 22. Providing. If detector 44 indicates that the co-channel interfering NTSC signal is not sufficiently complete, except when data area synchronization information F and data segment synchronization information S are recovered by data synchronization detector circuit 18. At other times, the controller 28 transmits a second state of the multiplexer control signal to the multiplexer 261 side. This condition is used by the multiplexer 261 to reproduce the intermediate code decoding result transmitted from the odd level data slicer 22 into its output signal. If the detector 44 indicates that the co-channel interfering NTSC signal is sufficiently complete to cause an uncorrectable error included in the data slicing performed by the data slicer 22, then data area synchronization information F and data segment synchronization At other times except when the information S is recovered by the data synchronization detector circuit 18, the controller 28 transmits a third state of the multiplexer control signal to the multiplexer 261 side. This condition is used by the multiplexer 261 to reproduce, as its output signal, the post coded code decoding result passed from the second linear combiner 262 as the second linear combination result.

FIG. 2 is an embodiment of the present invention, in a form that the co-channel interfering NTSC signal detector 44 may take. The subtractor 441 separately combines the intermediate code decoding result delivered from the odd level data slicer 22 and the post coded code decoding result passed as the second linear combination result from the second linear combiner 262. If the co-channel interfering NTSC signal is negligible and the random noise contained in the baseband I channel signal is negligible, these virtual, post coded code decoding results will be similar. Thus, the output signal difference from subtractor 441 will be less. However, if the co-channel interfering NTSC signal is a significant amount, the output signal difference from subtractor 441 will generally not be small. But sometimes the difference in signal is better.

The method for measuring the energy contained in the output signal difference from the subtractor 441 is to square the difference value of the output signal with the squarer 442, and the squarer response over a short interval with the average value circuit 443. It is obtained by determining the average value of. The squarer 442 is performed using ROM. The average circuit 443 is performed using a delay line memory for storing some suitable digital samples and an adder that adds the digital samples currently stored in the delay line memory. The average value of the energy distributed in the short interval included in the output signal difference obtained in the subtractor 441, determined by the average value circuit 443, is connected to the digital comparator to support the threshold detector 444. The threshold value of the threshold detector 444 does not exceed the difference between the average value of the short intervals included in the random noise accompanying the intermediate code decoding result and the post coded code decoding result value applied to the subtractor 441. Big enough The threshold is exceeded if the co-channel interfering NTSC signal is large enough to make it impossible to correct the errors included in the data slicing performed by the data slicer 22. The threshold detector 444 allows the controller 28 to indicate whether the threshold is exceeded.

Another embodiment of the squarer 442 may be replaced by another circuit, for example an absolute value circuit, that represents an absolute measurement of deviation between the first and second code decoding measurement results. For fast calculations, ROMs can be used to implement such absolute value circuits.

FIG. 3 details a portion of the digital TV signal receiver block configuration of FIG. 1 using 120 NTSC cancellation comb filters 20 and 126 post coding comb filters 26. FIG. Subtractor 1202 acts as the first linear combiner in NTSC rejection comb filter 120 and modulo-8 adder 1262 acts as the second linear combiner in post-coded comb filter 126. The NTSC cancellation comb filter 120 represents a delay of 12 code intervals using the first delay device 1201, and the post coded comb filter 126 uses 12 delays using the second delay device 1263. Indicates the delay of the code interval. The 12 code delays represented by each of the delay devices 1201 and 1263 are close to one cycle delay of the analog TV video carrier result at 59.75 times the horizontal scanning frequency f _H of the analog TV. The 12-signal delay is close to 5 cycles of analog TV color difference subcarrier results when 287.75 times f _H. The 12-signal delay is close to 6 cycles of analog TV color difference subcarrier results when 345.75 times f _H. This means that the differentially combined response of the subtractor 1202 to the audio carrier, video carrier, and frequency is close to the color subcarrier differentially delayed by the first delay device 1201 trying to block co-channel interference. Because. However, in the portion of the video signal where the end crosses the horizontal scanning line, the degree of correlation of the analog TV video signal at a distance in the horizontal space direction is very low.

1261 types of the multiplexer 261 are controlled by a multiplexer control signal, which in most cases is determined when NTSC co-channel interference is insufficient to correct the errors contained in the output signal from the data slicer 22. It is in a two state, and in most cases where it is determined that NTSC co-channel interference is sufficient to make it impossible to correct an error contained in the output signal from the data slicer 22, it is in the third state. The multiplexer 1261 becomes a control signal in a third state, is fed back to the sum of modulo-8 of the adder 1262, and delayed by 12 code intervals in the delay device 1263 to adder 1262. Becomes the singer of. This is a modular accumulation process where a single error propagates as a driving error, and the error is repeated in all 12 code intervals. The driving error included in the post coded code decoding result from the post coding comb filter 126 is determined by the multiplexer 1261 at the beginning of each data segment as long as the entire data segment includes area synchronization. It is shortened when it is in the first state for the two code intervals. When the control signal is in the first state, the multiplexer 1261 reproduces the output signal abnormal code decoding result delivered from the memory in the controller 28. The driving error stops when the abnormal code decoding result is led to the output signal of the multiplexer 1261. Since there is a 4 + 69 (12) code / data segment, the abnormal code decoding result slips back four code sections in the phase of each data segment, so that no driving error lasts longer than three data segments.

FIG. 4 details the block configuration of a portion of the digital TV signal receiver of FIG. 1 using 220 NTSC cancellation comb filters 20 and 226 post coding comb filters 26. FIG. The NTSC cancellation comb filter 220 represents a delay of six code intervals using the first delay device 2201, and the post-coded comb filter 226 uses six delays using the second delay device 2263. Indicates the delay of the code interval. The six code delays represented by each of the delay devices 2201 and 2263 are close to 0.5 cycle delays of the analog TV video carrier result at 59.75 times the horizontal scanning frequency f _H of analog TV, and 287.75 of f _H. When doubled, it is close to 2.5 cycles of analog TV color difference subcarrier results, and at 345.75 times of f _H , it is close to three cycles of analog audio carrier results. Adder 2202 acts as a first linear combiner in NTSC rejection comb filter 220, and modulo-8 subtractor 2262 acts as a second linear combiner in post-coded comb filter 226. Since the delay shown by the delay devices 2201 and 2263 is shorter than the delay shown in the delay devices 1201 and 1263, even if the frequency is close to zero converted from the analog TV carrier frequency, it becomes a narrow band and a subtractor. Rather than the good correlation in the signal differentially combined by 1202, the anti-correlation in the signal additionally combined by the adder 2202 is better. Acoustic carrier blocking is weaker in the NTSC cancellation comb filter 220 than in the NTSC cancellation comb filter 120 response. However, if the acoustic carrier of the same channel interfering with the analog TV signal is cut off in the surface acoustic filtering or the acoustic trap in the intermediate frequency amplifier chain 12, the less sound cancellation of the comb filter 220 is not a problem. While using the NTSC cancellation comb filter 220 of FIG. 4 rather than the NTSC cancellation comb filter 120 of FIG. 3, the response to the sync tip is removed. Thus, error correction enhancements in trellis decoding and Reed-Solomon coding are in fact decreasing.

2261 types of the multiplexer 261 are controlled by a multiplexer control signal, which in most cases is determined when NTSC co-channel interference is insufficient to correct the errors contained in the output signal from the data slicer 22. It is in a two state, and in most cases where it is determined that NTSC co-channel interference is sufficient to make it impossible to correct an error contained in the output signal from the data slicer 22, it is in the third state. The multiplexer 2221 becomes a control signal in a third state, fed back to the sum of modulo-8 of the adder 2262, and delayed by six code intervals in the delay device 2263 to adder 2262. Becomes the singer of. This is a modular accumulation process in which a single error propagates as a driving error, and the error is repeated in all six code intervals. The driving error included in the post coded code decoding result from the post coding comb filter 226 is determined by the multiplexer 2221 at the beginning of each data segment as long as the entire data segment includes area synchronization. It is shortened when it is in the first state for the two code intervals. When the control signal is in the first state, the multiplexer 2221 reproduces the output signal abnormal code decoding result delivered from the memory in the controller 28. The driving error stops when the abnormal code decoding result is led to the output signal of the multiplexer 2221. Since there is a 4 + 138 (6) code / data segment, the abnormal code decoding result slips back four code intervals in the phase of each data segment, so that no driving error lasts longer than the two data segments. Even if the driving error is repeated more frequently and affects the 12 inserted trellis codes in duplicate, the possibility of substantially continuing the cycle of driving error included in the post coding comb filter 226 is maintained. Less than for the coding comb filter 126.

FIG. 5 details a portion of the digital TV signal receiver of FIG. 1 using 320 NTSC canceling comb filters 20 and 326 post coding comb filters 26. FIG. The NTSC cancellation comb filter 320 uses a first delay device 3201 which exhibits a delay of 1368 code intervals, the delay of which is substantially equal to the interval of two horizontal scan lines of the analog TV signal, and also the post coding. The comb filter 326 represents the delay using the second delay device 3263. The first linear combiner included in the NTSC removal comb filter 320 is a subtractor 3202, and the second linear combiner included in the post coded comb filter 326 is a modulo-8 adder 3262.

3261 types of the multiplexer 261 are controlled by a multiplexer control signal, where in most cases it is determined that NTSC co-channel interference is insufficient to correct an error contained in the output signal from the data slicer 22, In the second state, the NTSC co-channel interference is found to be in the third state in most cases where it is determined that the error included in the output signal from the data slicer 22 is not correctable. The DTV receiver preferably has circuitry for detecting a change between scan lines intersected in the NTSC co-channel interference, such that the controller 28 causes the multiplexer 3331 to be in a third state under such conditions. Withhold it.

The multiplexer 3221 becomes a control signal in a third state, fed back to the sum of modulo-8 of the adder 3262, and delayed to 1368 code intervals in the delay device 3263 to adder 3262. Becomes the singer of. This is a modular accumulation process where a single error propagates as a driving error, and the error is repeated in all 1368 code intervals. Since the length of this code code is longer than the length of a single block of the Reed-Solomon code, the single running error is easily corrected during Reed-Solomon decoding. The driving error included in the post coded code decoding result from the post coding comb filter 326 is, like the four code intervals at the beginning of each data segment, the entire interval of the area synchronization included in each data segment. Over, the multiplexer 3221 is shortened by placing it in a first state. When the control signal is in the first state, the multiplexer 3331 reproduces the abnormal code decoding result received from the memory of the controller 28 as an output signal. The error driving is stopped by using the abnormal code decoding result as an output signal of the multiplexer 3331. The 16.67 x10 ^-6 second period of the NTSC video region exhibits phase deviation over the 24.19 x10 ^-6 second period of the DTV data region so that the DTV data segment with region synchronization eventually scans the entire NTSC frame picture. NTSC frame images each having 684 code intervals have 525 lines, totaling 359,100 code intervals. This is somewhat less than 432 times the 832 code intervals contained in the DTV data segment containing region synchronization. One inferred fact is that if the duration of a driving error is longer than 432, the DTV data segment contains region synchronization. While doing so, the multiplexer 3221 reproduces the abnormal code decoding, causing the data area to be lost. In addition, a phase deviation occurs between the data segment for the start code group using the abnormal code decoding result and the NTSC video scan line. 359,100 code intervals corresponding to 89,775 times the four code intervals included in the code start group are measured, which are scanned values for 89,775 consecutive data segments. One fact that is inferred from the DTV data area / 313 data segments is that if the duration of the driving error is longer than 287, the multiplexer 3221 reproduces the abnormal code decoding while the code start group is in progress and the data area is lost. Will be. Since the two driving error prevention methods are independent of each other, it is unlikely that the driving duration of the error will be longer than 200 or the data area. In addition, if the degree of NTSC co-channel interference is lowered when the driving error is cycled, a condition is formed in which the multiplexer 3221 reproduces the response of the data slicer 22 as an output signal, so that the error is different. This can be corrected faster than using.

5 shows demodulation derived from the response to analog TV horizontal sync pulses, just as NTSC cancellation comb filter 32 equalizes the pulse by blocking a large amount of demodulation results derived from the response to analog TV vertical sync pulses. An example of blocking the results is shown.

These results are co-channel interference with a large amount of energy. The NTSC rejection comb filter 320 blocks colors that are not related to the video content, except for the difference between the scan line and the scan line included in the video content of the analog TV signal over two scan line periods. When not blocked in the tracking comb filter included in the code synchronization and equalization circuit 16, the FM audio carrier of the analog TV signal is cut off. Most analog TV color burst results are also blocked in the NTSC rejection comb filter 320 response. Further, filtering by the NTSC cancellation comb filter 320 is perpendicular to the NTSC interference cancellation made in the trellis decoding procedure.

FIG. 6 details a portion of the digital TV signal receiver of FIG. 1 using 420 NTSC cancel comb filters 20 and 426 post coded comb filters 26. The NTSC cancellation comb filter 420 uses a first delay device 4201 which exhibits a delay of 179,208 code intervals, the delay of which is substantially equal to the period of 262 horizontal scan lines of the analog TV signal, and also the post coding. The comb filter 426 represents a delay using the second delay device 4421. A subtractor 4202 acts as a first linear combiner included in the NTSC rejection comb filter 420, and a modulo-8 adder 4426 acts as a second linear combiner included in the post coding comb filter 426. do.

The 4261 types of the multiplexer 261 are controlled by a multiplexer control signal, in which case most of the NTSC co-channel interference is determined to be insufficient so that an error contained in the output signal from the data slicer 22 cannot be corrected. In the second state, the NTSC co-channel interference is found to be in a third state in most cases where it is determined that the error included in the output signal from the data slicer 22 is not correctable. The DTV receiver preferably has circuitry for detecting a change between areas and areas included in the NTSC co-channel interference, so that the controller 28 is in such a condition that the multiplexer 4401 is in a third state. Holds to be placed.

The multiplexer 4421 becomes a control signal in a third state, fed back to the sum of modulo-8 of the adder 4426, delayed by the delay device 4403 to 179,208 code intervals, and added to the adder 4426. Becomes the singer of. This is a modular accumulation process in which a single error propagates as a driving error, and the error is repeated in all 179,208 code intervals. Since the length of this code code is longer than the length of a single block of the Reed-Solomon code, the single running error is easily corrected during Reed-Solomon decoding. The driving error included in the post coded code decoding result from the post coding comb filter 426 is, as in the four code intervals at the beginning of each data segment, the entire interval of the area synchronization included in each data segment. Over, the multiplexer 4421 is shortened by placing it in a first state. When the control signal is in the first state, the multiplexer 4401 reproduces the abnormal code decoding result received from the memory of the controller 28 as an output signal. The error driving is stopped by using the abnormal code decoding result as an output signal of the multiplexer 4421. The maximum value of the data area required for removing the driving error included in the output signal of the multiplexer 4421 is substantially the same as the value required for removing the driving error included in the multiplexer 3331. However, the number of repeated errors within this period is lowered by factor 131.

An embodiment in which the NTSC cancellation comb filter 420 of FIG. 6 blocks most of the demodulation results derived from the analog TV vertical sync pulse response, just as it blocks all the demodulation results derived from the analog TV horizontal sync pulse response. It is shown. These results are co-channel interference with high energy. In addition, the NTSC rejection comb filter 420 blocks the results occurring in the video content of the analog TV signal, not inter-region or inter-line changes, and removes those horizontal spatial frequencies or color-free stop patterns. Also, most of the results of analog TV color bursts are blocked in the response of the NTSC cancellation comb filter 420.

FIG. 7 details a portion of the digital TV signal receiver of FIG. 1 using 520 NTSC canceling comb filters 20 and 526 post coding comb filters 26. FIG. The NTSC cancellation comb filter 520 uses a first delay device 5201 which exhibits a delay of 718,200 code intervals, the delay of which is substantially equal to two frame periods of the analog TV signal, and also the post coded comb filter. 526 denotes a delay using the second delay apparatus 5251. A subtractor 5202 acts as a first linear combiner included in the NTSC rejection comb filter 520, and a modulo-8 subtractor 5262 acts as a second linear combiner included in the post coding comb filter 526. do.

5261 types of the multiplexer 261 are controlled by a multiplexer control signal, in which case most of the NTSC co-channel interference is determined to be insufficient to correct an error included in the output signal from the data slicer 22, In the second state, the NTSC co-channel interference is found to be in a third state in most cases where it is determined that the error included in the output signal from the data slicer 22 is not correctable. The DTV receiver preferably has circuitry for detecting a change between interframes included in the NTSC co-channel interference, such that the controller 28 causes the multiplexer 5201 to be in a third state under such conditions. Withhold.

The multiplexer 5221 becomes a control signal in a third state, fed back to the sum of modulo-8 of the adder 5262, and delayed by 718,200 code intervals in the delay device 5203 to adder 5262. Becomes the singer of. This is a modular accumulation process where a single error propagates as a driving error, and the error is repeated in all 718,200 code intervals. Since the length of this code code is longer than the length of a single block of the Reed-Solomon code, the single running error is easily corrected during Reed-Solomon decoding. The driving error included in the post coded code decoding result from the post coding comb filter 526 is, as in the four code intervals at the beginning of each data segment, the entire interval of the area synchronization included in each data segment. Over, the multiplexer 5201 is shortened by placing it in a first state. When the control signal is in the first state, the multiplexer 5221 reproduces the abnormal code decoding result received from the memory of the controller 28 as an output signal. The error driving is stopped by using the abnormal code decoding result as an output signal of the multiplexer 5221. The maximum value of the data area required for removing the driving error included in the output signal of the multiplexer 5251 is substantially the same as the value required for removing the driving error included in the multiplexer 3331. However, the number of repeated errors within this period is lowered by factor 525.

7 shows an embodiment that blocks all demodulation results derived from analog TV vertical sync pulse response, as does NTSC cancellation comb filter 520 of FIG. 7 blocks all demodulation results derived from analog TV horizontal sync pulse response. have. These results are co-channel interference with high energy. In addition, the NTSC rejection comb filter 520 blocks the results from the video content of the analog TV signal, rather than a change over two frames, to remove a still pattern independent of those spatial frequencies or colors. Also all the results of the analog TV color burst are cut off in response to the NTSC cancellation comb filter 520.

If you are in the field of TV system design, you can see the different characteristics of correlation and anti-correlation shown in Figures 3-7 in analog signals that can be used to design other types of NTSC rejection comb filters. will be. The use of the two dependent NTSC cancellation comb filters is already known, raising the baseband signal at the 2N level to the (8N-1) data level. Although such a filter has a drawback of limiting a signal-to-noise ratio for random noise interference with code decoding, there is an urgent need for solving the problem of malicious co-channel interference.

FIG. 8 shows the structure of the multiplexer 261 of FIG. 1 in more detail. In this circuit, the abnormal code decoding result is generated and applied to the multiplexer 261. FIG. The multiplexer 261 selectively reads a 3-bit wide output bus 2610 from the multiplexer 261, including the output buffer registers of the ROM 46, ROM 48, ROM 50. The multiplexer 261 further includes a three-phase buffer 2611 to deliver the 3-bit optical output of the multiplexer 2612 to the output bus 2619 while no abnormal code decoding results are generated. The multiplexer 261's response to the NTSC co-channel interference detector 44 is zero, which is such that the amplitude of NTSC co-channel interference cannot correct the error contained in the intermediate code decoding result passed in the data slicer 22. The intermediate code decoding result is reproduced to be an input signal of the three-phase data buffer 2611. The multiplexer 261's response to the NTSC co-channel interference detector 44 is 1, such that the amplitude of the NTSC co-channel interference cannot be corrected in the error contained in the intermediate code decoding result passed in the data slicer 22. The precoded code decoding result is reproduced from the second linear combiner 262 to be an input signal of the three-phase data buffer 2611.

Circuitry for delivering the ideal code decoding result to the output bus 2610 includes an address counter 54, an address counter 54 for addressing the ROM clocks 46, 48, 50, a code clock generator 52, and a ROM 46, 48, 50. Jam reset circuit 56 for resetting 54, ROM 46, 48, 50 to NOR gate 92 which controls address decoders 60, 62, 64 and three-phase buffer 2611 for generating a readable signal. It is composed. The address counter 54 receives the code decoding rate at the code clock generator 52 to count input pulses. Thus, addresses for each of the symbols in one data frame are successively assigned. Appropriate portions of these addresses are taken as input addresses of the ROMs 46, 48, and 50. The jam reset circuit 56 resets the counter 54 so that the data area synchronization information F and the data segment synchronization information S are set to an appropriate value so as to be quickly recovered by the data synchronization detection circuit 18 of FIG. The configuration of the counter 54 preferably consists of a group counting more significant bits in data segment number / data frame and a group counting less significant bits in data segment number / data frame. This configuration simplifies the design of the jam reset circuit 54, reduces the bit width of the input signal applied to the address decoders 60, 62, 64, and facilitates the ROM 46, 48, 50 to some address of the counter 54. Addressed, the bit width of ROM addressing can be reduced.

The ROM 46 stores the ideal sign decoding result for the odd area sync segment and is selectively enabled for receiving 1 at the address decoder 60. The ROM 46 is addressed by a group that counts less significant bits in data segment number / data frame, and the address decoder 60 responds to a group that counts more significant bits in data segment number / data frame. The address decoder 60 becomes 1 only when the data segment portion of the address delivered by the address counter 54 coincides with the address of the odd area sync segment.

The ROM 48 stores the ideal sign decoding result for the even region sync segment and is selectively enabled for receiving 1 at the address decoder 62. The ROM 48 is addressed by a group that counts less significant bits in data segment number / data frame, and the address decoder 62 responds to a group that counts more significant bits in data segment number / data frame. The address decoder 62 becomes 1 only when the data segment portion of the address delivered by the address counter 54 matches the address of the even-region sync segment.

The ROM 50 stores the ideal code decoding result for the start code group at the beginning of each sync segment, and the value received by reading 1 from the address decoder 64 is selectively enabled. The ROM 50 responds to two meaningless bits of the counter 54 output, and the address decoder 64 responds to a group that counts less significant bits in data segment number / data frame. The address decoder 64 becomes 1 only if the data segment portion of the data code / address delivered by the address counter 54 matches some address of the start code group.

The NOR gate 92 receives the response of the address decoders 60, 62 and 64 at one point of each of the three input connections. When an ideal sign decoding result is obtained, one of the address decoders 60, 62, 64 passes 1 as its output signal, and the condition for the NOR gate 92 to respond to 0 in the 3-phase data buffer 2611 is Is formed. In this condition, the three-phase data buffer 2611 imposes a high power impedance on the bit line of the data bus 2610 such that a signal of the multiplexer 2611 cannot be transferred from the multiplexer 2612 to the 3 bit wideband data bus 2610. In the data segment portion for an unpredictable ideal code decoding result, none of the address decoders 60, 62, 64 carry a 1 as an output signal, and the NOR gate 92 has a value of 1 in the three-phase data buffer 2611. Conditions for responding are established. In this condition, the three-phase data buffer 2611 imposes a low power impedance on the bit line of the data bus 2610 so that the signal of the multiplexer 2612 is transferred to the three bit wideband data bus 2610.

Figure 9 shows a modified digital TV signal receiver, as described above, such as the operation of multiple code decoders in parallel, using each even-level data slicer in accordance with the prior art. Another type of NTSC rejection comb filter at this time is each post-coded comb filter that compensates for the precoding announced prior to the NTSC rejection comb filter. The even-level data slicer A24 converts the first type of NTSC cancellation comb filter A20 response into a precoded sign decoding result for application to the first type of post coding comb filter A26. The even level data slicer B24 converts the second type of NTSC cancellation comb filter B20 response into a precoded sign decoding result for application to the second type of post coding comb filter B26. The even level data slicer C24 converts the third type of NTSC cancellation comb filter C20 response into a precoded sign decoding result for application to the third type of post coding comb filter C26. A, B, and C, which are preceded by the component numbers of FIG. 9, distinguish 1,2,3,4,5 corresponding to some of the receivers applied in FIGS.

The code decoding selection circuit 66 of Fig. 9 applies the optimal value of the modified code decoding to the trellis decoding circuit 34 from the abnormal code decoding result value, selects from the intermediate code decoding result received from the data slicer 22, and post-codes it. The various post coded sign coding results received from the comb filters A26, B26, C26 are shown. The optimal value of the sign decoding result is used to add and correct the post-coding filters A26, B26, C26.

The NTSC elimination comb filter A20 and the post-coded comb filter A26 circuit improve the NTSC elimination comb filter 520 and the post coded comb filter 526 of FIG. 7. Therefore, this requires memory to be considered in terms of cost, since 718,200 codes must be stored in each of the two-video frame delay devices 5201 and 5263. However, the storage of the two-video frame delay apparatus 5201 is used when executing the delay apparatuses 4201, 3201, 2201, and 1201 for delaying a short section. The storage location in the two-video frame delay can also be used to realize shorter delay devices 4403, 3263, 2263, 1263.

The high energy demodulation results in response to analog TV sync pulses, equalization pulses, and color bursts are all blocked when the NTSC cancellation comb filter A20 additionally combines alternating video frames. In addition, the results derived from the video content of the analog TV signal, which are not converted over two frames, are blocked to remove their spatial frequency or color-independent stop pattern.

The problem of blocking demodulation results that should be considered first is those demodulation results that occur as a difference between a frame and a frame at a particular pixel point in the analog TV signal image. These demodulation results can be blocked by internal frame filtering methods. The NTSC elimination comb filter B20 and the post coded comb filter B26 circuit are selected to block the remaining demodulation results dependent on the correlation in the horizontal direction, and the NTSC elimination comb filter C20 and the post coded comb filter C26 circuit are arranged in the vertical direction. It is chosen to block the remaining demodulation results that depend on the correlation of. Let's look at how such a design performs better.

If the same channel acoustic carrier that interferes with the analog TV signal is not blocked by surface acoustic wave filtering or acoustic trap in the intermediate frequency amplifier chain 12, the NTSC cancellation comb filter B20 and the post coded comb filter B26 circuit are The NTSC rejection comb filter 120 and the post-coded comb filter 126 of this type are selected. If the same channel acoustic carrier that interferes with the analog TV signal is interrupted by surface acoustic wave filtering or acoustic trap in the intermediate frequency amplifier chain 12, then the NTSC cancellation comb filter B20 and the post coded comb filter B26 circuit are: The NTSC rejection comb filter 220 and the post coded comb filter 226 circuit of FIG. 4 are selected. The reason is that the anti-correlation between video components having six code intervals apart from each other is better than the correlation between video components having 12 code intervals separated from each other.

Only one of two, whether to select a temporally close scan line in the same area or a spatially close scan line in the area combined with the current scan line included in the NTSC removal comb filter C20, must be selected. This makes the best choice for the NTSC cancellation comb filter C20 and the post coded comb filter C26 circuit simpler. In general, it is better to select scan lines that are close in time in the same region because the jump cut between regions reduces the NTSC removal by the comb filter C20. As such, the NTSC cancellation comb filter C20 and the post coded comb filter C26 circuit are of the same type as the NTSC cancellation comb filter 320 and the post coded comb filter 326 circuit of FIG. In another alternative, the NTSC cancellation comb filter C20 and the post coded comb filter C26 circuit are of the same type as the NTSC cancellation comb filter 420 and the post coded comb filter 426 circuit of FIG.

The digital receiver device of FIG. 9 is modified for use in the additional parallel data slicing operation of the present invention, and the performance of each is performed by cascading respective NTSC cancellation comb filters, even level data slicers, and post coding comb filters. do. While two additional parallel data slicing operations are shown in FIG. 9, a variation of the parallel data slicing operation continues to optimize the measurement of the modified sign decoding result.

In the better embodiment of Fig. 9, the code decoding selection circuit 66 selects the abnormal decoding result as the final decoding result if possible. If NTSC cochannel interference is actually measured, the difference between the intermediate code decoding result and the plurality of precoded code decoding results is considered to be due to NTSC cochannel interference. Thus, if anomalous decoding result values are not available, the final code decoding result selection by the code decoding selection circuit 66 relies on a comparison of a plurality of post coded code decoding results with another, and a comparison with the intermediate decoding result. can do.

The above is preferable for the initial measurement because the NTSC co-channel interference actually obtains because the difference between these multiple code decoding results can occur during the formation of noise intervention conditions, where the white noise level is actually The condition is sufficient to cause more error in the post coded sign decoding result than in the intermediate decoding result. The fact that NTSC co-channel interference is present can be ascertained using US Pat. No. 5,594,496 or similar techniques using other NTSC co-channel interference cancellation filters while region synchronization information is occurring. However, first of all, the intensity of NTSC co-channel interference should be monitored in continuous real time to see if the change in NTSC co-channel interference level is fading or the change in video content is counted.

9 illustrates the use of an intercarrier signal for NTSC interference detection in a digital TV receiver having US patent application Ser. No. 08 / 821,945, filed Mar. 21, 1997, for the 4.5 MHz intercarrier included in the NTSC co-channel interference. The monitoring which measured the level is shown. The DTV signal is converted to IF at device termination 10 and applied to a quasi-parallel type IF amplifier chain 68 for NTSC signals. The amplifying step in the IF amplifier chain 68 for an NTSC signal obtains a substantially linear gain in response to a similar amplifying step in the IF amplifier chain 12 for a DTV signal, and in amplifying step in the IF amplifier chain 12. It has the same automatic gain control as the corresponding one. The frequency selectivity of the IF amplifier chain 68 is focused on within ± 250 KHz of the NTSC audio carrier or within ± 250 KHz of the NTSC video carrier. The filtering procedure for measuring the frequency selectivity of the IF amplifier chain 68 is performed using SAW filtering in a UHF IF amplifier when using multiple conversion receiver circuits. The response of the IF amplifier chain 68 is applied to an intercarrier detector 70, which heterodynates the NTSC carrier using a modulated NTSC video carrier as the inherent carrier, and has an intercarrier acoustic intermediate frequency signal having a carrier frequency of 4.5 MHz. Create The intercarrier acoustic IF signal is amplified by an intercarrier acoustic IF amplifier 72, which applies a 4.5 MHz IF amplifier 72 to the intercarrier amplitude detector 74. The response of the amplitude detector 74 is applied to the threshold detector 76. If the NTSC co-channel interference is of sufficient strength to cause an error in the data slicing performed by the data slicer 22, the threshold at the threshold detector 76 is exceeded. The threshold detector 76 provides the code decoding selection circuit 66 to indicate whether the threshold is exceeded. If the indication indicates that the NTSC co-channel interference is of sufficient strength to cause an error in the data slicing performed by the data slicer 22, the indication indicates that the code decoding selection circuit 66 is intermediate from the data slicer 22. By selecting the code decoding result value as the final code decoding result value, the abnormal code decoding result value is not possible in the current code interval. The time constant at the intercarrier amplitude detector 74 should be carefully chosen if finding the best performance. Since the separated code decoding error is correctable, short pulse cancellation of the output signal from the intercarrier amplitude detector 74 with a fast time constant preferentially generates a control signal for selective switching of the final code decoding result from the intermediate code decoding result. The code decoding result is post coded.

Various circuit arrangements are possible for deriving an intercarrier signal having a heterodyne between the audio and video carriers of a co-channel interfering analog TV signal. The number of such arrays is shown in US patent application Ser. No. 08 / 821,945.

10A and 10B show in more detail the circuitry included in the code decoding selection circuit 66 for selecting the final code decoding result.

FIG. 10 is a combined diagram showing how FIG. 10A and FIG. 10B are arranged to provide a complete block diagram of the code decoding selection circuit 66. FIG. The code decoding selection circuit 66 has a 3-bit wide output data bus 78, which travels from the bottom of FIG. 10A to the bottom of FIG. 10B, similar to that shown in FIG. 1, a data assembler 30, a data interleaver. 32, the trellis decoder circuit 34, the data de-interleaver 36, the byte forming circuit 38, the Reed-Solomon decoder circuit 40, and the data de-randomizer 42 are cascaded. FIG. 10A is a circuit similar to that of FIG. 8, and reads the abnormal code decoding result from the ROMs 46, 48 and 50 and applies it to the output data bus 78. FIG.

Fig. 10B shows an optimum measurement selection circuit for selecting the final code decoding result in the time interval when the abnormal code decoding result is impossible. That is, the time interval between data segments or data area sync codes is provided in the DTV signal. The three-phase data buffer 2611 and the multiplexer 2612 of FIG. 8 are replaced by three-phase data buffers O80, A80, B80, and C80 in the optimum measurement selection circuit of FIG. 10B. The three-phase data buffer O80 is a condition under which the response of the AND gate O82 becomes a logic one, so that the intermediate code decoding result is transmitted from the odd level data slicer 22 of FIG. 9 on the output data bus 78. The three-phase data buffer A80 is a condition that the response of the AND gate A82 is a logic one, causing the post coded code decoding result to be transferred from the post coding comb filter A26 of FIG. 9 on the output data bus 78. The three-phase data buffer B80 is a condition that the response of the AND gate B82 is a logic one, causing the post coded code decoding result to be transferred from the post coding comb filter B26 of FIG. 9 on the output data bus 78. The three-phase data buffer C80 is a condition that the response of the AND gate C82 is a logic one, causing the post coded code decoding result to be transferred from the post coding comb filter C26 of FIG. 9 on the output data bus 78. The NOR gate 58 of Fig. 10A transfers its response to the AND gates O82, A82, B82, and C82 as their respective inputs, and thus the three-phase buffers O80, A80, B80, and C80 are abnormal. The condition of low power impedance in the bit line of the output data bus 78 is only present if a sign decoding result is not present on the data bus 78 by one of the three phase output buffers of the ROM 46, 48, 50. Is formed.

The output signal from threshold detector 76 of FIG. 9 becomes logical 1 when the NTSC co-channel interference has sufficient strength to cause an error in the data slicing performed by data slicer 22. The output signal from threshold detector 76 of FIG. 9 is applied as an input signal of each AND gate (A82, B82, C82), so that the three-phase buffers (A80, B80, C80) have the NTSC co-channel interference the data. Only when there is sufficient strength to cause an error in the data slicing performed by slicer 22, a condition is established in the bit line of the output data bus 78 which indicates a low power impedance. The output signal from threshold detector 76 of Fig. 9 is complemented before being applied as an input signal to AND gate O82, so that the three-phase buffer O80 causes the NTSC co-channel interference to be performed by the data slicer 22. Only when there is insufficient strength enough to cause an error in slicing, a condition is established in the bit line of the output data bus 78 which indicates a low power supply impedance.

If the output signal from the threshold detector 76 is logical 1, indicating that the NTSC co-channel interference of Fig. 9 has insufficient strength to cause an error in the data slicing performed by the data slicer 22, How to select among the post coded code decoding result values applied from the post coding comb filters A26, B26, and C26 in FIG. The post coded code decoding result, which deviates mostly from the intermediate code decoding result in the absolute interval, is considered to indicate such an absolute deviation, because the comb filter in suppressing the NTSC co-channel interference artifact component This is because using the post coded code decoding result is more effective than using other post coded code decoding result. Thus, the difference between the intermediate code decoding results is conveyed by the data slicer, and the post coded code decoding result is determined by the post coding comb filter A26, which is determined by the digital subtractors A84, B84 and C84 in Fig. 10B. B26, C26). The absolute values of these difference values are determined by the absolute value circuits A86, B86, C86 and passed back from the intermediate code decoding result passed by the data slicer 22. The post-coded comb filters A26, B26, Determine the absolute deviation of the post coded code decoding result passed from C26). The absolute value circuits A86, B86 and C86 use ROM to achieve faster computational speed than adding bit complement and 1. Using ROM to perform both subtraction and absolute value processing can achieve faster computation speed.

Fig. 10B shows an optimal measurement selection circuit comprising digital comparators A88, B88 and C88, the three-phase buffers O80, A80, B80 and C80 and the AND gates O82, A82, B82 and C82. The digital comparator A88 is an absolute deviation from the intermediate code decoding result of the post coded code decoding result delivered by the post coding comb filter A26, and the post coded code decoding transmitted by the post coding comb filter B26. Determining whether the result is equal to or exceeding the absolute deviation from the intermediate code decoding result, provides a logical one if equal and a logical zero if exceeded The digital comparator B88 is an absolute deviation from the intermediate code decoding result of the post coded code decoding result delivered by the post coding comb filter B26, and the post coded code decoding transmitted by the post coding comb filter C26. Determine if the absolute deviation from the result of the intermediate code decoding result is exceeded, providing logic 1 if exceeded, and logic 0 if not exceeded. The digital comparator C88 is an absolute deviation from the intermediate code decoding result of the post coded code decoding result delivered by the post coding comb filter C26, and the post coded code decoding transmitted by the post coding comb filter A26. Determine if the absolute deviation from the result of the intermediate code decoding result is exceeded, providing logic 1 if exceeded, and logic 0 if not exceeded.

The response of AND gate A82 is logical 1 such that the three-phase buffer A80 forms a condition to assert the post coded code decoding result conveyed by the post coding comb filter A26 on the output data bus 78. In order to ensure that the digital comparator A88 is the absolute deviation from the intermediate code decoding result of the post coded code decoding result delivered by the post coding comb filter A26, the digital coder A88 delivered by the post coding comb filter B26. It must be determined whether a post coded code decoding result is equal to or exceeds an absolute deviation from the intermediate code decoding result, and at the same time the digital comparator C88 is used to determine the result of the post coded code decoding result conveyed by the post coding comb filter C26. Absolute deviation from the intermediate code decoding result It determines that it does not exceed the absolute displacement of the post from the said code of the code decoding intermediate result code decoding results transmitted by the post-coding comb filter A26. The response of AND gate B82 is logical 1 such that the three-phase buffer B80 forms a condition to assert the post coded code decoding result conveyed by the post coding comb filter B26 on the output data bus 78. In order to ensure that the digital comparator B88 is an absolute deviation from the intermediate code decoding result of the post coded code decoding result delivered by the post coding comb filter B26, the digital coder B88 delivered by the post coding comb filter C26. It must be determined whether the absolute deviation of the post coded code decoding result from the intermediate code decoding result is exceeded, and at the same time the digital comparator A88 is the intermediate code of the post coded code decoding result conveyed by the post coding comb filter A26. The absolute deviation from the decoding result is the post coding It determines whether the absolute deviation of the post from the said code of the code decoding intermediate result code decoding result by the transmission filter B26 and sure or exceeds the same. The response of the AND gate C82 is logical 1 such that the three-phase buffer C80 forms a condition to assert the post coded code decoding result carried by the post coding comb filter C26 on the output data bus 78. In order to ensure that the digital comparator C88 is an absolute deviation from the intermediate code decoding result of the post coded code decoding result delivered by the post coding comb filter C26, the digital coder C88 delivered by the post coding comb filter A26. It must be determined whether a post coded code decoding result is equal to or exceeds an absolute deviation from the intermediate code decoding result, and at the same time the digital comparator B88 is used to determine the result of the post coded code decoding result conveyed by the post coded comb filter B26. Absolute deviation from the intermediate code decoding result It determines that it does not exceed the absolute displacement of the post from the said code of the code decoding intermediate result code decoding results transmitted by the post-coding the comb filter C26. Only one comparator of the comparators A88, B88, and C88 (comparator A88 in Fig. 10B) applies logic 1 when the two inputs have the same value, so that among the three-phase data buffers A80, B80, and C80, Neither of these avoids the condition in which the absolute value circuits A86, B86, C86 cannot all drive the bit line of the output data bus 78 from the low power impedance due to the absolute deviation delivered from the same.

As described above through an embodiment of the present invention, the digital TV signal receiver provides a digital TV signal detecting device, M single comb filters to respond to the series of 2N-level codes, and each code decoding measurement result value. A plurality of code decoders for generating a plurality of code decoders, an optimum measurement selection circuit for selecting an optimum value from each code decoding measurement result, a trellis decoder circuit for generating an internal error correction decoding result in response to a final code decoding result, and an internal error correction decoding A Reed-Solomon decoder circuit that generates an external error correction decoding result in response to the bytes of the result, an averager for generating the average of absolute values, a multiplexer connected to generate the final coded decoding result, and a sync code is generated Threshold Detection that Generates Final Code Decoding Result Between Intervals And a squarer for representing deviations between the first and second code decoding measurement results and representing the squared results as the absolute values of these deviations. According to the present invention, the final code decoding result suppresses NTSC co-channel interference rather than a measurement selected from an intermediate code decoding result obtained by data slicing a base code code code without using a comb filter to suppress NTSC co-channel interference. To do this, the comb filtering is performed based on the measurement value selected from the post coded code decoding result obtained. The decision is obtained by comparing each post coded code decoding result corresponding to the intermediate code decoding result over each data segment. From the corresponding value of the intermediate code decoding result, a substantial deviation of the post coded code decoding result is considered to occur, due to the appearance of an NTSC artifact component in the baseband, whereby the post coded code decoding result is intermediate. It is preferentially selected over the decoding result value and included in the final code decoding result value, otherwise it indicates that the selection is wrong.

In the field of digital TV systems applied to digital TV systems used for terrestrial broadcasting in the United States, co-channel interference with analog TV signals of other standards than NTSC such as the PAL standard will emerge. The present invention has a simple design suitable for such co-channel interference, which can be easily modified.

Claims (15)

For digital TV signal receivers: A digital TV signal detection apparatus for providing a series of 2N-level codes each having a code interval of a constant length of time, wherein N represents a positive integer, and the series of 2N-level codes represents a co-channel interfering analog TV signal. Easy to be accompanied by an artifact component of the symbols, the codes are grouped into successive data segments having a header of each data segment sync code, the data segment comprising a data region sync code for converting from a data region to a data region A digital TV signal detection device providing to be grouped into a continuous data area having an initial data segment of each data area; A circuit for providing M single comb filters to respond to the series of 2N-level codes, wherein each single comb filter response is accompanied by less artifact components of the same channel interfering analog TV signal than the series of 2N-level codes. Prone circuits; A plurality of code decoders for generating respective code decoding measurement result values, wherein a first code decoder of the plurality of code decoders responds to the series of 2N-level codes and generates a first code decoder measurement result value; Each responds to one of the M single comb filters respectively, and each code decode measurement result is completely post coded to each matched filtering, from the respective code decode measurement result to the A plurality of code decoders for obtaining a response of each of the single comb filters of M, the other of the plurality of code decoders including a second code decoder to generate a second code decoding result measured; A circuit for detecting whether a deviation exists between the first and second code decoding measurement results; Selecting an optimum value from each of the code decoding measurement results to generate a final code decoding result value between the intervals in which a sync code occurs, and selecting the optimum value from the first code decoding result value of another code decoding measurement result value. An optimum measurement selection circuit that depends on the deviation of the; Digital television signal receiver, characterized in that consisting of. The method of claim 1, The digital TV signal receiver is: A trellis decoder circuit for generating an internal error correction decoding result in response to the final code decoding result; A Reed-Solomon decoder circuit for generating an external error correction decoding result in response to the bytes of the internal error correction decoding result; Digital television signal receiver, characterized in that consisting of. The method of claim 1, A digital TV signal receiver, characterized in that M is one. The method of claim 3, The optimal measurement selection circuit of the digital TV signal receiver is: A multiplexer coupled to provide a selectability between the first and second code decoding measurement results, so as to generate the final code decoding result value in a section in which the sync code occurs; Circuitry for indicating an absolute value for deviation between the first and second code decoding measurement results; An averager for generating an average of the absolute values; In response to the average value of the squared result, exceeding a predetermined threshold, forming a condition for the multiplexer to select the second code decoding measurement result value so that the final code decoding between intervals in which the sync code occurs A threshold detector that generates a result value, and otherwise, causes the multiplexer to form a condition for selecting the first code decoding result value to generate the final code decoding result value between intervals in which the sync code occurs; Digital television signal receiver, characterized in that consisting of. The method of claim 3, The optimal measurement selection circuit of the digital TV signal receiver is: A multiplexer coupled to provide a selectability between the first and second code decoding measurement results, so as to generate the final code decoding result value in a section in which the sync code occurs; A squarer for representing a deviation between the first and second code decoding measurement results and a squared result as an absolute value of these deviations; An averager for generating an average of the squared results; In response to the average value of the squared result, exceeding a predetermined threshold, forming a condition for the multiplexer to select the second code decoding measurement result value so that the final code decoding between intervals in which the sync code occurs A threshold detector that generates a result value, and otherwise, causes the multiplexer to form a condition for selecting the first code decoding result value to generate the final code decoding result value between intervals in which the sync code occurs; Digital television signal receiver, characterized in that consisting of. The method of claim 5, The digital TV signal receiver is: A trellis decoder circuit for generating an internal error correction decoding result in response to the final code decoding result; A Reed-Solomon decoder circuit for generating an external error correction decoding result in response to the bytes of the internal error correction decoding result; Digital television signal receiver, characterized in that consisting of. The method of claim 1, M is at least two, digital television signal receiver. The method of claim 7, wherein The digital TV signal receiver is: Determine whether the co-channel interfering analog TV signal is currently at a level sufficient to cause an error in the first code decoding measurement result, and the co-channel interfering analog TV signal causing an error in the first code decoding measurement result The optimal measurement selection, when indicating that the level is not sufficient, indicating a first indication, and indicating a second indication when the co-channel interfering analog TV signal is determined to be at a level sufficient to cause an error in the first code decoding measurement result. A circuit within the circuit; Circuitry for independently selecting the optimum value from the current first code decoding measurement result value in response to the first indication and generating the last code decoding result value in a section in which the sync code occurs; Digital television signal receiver, characterized in that consisting of. The method of claim 8, The optimal measurement selection circuit of the digital TV signal receiver is: Determining the deviation of the other code decoding measurement result having the maximum absolute value from the first code decoding measurement result value, so that the other code decoding measurement result value is likely to cause an error by at least the artifact component of the co-channel interfering analog TV signal. Circuitry for generating an indication indicative of; The other code decoding measurement result is in response to an indication indicating that an error component is likely to be caused by an artifact component of the co-channel interfering analog TV signal, and when the second indication is provided at the same time, the optimal value is changed to the other code. Circuitry for selecting from a decoding result value, indicating that at least an artifact component of said co-channel interfering analog TV signal is likely to cause an error, and generating said final code decoding result value between intervals in which said sync code occurs; Digital television signal receiver, characterized in that consisting of. The method of claim 7, wherein The digital TV signal receiver is: Circuitry for deriving an intercarrier signal from a heterodyne between the audio and video carrier of the co-channel interfering analog TV signal; The case where the amplitude of such an intercarrier signal exceeds a predetermined level is detected so that the co-channel interfering analog TV signal causes an error in the first code decoding measurement result generated by the first code decoder. An indication indicating that it is sufficient; otherwise, the co-channel interfering analog TV signal is present within the optimal measurement selection circuitry indicating that it is not sufficient to cause an error in the first code decoding measurement result. Circuit; Circuitry for selecting said optimal value independently from said first code decoding measurement result in response to an indication that said co-channel interfering analog TV signal is not sufficient to cause an error in said first code decoding measurement result; Digital television signal receiver, characterized in that consisting of. The method of claim 10, The optimal measurement selection circuit in the digital TV signal receiver is: Determining the deviation of the other code decoding measurement result having the maximum absolute value from the first code decoding measurement result value, so that the other code decoding measurement result value is likely to cause an error by at least the artifact component of the co-channel interfering analog TV signal. Circuitry for generating an indication indicative of; The other code decoding measurement result is in response to an indication indicating that an error component is likely to be caused by an artifact component of the co-channel interfering analog TV signal, and when the second indication is provided at the same time, the optimal value is changed to the other code. Circuitry for selecting from a decoding result value, indicating that at least an artifact component of said co-channel interfering analog TV signal is likely to cause an error, and generating said final code decoding result value between intervals in which said sync code occurs; Digital television signal receiver, characterized in that consisting of. Digital TV Signal Receivers are: A digital TV signal detection apparatus for providing a series of 2N-level codes each having a code interval of a constant length of time, wherein N represents a positive integer, and the series of 2N-level codes represents a co-channel interfering analog TV signal. Easy to be accompanied by an artifact component of the symbols, the codes are grouped into successive data segments having a header of each data segment sync code, the data segment comprising a data region sync code for converting from a data region to a data region A digital TV signal detection device providing to be grouped into a continuous data area having an initial data segment of each data area; A first code decoder for generating a first code decoding measurement result and responding to the series of 2N-level codes; A circuit in which a plurality of single comb filters respond to the series of 2N-level codes, wherein each single comb filter response is less likely to be accompanied by an artifact component of a co-channel interfering analog TV signal than the series of 2N-level codes. Circuitry to provide; Generating a respective code decoding measurement result in response to each of the plurality of single comb filters, wherein each code decoding measurement result is completely post coded to each matched filtering, from the respective code decoding measurement result A respective code decoder to obtain a response of each of the single comb filters; Circuitry for deriving an intercarrier signal from a heterodyne between the audio and video carrier of the co-channel interfering analog TV signal; The case where the amplitude of such an intercarrier signal exceeds a predetermined level is detected so that the co-channel interfering analog TV signal causes an error in the first code decoding measurement result generated by the first code decoder. Circuitry for indicating that it is sufficient and for indicating that said co-channel interfering analog TV signal is not sufficient to cause an error in said first code decoding measurement result; Detect the presence of a synchronization code in the series of 2N-level codes, provide an indication that the synchronization code is in the series of 2N-level codes, and lack the synchronization code in the series of 2N-level codes A circuit providing an indication indicative of; Circuitry for detecting that a synchronization code is present in the series of 2N-level codes and generating an abnormal code decoding result for the synchronization code; Determine that the code decoding measurement result currently has a maximum absolute deviation from the first code decoding measurement result than the first code decoding measurement result, such that the other code decoding measurement result is at least the same channel interference analog TV signal Circuitry for generating an indication indicating that an artifact is likely to be caused by an artifact component of the circuitry; Provide a multiplexer with a final coded decoding result of reproducing one of a plurality of currently selected input signals, the plurality of input signals including the first coded decoding measurement result, the respective other coded decoding measurement result value and the abnormal code A multiplexer for providing a decoding result; The multiplexer further comprises: a multiplexer responsive to an indication that a condition for providing the final code decoding result value, which reproduces the abnormal code decoding result value, is formed, indicating that a synchronization code is present in the series of 2N-level codes; The multiplexer is further configured to provide a condition for providing the final coded decoding result of the reproduction of the first coded decoding result, indicating that there is no synchronization code in the series of 2N-level codes, and the same channel interference. A multiplexer simultaneously providing an indication that a sign decoding analog TV signal is not sufficient to cause an error in said first sign decoding measurement result; The multiplexer is further configured to provide a condition for providing the final coded decoding result of reproducing the other coded decoding measurement result, optionally causing a minimum error by the artifact component of the co-channel interfering analog TV signal, wherein the series of 2N In response to simultaneously providing said indication that a sync code is present in a level code and an indication that said co-channel interfering analog TV signal is sufficient to cause an error in said first code decoding measurement result, said other A multiplexer providing an indication that the sign decoding measurement result is caused by a minimal component by an artifact component of the co-channel interfering analog TV signal; Digital television signal receiver, characterized in that consisting of. The method of claim 12, The digital TV signal receiver is: A trellis decoder circuit for generating an internal error correction decoding result in response to the final code decoding result; A Reed-Solomon decoder circuit for generating an external error correction decoding result in response to the bytes of the internal error correction decoding result; Digital television signal receiver, characterized in that consisting of. The digital TV signal receiver is: A digital TV signal detection apparatus for providing a series of 2N-level codes each having a code interval of a constant length of time, wherein N represents a positive integer, and the series of 2N-level codes represents a co-channel interfering analog TV signal. Easy to be accompanied by an artifact component of the symbols, the codes are grouped into successive data segments having a header of each data segment sync code, the data segment comprising a data region sync code for converting from a data region to a data region A digital TV signal detection device providing to be grouped into a continuous data area having an initial data segment of each data area; A first code decoder for generating a first code decoding measurement result and responding to the series of 2N-level codes; A circuit in which a plurality of single comb filters respond to the series of 2N-level codes, wherein each single comb filter response is less likely to be accompanied by an artifact component of a co-channel interfering analog TV signal than the series of 2N-level codes. Circuitry to provide; A first post-coded comb filter in response to the first comb filter response to produce a second code decode measurement result, and to provide the second code decode measurement result in a filter response matched to the first comb filter response. A second code decoder comprising a; A second post-coded comb filter in response to the second comb filter response to generate a third code decode measurement result, and to provide the third code decode measurement result in a filter response matched to the second comb filter response. A third code decoder comprising a; A third post-coded comb filter in response to the third comb filter response to produce a fourth code decode measurement result, and to provide the fourth code decode measurement result in a filter response matched to the third comb filter response. A fourth code decoder comprising a; Circuitry for deriving an intercarrier signal from a heterodyne between the audio and video carrier of the co-channel interfering analog TV signal; The case where the amplitude of such an intercarrier signal exceeds a predetermined level is detected so that the co-channel interfering analog TV signal causes an error in the first code decoding measurement result generated by the first code decoder. Circuitry for indicating that it is sufficient and for indicating that said co-channel interfering analog TV signal is not sufficient to cause an error in said first code decoding measurement result; Detect the presence of a synchronization code in the series of 2N-level codes, provide an indication that the synchronization code is in the series of 2N-level codes, and lack the synchronization code in the series of 2N-level codes A circuit providing an indication indicative of; Circuitry for detecting that a synchronization code is present in the series of 2N-level codes and generating an abnormal code decoding result for the synchronization code; It is determined whether the second, third, and fourth code decoding measurement result values currently have a maximum deviation absolute value from the first code decoding measurement result value, so that the second, third and fourth code decoding measurement result values are at least equal. Means for generating an indication that an error is likely caused by an artifact component of the channel interference analog TV signal; Provide a multiplexer with a final code decoding result of reproducing one of a plurality of currently selected input signals, wherein the plurality of input signals comprise the first code decoding measurement result, the second code decoding measurement result, and the third code decoding measurement result A multiplexer including the fourth code decoding measurement result value and the abnormal code decoding result value; The multiplexer further comprises: a multiplexer responsive to an indication that a condition for providing the final code decoding result value, which reproduces the abnormal code decoding result value, is formed, indicating that a synchronization code is present in the series of 2N-level codes; The multiplexer is further configured to provide a condition for providing the final coded decoding result of the reproduction of the first coded decoding result, indicating that there is no synchronization code in the series of 2N-level codes, and the same channel interference. A multiplexer responsive to simultaneously providing an indication that a sign decoding analog TV signal is not sufficient to cause an error in said first sign decoding measurement result; The multiplexer is configured to selectively reproduce the second code decoding result and provide the final code decoding result, an indication that a synchronization code is present in the series of 2N-level codes, and the same channel interference An indication that the analog TV signal is at a level sufficient to cause an error in the first code decoding measurement result value and the second code decoding measurement result value cause a minimum error due to an artifact component of the co-channel interfering analog TV signal. A multiplexer responsive to concurrently providing an indication indicating a likeness; The multiplexer is configured to selectively reproduce the third code decoding result value to provide the final code decoding result value, an indication that a synchronization code is present in the series of 2N-level codes, and the same channel interference An indication that the analog TV signal is at a level sufficient to cause an error in the first code decoding measurement result value, and the third code decoding measurement result value cause a minimum error due to an artifact component of the co-channel interfering analog TV signal. A multiplexer responsive to concurrently providing an indication indicating a likeness; The multiplexer is configured to selectively reproduce the fourth code decoding result and provide the final code decoding result, an indication that a synchronization code is present in the series of 2N-level codes, and the same channel interference An indication that the analog TV signal is at a level sufficient to cause an error in the first code decoding measurement result value, and the fourth code decoding measurement result value cause a minimum error by an artifact component of the co-channel interfering analog TV signal. A multiplexer responsive to concurrently providing an indication indicating a likeness; Digital television signal receiver, characterized in that consisting of. The method of claim 14, The digital TV signal receiver is: A trellis decoder circuit for generating an internal error correction decoding result in response to the final code decoding result; A Reed-Solomon decoder circuit for generating an external error correction decoding result in response to the bytes of the internal error correction decoding result; Digital television signal receiver, characterized in that consisting of.