JPWO2014128947A1 - AC motor control device - Google Patents

Abstract

An object of the present invention is to provide a highly accurate method for detecting a current differential value, and to realize position sensorless control of an AC motor using this method. A current detection circuit unit for detecting the current value of the alternating current; an integration circuit to which the alternating current detected by the current detection circuit unit is input; and a motor voltage calculation circuit unit, wherein the integration circuit includes the current detection circuit unit. The difference between the AC current detected in step 1 and the integration output value of the integration circuit is calculated as an integration input value, and the motor voltage calculation circuit unit calculates the voltage phase of the inverter based on the integration input value and the integration output value. AC motor control device.

Description

  The present invention relates to an AC motor control device, and more particularly to position sensorless control of an AC motor.

  Position sensorless control is a technique for driving an AC motor robustly and inexpensively. Position sensorless control is a technique for estimating the rotor phase of an AC motor from voltage or current without directly attaching a position sensor. Since position sensors are vulnerable to high temperature environments and chemical loads, position sensorless control is particularly effective in automotive applications where the temperature is high or compressor applications that are exposed to refrigerant. Further, the size of the control device can be reduced by omitting the position sensor.

  The accuracy of rotor phase estimation during position sensorless control affects efficiency and control response. If the estimation accuracy is low, they may not only decrease but also result in step-out / stop. In order to prevent this, it is important to obtain high estimation accuracy.

  In Patent Document 1, the rotor phase is estimated based on the differential value (current differential value) of the three-phase AC of the AC motor. The current differential value is proportional to the reciprocal of the inductance of the AC motor, and the inductance changes periodically according to the rotor phase. For this reason, there is a correlation between the current differential value and the rotor phase via the inductance, and phase estimation based on this is possible. However, in Patent Document 1, the current differential value is obtained by dividing the current detection value by the detection interval. For this reason, there is a problem that the shorter the detection interval, the more susceptible to noise during current detection during division. Further, if the detection interval is lengthened in order to prevent this, there arises a problem that the response of the control system cannot be improved.

  In Patent Document 2, a current differential value is obtained using a differential circuit composed of an operational amplifier, a capacitor, and a resistor. In this method, since the current differential value can be obtained directly, there is no problem even if the detection interval is shortened. However, since the noise of the differentiation circuit itself is unavoidable, measures such as the use of highly accurate elements or the insertion of low-pass filters are necessary. Since these measures include experimental adjustment factors, there is a problem that the development period is prolonged.

JP 2005-328635 A JP 2002-159199 A

  An object of the present invention is to provide a highly accurate method for detecting a current differential value, and to realize position sensorless control of an AC motor using this method.

  In order to solve the above problems, an AC motor control device according to the present invention is an AC motor control device that controls an inverter that supplies an AC current to a motor, and the current value of the AC current is a current detection circuit unit. An integration circuit to which the alternating current detected by the current detection circuit unit is input and a motor voltage calculation circuit unit, and the integration circuit integrates the alternating current detected by the current detection circuit unit and the integration circuit The difference between the output values is calculated as an integral input value, and the motor voltage calculation circuit unit calculates the voltage phase of the inverter based on the integral input value and the integral output value.

  According to the present invention, the current differential value can be detected with high accuracy. For this reason, the estimation accuracy of the rotor phase can be improved.

1 is a configuration diagram of an AC motor control device according to Embodiment 1. FIG. Voltage / current vector diagram. The transfer function block diagram of an integral input value and an integral output value. The operation | movement waveform diagram in this invention. The block diagram of a motor voltage calculating means. The waveform diagram of inductance L and integral input value Iin with respect to phase. The block diagram of the control apparatus of the alternating current motor in Example 2. FIG. The block diagram of the control apparatus of the alternating current motor in Example 3. FIG. Operation waveform diagram. FIG. 6 is a configuration diagram of Example 5. FIG. 10 is a configuration diagram of Example 6.

  Embodiments of the present invention will be described below with reference to the drawings.

  The control apparatus of Example 1 is demonstrated using FIG. 1 thru | or FIG. FIG. 1 is a configuration diagram of a control device for an AC motor 1 according to the first embodiment of the present invention.

  The AC motor 1 is applied with three-phase AC voltages Vu, Vv, and Vw, so that three-phase AC currents Iu, Iv, and Iw flow and outputs torque τ. The basic characteristics of the AC motor 1 will be described. FIG. 2 shows a vector diagram of voltage and current. The U axis indicates the direction of magnetic flux of the U-phase coil. The d-axis is an axis obtained by rotating the U-axis by the rotor phase θd. The rotation of the AC motor 1 is rotation of the d axis, and the rotation speed, that is, the electrical angular frequency is ω.

  The motor voltage Vuvw is a voltage applied to the AC motor 1, the magnitude of which is the voltage amplitude V1, and the phase based on the U axis or d axis is the U axis voltage phase φu and d axis voltage phase φd, respectively. . A component of the motor voltage Vuvw in the U-axis direction is a U-phase voltage Vu. The same applies to the V phase and W correlation. As the U-axis voltage phase φu of the motor voltage Vuvw advances, a rotating magnetic field is generated, and the rotor phase θd advances so as to follow it.

  The DC power source 2 applies a DC voltage VDC to the inverter 3. The inverter 3 converts the DC voltage VDC into a three-phase AC voltage Vu, Vv, Vw and applies it to the AC motor 1. The current detection circuit unit 4 detects three-phase alternating currents Iu, Iv, and Iw.

  The inverter control circuit unit 5 includes a differential amplifier circuit 51, an integration circuit 52, a motor voltage calculation circuit unit 53, and a PWM signal generation circuit unit 54. Hereinafter, components of the inverter control circuit unit 5 will be described.

  The differential amplifier circuit 51 includes resistors 511 to 514 and an operational amplifier 515. Thereby, the difference between the three-phase alternating currents Iu, Iv, Iw and the integration output value Iout of the integration circuit 52 can be output. This difference is input to the integration circuit 52 as an integration input value Iin. Here, it is desirable that the differential amplifier circuit 51 includes three each corresponding to the three-phase alternating currents Iu, Iv, and Iw (not shown in the drawing). The gain of the differential amplifier circuit 52 can be set by resistors 511 to 514. Here, the gain is set to 1 for simplicity. At this time, the behavior of the differential amplifier circuit 51 is expressed by (Equation 1).

  The integrating circuit 52 includes a resistor 521, a capacitor 522, and an operational amplifier 523. Thereby, the integration circuit 52 can integrate the integration input value Iin and output the integration output value Iout. The integral gain of the integrating circuit 52 is an inverse sign value of the reciprocal of the product RC of the resistance value R of the resistor 521 and the capacitor capacity C of the capacitor 522. Thus, the behavior of the integrating circuit 52 is expressed by (Equation 2).

Where s: Laplace operator

  The physical meaning of the integral input value Iin and the integral output value Iout will be described. A transfer function block diagram obtained from (Equation 1) and (Equation 2) is shown in FIG. FIG. 3 shows a type 1 control system whose behavior is expressed by (Equation 3) obtained from (Equation 1) and (Equation 2).

T: Time constant (T = RC)

  It is known that the type 1 control system is stable regardless of the time constant T. Therefore, the integral output value Iout follows the U-phase current Iu and is constantly equal to the U-phase current Iu. Further, if the integral output value Iout is the U-phase current Iu, the integral input value Iin is an opposite sign value of the differential value of the U-phase current Iu. By using the type 1 control system in this way, the differential operation can be performed only by the integrating circuit 52. Since the operation of the integration circuit 52 is stable, unlike the conventional invention using a division or differentiation circuit, the present invention does not require any special noise countermeasure.

  FIG. 4 shows an operation waveform diagram. Although the U-phase voltage Vu is ideally a sine wave, it is actually a rectangular wave with an amplitude of ± VDC by PWM control of the inverter 3. Although the U-phase current Iu is ideally a sine wave, it is actually a triangular wave because it is affected by the rectangular wave of the U-phase voltage Vu. The lower part of FIG. 4 is an enlarged view of operation waveforms of the integral input value Iin and the integral output value Iout at that time. When PWM control switching occurs, the U-phase current Iu changes sharply. At this time, the integrated output value Iout follows the U-phase current Iu with a time constant T, and becomes equal to the U-phase current Iu when the time constant T has sufficiently passed. Further, the integral input value Iin changes in steps and converges to a constant value with a time constant T. The constant value at this time is the reverse sign value -dIu / dt of the U-phase current differential value dIu / dt.

  The configuration of the motor voltage calculation circuit unit 53 is shown in FIG. The motor voltage calculation circuit unit 53 includes an effective value calculation circuit unit 531, a gain K1 multiplier 532, a gain K2 multiplier 533, and a table conversion unit 534. Thereby, the motor voltage calculation circuit unit 53 calculates the voltage amplitude V1 and the U-axis voltage phase φu based on the integral input value Iin and the integral output Iout. Hereinafter, components of the motor voltage calculation circuit unit 53 will be described.

  The effective value calculation circuit unit 531 calculates the effective value Iout0 of the integrated output value Iout. Since the integrated output value Iout is equal to the U-phase current Iu in the long term, the integrated output effective value Iout0 is equal to the U-phase current effective value Iu0 in FIG.

  The gain K1 multiplier 532 and the gain K2 multiplier 533 calculate the voltage amplitude V1 by (Expression 4) using the positive gains K1 and K2.

However, ω *: Electrical angular frequency command

By (Equation 4), the voltage amplitude V1 can be adjusted in accordance with the increase / decrease of the U-phase current effective value Iu0 and the electrical angular frequency command ω *. The gains K1 and K2 may be variable. Further, the voltage amplitude V1 may be calculated from the characteristic value of the AC motor 1 based on the vector control theory.

  The table conversion means 534 converts the U phase voltage Vu and the integral input value Iin into a phase estimated value θdc. FIG. 6 shows waveform diagrams of the inductance L, the integral input value Iin, and the U-phase voltage Vu with respect to the phase. First, the inductance L is a periodic function of phase. Next, since the integral input value Iin is the reverse sign value of the current differential value, it is expressed by (Equation 5).

(Equation 5) shows that the integral input value Iin is inversely proportional to the inductance L, and the sign is switched in synchronization with the U-phase voltage Vu. In FIG. 6, the envelopes of the integral input Iin when the U-phase voltage Vu is + VDC and −VDC are + Iin and −Iin, respectively. The table conversion means 534 can store the envelopes and estimate the rotor phase θd by referring to the U-phase voltage Vu and the integral input value Iin. The d-axis voltage phase φd is added to the phase estimation value θdc to obtain the U-axis voltage phase φu.

  The above is the description of the components of the motor voltage calculation circuit unit 53.

  The PWM signal generation circuit unit 54 generates the PWM signal PWMuvw according to the voltage amplitude V1 and the U-axis voltage phase φu. Thereby, the switching element of the inverter 3 is PWM-controlled.

  The PWM signal generation circuit unit 54 can also adjust the switching period to be longer than the time constant T in order to improve the detection accuracy of the current differential value. In this case, in FIG. 4, the followability of the integrated output value Iout is ensured, so that a detection error of the current differential value can be prevented.

  The above is the description of the components of the inverter control circuit unit 5.

  In the present invention, the current differential value can be obtained by using a stable type 1 control system using the integrating circuit 52. Since an integration circuit is used, there is an advantage that noise is not generated as in the case of using a differentiation circuit.

  The current detection circuit unit 4 and the differential amplifier circuit 51, the differential amplifier circuit and the integration circuit 52, and the integration circuit 52 and the motor voltage calculation circuit unit 53 need not always be connected. For example, when the inertia of the AC motor 1 is large and the change in the electrical angular frequency ω is slow, the calculation period of the motor voltage calculation circuit unit 53 is set to be long, and it is only necessary to connect only immediately before executing the calculation. That is, the motor voltage calculation circuit unit 53 is started up a predetermined time before the calculation of the voltage phase of the inverter 3 based on the integral input value Iin and the integral output value Iout, and the integration circuit 52 is driven after the calculation is completed. A cut-off mode for cutting off the supply of current is provided, and the cut-off mode is repeated periodically according to the electrical angular frequency.

  As a result, the current flowing through the differential amplifier circuit 51 and the integrating circuit 52 can be minimized. For this reason, the overall efficiency of the AC motor 1 including the loss of the inverter control circuit section 5 can be improved.

  The differential amplifier circuit 51 and the integrating circuit 52 are not necessarily analog circuits. By using a high-speed A / D converter and an FPGA, the differential amplifier circuit 51 and the integration circuit 52 can be configured as high-speed digital circuits. As a result, robustness against noise can be improved.

  FIG. 7 is a configuration diagram of the second embodiment. However, the same points as in the first embodiment are omitted. In the second embodiment, the inverter control circuit unit 5 includes a switching unit 55, and includes a plurality of differential amplifier circuits 51A and 51B, and integrating circuits 52A and 52B. The integral gains of the integrating circuits 52A and 52B are “−1 / (R1 · C1)” and “−1 / (R2 · C2)”, respectively, as in the first embodiment. The time constants are T1 = R1 · C1 and T2 = R2 · C2, respectively. Here, T1> T2.

  The switching unit 55 switches the input destination of the U-phase current Iu based on the PWM carrier frequency or the modulation factor of the PWM signal generation unit 55. It will be described that the detection accuracy of the current differential value can be improved.

  In FIG. 4, when the PWM carrier frequency is high, the period of the triangular wave of the U-phase current Iu is shortened. When the period of the triangular wave is shorter than the time constant T, the followability of the integrating circuit 52 is lowered, and the current differential value detection accuracy is also lowered. Therefore, it is conceivable to improve the follow-up performance of the integrating circuit 52 by reducing the time constant T. However, if the time constant T is made extremely small, the rise of the integral input value Iin becomes steep, which may cause surge voltage or noise. Therefore, the switching means 55 switches the input destination of the U-phase current Iu from the differential amplifier circuit 51A to the differential amplifier circuit 51B only when the PWM carrier frequency is high. The same is true when the modulation rate is low.

  FIG. 8 is a configuration diagram of the third embodiment. However, the same points as in the first embodiment are omitted. In the third embodiment, the AC motor 1 is a switched reluctance motor, and the inverter 3 is an asymmetric half-bridge type. In this case, each phase winding of the AC motor 1 can be energized independently, and can be driven by sequentially switching the energized phases.

  In the present embodiment, the inverter control circuit unit 5 switches the energized phase of the inverter 3 based on the timing when the absolute value of the integral input value Iin becomes equal to or less than a predetermined value. This explains that position sensorless control is realized.

  FIG. 9 shows an operation waveform diagram. The U-phase inductance Lu and the V-phase inductance Lv of the AC motor 1 are periodic functions of the rotor phase θd (the W phase is omitted). Here, when the U-phase current Iu flows through the AC motor 1, the torque τ of (Equation 6) is generated in the U-phase.

(Formula 6) is the same for the V phase or the W phase. As can be seen from (Equation 6), in order to obtain a positive torque τ, the U-phase current Iu needs to flow in a section where the slope of the U-phase inductance Lu is positive, that is, the section Lup in FIG. Conversely, in a section where the slope of the U-phase inductance Lu is negative, that is, a section LuM in FIG. 9, the U-phase current Iu flows and a negative torque τ is generated. Therefore, during power running, it is necessary to shift the energized phase from the U phase to the V phase before reaching the section LuM. The same applies to the transition from the V phase to the W phase and from the W phase to the U phase.

  In position sensorless control, a method for determining the timing for switching the energized phase is important. In the present invention, attention is paid to the behavior of the integral input value Iin in the process from the section LuP to the section LuM. Since the amplitude of the integral input value Iin is the reciprocal of the U-phase inductance Lu as can be seen from (Equation 5), it decreases as the rotor phase θd advances in the interval LuP. Therefore, when the absolute value of the integral input value Iin is equal to or less than the threshold value Iin *, the inverter control circuit unit 5 ends the energization of the U phase. Actually, since there is a delay of the switching element, the energization of the U phase is terminated with a delay of the phase θsw. The threshold value Iin * is obtained by substituting the DC voltage VDC and the U-phase inductance Lu into (Equation 5). After the end of energization of the U phase, energization of the V phase is started with a delay of the phase θuv. The phase θuv is adjusted based on the required torque τ. Position sensorless control is realized by sequentially switching the energized phases in this way.

  The above is the case of power running. At the time of regeneration, since it is sufficient to energize in the interval LuM, if the absolute value | Iin | of the integral input value Iin is equal to or greater than the threshold value Iin *, the energization of the U phase may be terminated. The same applies to the other phases.

  In the present embodiment, position sensorless control is realized by using a timing at which the absolute value | Iin | of the integral input value Iin is equal to or less than the threshold value Iin *. The feature of this embodiment is that it can be easily implemented because it is a simple algorithm.

  The AC motor 1 is not limited to a switched reluctance motor, and the inverter 3 is not limited to an asymmetric half bridge type. The present invention is applicable if each phase winding of AC motor 1 is independent (has no neutral point) and inverter 3 can switch the energized phase.

  FIG. 10 is a configuration diagram of the fifth embodiment. However, the same points as in the first embodiment are omitted. In the fourth embodiment, the differential amplifier circuit 51 and the integrating circuit 52 are modularized, and a type 1 control system module 56 is provided which receives the U-phase current Iu and outputs the integrated input value Iin and the integrated output value Iout.

  It will be described that modularization of the differential amplifier circuit 51 and the integrating circuit 52 is effective. The U-phase current Iu in FIG. 1 is the output of the current detection circuit unit 4 itself. Since this is installed in the vicinity of the inverter 3, it is considered that it is affected by noise generated during switching. The type 1 control system module 56 can output an integrated output value Iout in which noise is suppressed by the type 1 control system of FIG. 3 and can output an integrated input value Iin for estimating the rotor phase θd. That is, the type 1 control system module 56 has two functions of noise suppression and position estimation with only one type 1 control system electric circuit. For this reason, it can contribute to size reduction of the control apparatus of AC motor 1.

  FIG. 11 is a configuration diagram of the sixth embodiment. However, the same points as in the first embodiment are omitted. In the sixth embodiment, the wheel 58 of the vehicle 57 is driven by the AC motor 1.

  In the present invention, the AC motor 1 is subjected to position sensorless control based on the current differential value. The current differential value is determined by the DC voltage VDC and the inductance L as shown in (Equation 5), which can be detected regardless of the speed range. For this reason, it is possible to control the AC motor 1 from the stop time to the high speed range, whereby the wheels 58 can be driven in a wide speed range.

DESCRIPTION OF SYMBOLS 1 ... AC motor 2 ... DC power supply 3 ... Inverter 4 ... Current detection circuit part 5 ... Inverter control circuit part 51, 51A, 51B ... Differential amplifier circuit 511-514 ... Resistor 515 ... Operational amplifier 52, 52A, 52B ... Integration circuit 521... Resistor 522. Capacitor 523... Operational amplifier 53... Motor voltage arithmetic circuit unit 531... Effective value arithmetic circuit unit 532... Gain K1 multiplier 533. Switching means 56 ... 1 type control system module 57 ... vehicle 58 ... wheel
VDC ... DC voltage
Vu, Vv, Vw… U phase voltage, V phase voltage, W phase voltage
V1 ... Motor voltage
Iu, Iv, Iw… U phase current, V phase current, W phase current
Iin ... Integral input value
Iout ... Integral output value
Iout0 ... Integral output effective value φu ... U-axis voltage phase φd ... d-axis voltage phase ω ... Electrical angular frequency ωr * ... Electrical angular frequency command
PWMuvw ... PWM signal θd ... Rotor phase θdc ... Phase estimate τ ... Torque
R ... resistance
C: Capacitor capacity
T ... Time constant
L ... Inductance (Subscript indicates phase)
s… Laplace operator

In order to solve the above-described problem, as an example , an AC motor control device according to the present invention is an AC motor control device that controls an inverter that supplies an AC current to the motor, and detects a current value of the AC current. A circuit unit, an integration circuit to which an alternating current detected by the current detection circuit unit is input, and a motor voltage calculation circuit unit, wherein the integration circuit detects the alternating current detected by the current detection circuit unit and the integration The difference between the integrated output values of the circuit is calculated as an integrated input value, the motor voltage calculating circuit unit calculates the voltage phase of the inverter based on the integrated input value and the integrated output value, and the motor voltage calculating circuit unit Is started before a predetermined time for calculating the voltage phase of the inverter based on the integral input value and the integral output value, and after the calculation is completed, the drive current to the integrating circuit It has a shut-down mode to shut off the supply, periodically repeating the blocking mode in response to an electrical angular frequency.

Claims (7)

An AC motor control device for controlling an inverter that supplies an AC current to a motor,
The current value of the alternating current is a current detection circuit unit,
An integration circuit to which an alternating current detected by the current detection circuit unit is input;
A motor voltage calculation circuit unit,
The integration circuit calculates the difference between the AC current detected by the current detection circuit unit and the integration output value of the integration circuit as an integration input value,
The motor voltage calculation circuit unit is an AC motor control device that calculates the voltage phase of the inverter based on the integral input value and the integral output value. An AC motor control device according to claim 1,
The motor voltage calculation circuit unit is an AC motor control device that controls the voltage amplitude of the inverter based on the integrated output value. A control device for an AC motor according to claim 1 or 2,
The control apparatus of the AC motor which makes the switching period of the said inverter longer than the response time constant of the said integrated output value with respect to the alternating current detected by the said current detection circuit part. A control device for an AC motor according to any one of claims 1 to 3,
The motor voltage calculation circuit section is started before a predetermined time for calculating the voltage phase of the inverter based on the integral input value and the integral output value, and the drive current is supplied to the integration circuit after the calculation is completed. Has a blocking mode to block
An AC motor control device that periodically repeats the cut-off mode according to an electrical angular frequency. A control device for an AC motor according to any one of claims 1 to 4,
The integrating circuit is a control device for an AC motor constituted by a digital circuit. A control device for an AC motor according to any one of claims 1 to 5,
A plurality of the integration circuits are provided,
The integration gains of the plurality of integration circuits are different from each other,
An AC motor control device that switches the integration circuit for inputting the difference between circuits of the plurality of integration circuits based on a carrier frequency or a modulation rate of the inverter. The control device for an AC motor according to any one of claims 1 to 6,
A plurality of windings provided in the motor are independently connected to the inverter, and control of an AC motor that switches an energized phase of the inverter based on a timing at which the absolute value of the integral input value is greater than or less than a predetermined value. apparatus.