JPH01240020A - Simultaneous bidirectional fm transmitter-receiver - Google Patents

Abstract

PURPOSE:To prevent beat generating, to miniaturize a circuit and to save power by forming a first intermediate frequency signal corresponding to the difference of a transmitting/receiving frequency in a first mixer, forming a base band signal to have the phase difference of 90 degrees in a second mixer and demodulating the base band signals by an FM direct converting system. CONSTITUTION:The output of a PLL synthesizer for transmitting PLL2 is connected to the local input terminal of a first mixer 4. Besides, a PLL synthesizer PLL3 of a fixed channel is used as a second local oscillator. Consequently, a first IF frequency becomes 45MHz which is the difference of a transmitting frequency and a receiving frequency. In second misers 6a and 6b, the output signal of the PLL synthesizer for second local oscillation PLL3 and a 45MHz differential signal are frequency-mixed and converted into a base band signal. Consequently, the FM modulating component of a first local oscillating signal is cancelled by the FM modulating component of a second oscillating signal. Thus, spurious emission and the generation of a beat as an unnecessary interfering wave can be prevented, moreover, a circuit scale and the consumption power of a circuit can bed reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION "Field of Industrial Application" The present invention relates to a simultaneous two-way FM transceiver suitable for application to cordless telephones and the like.

"Prior Art" In recent years, cordless telephones, which can be freely carried over short distances such as indoors, are becoming popular.

A cordless telephone transmits the voice signals sent and received from the phone simultaneously between a handset (mobile device) and a base device (fixed device).The base device is fixedly installed indoors and connected to a telephone line. ing.

For example, in Europe, the wireless transmission frequency band used for cordless phones is 95.
9 to 960 MHz, the transmission frequency from the handset is 914
~915MHz. That is, the transmission/reception frequency interval is 45 MHz. Furthermore, 40 channels are allocated at 25 kHz intervals. The MCA (multi-channel access) method is adopted for channel operation, and the base unit has a built-in channel control circuit that automatically selects empty channels and performs simultaneous bidirectional transmission.

FIG. 2 is a block diagram showing the circuit configuration of a wireless section of a conventional handset.

First, the receiving section will be explained.

A received signal input to the antenna terminal ANT is demultiplexed by a duplexer l and input to an RF (high frequency) amplifier 2. The output of the RF amplifier 2 is in the receiving frequency band (9
59 to 960 MHz), and is input to the RF input terminal of the first mixer 4.

The local oscillator input terminal of the first mixer 4 is connected to a receiving P which will be described later.
The output signal of the voltage controlled oscillator (VCO) 12 constituting the LL synthesizer PLL l is input, frequency mixing is performed, and from the output, only a predetermined frequency component is extracted by the 11th F (intermediate frequency) band pass filter 5. The 11th F' signal is obtained. Their frequencies are
For example, if the center frequency of the receiving channel is 959.5M
Hz, the output frequency of VCO12 is 948.8M
Hz, and the IO, MF'th frequency/number of 7 MHz is obtained.

The first IF signal is input to the RF input terminal of the second mixer 6a, and the IO signal, which is the center frequency of the 11F frequency, is input from the second local oscillation circuit 11 to the local input terminal of the second mixer 6a.
7M) Iz crystal oscillation signal is manually input. That is, in the second mixer 6a, the first IP signal and IO, 7MHz
A baseband signal is obtained as a result of mixing with a crystal oscillation signal. Further, the crystal oscillation signal of the second local oscillation circuit 11 is phase-shifted by 90 degrees by a 90-degree phase shifter SHI FT, and is input to the local oscillation input terminal of the second mixer 6b, and then input to the RF input terminal of the second mixer 6b. The 11th F signal is manually operated. Thereby, a baseband signal having a phase different by 90 degrees from the baseband signal output from the second mixer 6a is obtained at the output end of the second mixer 6b. The baseband signals output from the second mixers 6a and 6b are filtered by low-pass filters 7a and 7b, respectively, to remove interfering signals such as adjacent channels and filter only the baseband signal components. The baseband signals thus filtered are input to the detector 9 after being amplified by IP amplifiers 158a and 8b, respectively. Here, the configuration of the detector 9 is shown in FIG. In the figure, DI and D2 are differentiators, M l
, M2 is a multiplier, and A1 is an adder. In the illustrated configuration, the baseband signal supplied to the input terminal has a phase difference of 90 degrees as described above, and therefore is expressed as a cos δω and b sin δω as shown in FIG. Therefore, the output signals of the differentiators DI and D2 are (-aδω sin δω) and (bδω cos δω). As a result, the output signals of the multipliers M 1 and M 2 are (abδω cos ”δω) and (−abδ
ω sin' δω), and when this is added by adder AI, cos' + sin' = l, so
The addition result is abδω, and it can be seen that detection is performed.

The detection output of the wave detector 9 is amplified by a low frequency (AF) amplifier 10 and supplied to the speaker of the slave device.

Next, the transmitter will be explained.

The output signal of the VCO 17 that constitutes the transmitting PLL synthesizer PLL2, which will be described later, is 45M! higher than the receiving frequency! −
1z lower value, for example 914.5MHz, and RFf
It is amplified by a power amplifier 26 and is then manually applied to the duplexer I.

The output power of the RF power amplifier 26 is transmitted to the antenna terminal ANT as a transmission signal by the demultiplexing operation of the duplexer 1 without being transmitted to the receiving circuit section.

Next, the reception PLL synthesizer PLLI and the transmission PLL synthesizer PLL2 will be explained.

The receiving PLL synthesizer PLL 1 includes a VCO 12,
Prescaler 13, variable frequency divider I4, phase comparator 15,
and a low-pass filter 16, and the phase comparator 15 is supplied with a reference frequency signal (frequency fRgr) obtained by dividing the output horn of the reference oscillator 22 by a fundamental frequency divider 23.

The basic frequency r*tv is, for example, 1.5625 kHz, the frequency division ratio P of the prescaler 13 is 16, and the variable frequency divider 1
If the frequency division ratio Nl of 4 is 37952, the output frequency fvco of the VCO 12 is fvco= rRErX PxN
It is locked at a frequency of +-948.8 MHz. For example, by changing the frequency division ratio N1 between 37932 and 37972, the output frequency fv of the VCO 12 can be adjusted.
co varies in 25 kHz steps between 948.3 and 949.3 MHz.

In this way, the receiving PLL synthesizer PLL I can perform its intended function.

The transmitting PLL synthesizer PLL2 includes a VCO 17, a prescaler 18, a variable frequency divider 19, a phase comparator 20, and a low-pass filter 21.
0 includes the above-mentioned receiving PL L synthesizer PLL.
A reference frequency signal used in common with 1 is supplied.

Here, the fundamental frequency fnev is 1.5625 kl-1
z.

When the frequency division ratio P of the prescaler 18 is 16 and the frequency division ratio N of the variable frequency divider 19 is 36580, the output frequency fvco of the VCO 17 is rvco= fRtrX P X N
It is locked at a frequency of t=914.5MHz. In addition, the frequency division ratio N is the frequency division ratio N, (37932 to 3
7972) by (+800-428), that is, between 36560 and 36600, the output frequency r vco of VCOI7 is set to 914 to
It can be varied between 915 MHz in 25 kHz steps. Note that the above value 1800 corresponds to a difference of 45 MHz between the transmitting and receiving frequencies, and the value 428 corresponds to the 11th frequency 10
．． This value corresponds to 7Ml-1z.

The output signal MOD IN of the microphone of the handset (not shown) is supplied to the modulation input terminal MOD of C017 of the transmitting PLL synthesizer PLL2 via the AF amplifier 25, and FM modulation is performed by this modulation input signal MODrH. It is becoming more and more popular.

Note that since the time constant of the low-pass filter 21 is set to be sufficiently larger than the period of the lowest frequency of the modulation signal,
The output center frequency fout of the modulated wave output from VCOl 7 is rou, = fRgpX P X N 2
A modulated FM wave can be obtained while being under PLL control.

Next, the channel control circuit 24 is a circuit that selects an empty channel during transmission and reception, and receives a signal proportional to the received signal level from the IF amplifier 8b, and receives demodulated data from the AF amplifier IO.

The output signal of the channel control circuit 24 is sent to a receiving PLL.
It is connected to the variable frequency divider 14 and the variable frequency divider 19 that determine the frequency (channel) of the synthesizer PLL I, the transmission PL, and the L synthesizer PLL2, and has a frequency division ratio N, -N,
-(1800-428), the channel to be used is selected.

The above explanation regarding FIG. 2 indicates that the receiving frequency band is 959
~960MHz, and the transmission frequency band is 914-91
This is an explanation of the operation of the slave unit, which is 5MHz, but regarding the operation of the base unit, the receiving frequency band and the transmitting frequency band are simply swapped, and the circuit configuration is the same as the second one. Same as the figure.

That is, in a configuration similar to that shown in FIG.
Output frequency of VCOI 2 of synthesizer PLL 1,
That is, the frequency division ratio N of the variable frequency divider 14 that determines the local oscillation frequency of the first mixer 4 is 36132 to 36172.
By changing the receiving frequency between 914 and 915 MI (IO, 7 MHz lower than z), the local oscillation frequency of the variable frequency divider 19 that determines the output frequency of the VCo 18 of the transmitting PLL synthesizer PLL2 , 383
Between 60 and 38400, N t = N I+
While maintaining the relationship of (1800+428), it is changed by the channel control circuit 24 to control the selection of the operating channel.

As a result, the base unit can receive frequency bands from 914 to 91.
At 5 MHz, the transmission frequency band is 959 to 960 MHz, and simultaneous bidirectional transmission is performed with the handset.

"Problem to be Solved by the Invention" By the way, the above-mentioned conventional device had the following drawbacks.

■ Since the first local oscillation frequency of the receiving synthesizer PLL I is a frequency component close to the transmitting and receiving frequency, spurious emissions as unnecessary interference waves occur.

■ Two PLL synthesizer PLLs for transmitting and receiving
1. Since PLL2 is required, the circuit scale is large (two prescalers, variable frequency dividers, etc. are required), and the circuit current consumption is accordingly large.

(2) When the transmission frequency component mixes into the first mixer 4 of the own machine, a beat is generated due to high-order distortion between the transmission frequency component and the first local oscillation frequency component. In order to reduce this beat, a high isolation value is required between the transmitting terminal and receiving terminal of the duplexer I. For this reason, the duplexer l becomes large, and the insertion loss between the antenna terminal ANT and the transmitting terminal and between the antenna terminal ANT and the receiving terminal becomes large.

The present invention was made against this background, and an object of the present invention is to provide a simultaneous bidirectional FM transceiver that solves the above problems.

"Means for Solving the Problems" In order to solve the above problems, the present invention provides a first PLL synthesizer that outputs a first modulated wave that is frequency-modulated with a low frequency signal to be transmitted; a second PLL synthesizer that outputs a second modulated wave that is frequency-modulated with a frequency signal; and a part of the first modulated wave and the high-frequency received signal are manually output to output a first intermediate frequency signal. a first mixer that mixes the first intermediate frequency signal and the second modulated wave, and generates a 90 degree phase shifted wave of the first intermediate frequency signal and the second modulated wave. and a second mixer capable of mixing the first modulated wave with the second mixer, and transmits the first modulated wave and demodulates the low frequency received signal from the output signal of the second mixer by an FM direct conversion method. Features.

"Operation" According to the above configuration, the received signal and the first modulated wave are mixed in the first mixer, and the first intermediate frequency signal corresponding to the difference between the transmission and reception frequencies is obtained. Further, in the second mixer, the second modulated wave and the first intermediate frequency signal, and the 90 degree phase shifted signal of the second modulated wave and the first intermediate frequency signal are mixed, respectively. A baseband signal with a variable phase difference is obtained. Then, these baseband signals are demodulated using the FM direct conversion method.

In this way, since the first modulated signal corresponding to the transmission signal is used as the first local oscillation signal, no beat occurs between the transmission signal and the first local oscillation signal. Therefore, the duplexer can be made smaller.

Furthermore, the second PLL synthesizer that generates the second modulated wave (second local oscillation signal) does not require a variable frequency divider,
And because it operates at a relatively low frequency (45MHz),
It is possible to achieve miniaturization and power saving.

Furthermore, the frequency of the second PLL synthesizer is
Since it is far away from the transmitting and receiving frequencies and has a relatively low frequency, there is little spurious emission as unnecessary interference waves.

Furthermore, since the first modulated wave and the second modulated wave are FM-modulated by the low-frequency signal to be transmitted, the FM by the low-frequency signal included in the first intermediate frequency signal is
The frequency shift amount cancels out the FM frequency shift amount of the second modulated wave in the second mixer. Therefore, the amount of frequency deviation of the second intermediate frequency signal remains unchanged from the conventional one, and it is possible to avoid the adverse effects caused by an increase in the amount of frequency deviation (details of this will be described later).

Embodiment FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention.

In this figure, the same parts as in the conventional example shown in FIG. 2 are denoted by the same reference numerals, and the explanation thereof will be omitted.

The main difference between this embodiment and the conventional example is that the components 17 to
21 is connected to the local oscillator input terminal of the first mixer 4, and a fixed channel PLL synthesizer PLL3 is used as the second local oscillator instead of a crystal oscillation circuit. This is what I used.

Therefore, the first IP frequency is 45 MHz, which is the difference between the transmitting frequency and the receiving frequency. For this reason, a band-pass filter that passes the 45 MHz band is used as the band-pass filter 5 after the first mixer 4. The first IF signal output from the bandpass filter 5 is input to second mixers 6a and 6b. In the second mixer 6a, a second local oscillation PLL synthesizer P
The output signal of LL3 and the above 45MHz difference signal are frequency mixed and converted into a baseband signal, and the second mixing ag
6b, the second local oscillation PLL synthesizer P
The output signal of LL3 and the 45 MHz difference signal are frequency mixed and converted into a baseband signal.

The second local oscillation PLL synthesizer PLL3 has V
It is composed of COI02, fixed frequency divider 103 (frequency division ratio R), phase comparator 104, and low-pass filter +05, and the reference frequency is obtained from the reference oscillator 22 and reference frequency divider 23, which are common to the transmitting PLL synthesizer PLL 2. signal(
1,5625kHz signal).

Here, for example, the frequency division ratio R of the fixed frequency divider 103 is set to 28
800, the output frequency fvco of the VCO 102 is −fvco” f*irX R =1.5625x28800 =45MI(z). That is, it is locked at the center frequency of the 11th F frequency.

In addition, the first local oscillator signal (the modulated wave of ■) and the second local oscillator signal
The local oscillator signal (second modulated wave) is FM modulated by the low frequency signal to be transmitted, but the FM modulation component contained in the first local oscillator signal is included in the second local oscillator signal. This is canceled out by the FM modulation component, so that the influence of FM modulation by the low frequency signal does not appear on the second IF signal.

That is, the low frequency signal to be transmitted, which is supplied to VCOI7 (tth VCO) (7) modulation input terminal MOD, is
It is also input to the modulation input terminal MOD of the CO 102, and the V
The COI02 is FM modulated to cancel out the FM frequency shift due to the low frequency signal to be transmitted.

Note that the time constant of the low-pass filter 105 is set to be sufficiently larger than the period of the lowest frequency of the modulation signal.
ul is PLL-controlled to f (Jul = fREFX R) so that an FM modulated wave can be obtained.

According to such a configuration, the maximum frequency deviation amount of the received signal in the first IF signal (45 MHz) output from the first mixer 4 is the FM frequency deviation amount of the received signal and the FM frequency deviation of the self-transmitted signal. This is the sum of the amount of frequency deviation, and the deviation m is twice that of the conventional example.

However, as mentioned above, the second local oscillator signal is
Since the frequency deviation included in the 11th F signal is canceled by M modulation, the maximum frequency deviation of the received signal in the baseband signal is 1 by a large amount of the FM frequency deviation of the received signal, as in the conventional example. This will happen.

Therefore, it is not necessary to widen the passbands of the low-pass filters 7a and 7b, and the same passband as in the conventional example can be used, making it possible to perform demodulation without deteriorating reception performance characteristics.

There is also a conventional method (not shown) in which a fixed oscillator without FM modulation is used as the second local oscillator, and the own modulation signal is added to the FM demodulated signal with the opposite polarity.In this method, the passband of the low-pass filter is It is necessary to widen the band to twice the amount of FM frequency deviation, and as a result, reception performance such as adjacent channel selectivity deteriorates.
The present invention can solve such problems.

"Effects of the Invention" As explained above, the present invention uses the first modulated wave corresponding to the transmission signal as the first local oscillator signal on the receiving side, and also uses the first modulated wave corresponding to the transmission signal to receive similar modulation by the transmission signal. Since the second modulated wave is used as the second local oscillator signal and the above modulation by the transmission signal cancels each other out, the following effects can be achieved.

1. Since the frequency of the second local oscillation PLL synthesizer is far from the transmitting and receiving frequency and is a low frequency component,
Spurious emissions as unnecessary interference waves are reduced.

2. Only one PLL synthesizer for radio frequency with a variable frequency divider is required for transmission, and the second local oscillation PLL synthesizer has a low fixed frequency, so the circuit size is smaller than that of the conventional one. Also, circuit/l! The current cost is small.

3. Since the transmission frequency component and the first local oscillation frequency component of the own device are the same, beats due to high-order distortion do not occur as in conventional examples, and the required isolation between the transmission terminal and reception terminal of the duplexer is It can be small, making the duplexer smaller and reducing insertion loss.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention.
FIG. 2 is a block diagram showing the circuit configuration of a radio section of a handset of a conventional cordless telephone, and FIG. 3 is a block diagram showing the configuration of the detector shown in FIG. 3. 4...First mixer, 6U, 6b...
2nd mixer, 9...detector (detector using FM direct conversion method), PLL2... transmitting PLL synthesizer (first PLL synthesizer), PLL3...
...Second local oscillation PLL synthesizer (second PI, L synthesizer).

Claims (1)

[Claims] A first PLL synthesizer that outputs a first modulated wave that is frequency-modulated with a low-frequency signal to be transmitted; and a second modulated wave that is frequency-modulated with the low-frequency signal. a second PLL synthesizer that inputs a portion of the first modulated wave and a high frequency reception signal and outputs a first intermediate frequency signal; and the second modulated wave, and a second mixer that mixes the first intermediate frequency signal and the 90 degree phase shifted wave of the second modulated wave, 1. A simultaneous bidirectional FM transceiver, characterized in that it transmits one modulated wave and demodulates a low frequency received signal from the output signal of the second mixer by an FM direct conversion method.