JP7457289B2 - radar equipment - Google Patents

Description

The present disclosure relates to a radar device.

In recent years, radar devices that use short-wavelength radar transmission signals, including microwaves and millimeter waves, which provide high resolution, have been studied. In addition, to improve outdoor safety, there is a demand for the development of radar devices that can detect objects (targets) including pedestrians in a wide-angle range in addition to vehicles.

For example, a pulse radar device that repeatedly transmits pulse waves is known as a radar device. The received signal of a wide-angle pulse radar that detects vehicles/pedestrians in a wide-angle range is a signal that is a mixture of multiple reflected waves from nearby targets (e.g., vehicles) and distant targets (e.g., pedestrians). For this reason, (1) the radar transmitter is required to be configured to transmit pulse waves or pulse-modulated waves with autocorrelation characteristics that result in low range side lobes (hereinafter referred to as low range side lobe characteristics), and (2) the radar receiver is required to be configured to have a wide reception dynamic range.

There are two possible configurations for a wide-angle radar device:

The first method is to transmit radar waves by scanning pulse waves or modulated waves mechanically or electronically using a directional beam with a narrow angle (beam width of several degrees). The configuration is such that the reflected waves are received using the With this configuration, many scans are required to obtain high resolution, which deteriorates the ability to follow a target that moves at high speed.

The second method is to receive reflected waves using an array antenna consisting of multiple antennas (antenna elements), and estimate the arrival angle of the reflected waves using a signal processing algorithm based on the reception phase difference with respect to the antenna spacing. (DOA) estimation). In this configuration, even if the scanning interval of the transmitted beam in the radar transmitter is thinned out, the angle of arrival can be estimated in the radar receiver, so the scanning time can be shortened, and the tracking performance is better than in the first configuration. will improve. For example, direction of arrival estimation methods include Fourier transform based on matrix operations, Capon method and LP (Linear Prediction) method based on inverse matrix operations, or MUSIC (Multiple Signal Classification) and ESPRIT (Estimation of Signal Parameters) based on eigenvalue operations. via Rotational Invariance Techniques).

In addition, as a radar device, in addition to the radar receiving section, the radar transmitting section is also equipped with multiple antennas (array antennas), and beam scanning is performed by signal processing using the transmitting and receiving array antenna (sometimes called a MIMO radar). has been proposed (for example, see Non-Patent Document 2).

Special Publication No. 2011-526370

Budisin, S.Z., "New complementary pairs of sequences," Electron. Lett., 1990, 26, (13), pp.881-883 Jian Li, Stoica, Petre, "MIMO Radar with Colocated Antennas," Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007

Incidentally, in order to increase the directivity gain of the array antenna, a sub-array antenna may be used in which each antenna element (hereinafter referred to as an array element) constituting the array antenna is further composed of a plurality of antenna elements.

It is difficult to arrange the elements of the array antenna at intervals narrower than the size of the array elements. However, when using a sub-array antenna configuration, the size of the array element increases, so it is necessary to widen the spacing between the sub-array antennas, and there is a possibility that grating lobes will occur on the directivity pattern of the array antenna.

One aspect of the present disclosure provides a radar device that can suppress the occurrence of unnecessary grating lobes and achieve a desired directivity pattern, even in the case of a sub-array antenna configuration.

A radar device according to an aspect of the present disclosure includes a transmitting array antenna, a receiving array antenna, a radar transmitter that transmits a radar signal using the transmitting array antenna, and a reflected wave signal obtained by reflecting the radar signal on a target. a radar receiving unit that receives using the receiving array antenna, the transmitting array antenna including a plurality of transmitting antennas, the receiving array antenna including a plurality of receiving antennas, and the plurality of transmitting antennas. are arranged at different positions in a first direction, the plurality of receiving antennas are arranged at different positions in the first direction, and the interval between two adjacent transmitting antennas among the plurality of transmitting antennas is The distance between two adjacent receiving antennas among the plurality of receiving antennas is one wavelength or more in the first direction, and the distance between the two adjacent transmitting antennas is one wavelength or more in the first direction. The absolute value of the difference between the antenna spacing and the spacing between the two adjacent receiving antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction, and at least one of the receiving antennas includes a plurality of antenna elements arranged in the first direction.
A radar device according to another aspect of the present disclosure includes a first array antenna, a second array antenna, and transmits a radar signal using one of the first array antenna and the second array antenna. a radar transmitting section; a radar receiving section that receives a reflected wave signal obtained by reflecting the radar signal from a target using the other of the first array antenna and the second array antenna; The array antenna includes a plurality of first antennas, each of the plurality of first antennas is arranged at a different position in a first direction, and the second array antenna includes a plurality of second antennas. each of the plurality of second antennas is arranged at a different position in the first direction, and the interval between two adjacent first antennas among the plurality of first antennas is equal to the distance between the first antennas. The distance between two adjacent second antennas among the plurality of second antennas is one wavelength or more in the first direction, and the distance between two adjacent second antennas is one wavelength or more in the first direction. The absolute value of the difference between the distance between the first antenna and the distance between the two adjacent second antennas, in the first direction, is 0.5 wavelength or more and 0.75 wavelength or less, At least one of the plurality of first antennas and the plurality of second antennas includes a plurality of antenna elements arranged in the first direction.

Note that these comprehensive or specific aspects may be realized by a system, a method, an integrated circuit, a computer program, or a recording medium, and any of the systems, devices, methods, integrated circuits, computer programs, and recording media may be implemented. It may be realized by any combination.

According to one aspect of the present disclosure, even in the case of a sub-array antenna configuration, it is possible to suppress the occurrence of unnecessary grating lobes and achieve a desired directivity pattern.

Further advantages and advantages of one aspect of the disclosure will become apparent from the specification and drawings. Such advantages and/or effects may be provided by each of the several embodiments and features described in the specification and drawings, but not necessarily all are provided in order to obtain one or more of the same features. There isn't.

Diagram showing a configuration example of a subarray element Diagram showing an example of the configuration of an array antenna consisting of sub-array elements A block diagram showing the configuration of a radar device according to an embodiment of the present disclosure A diagram showing an example of a radar transmission signal according to an embodiment of the present disclosure FIG. 13 is a block diagram showing another configuration of the radar transmission signal generator according to the embodiment of the present disclosure. A diagram showing an example of transmission timing and measurement range of a radar transmission signal according to an embodiment of the present disclosure A diagram showing antenna arrangements of a transmitting array, a receiving array, and a virtual receiving array according to an embodiment of the present disclosure. A diagram showing a directivity pattern according to an embodiment of the present disclosure A diagram showing antenna arrangements of a transmitting array, a receiving array, and a virtual receiving array according to variation 1 of an embodiment of the present disclosure. FIG. 13 is a diagram showing a horizontal directivity pattern according to a first variation of an embodiment of the present disclosure; A diagram showing a vertical directivity pattern according to variation 1 of an embodiment of the present disclosure A diagram showing antenna arrangements of a transmitting array, a receiving array, and a virtual receiving array according to variation 2 of an embodiment of the present disclosure. A diagram showing a horizontal directivity pattern according to variation 2 of an embodiment of the present disclosure A diagram showing a vertical directivity pattern according to variation 2 of an embodiment of the present disclosure A diagram showing antenna arrangements of a transmitting array, a receiving array, and a virtual receiving array according to variation 3 of an embodiment of the present disclosure. A diagram showing a horizontal directivity pattern according to variation 3 of an embodiment of the present disclosure A diagram showing an example diagram showing a vertical directivity pattern according to variation 3 of an embodiment of the present disclosure

[How this disclosure came to be made]
FIG. 1A shows an example of an antenna element having a subarray configuration (hereinafter also referred to as a subarray element). The sub-array element shown in FIG. 1A is composed of four 2×2 antenna elements. Furthermore, in the example shown in FIG. 1A, the size of the subarray element is 0.8 wavelength in both the horizontal and vertical directions.

Figure 1B shows an example of an array antenna consisting of four subarray elements, each of which is shown in Figure 1A, arranged in series. As shown in Figure 1B, the size of each subarray element is 0.8 wavelengths (see Figure 1A), so the spacing between the subarray elements must be at least one wavelength.

For example, the array element spacing (desired element spacing) for preventing grating lobes from occurring within ±90° of the main lobe is 0.5 wavelengths. In the array antenna shown in FIG. 1B, the element spacing of the subarray elements is approximately one wavelength or more, making it difficult to set the desired element spacing, and grating lobes will occur within ±90° of the main lobe.

As described above, when the size of the subarray element is 0.5 wavelength or more, it may be difficult to set the element interval of the array antenna to 0.5 wavelength. Therefore, unnecessary grating lobes are generated within the range of ±90° of the main lobe, and a virtual image is generated during angle measurement, causing false detection.

Here, Patent Document 1 discloses an array antenna configuration using sub-array elements with a width d=about 1 wavelength. In Patent Document 1, the element spacing of transmitting antennas Tx0 and Tx1 is 6 wavelengths, and the element spacing of receiving antennas RX0, RX1, RX2, and RX3 is 1.5 wavelengths ± (λ/8) (λ represents 1 wavelength. ). Furthermore, in Patent Document 1, a radar transmission signal is transmitted by switching the transmission antennas Tx0 and Tx1 in a time division manner, and for the radar transmission signal transmitted from each transmission antenna Tx0 and Tx1, the reception antenna RX0, RX1, RX2, It has a configuration that acquires the received signal with RX3.

With such a configuration, a phase change due to a change in the position of the transmitting antenna is superimposed on the received signal acquired by the receiving array antenna, so that an effect of virtually increasing the aperture length of the receiving antenna can be obtained. Hereinafter, a virtual receiving array antenna whose effective aperture length is increased by the arrangement of antenna elements in the transmitting/receiving array antenna will be referred to as a "virtual receiving array."

However, in Patent Document 1, the element spacing of the receiving array antenna is 1.5 wavelengths ± λ/8, so grating lobes occur in a direction shifted by about 40° from the main beam direction.

One aspect of the present disclosure suppresses the generation of unnecessary grating lobes and achieves a desired directivity pattern even when using array elements in a subarray configuration.

Hereinafter, embodiments according to one aspect of the present disclosure will be described in detail with reference to the drawings. Note that in the embodiments, the same components are given the same reference numerals, and the description thereof will be omitted since it is redundant.

[Configuration of radar device]
FIG. 2 is a block diagram showing the configuration of radar device 10 according to this embodiment.

The radar device 10 includes a radar transmitter 100, a radar receiver 200, and a reference signal generator 300.

The radar transmitter 100 generates a high-frequency radar signal (radar transmission signal) based on the reference signal received from the reference signal generator 300. The radar transmitter 100 then transmits the radar transmission signal at a predetermined transmission period using a transmission array antenna composed of multiple transmission antennas 106-1 to 106-Nt.

The radar receiver 200 receives a reflected wave signal, which is a radar transmission signal reflected by a target (not shown), using a receiving array antenna consisting of multiple receiving antennas 202-1 to 202-Na. The radar receiver 200 processes the reflected wave signal received by each antenna 202 using a reference signal received from a reference signal generator 300, and detects the presence or absence of a target and estimates its direction. The target is an object to be detected by the radar device 10, and includes, for example, a vehicle or a person.

The reference signal generation section 300 is connected to each of the radar transmission section 100 and the radar reception section 200. The reference signal generation section 300 commonly supplies a reference signal as a reference signal to the radar transmission section 100 and the radar reception section 200, and synchronizes the processing of the radar transmission section 100 and the radar reception section 200.

[Configuration of radar transmitter 100]
Radar transmitting section 100 includes radar transmitting signal generating sections 101-1 to 101-Nt, transmitting radio sections 105-1 to 105-Nt, and transmitting antennas 106-1 to 106-Nt. That is, the radar transmitting section 100 has Nt transmitting antennas 106, and each transmitting antenna 106 is connected to an individual radar transmitting signal generating section 101 and a transmitting radio section 105, respectively.

The radar transmission signal generator 101 generates a timing clock by multiplying the reference signal received from the reference signal generator 300 by a predetermined number, and generates a radar transmission signal based on the generated timing clock. The radar transmission signal generator 101 then repeatedly outputs the radar transmission signal at a predetermined radar transmission period (Tr). The radar transmission signal is expressed as _rz (k,M)= _Iz (k,M)+ _jQz (k,M), where z represents a number corresponding to each transmitting antenna 106, and z=1,...,Nt. In addition, j represents an imaginary unit, k represents a discrete time, and M represents the ordinal number of the radar transmission period.

Each radar transmission signal generation section 101 includes a code generation section 102, a modulation section 103, and an LPF (Low Pass Filter) 104. Each component in the radar transmission signal generation section 101-z corresponding to the z-th (z=1, . . . , Nt) transmission antenna 106 will be described below.

Specifically, the code generation unit 102 generates a code a(z) _n (n=1, . . . , L) (pulse code) of a code sequence with a code length L for each radar transmission period Tr. Codes a(z) _n (z=1, . . . , Nt) generated in each code generation unit 102-1 to 102-Nt are codes that have low correlation or no correlation with each other. Examples of code sequences include Walsh-Hadamard codes, M-sequence codes, and Gold codes.

The modulation unit 103 performs pulse modulation (amplitude modulation, ASK (Amplitude Shift Keying), pulse shift keying) or phase modulation (Phase Shift Keying) on the code a(z) _n received from the code generation unit 102 to generate a modulated signal. is output to the LPF 104.

LPF 104 outputs signal components below a predetermined limited band of the modulated signal received from modulation section 103 to transmission radio section 105 as a baseband radar transmission signal.

The z-th (z=1,...,Nt)-th transmission radio section 105 performs frequency conversion on the baseband radar transmission signal output from the z-th radar transmission signal generation section 101 to obtain a carrier frequency ( A radar transmission signal in the Radio Frequency (RF) band is generated, amplified to a predetermined transmission power P [dB] by a transmission amplifier, and output to the z-th transmission antenna 106.

The zth (z=1, ..., Nt) transmitting antenna 106 radiates the radar transmission signal output from the zth radio transmission unit 105 into space.

FIG. 3 shows radar transmission signals transmitted from the Nt transmitting antennas 106 of the radar transmitter 100. A pulse code sequence of code length L is included in the code transmission period Tw. In each radar transmission period Tr, a pulse code sequence is transmitted during a code transmission period Tw, and the remaining period (Tr-Tw) is a no-signal period. By performing pulse modulation using No samples per pulse code (a(z) _n ), a signal of Nr (=No×L) samples is generated within each code transmission period Tw. is included. That is, the sampling rate in modulation section 103 is (No×L)/Tw. Further, it is assumed that the no-signal period (Tr-Tw) includes Nu samples.

Note that the radar transmitting section 100 may include a radar transmitting signal generating section 101a shown in FIG. 4 instead of the radar transmitting signal generating section 101. The radar transmission signal generation section 101a does not have the code generation section 102, modulation section 103, and LPF 104 shown in FIG. 2, but instead includes a code storage section 111 and a DA conversion section 112. The code storage unit 111 stores in advance the code sequences generated by the code generation unit 102 (FIG. 2), and sequentially reads out the stored code sequences cyclically. The DA conversion unit 112 converts the code series (digital signal) output from the code storage unit 111 into an analog signal.

[Configuration of radar receiving section 200]
In FIG. 2, the radar receiving unit 200 includes Na receiving antennas 202, forming an array antenna. Further, the radar receiving section 200 includes Na antenna system processing sections 201-1 to 201-Na and a direction estimation section 214.

Each receiving antenna 202 receives a reflected wave signal that is a radar transmission signal reflected by a target (object), and outputs the received reflected wave signal to the corresponding antenna system processing section 201 as a received signal.

Each antenna system processing section 201 includes a reception radio section 203 and a signal processing section 207.

The reception radio section 203 includes an amplification section 204, a frequency converter 205, and a quadrature detector 206. The reception radio section 203 generates a timing clock obtained by multiplying the reference signal received from the reference signal generation section 300 by a predetermined number, and operates based on the generated timing clock. Specifically, the amplifier 204 amplifies the received signal received from the receiving antenna 202 to a predetermined level, the frequency converter 205 converts the frequency of the received signal in the high frequency band to the baseband band, and the quadrature detector 206 amplifies the received signal in the baseband band. A received band signal is converted into a received baseband signal including an I signal and a Q signal.

The signal processing unit 207 has AD conversion units 208 and 209 and separation units 210-1 to 210-Nt.

The I signal from the quadrature detector 206 is input to the AD converter 208 , and the Q signal from the quadrature detector 206 is input to the AD converter 209 . The AD converter 208 converts the I signal into digital data by sampling the baseband signal including the I signal in discrete time. The AD converter 209 performs discrete time sampling on the baseband signal including the Q signal, thereby converting the Q signal into digital data.

Here, in the sampling of the AD converters 208 and 209, Ns discrete samples are performed per time Tp (=Tw/L) of one subpulse in the radar transmission signal. That is, the number of oversamples per subpulse is Ns.

In the following explanation, the I signal Ir(k, M) and the Q signal Qr(k, M) are used to describe the discrete time of the M-th radar transmission period Tr[M] as the output of the AD converters 208 and 209. The baseband received signal at k is expressed as a complex signal x(k, M)=Ir(k, M)+jQr(k, M). In addition, in the following, the discrete time k is based on the timing at which the radar transmission cycle (Tr) starts (k=1), and the signal processing unit 207 uses the sample point k before the radar transmission cycle (Tr) ends. Operates periodically until = (Nr + Nu)Ns/No. That is, k=1,...,(Nr+Nu)Ns/No. Here, j is an imaginary unit.

The signal processing unit 207 includes Nt separators 210, the number of which is equal to the number of systems of the transmitting antennas 106. Each separator 210 has a correlation calculation unit 211, an adder 212, and a Doppler frequency analysis unit 213. The configuration of the zth (z=1, ..., Nt) separator 210 will be described below.

The correlation calculation unit 211 receives discrete sample values x(k, M) including the discrete sample values Ir(k, M) and Qr(k, M) received from the AD conversion units 208 and 209 for each radar transmission period Tr, A correlation calculation is performed with a pulse code a(z) _n of code length L (where z=1,...,Nt, n=1,...,L) transmitted by the radar transmitter 100. For example, the correlation calculation unit 211 performs a sliding correlation calculation between the discrete sample value x(k, M) and the pulse code a(z) _n . For example, the correlation calculation value AC _(z) (k, M) of the sliding correlation calculation at discrete time k in the M-th radar transmission period Tr[M] is calculated based on the following equation.

In the above formula, the asterisk (*) represents a complex conjugate operator.

The correlation calculation unit 211 performs correlation calculation over a period of k=1, . . . , (Nr+Nu)Ns/No, for example, according to equation (1).

Note that the correlation calculation unit 211 is not limited to the case where correlation calculation is performed for k=1, ..., (Nr+Nu)Ns/No, but can perform measurement according to the existence range of the target to be measured by the radar device 10. The range (ie, the range of k) may be limited. Thereby, in the radar device 10, it is possible to reduce the amount of calculation processing performed by the correlation calculation section 211. For example, the correlation calculation unit 211 may limit the measurement range to k=Ns(L+1),...,(Nr+Nu)Ns/No-NsL. In this case, as shown in FIG. 5, the radar device 10 does not perform measurement in the time interval corresponding to the code transmission interval Tw.

As a result, in the radar device 10, even if the radar transmission signal goes around directly to the radar reception unit 200, the processing by the correlation calculation unit 211 is not performed during the period during which the radar transmission signal goes around (at least a period less than τ1). Therefore, it is possible to perform measurements that eliminate the effects of wraparound. In addition, when limiting the measurement range (range of k), the measurement range (range of k) is similarly limited for the processing of the addition unit 212, Doppler frequency analysis unit 213, and direction estimation unit 214, which will be described below. All you have to do is apply the processing that was done. Thereby, the amount of processing in each component can be reduced, and the power consumption in the radar receiving section 200 can be reduced.

The adding unit 212 performs radar transmission a predetermined number of times (Np times) using the correlation calculation value AC _(z) (k, M) received from the correlation calculation unit 211 at every discrete time k of the M-th radar transmission period Tr. The correlation calculation value AC _(z) (k, M) is added (coherent integration) over a period of period Tr (Tr×Np). Addition (coherent integration) processing of the number of additions Np over a period (Tr×Np) is expressed by the following equation.

Here, CI _(z) (k, m) represents an added value of correlation calculation values (hereinafter also referred to as a correlation added value), Np is an integer value of 1 or more, and m is an integer value of 1 or more. It is an integer of 1 or more indicating the ordinal number of the number of additions when the number of additions Np is one unit. Also, z=1,...,Nt.

The adder 212 performs addition Np times, with the output of the correlation calculation unit 211 obtained in units of the radar transmission period Tr as one unit. In other words, the adder 212 calculates the correlation value CI _(z) (k,m) for each discrete time k by adding the correlation calculation values AC ₍ z)(k,Np(m-1)+1) to AC _(z) (k,Np×m) at the same timing for the discrete time k as one unit. As a result, the adder 212 can improve the SNR of the reflected wave signal in a range where the reflected wave signal from the target has a high correlation due to the effect of adding the correlation calculation values Np times. Therefore, the measurement performance for estimating the arrival distance of the target can be improved.

Note that in order to obtain an ideal addition gain, a condition is required in which the phase components of the correlation calculation values are aligned within a certain range in the addition interval of Np times of addition of the correlation calculation values. That is, the number of additions Np is preferably set based on the assumed maximum moving speed of the target to be measured. This is because the higher the assumed maximum speed of the target, the greater the amount of fluctuation in the Doppler frequency included in the reflected wave from the target, and the shorter the time period in which there is a high correlation. In this case, since the number of additions Np is a small value, the gain improvement effect due to the addition in the adding section 212 becomes small.

The Doppler frequency analysis unit 213 performs coherent integration by aligning the timing of the discrete time k, using CI _(z) (k,Nc(w-1)+1) to CI _(z) (k,Nc×w), which are the Nc outputs of the adder unit 212 obtained at each discrete time k, as one unit. For example, the Doppler frequency analysis unit 213 performs coherent integration after correcting the phase fluctuation Φ(fs)=2πfs(Tr×Np)ΔΦ corresponding to 2Nf different Doppler frequencies fsΔΦ, as shown in the following equation.

Here, FT_CI _(z) ^Nant (k,fs,w) is the wth output of the Doppler frequency analysis unit 213 and indicates the coherent integration result of the Doppler frequency fsΔΦ at the Nantth discrete time k in the antenna system processing unit 201. Here, Nant=1 to Na, fs=-Nf+1,...,0,...,Nf, k=1,...,(Nr+Nu)Ns/No, w is an integer equal to or greater than 1, and ΔΦ is a phase rotation unit.

As a result, each antenna system processing unit 201 calculates coherent integration results corresponding to 2Nf Doppler frequency components at each discrete time k, FT_CI _(z) ^Nant (k, -Nf+1,w),..., FT_CI _{( z)} ^Nant (k, Nf-1, w) is obtained every multiple Np×Nc periods (Tr×Np×Nc) of the radar transmission period Tr. Note that j is an imaginary unit, and z=1,...,Nt.

When ΔΦ=1/Nc, the processing of the Doppler frequency analysis section 213 described above is to perform discrete Fourier transform (DFT) on the output of the addition section 212 at a sampling interval Tm=(Tr×Np) and a sampling frequency fm=1/Tm. This is equivalent to what is being processed.

Further, by setting Nf to a number that is a power of 2, the Doppler frequency analysis unit 213 can apply fast Fourier transform (FFT) processing, and the amount of calculation processing can be greatly reduced. At this time, if Nf>Nc, by performing zero-filling processing to set CI _(z) (k, Nc(w-1)+q) = 0 in the region where q>Nc, FFT processing can be applied, and the amount of calculation processing can be greatly reduced.

Further, the Doppler frequency analysis unit 213 may perform a process of sequentially calculating the product-sum calculation shown in the above equation (3) without performing the FFT process. In _other words, the Doppler frequency analysis unit 213 calculates fs The coefficient exp[-j2πf _s T _r N _p qΔφ] corresponding to =-Nf+1,...,0,...,Nf-1 may be generated and the sum-of-products operation may be performed sequentially. Here, q=0 to Nc-1.

In the following description, the w-th output FT_CI _(z) ¹ (k,fs,w), FT_CI(z)2(k,fs,w), ... ^, FT_CI _{(z) Na} ⁽ k,fs,w) obtained by performing similar processing in each of the Na antenna system processors 201 is expressed as a virtual receiving array correlation vector h(k,fs,w) as shown in the following equation. The virtual receiving array correlation vector h(k,fs,w) includes Nt x Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na. The virtual receiving array correlation vector h(k,fs,w) is used to explain the process of estimating the direction of a reflected wave signal from a target based on the phase difference between the receiving antennas 202, which will be described later. Here, z=1, ...,Nt, and b=1, ...,Na.

The processing in each component of the signal processing section 207 has been described above.

The direction estimation unit 214 calculates an array correction value h_cal for the virtual reception array correlation vector h (k, fs, w) of the w-th Doppler frequency analysis unit 213 output from the antenna system processing units 201-1 to 201-Na. _[y] is used to calculate a virtual receiving array correlation vector _{h_after_cal} (k, fs, w) that corrects the phase deviation and amplitude deviation between the antenna system processing units 201. The virtual receiving array correlation vector h _{_after_cal} (k, fs, w) is expressed by the following equation. Note that y=1,...,(Nt×Na).

Then, the direction estimation unit 214 uses the virtual reception array correlation vector h _{_after_cal} (k, fs, w) to perform direction estimation processing in the horizontal direction and the vertical direction based on the phase difference of the reflected wave signals between the reception antennas 202. I do. The azimuth estimating unit 214 calculates a spatial profile by varying the azimuth direction θ and the elevation direction Φ in the direction estimation evaluation function value P(θ, φ, k, fs, w) within a predetermined angular range, and calculates the spatial profile using the calculated spatial profile. A predetermined number of maximum peaks are extracted in ascending order of magnitude, and the azimuth and elevation directions of the maximum peaks are used as the estimated direction of arrival.

Note that there are various evaluation function values P(θ, φ, k, fs, w) depending on the direction of arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Reference Non-Patent Document 1 may be used.

(Reference non-patent document 1) Direction-of-arrival estimation using signal subspace modeling Cadzow, J.A.; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28 , Issue: 1 Publication Year: 1992 , Page(s): 64 - 79

For example, the beamformer method can be expressed as follows. Other methods such as Capon and MUSIC are also applicable.

Here, the superscript H is the Hermitian transpose operator, and a(θ _u , φ _v ) represents the direction vector of a virtual receiving array with respect to an arriving wave with an azimuth direction of θ _u and an elevation direction of φ _v .

As described above, the direction estimation unit 214 outputs the calculated w-th direction of arrival estimated value, discrete time k, Doppler frequency fsΔΦ, and angle θ _u as the radar positioning result.

Here, _the direction vector a ₍ θ _u , φ _v ) is expressed as (Nt × Na ) is the next column vector. The complex response a(θ _u , φ _v ) of the virtual receiving array represents a phase difference calculated geometrically based on the element spacing between antennas.

Further, θ _u is a value obtained by changing the azimuth range within which direction of arrival estimation is performed by a predetermined azimuth interval β ₁ . For example, θ _u is set as follows.
θ _u = θmin + uβ ₁ , u=0,…, NU
NU=floor[(θmax-θmin)/β ₁ ]+1
Here, floor(x) is a function that returns the largest integer value that does not exceed the real number x.

Further, φ _v is the elevation angle range within which the direction of arrival is estimated, which is varied by a predetermined elevation angle interval β ₂ . For example, φ _v is set as follows.
φ _v = φmin + vβ ₂ , v=0,…, NV
NV=floor[(φmax-φmin)/β ₂ ]+1

In this embodiment, it is assumed that the direction vector of the virtual reception array is calculated in advance based on the virtual reception array arrangement VA#1,..., VA#(Nt×Na), which will be described later. The elements of the direction vector of the virtual reception array represent phase differences calculated geometrically at element intervals between antennas in the order of virtual reception array arrangement numbers VA#1, . . . , VA#(Nt×Na), which will be described later.

Moreover, the above-mentioned time information k may be converted into distance information and output. When converting time information k into distance information R(k), the following equation may be used. Here, Tw represents the code transmission period, L represents the pulse code length, and C ₀ represents the speed of light.

Further, the Doppler frequency information (fsΔΦ) may be converted into a relative velocity component and output. When converting the Doppler frequency fsΔΦ into the relative velocity component vd(fs), the following equation can be used. Here, λ is the wavelength of the carrier frequency of the RF signal output from the transmitter radio section 107.

[Antenna arrangement in radar device 10]
The arrangement of Nt transmitting antennas 106 and Na receiving antennas 202 in radar device 10 having the above configuration will be explained.

Figure 6 shows the antenna arrangement of a transmitting array consisting of Nt = 2 transmitting antennas 106 (Tx#1, Tx#2), the antenna arrangement of a receiving array consisting of Na = 3 receiving antennas 202 (Rx#1, Rx#2, Rx#3), and the antenna arrangement of a virtual receiving array (number of elements: Nt × Na = 6) constructed based on these transmitting and receiving array antennas.

Each of transmitting antenna 106 and receiving antenna 202 is configured using a subarray element including two antenna elements.

Further, the size (width) of the subarray element is D _subarry , and the desired antenna element spacing at which grating lobes do not occur in the radar detection angle range is De. In FIG. 6, the subarray element size D _subarry is larger than the desired antenna element spacing De (D _subarry > De). Note that a value of 0.5 wavelength or more and 0.75 wavelength or less is used as the desired antenna element spacing De.

Further, let Dt be the sub-array element spacing of the transmitting array antenna, and Dr be the sub-array element spacing of the receiving array antenna. For example, in FIG. 6, the subarray element spacing Dt of the transmitting array antenna is 1.5λ (1.5 wavelengths), and the subarray element spacing Dr of the receiving antenna is 1λ (1 wavelength). In other words, the subarray element spacing Dt, Dr is approximately one wavelength (λ) or more.

In this embodiment, the subarray element size D _subarry is wider than the desired antenna element spacing De at which grating lobes do not occur in the radar detection angle range (D _subarry > De). In this case, the transmitting array and the receiving array are arranged so that the relationship shown in the following equation is satisfied between the sub-array element spacing Dt of the transmitting array antenna and the sub-array element spacing Dr of the receiving array antenna.
｜Dt - Dr | = De (10)

That is, the absolute value of the difference between the sub-array element spacing Dt of the transmitting array antenna and the sub-array element spacing Dr of the receiving array antenna is the same as the desired antenna element spacing De.

FIG. 6 shows, as an example, the case where De=λ/2, the sub-array element spacing Dt of the transmitting array antenna is 1.5λ, and the sub-array element spacing Dr=λ of the receiving array antenna.

In this case, as shown in FIG. 6, the element spacing near the center (other than the ends) of the virtual receiving array becomes the desired antenna element spacing De (=|Dt-Dr|=λ/2). That is, the virtual receiving array provides an array arrangement in which grating lobes do not occur within the radar detection angle range.

FIG. 7 shows a directivity pattern (Fourier beam pattern; main beam: 0° direction) in the transmitting/receiving array antenna arrangement shown in FIG. 6 (in the case of De=0.5λ, Dt=1.5λ, Dr=λ). As shown in FIG. 7, it can be seen that no grating lobes are generated in the angular range of ±90° from the main beam direction.

In this way, in this embodiment, the difference (absolute value) between the element spacing of the transmitting array antenna made up of the transmitting antenna 106 and the element spacing of the receiving array antenna made up of the receiving antenna 202 is determined by the grating lobe. The transmitting antenna 106 and the receiving antenna 202 are arranged so that the spacing between the elements is equal to the desired element spacing at which no error occurs.

By doing so, it is possible to set the element spacing of the virtual receiving array configured according to the arrangement relationship of the transmitting antenna 106 and the receiving antenna 202 to a desired element spacing at which grating lobes do not occur. This makes it possible to eliminate false detections due to grating lobes when the direction estimation unit 214 performs direction estimation processing.

Therefore, according to this embodiment, even when using an array element having a subarray configuration, it is possible to suppress the generation of unnecessary grating lobes and realize a desired directivity pattern.

Note that FIG. 6 shows an example of a configuration in which the array antennas are arranged linearly in the horizontal direction to estimate the direction of arrival in the horizontal direction. However, in this embodiment, even when the array antennas are arranged linearly in the vertical direction to estimate the direction of arrival in the vertical direction, it is possible to similarly arrange a virtual receiving array with the desired element spacing in the vertical direction in such a way that no grating lobes occur.

(Variation 1)
In variation 1, a case will be described where direction-of-arrival estimation is performed in both the horizontal and vertical directions.

Transmitting array elements or receiving array elements are arranged in two dimensions, vertically and horizontally.

FIG. 8 shows the antenna arrangement of a transmitting array consisting of Nt = 6 transmitting antennas 106 (Tx#1 to Tx#6), Na = 3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3), and the antenna arrangement of a virtual receiving array (number of elements: Nt×Na=18) constructed based on these transmitting and receiving array antennas.

In FIG. 8, the transmission array has two subarray elements arranged in a two-dimensional manner, two in the horizontal direction and three in the vertical direction.

Further, in FIG. 8, the size of the sub-array element in the horizontal direction is set to D _subarry , and the size of the sub-array element in the vertical direction is set to be less than or equal to De. That is, the size of the antenna element is larger than the desired antenna element spacing De in the horizontal direction and less than or equal to the desired antenna element spacing De in the vertical direction.

In FIG. 8, as an example, the desired antenna element spacing De=λ/2, the horizontal sub-array element spacing Dt of the transmitting array antenna is 1.5λ, and the vertical element spacing of the transmitting array antenna is De. Further, it is assumed that the horizontal subarray element spacing Dr=λ of the receiving antenna.

In this case, as shown in FIG. 8, the absolute value of the difference in the horizontal direction between the subarray element spacing Dt of the transmitting array antenna and the subarray element spacing Dr of the receiving array antenna is the same as the desired antenna element spacing De. Also, as shown in FIG. 8, in the vertical direction, the element spacing of the transmitting array antenna is the same as the desired antenna element spacing De.

As a result, as shown in FIG. 8, the element spacing near the center (other than the edges) of the virtual reception array in the horizontal direction becomes the desired antenna element spacing De (=|Dt-Dr|=λ/2). .

Further, as shown in FIG. 8, in the vertical direction, the element spacing of the virtual receiving array becomes a desired antenna element spacing De, similar to the vertical element spacing of the transmitting array.

In other words, the virtual receiving array provides an array arrangement in which no grating lobes occur within the radar detection angle range in either the horizontal or vertical direction.

When estimating the direction of arrival in the horizontal and vertical directions in the direction estimation unit 214, the azimuth direction _θu and elevation angle direction _φv are varied as shown in the following equation to calculate a direction estimation evaluation function value P( _θu , _φv , k, fs, w), and the azimuth direction and elevation angle direction which give the maximum value are determined as the direction of arrival estimated value DOA(k, fs, w).

Here, u=1,...,NU. Note that arg max P(x) is an operator whose output value is the value in the domain where the function value P(x) is maximum.

Note that there are various evaluation function values P (θ _u , φ _v , k, fs, w) depending on the direction of arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Reference Non-Patent Document 1 mentioned above may be used. For example, the beamformer method can be expressed as follows. Other methods such as Capon and MUSIC are also applicable.

Here, the superscript H is the Hermitian transposition operator. Further, a(θ _u , φ _v ) indicates a direction vector for the arriving wave in the azimuth direction θ _u and the elevation direction φ _v .

Figures 9A and 9B show the horizontal and vertical directivity patterns (Fourier beam pattern; main beam: 0° direction) of the transmitting and receiving array antenna arrangement shown in Figure 8 (when De=0.5λ, Dt=1.5λ, Dr=1λ).

As shown in FIG. 9A, it can be seen that no grating lobes are generated in the angular range of ±90° from the main beam direction in the horizontal direction. Further, as shown in FIG. 9B, it can be seen that a beam pattern is formed in which no grating lobes occur even in the vertical direction.

By using such an arrangement of the transmitting and receiving array antennas, it is possible to eliminate the occurrence of false detections due to grating lobes in both the horizontal and vertical directions when performing direction estimation processing in the direction estimation unit 214.

Therefore, according to variation 1, even when using array elements arranged two-dimensionally in a subarray configuration, it is possible to suppress the generation of unnecessary grating lobes and realize a desired directivity pattern.

In addition, in FIG. 8, the case where the horizontal size of the subarray element is D _subarry (>De) has been explained, but in variation 1, the case where the vertical size of the subarray element is D _subarry (>De) is explained. can be similarly applied. In this case, in the vertical arrangement of the transmitting array, the difference (absolute value) between the element spacing of the transmitting array antenna and the element spacing of the receiving array antenna is equal to the desired element spacing at which grating lobes do not occur. All you need to do is arrange a transmitting array.

(Variation 2)
In variation 2, another example will be described in which direction of arrival estimation is performed in both the horizontal and vertical directions.

Specifically, in the transmitting array antenna, when the horizontal element spacing is Dt (>De) and the vertical element spacing is the desired antenna element spacing De, in the transmitting array antenna, adjacent elements in the vertical direction, Two subarray element arrays arranged in a straight line in the horizontal direction are shifted in the horizontal direction by an interval equal to the desired antenna element interval De.

FIG. 10 shows the antenna arrangement of a transmitting array consisting of Nt = 6 transmitting antennas 106 (Tx#1 to Tx#6), Na = 3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3), and the antenna arrangement of a virtual receiving array (number of elements: Nt×Na=18) constructed based on these transmitting and receiving array antennas.

In Figure 10, the transmitting array has subarray elements arranged two-dimensionally, two in the horizontal direction and three in the vertical direction.

Further, in FIG. 10, the size of the subarray element in the horizontal direction is set as D _subarry , and the size of the subarray element in the vertical direction is set as De or less. That is, the size of the antenna element is larger than the desired antenna element spacing De in the horizontal direction and less than or equal to the desired antenna element spacing De in the vertical direction.

In FIG. 10, as in FIG. 8, the desired antenna element spacing De=λ/2, the horizontal subarray element spacing Dt of the transmitting array antenna is 1.5λ, and the vertical element spacing of the transmitting array antenna is De. . Further, it is assumed that the sub-array element spacing in the horizontal direction of the receiving array antenna is Dr=λ.

As with variation 1 (Figure 8), as shown in Figure 10, the absolute value of the difference between the subarray element spacing Dt of the transmitting array antenna and the subarray element spacing Dr of the receiving array antenna in the horizontal direction is the same as the desired antenna element spacing De. Also, as shown in Figure 10, in the vertical direction, the element spacing of the transmitting array antenna is the same as the desired antenna element spacing De.

Furthermore, in FIG. 10, the transmitting antennas 106 (vertically adjacent transmitting antennas 106) that are spaced apart by the antenna element spacing De in the vertical direction of the transmitting array antenna are arranged shifted by the same distance as the antenna element spacing De in the horizontal direction. Ru. In other words, in the transmitting array antenna, two subarray element arrays that are adjacent in the vertical direction and arranged in a straight line in the horizontal direction are arranged horizontally shifted by the same interval as the desired element interval.

For example, what is the arrangement of transmitting antennas Tx#1 and Tx#2 shown in FIG. 10 (i.e., subarray element arrangement; the same applies hereinafter) and the arrangement of transmitting antennas T#3 and Tx#4 that are vertically adjacent to this arrangement? , are arranged with the same spacing as the antenna element spacing De. Similarly, the array of transmitting antennas Tx#3 and Tx#4 and the array of transmitting antennas T#5 and T#6 vertically adjacent to the array are shifted horizontally by the same distance as the antenna element spacing De. It is located.

In Figure 10, in the horizontal direction, the element spacing near the center (other than the ends) of the virtual receiving array is the desired antenna element spacing De (= |Dt-Dr| = λ/2). Also, as shown in Figure 10, in the vertical direction, the element spacing of the virtual receiving array is the desired antenna element spacing De, just like the vertical element spacing of the transmitting array. In other words, with the virtual receiving array, an array arrangement can be obtained in which no grating lobes occur within the radar detection angle range.

Furthermore, as shown in FIG. 10, in the vertical direction of the virtual reception array, the arrangement of the array elements in the center (second stage) is compared with the arrangement of array elements in the other array elements (first and third stages). , and are shifted horizontally. As a result, in FIG. 10, compared to variation 1 (FIG. 8), the antenna elements are spaced closer together in the two-dimensional plane in which the virtual reception array is arranged. This makes it possible to reduce the sidelobe level in the virtual reception array.

11A and 11B show the horizontal and vertical directivity patterns (Fourier beam patterns. Main beam: 0° direction).

As shown in FIG. 11A, it can be seen that no grating lobes are generated in the angular range of ±90° from the main beam direction in the horizontal direction. Furthermore, as shown in FIG. 11B, it can be seen that a beam pattern is formed in which no grating lobes occur even in the vertical direction.

Furthermore, compared to variation 1 (Figure 9A), it can be seen that the side lobe level is reduced in the horizontal directivity pattern, as shown in Figure 11A.

By using such an arrangement of the transmitting and receiving array antennas, it is possible to eliminate the occurrence of false detections due to grating lobes and side lobes in both the horizontal and vertical directions when performing direction estimation processing in the direction estimation section 214. can.

Therefore, according to variation 2, even when using array elements arranged two-dimensionally in a subarray configuration, it is possible to suppress the generation of unnecessary grating lobes and the side lobe level, and realize a desired directivity pattern. can.

(Variation 3)
In variation 3, another example will be described in which direction of arrival estimation is performed in both the horizontal direction and the vertical direction.

Specifically, in the transmitting array antenna, the spacing between subarray element arrays that are adjacent in the vertical direction and aligned in a straight line in the horizontal direction is the desired antenna element spacing De multiplied by a constant α, and two subarray element arrays that are adjacent in the vertical direction and aligned in a straight line in the horizontal direction are shifted in the horizontal direction by a spacing that is the desired antenna element spacing De multiplied by a constant β.

FIG. 12 shows the antenna arrangement of a transmitting array consisting of Nt = 6 transmitting antennas 106 (Tx#1 to Tx#6), Na = 3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3), and the antenna arrangement of a virtual receiving array (number of elements: Nt×Na=18) constructed based on these transmitting and receiving array antennas.

In Figure 12, the transmitting array has subarray elements arranged two-dimensionally, two in the horizontal direction and three in the vertical direction.

Further, in FIG. 12, the size of the sub-array element in the horizontal direction is set as D _subarry , and the size of the sub-array element in the vertical direction is set to be less than or equal to De. That is, the size of the antenna element is larger than the desired antenna element spacing De in the horizontal direction and less than or equal to the desired antenna element spacing De in the vertical direction.

Similar to FIG. 8, in FIG. 12, the desired antenna element spacing De=λ/2, the horizontal sub-array element spacing Dt of the transmitting array antenna is 1.5λ, and the horizontal sub-array element spacing Dr=λ of the receiving array antenna. shall be. Further, it is assumed that the sub-array element spacing in the horizontal direction of the receiving array antenna is Dr=λ.

Similar to variations 1 and 2 (FIGS. 8 and 10), as shown in FIG. 12, the absolute value of the difference between the sub-array element spacing Dt of the transmitting array antenna and the sub-array element spacing Dr of the receiving array antenna in the horizontal direction is , is the same as the desired antenna element spacing De.

On the other hand, as shown in FIG. 12, in the vertical direction, the element spacing of the transmitting array antenna is the spacing αDe obtained by multiplying the desired antenna element spacing De by a constant α.

In addition, in FIG. 12, transmitting antennas 106 separated by an element spacing αDe in the vertical direction of the transmitting array antenna (transmitting antennas 106 adjacent in the vertical direction) are arranged with a horizontal spacing βDe offset, which is the desired antenna element spacing De multiplied by a constant β. In other words, in the transmitting array antenna, two subarray element arrays that are adjacent in the vertical direction and aligned in a straight line in the horizontal direction are arranged with a horizontal spacing offset β times the desired element spacing.

For example, the arrangement of transmitting antennas Tx#1 and Tx#2 shown in FIG. 12 and the arrangement of transmitting antennas T#3 and Tx#4 that are vertically adjacent to the arrangement are shifted by an interval βDe. Similarly, the array of transmitting antennas Tx#3 and Tx#4 and the array of transmitting antennas T#5 and T#6 that are vertically adjacent to the array are shifted by an interval βDe in the horizontal direction.

For example, α=(3) ^0.5 /2≈0.866 and β=0.5.

In FIG. 12, in the horizontal direction, the element spacing near the center (other than the ends) of the virtual receiving array is the desired antenna element spacing De (=|Dt-Dr|=λ/2).

Further, as shown in FIG. 12, the element spacing of the virtual receiving array in the vertical direction is αDe (=(3) ^0.5 De), similar to the vertical element spacing of the transmitting array.

That is, the virtual reception array provides an array arrangement in which grating lobes do not occur within the radar detection angle range.

Furthermore, as shown in FIG. 12, in the vertical direction of the virtual receiving array, the arrangement of the array elements in the center (second stage) is compared with the arrangement of array elements in the other array elements (first and third stages). Then, they are arranged shifted horizontally by βDe (=0.5De).

As a result, in FIG. 12, similar to variation 2 (FIG. 10), the spacing between antenna elements in the two-dimensional plane in which the virtual receiving array is arranged is closer than in variation 1 (FIG. 8). This allows the virtual receiving array to reduce the side lobe level.

Here, as shown in FIG. 12, near the center of the virtual receiving array, the distance between three adjacent antenna elements in the two-dimensional plane on which the virtual receiving array is arranged becomes a desired antenna element spacing De. In other words, the straight line connecting three adjacent array elements in the two-dimensional plane where the virtual reception array is arranged forms an equilateral triangle with one side having the antenna element spacing De. Since the equilateral triangular grating arrangement has higher grating lobe suppression performance than the rectangular grating arrangement with the same aperture length, it is possible to further reduce the levels of grating lobes and side lobes compared to variation 2.

In other words, the constants α and β should be set so that the element spacing between three adjacent array elements in the vertical and horizontal directions is the desired antenna element spacing De (an equilateral triangle with one side De).

13A and 13B show the horizontal and vertical directions of the transmitting/receiving array antenna arrangement (De=0.5λ, Dt=1.5λ, Dr=1λ, α=(3) ^0.5 /2, β=0.5) shown in FIG. 12. Directivity patterns (Fourier beam patterns. Main beam: 0° direction) are shown.

As shown in FIG. 13A, it can be seen that no grating lobes are generated in the angular range of ±90° from the main beam direction in the horizontal direction. Furthermore, as shown in FIG. 13B, it can be seen that a beam pattern is formed in which no grating lobes occur even in the vertical direction.

Furthermore, when compared with variation 1 (FIG. 9A), it can be seen that the sidelobe level is reduced in the horizontal directivity pattern, as shown in FIG. 13A.

Also, compared to variation 2 (Fig. 11A), as shown in Fig. 13A, the sidelobe level appearing in the direction closest to the main lobe (±30° direction in Fig. 13A) in the horizontal directivity pattern is It can be seen that this has been reduced.

By using such an arrangement of the transmitting and receiving array antennas, it is possible to eliminate the occurrence of false detections due to grating lobes and side lobes in both the horizontal and vertical directions when performing direction estimation processing in the direction estimation unit 214.

Therefore, according to variation 3, even when using array elements arranged two-dimensionally in a subarray configuration, it is possible to suppress the generation of unnecessary grating lobes and the side lobe level, and realize a desired directivity pattern. can.

The above describes one embodiment of the present disclosure.

Note that the operations according to the above embodiment and each modification may be combined as appropriate.

Furthermore, in the above embodiment, the case where the number of transmitting antennas 106 Nt=2 or 3 and the number Na of receiving antennas 202=3 is illustrated. However, the number Nt of transmitting antennas 106 and the number Na of receiving antennas 202 are not limited to these numbers.

Further, in the above embodiment, the case where the transmitting antenna 106 and the receiving antenna 202 are sub-array elements consisting of two antenna elements has been described. It may be composed of more than one element.

Furthermore, in variations 1 to 3 of the above embodiment, the case has been described in which the transmitting array antenna is arranged two-dimensionally in the horizontal and vertical directions, and the receiving array antenna is arranged one-dimensionally in the horizontal direction. However, in the present disclosure, the receiving array antenna may be arranged two-dimensionally, and the transmitting array antenna may be arranged one-dimensionally. In this case, the arrangement of the sub-array elements in the transmitting array antenna described above may be applied to the arrangement of the sub-array elements in the receiving array antenna.

In addition, in the above embodiment, a case has been described in which the size of the antenna elements is larger than the desired antenna element spacing De in the horizontal direction and is equal to or smaller than the desired antenna element spacing De in the vertical direction. However, the size of the antenna elements may be larger than the desired antenna element spacing De in the vertical direction and is equal to or smaller than the desired antenna element spacing De in the horizontal direction. In this case, the arrangement of the subarray elements in the above-mentioned transmitting and receiving array antennas may be switched between the horizontal and vertical directions.

Further, in the above embodiment, a case has been described in which a coded pulse radar is used, but the present disclosure is also applicable to a radar system using a frequency-modulated pulse wave such as a chirp pulse radar.

Furthermore, in the radar device 10 shown in FIG. 2, the radar transmitting section 100 and the radar receiving section 200 may be individually arranged at physically separate locations.

Note that in the radar apparatus, a configuration is shown in which the radar transmitter sends out different code-division multiplexed transmission signals from a plurality of transmitting antennas, and the radar receiver separates each transmission signal and performs reception processing. The configuration of the radar device is not limited to this, but a radar transmitter transmits different frequency-division multiplexed transmission signals from a plurality of transmitting antennas, and a radar receiver separates each transmission signal and performs reception processing. It may be a configuration. Similarly, the configuration of the radar device may be such that the radar transmitter transmits time-division multiplexed transmission signals from a plurality of transmitting antennas, and the radar receiver performs reception processing, similar to the above embodiment. You can get the following effect.

Although not shown, the radar device 10 includes, for example, a CPU (Central Processing Unit), a storage medium such as a ROM (Read Only Memory) that stores a control program, and a working memory such as a RAM (Random Access Memory). . In this case, the functions of each section described above are realized by the CPU executing a control program. However, the hardware configuration of the radar device 10 is not limited to this example. For example, each functional unit of the radar device 10 may be realized as an integrated circuit (IC). Each functional unit may be individually integrated into one chip, or may include a part or all of the functional units into one chip.

<Summary of this disclosure>
The radar device of the present disclosure includes a radar transmitter that transmits a radar signal at a predetermined transmission cycle using a transmitting array antenna, and a receiving array antenna that receives a reflected wave signal obtained by reflecting the radar signal on a target. a radar receiving section, the transmitting array antenna and the receiving array antenna each include a plurality of sub-array elements, and the plurality of sub-array elements include a first one in the transmitting array antenna and the receiving array antenna. each subarray element includes a plurality of antenna elements, the size of the subarray element is larger than a desired antenna element spacing in the first direction, and the subarray element of the transmitting array antenna The absolute value of the difference between the spacing and the subarray element spacing of the receiving array antenna is the same as the desired antenna element spacing.

Further, in the radar device of the present disclosure, the desired antenna element spacing is 0.5 wavelength or more and 0.75 wavelength or less.

Further, in the radar device of the present disclosure, in either the transmitting array antenna or the receiving array antenna, the plurality of sub-array elements are further arranged in a second direction orthogonal to the first direction, and If the size of the sub-array element is larger than the desired antenna element spacing in the first direction and less than or equal to the desired antenna element spacing in the second direction, the size of the sub-array element is larger than the desired antenna element spacing in the first direction, The absolute value of the difference between the subarray element spacing and the subarray element spacing of the receiving array antenna is the same as the desired antenna element spacing, and in the second direction, the subarray element spacing is equal to the desired antenna element spacing. Same as interval.

In addition, in the radar device disclosed herein, the subarray elements arranged in the second direction are arranged in the first direction with a shifted distance equal to the desired antenna element spacing.

Further, in the radar device of the present disclosure, in either the transmitting array antenna or the receiving array antenna, the plurality of sub-array elements are further arranged in a second direction orthogonal to the first direction, and If the size of the sub-array element is larger than the desired antenna element spacing in the first direction and less than or equal to the desired antenna element spacing in the second direction, the size of the sub-array element is larger than the desired antenna element spacing in the first direction, The absolute value of the difference between the sub-array element spacing and the sub-array element spacing of the receiving array antenna is the same as the desired element spacing, and in the second direction, the sub-array element spacing is equal to the desired antenna element spacing. ((√3)/2) times the length, and the plurality of subarray elements arranged in the second direction have a length (1/2) of the desired antenna element spacing in the first direction. The double spacing is shifted and placed.

Although various embodiments (variations) have been described above with reference to the drawings, it goes without saying that the present disclosure is not limited to such examples. It is clear that those skilled in the art can come up with various changes or modifications within the scope of the claims, and these naturally fall within the technical scope of the present disclosure. Understood. Further, each component in the above embodiments (variations) may be arbitrarily combined without departing from the spirit of the disclosure.

In each of the above embodiments, the present disclosure has been described as being configured using hardware, but the present disclosure can also be realized using software in conjunction with hardware.

Furthermore, each functional block used in the description of each of the above embodiments is typically realized as an LSI, which is an integrated circuit. The integrated circuit may control each functional block used in the description of the above embodiments and may include inputs and outputs. These may be integrated into one chip individually, or may be integrated into one chip including some or all of them. Although it is referred to as an LSI here, it may also be called an IC, system LSI, super LSI, or ultra LSI depending on the degree of integration.

Further, the method of circuit integration is not limited to LSI, and may be implemented using a dedicated circuit or a general-purpose processor. After the LSI is manufactured, an FPGA (Field Programmable Gate Array) that can be programmed or a reconfigurable processor that can reconfigure the connections or settings of circuit cells inside the LSI may be used.

Furthermore, if an integrated circuit technology that replaces LSI emerges due to advancements in semiconductor technology or other derivative technologies, it is natural that functional blocks may be integrated using that technology. Possibilities include the application of biotechnology.

The present disclosure is suitable as a radar device that detects a wide angle range.

10 Radar device 100 Radar transmitter 200 Radar receiver 300 Reference signal generator 400 Control unit 101, 101a Radar transmission signal generator 102 Code generator 103 Modulator 104 LPF
105 Transmission radio section 106 Transmission antenna 111 Code storage section 112 DA conversion section 201 Antenna system processing section 202 Reception antenna 203 Reception radio section 204 Amplifier 205 Frequency converter 206 Quadrature detector 207 Signal processing section 208, 209 AD conversion section 210 Separation section 211 Correlation calculation unit 212 Addition unit 213 Doppler frequency analysis unit 214 Direction estimation unit

Claims (10)

a transmitting array antenna;
receiving array antenna,
a radar transmitting unit that transmits radar signals using the transmitting array antenna;
a radar receiving unit that receives a reflected wave signal obtained by reflecting the radar signal from a target using the receiving array antenna;
Equipped with
The transmitting array antenna includes a plurality of transmitting antennas,
The receiving array antenna includes a plurality of receiving antennas,
The plurality of transmitting antennas are arranged at different positions in a first direction,
The plurality of receiving antennas are arranged at different positions in the first direction,
An interval between two adjacent transmitting antennas among the plurality of transmitting antennas is one wavelength or more in the first direction,
The distance between two adjacent receiving antennas among the plurality of receiving antennas is one wavelength or more in the first direction,
The absolute value of the difference between the interval between the two adjacent transmitting antennas and the interval between the two adjacent receiving antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction,
At least one of the plurality of transmitting antennas and the plurality of receiving antennas includes a plurality of antenna elements arranged in the first direction.
radar equipment. At least one of the plurality of transmitting antennas and the plurality of receiving antennas, each including a plurality of antenna elements arranged in the first direction, has a size in the first direction equal to the spacing between the two adjacent transmitting antennas. greater than the absolute value of the difference between the spacing between the two adjacent receiving antennas,
The radar device according to claim 1. At least one of the plurality of transmitting antennas and the plurality of receiving antennas, each including a plurality of antenna elements arranged in the first direction, has a size in the first direction equal to the spacing between the two adjacent transmitting antennas. smaller than the narrowest interval between the two adjacent receiving antennas;
The radar device according to claim 1 or 2. A virtual receiving array antenna formed by the transmitting array antenna and the receiving array antenna has a plurality of virtual receiving antennas,
At least one interval between two adjacent virtual reception antennas among the plurality of virtual reception antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction.
A radar device according to any one of claims 1 to 3. The wavelength is determined by the frequency of the radar signal.
A radar device according to any one of claims 1 to 4. A first array antenna;
A second array antenna; and
a radar transmitter that transmits a radar signal using one of the first array antenna and the second array antenna;
a radar receiving unit that receives a reflected wave signal, which is the radar signal reflected by a target, by using the other of the first array antenna and the second array antenna;
Equipped with
the first array antenna includes a plurality of first antennas;
each of the plurality of first antennas is disposed at a different position in a first direction;
the second array antenna includes a plurality of second antennas;
each of the plurality of second antennas is disposed at a different position in the first direction;
a distance between two adjacent first antennas among the plurality of first antennas is equal to or greater than one wavelength in the first direction;
a distance between two adjacent second antennas among the plurality of second antennas is equal to or greater than one wavelength in the first direction;
an absolute value of a difference between a distance between two adjacent first antennas and a distance between two adjacent second antennas is equal to or greater than 0.5 wavelengths and equal to or less than 0.75 wavelengths in the first direction;
At least one of the first and second antennas includes a plurality of antenna elements arranged in the first direction.
Radar equipment. At least one of the plurality of first antennas and the plurality of second antennas, each including a plurality of antenna elements arranged in the first direction, has a size in the first direction that is greater than an absolute value of a difference between a distance between two adjacent first antennas and a distance between two adjacent second antennas.
The radar device according to claim 6. At least one of the plurality of first antennas and the plurality of second antennas, each including a plurality of antenna elements arranged in the first direction, has a size in the first direction that is smaller than the narrowest interval between the interval between the two adjacent first antennas and the interval between the two adjacent second antennas.
The radar device according to claim 6 or 7. A virtual reception array antenna formed by the first array antenna and the second array antenna has a plurality of virtual reception antennas,
At least one interval between two adjacent virtual reception antennas of the plurality of virtual reception antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction.
The radar device according to any one of claims 6 to 8. the wavelength is determined by the frequency of the radar signal;
The radar device according to any one of claims 6 to 9.