JP7106441B2 - Control device and its error correction method - Google Patents

Description

The present invention relates to a control device and its error correction method, and more particularly to a control device and its error correction method suitable for accurately detecting a rotor rotation angle.

Patent Document 1 discloses a control device that calculates the rotation angle of a rotor portion of a resolver attached to the rotating shaft of a motor from an electrical signal output from the resolver and controls the motor based on the calculation result. there is

This control device generates a magnetic field in the resolver, for example by supplying a sinusoidal carrier signal to an excitation coil arranged in the stator part of the resolver. Here, the magnetic field generated in the resolver fluctuates according to the rotation of the rotor portion of the resolver. Two detection coils arranged in the stator portion of the resolver together with the excitation coil detect variations in the magnetic field generated in the resolver and output them as two-phase signals. This control device calculates the rotation angle of the rotor from the two-phase phase signals.

Here, it is known that the magnetic flux generated in the motor leaks to the excitation coil and the detection coil provided in the resolver and interlinks with these coils, thereby inducing an unintended voltage signal and generating magnetic noise. ing. Even if the resolver is not excited by the carrier signal, the magnetic noise is generated just by rotating the motor, and is superimposed on the two-phase phase signals, thereby reducing the detection accuracy of the rotor rotation angle. Therefore, this control device uses a bandpass filter to remove the magnetic noise contained in the composite signal of the two-phase phase signals.

Japanese Unexamined Patent Application Publication No. 2017-32480

However, in the control device disclosed in Patent Document 1, when the magnetic noise increases as the rotational speed of the motor (the number of rotations per predetermined time) increases, the combined signal of the two-phase phase signals saturates. Since it ends up (clipping) and harmonic components are generated, there is a problem that the detection accuracy of the rotor rotation angle is lowered. Other problems and novel features will become apparent from the description of the specification and the accompanying drawings.

According to one embodiment, the control device includes a first phase shifter that shifts the phase of the first phase signal among the first and second phase signals with different phases detected from the resolver excited by the carrier signal. a second phase shifter for shifting the phase of the second phase signal; the first phase signal phase-shifted by the first phase shifter; and the second phase signal phase-shifted by the second phase shifter. and a combiner that outputs a phase-modulated signal in which the carrier signal is modulated by the rotation angle of the rotor of the resolver, a band-pass filter that removes magnetic noise contained in the phase-modulated signal, a gain adjustment circuit that adjusts the gain of the combiner based on the rotation speed of the rotor.

Further, according to one embodiment, the error correction method of the control device includes first and second phase signals having different phases detected from a resolver excited by a carrier signal, and correcting the phase of the first phase signal as shifted by a first phase shifter, the phase of said second phase signal shifted by a second phase shifter, said first phase signal phase-shifted by said first phase shifter and phase-shifted by said second phase shifter; By combining the second phase signal using a combiner, a phase modulation signal in which the carrier signal is modulated by the rotation angle of the rotor of the resolver is output, and the magnetic noise included in the phase modulation signal is output. is removed and the gain of the combiner is adjusted based on the rotational speed of the rotor.

According to the above embodiment, it is possible to provide a control device capable of detecting the rotor rotation angle with high accuracy and an error correction method thereof.

It is a block diagram which shows the structural example of the control apparatus based on the outline|summary of embodiment. 1 is a diagram showing a configuration example of a motor control system equipped with a control device according to Embodiment 1; FIG. FIG. 4 is a diagram showing a specific configuration example of a phase shifter; FIG. 3 is a diagram showing a first specific configuration example of variable resistors provided in a combiner; FIG. 10 is a diagram showing a second specific configuration example of variable resistors provided in the combiner; FIG. 4 is a diagram showing a waveform of a phase modulated signal before passing through a bandpass filter; FIG. 4 is a diagram showing a waveform of a phase modulated signal after passing through a bandpass filter; FIG. 5 is a diagram showing the relationship between the rotation speed of the rotor and the ratio of magnetic noise to the phase modulation signal; FIG. 4 is a diagram showing an example of information about the rotational speed of the rotor and the corresponding gain of the combiner stored in the memory; FIG. 4 is a diagram showing the relationship between a carrier signal and a phase modulated signal; FIG. 4 is a diagram showing the relationship between a carrier frequency clock signal and a shaped phase modulated signal; 3 is a diagram showing a modification of the motor control system shown in FIG. 2; FIG. FIG. 3 is a diagram showing a first modified example of a control device used in the motor control system shown in FIG. 2; 3 is a diagram showing a second modification of the control device used in the motor control system shown in FIG. 2; FIG. FIG. 9 is a diagram showing a configuration example of a motor control system equipped with a control device according to Embodiment 2; FIG. 4 is a diagram showing an example of a carrier signal and first and second phase signals when there is no angular error component; FIG. 4 is a diagram showing an example of a carrier signal and first and second phase signals when there is an angular error component; FIG. 10 is a diagram showing a configuration example of a motor control system equipped with a control device according to Embodiment 3; FIG. 19 is a flow chart showing a method of obtaining information on the rotational speed of the rotor and the corresponding gain of the combiner by the control device shown in FIG. 18;

For clarity of explanation, the following descriptions and drawings are omitted and simplified as appropriate. In addition, each element described in the drawings as a functional block that performs various processes can be configured by a CPU (Central Processing Unit), a memory, and other circuits in terms of hardware, and a memory in terms of software. implemented by a program loaded in the Therefore, those skilled in the art will understand that these functional blocks can be realized in various forms by hardware only, software only, or a combination thereof, and are not limited to either one. In each drawing, the same elements are denoted by the same reference numerals, and redundant description is omitted as necessary.

(Overview of embodiment)
FIG. 1 is a block diagram showing the configuration of a control device according to an outline of an embodiment. In FIG. 1 , the control device 1 includes phase shifters 101 and 102 , a combiner 103 , a bandpass filter (BPF) 329 and a gain adjustment circuit 141 .

The control device 1 outputs a sinusoidal carrier signal having a carrier frequency fc to an exciting coil arranged in the stator portion of the resolver. A magnetic field is generated in the resolver by being excited by a sinusoidal carrier signal supplied to the excitation coil. Here, the magnetic field generated in the resolver fluctuates according to the rotation of the rotor portion of the resolver. Two detection coils arranged in the stator portion of the resolver together with the excitation coil detect variations in the magnetic field generated in the resolver 201 and output them as two-phase signals.

The phase shifter 101 is configured to have a pole with a frequency f1 lower than the carrier frequency fc, and shifts the phase of the phase signal (first phase signal) detected by one of the detection coils. The phase shifter 102 is configured to have a pole with a frequency f2 higher than the carrier frequency fc, and shifts the phase of the phase signal (second phase signal) detected by the other detection coil.

The combiner 103 combines the first phase signal phase-shifted by the phase shifter 101 and the second phase signal phase-shifted by the phase shifter 102 . As a result, a signal (phase-modulated signal) obtained by phase-modulating the carrier signal by the rotor rotation angle is obtained.

The bandpass filter 329 attenuates (removes) magnetic noise included in the phase modulated signal and outputs the signal. The magnetic noise is a voltage signal induced in a coil provided in the resolver 201 due to leakage of magnetic flux generated in the motor 202 and interlinking with the coil. This magnetic noise increases in proportion to the increase in rotational speed of the rotor.

For example, the rotation angle of the rotor portion of the resolver (that is, the rotation angle of the motor) can be calculated from the phase difference between the phase modulated signal from which the magnetic noise has been removed and the carrier signal. For example, the control device 1 controls the motor based on the calculated rotor rotation angle.

A gain adjustment circuit 141 adjusts the gain of the combiner 103 based on the rotation speed of the rotor. For example, the gain adjustment circuit 141 adjusts the gain of the synthesizer 103 so that it decreases as the rotation speed of the rotor increases. As a result, even when the magnetic noise increases as the rotational speed of the rotor increases, the combined signal output from the combiner 103 is not saturated (clipped), thereby suppressing the generation of harmonic components. .

Also, the gain adjustment circuit 141 adjusts the gain of the combiner 103 so that it increases as the rotation speed of the rotor decreases. As a result, when the magnetic noise decreases as the rotational speed of the rotor increases, the gain of the combiner 103 increases. Deterioration of the ratio (Signal to Noise ratio) is suppressed.

As described above, the control device 1 according to the outline of the embodiment adjusts the gain of the combiner 103 based on the rotation speed of the rotor, thereby suppressing magnetic noise while preventing deterioration of the SN ratio and generation of harmonic components. By removing it, the rotor rotation angle can be detected with high accuracy.

<Embodiment 1>
In Embodiment 1, a detailed configuration of the control device 1 described in the outline of the embodiment and a motor control system using the control device 1 will be described.

FIG. 2 is a diagram showing a configuration example of a motor control system equipped with the control device 1 according to the first embodiment. The motor control system shown in FIG. 2 includes a control device 1 , a motor 202 and a resolver 201 . The control device 1 also includes an analog circuit 300 , a counter circuit 400 , a microcomputer controller 500 and a power circuit 600 .

The control device 1 calculates the rotation angle of the rotor portion of the resolver 201 attached to the rotating shaft of the motor 202 from the electrical signal output from the resolver 201, and controls the motor 202 based on the calculation result.

For example, one microprocessor chip is constructed by forming the microcomputer controller 500 and the counter circuit 400 on one semiconductor substrate. Also, one analog front-end chip is configured by forming the analog circuit 300 on one semiconductor substrate. A single semiconductor package is constructed by sealing the microprocessor chip and the analog front-end chip with resin. Alternatively, one chip is constructed by forming the microcomputer controller 500, the counter circuit 400 and the analog circuit 300 on one semiconductor substrate. Then, one semiconductor package is constructed by sealing this one chip with resin.

The resolver 201 is a so-called transformer type resolver, and includes a rotor 204 , a stator 205 , an excitation coil 206 , and detection coils 207 and 208 . The rotor 204 is fixed to the rotating shaft 203 of the motor 202 and rotates as the motor 202 rotates. Stator 205 is provided to surround rotor 204 .

The excitation coil 206 and detection coils 207 and 208 are both arranged in the stator 205 portion. A magnetic field is generated in the resolver 201 by being excited by a sinusoidal carrier signal supplied from the control device 1 to the exciting coil 206 . Here, the magnetic field generated in the resolver 201 fluctuates as the rotor 204 rotates. The detection coils 207 and 208 detect variations in the magnetic field generated in the resolver 201 and output them as two-phase resolver detection signals (phase signals).

The detection coils 207 and 208 are arranged in different directions. As a result, the detection coils 207 and 208 can output fluctuations in the magnetic field generated in the resolver 201 as two-phase resolver detection signals having different phases. In this embodiment, an example will be described in which the arrangement direction of the detection coil 207 and the arrangement direction of the detection coil 208 are different from each other by 90°. Therefore, the detection coil 207 outputs a sinusoidal first resolver detection signal, while the detection coil 208 outputs a cosine wave second resolver detection signal.

In the present embodiment, a case where resolver detection signals of two phases are output by the resolver 201 is described as an example, but the present invention is not limited to this, and resolver detection signals of two or more phases are output. The rotor rotation angle may be detected by the above resolver detection signal. Further, in the present embodiment, the case where the excitation coil 206 and the detection coils 207 and 208 are both arranged in the stator 205 portion is described as an example, but the present invention is not limited to this. For example, the excitation coil 206 may be arranged in the rotor 204 portion and the detection coils 207 and 208 may be arranged in the stator 205 portion.

(Configuration of Analog Circuit 300)
Next, the configuration of the analog circuit 300 will be described.
The analog circuit 300 includes an excitation circuit 301, an amplifier circuit 331, differential amplifier circuits 105 and 106, phase shifters 101 and 102, a combiner 103, a bandpass filter 329, and an operational amplifier OP330.

The excitation circuit 301 generates a sine wave carrier signal with a carrier frequency fc from a square wave clock signal with a carrier frequency fc.

The amplifier circuit 331 amplifies the sinusoidal carrier signal generated by the excitation circuit 301 . Specifically, the amplifier circuit 331 includes an operational amplifier OP302, transistors TR303 and TR304, and diodes D305 and D306. The operational amplifier OP302 amplifies the potential difference between the sine wave carrier signal and the output signal of the amplifier circuit 331, and outputs the amplified potential difference to the bases of the transistors TR303 and TR304. The transistors TR303 and TR304 are both bipolar transistors and are push-pull connected. Diodes D305 and D306 are provided in parallel with transistors TR303 and TR304, respectively. The amplifier circuit 331 outputs the voltage of the node between the transistors TR303 and TR304 to the exciting coil 206 as the output signal of the amplifier circuit 331 .

A differential amplifier circuit 105 amplifies the potential difference of the resolver detection signal (differential signal) detected by the detection coil 207 . Also, the differential amplifier circuit 106 amplifies the potential difference of the differential signal (resolver detection signal) detected by the detection coil 208 . Therefore, for example, the differential amplifier circuit 105 outputs a sine wave signal, while the differential amplifier circuit 106 outputs a cosine wave signal. The phase difference between the output signals of the differential amplifier circuits 105 and 106 is ideally 90°, but may be substantially 90° in the range of 88° to 92°, for example.

Specifically, the differential amplifier circuit 105 is composed of resistance elements R311 to R314 and an operational amplifier OP315. The resistance element R311 is provided between the inverting input terminal of the operational amplifier OP315 and one terminal of the detection coil 207 . The resistive element R312 is provided between the non-inverting input terminal of the operational amplifier OP315 and the other terminal of the detection coil 207. The resistance element R313 is provided between the inverting input terminal and the output terminal of the operational amplifier OP315. The resistance element R314 is provided between the non-inverting input terminal of the operational amplifier OP315 and the ground. The output signal of operational amplifier OP315 is used as the output signal of differential amplifier circuit 105 .

Further, the differential amplifier circuit 106 is composed of resistance elements R316 to R319 and an operational amplifier OP320. A resistive element R316 is provided between the inverting input terminal of the operational amplifier OP320 and one terminal of the detection coil 208 . The resistive element R317 is provided between the non-inverting input terminal of the operational amplifier OP320 and the other terminal of the detection coil 208. The resistance element R318 is provided between the inverting input terminal and the output terminal of the operational amplifier OP320. The resistance element R319 is provided between the non-inverting input terminal of the operational amplifier OP320 and the ground. The output signal of operational amplifier OP320 is used as the output signal of differential amplifier circuit 106 .

Phase shifter 101 is configured to have a pole of frequency f1 lower than carrier frequency fc, and shifts the phase of the output signal (first phase signal) of differential amplifier circuit 105 . Phase shifter 102 is configured to have a pole of frequency f2 higher than carrier frequency fc, and shifts the phase of the output signal (second phase signal) of differential amplifier circuit 106 .

(Specific configuration example of phase shifters 101 and 102)
FIG. 3 is a diagram showing a specific configuration example of the phase shifter 101. As shown in FIG.
As shown in FIG. 3, the phase shifter 101 is an all-pass filter and includes an operational amplifier OP701, resistance elements R702 to R704, and a capacitor C705.

The output signal (first phase signal) of the differential amplifier circuit 105 is input to the inverting input terminal of the operational amplifier OP701 via the resistance element R702. The output signal (first phase signal) of the differential amplifier circuit 105 is input to the non-inverting input terminal of the operational amplifier OP701 via the resistance element R703. A resistance element R704 is provided between the inverting input terminal and the output terminal of the operational amplifier OP701. A capacitor C705 is provided between the non-inverting input terminal of the operational amplifier OP701 and the ground. The output signal of operational amplifier OP701 is used as the output signal of phase shifter 101 .

Since the configuration of the phase shifter 102 is the same as that of the phase shifter 101, its description is omitted. The configuration of the phase shifters 101 and 102 is not limited to the configuration shown in FIG. 3, and can be appropriately changed to another configuration that can shift the phase by a desired shift amount.

Since the phase shift and poles are determinable by the transfer function of the all-pass filter, the impedance of resistive element R703 and the capacitance of capacitor C705 are determined according to the desired phase shift and poles.

Specifically, the carrier frequency fc, the pole frequency f1 of the phase shifter 101, and the pole frequency f2 of the phase shifter 102 are f1=fc/n and f2=fc×n (where n is any positive real number ) is satisfied, the difference between the phase shift amount of the phase shifter 101 and the phase shift amount of the phase shifter 102 can be set to 90°.

For example, by setting the impedance of the resistance elements R702 to R704 to 100 kΩ and the capacitance of the capacitor C705 to 80 pF, a pole of f1=1.99 kHz can be obtained. Also, by setting the impedance of the resistance elements R702 to R704 to 100 kΩ and the capacitance of the capacitor C705 to 135 pF, f2=11.8 kHz pole can be obtained.

f1 = 1.99 kHz and f2 = 11.8 kHz satisfy the relationship f1 = fc/n and f2 = fc x n (n is any positive real number) with respect to the carrier frequency fc = 4.88 kHz. , the difference between the phase shift amount of the phase shifter 101 and the phase shift amount of the phase shifter 102 is 90°. The difference between the phase shift amounts of the phase shifters 101 and 102 is ideally 90°, but may be approximately 90° in the range from 88° to 92°, for example.

Returning to FIG. 2, the description is continued.
The combiner 103 combines the first phase signal phase-shifted by the phase shifter 101 and the second phase signal phase-shifted by the phase shifter 102 . As a result, a signal (phase-modulated signal) obtained by phase-modulating the carrier signal by the rotor rotation angle is obtained.

Specifically, the combiner 103 is composed of an input resistor R325, an input resistor R326, a feedback resistor R328, and an operational amplifier OP327. The input resistors R325 and R326 are provided between the inverting input terminal of the operational amplifier OP327 and the output terminals of the phase shifters 101 and 102, respectively. A feedback resistor R328 is provided between the inverting input terminal and the output terminal of the operational amplifier OP327. A non-inverting input terminal of the operational amplifier OP327 is connected to the ground.

The operational amplifier OP327 amplifies the combined signal of the first phase signal supplied through the input resistor R325 and the second phase signal supplied through the input resistor R326. The output signal of the operational amplifier OP327 is fed back to the inverting input terminal of the operational amplifier OP327 via the feedback resistor R328. The output signal of operational amplifier OP327 is used as the output signal of combiner 103 .

Here, both the input resistors R325 and R326 are configured so that their resistance values can be adjusted based on the control signal from the gain adjustment circuit 141. FIG.

For example, the resistance values of the input resistors R325 and R326 are adjusted to increase as the rotation speed of the rotor 204 increases. As a result, the gain of combiner 103 is reduced. On the other hand, the resistance values of the input resistors R325 and R326 are adjusted to decrease as the rotational speed of the rotor 204 decreases. As a result, the gain of combiner 103 is increased.

(First specific configuration example of input resistors R325 and R326)
FIG. 4 is a diagram showing a first specific configuration example of the input resistor R325 as an input resistor R325a.

As shown in FIG. 4, the input resistor R325a has m (m is an integer equal to or greater than 2) switch elements SW1_1 to SW1_m and m resistor elements R1_1 to R1_m.

Resistive elements R1_1 to R1_m are connected in series between the output terminal of phase shifter 101 and the inverting input terminal of operational amplifier OP327. The switch element SW1_1 is provided between the inverting input terminal of the operational amplifier OP327 and one end of the resistance element R1_1. The switch elements SW1_2 to SW1_m are provided between the inverting input terminal of the operational amplifier OP327 and the node between the resistance elements R1_1 to R1_m, respectively. Here, the gain adjustment circuit 141 turns on one of the switch elements SW1_1 to SW1_m, and turns off the other switch elements. That is, the gain adjustment circuit 141 switches the resistance value of the input resistor R325a.

A first specific configuration example of the input resistor R326 is the same as the case of the input resistor R325a, so description thereof will be omitted.

(Second specific configuration example of input resistors R325 and R326)
FIG. 5 is a diagram showing a second specific configuration example of the input resistor R325 as an input resistor R325b.

As shown in FIG. 5, the input resistor R325b has a switch element SW2_1 and a resistance element R2_1.

The resistance element R2_1 is provided between the output terminal of the phase shifter 101 and the inverting input terminal of the operational amplifier OP327. The switch element SW2_1 is provided in parallel with the resistance element R2_1. Here, on/off of the switch element SW2_1 is switched with the duty ratio determined by the gain adjustment circuit 141 . That is, the gain adjustment circuit 141 switches the resistance value of the input resistor R325b.

A second specific configuration example of the input resistor R326 is the same as the case of the input resistor R325b, so description thereof will be omitted.

Returning to FIG. 2, the description is continued.
The bandpass filter 329 attenuates and outputs the phase modulated signal outside the predetermined frequency range. Specifically, the bandpass filter 329 attenuates (removes) the magnetic noise included in the phase modulated signal and outputs the signal.

FIG. 6 is a diagram showing the waveform of the phase modulated signal before passing through the bandpass filter 329. In FIG. FIG. 7 is a diagram showing the waveform of the phase modulated signal after passing through the bandpass filter 329. In FIG. As can be seen from FIGS. 6 and 7, the bandpass filter 329 removes the magnetic noise contained in the phase modulated signal.

The operational amplifier OP330 constitutes a comparator, and shapes the phase modulated signal supplied from the synthesizer 103 through the bandpass filter 329 into a square wave.

(Configuration of Counter Circuit 400)
Next, the configuration of the counter circuit 400 will be described.
The counter circuit 400 includes a reference CLK circuit 401 , an excitation CLK circuit 402 , a phase difference counter 404 and a CLK synchronization circuit 403 .

The reference CLK circuit 401 generates a reference clock signal with a reference frequency and outputs the generated reference clock signal to the excitation CLK circuit 402 , the CLK synchronization circuit 403 and the phase difference counter 404 .

The excitation CLK circuit 402 frequency-divides the reference clock signal generated by the reference CLK circuit 401 , and outputs a square-wave clock signal having a carrier frequency obtained by frequency division to the excitation circuit 301 and the phase difference counter 404 .

The CLK synchronization circuit 403 synchronizes the shaped phase modulated signal with the shaped carrier signal by detecting the shaped phase modulated signal in synchronization with the reference clock signal. A signal (detection signal) detected by the CLK synchronization circuit 403 is input to the phase difference counter 404 and the position calculator 501 .

Phase difference counter 404 counts the phase difference between the phase modulated signal obtained by synchronous detection and the carrier signal with the resolution of the reference frequency, and outputs the counting result to position calculator 501 and three-phase converter 509 .

(Configuration of microcomputer controller 500)
Next, the configuration of the microcomputer controller 500 will be described.
The microcomputer controller 500 includes a position calculator 501, a serial communication device 502, a subtractor 503, a position gain calculator 504, a differentiation processor 505, a subtractor 506, a velocity gain calculator 507, and a torque calculator. 508, a three-phase converter 509, multipliers 510 to 512, a gain adjustment circuit 141, and a memory 142.

The position calculator 501 calculates a position detection value from the detection signal and the phase difference count result, and outputs the position detection value to the subtractor 503 and the differentiation processor 505 .

Serial communication device 502 receives a position instruction signal from the outside and outputs a position command value to subtractor 503 . Subtractor 503 subtracts the position command value from the position detection value and outputs the obtained position deviation to position gain calculator 504 .

A position gain calculator 504 calculates a target speed of the motor 202 by multiplying the position deviation by a predetermined position gain. A differential processor 505 differentiates the detection signal representing the rotational position to calculate the rotational speed of the motor 202 . Subtractor 506 subtracts the rotation speed from the target speed and outputs the obtained speed deviation to speed gain calculator 507 .

A speed gain calculator 507 calculates a torque command value by multiplying the speed deviation by the speed gain. A torque calculator 508 calculates a current command value to be supplied to each phase of the motor 202 from the torque command value. Three-phase converter 509 generates a three-phase signal from the phase difference count result, and outputs the three-phase signal to multipliers 510 to 512, respectively.

Multipliers 510 to 512 each multiply the current command value by the three-phase signal to generate a three-phase control signal and output the three-phase control signal to power circuit 600 . The power circuit 600 is an inverter that performs three-phase PWM (Pulse Width Modulation) control of the motor 202 based on three-phase control signals.

The gain adjustment circuit 141 adjusts the gain of the synthesizer 103 based on the rotational speed of the motor 202 (in other words, the rotational speed of the rotor 204) calculated by the differentiation processor 505. FIG.

FIG. 8 is a diagram showing the relationship between the rotation speed of the rotor 204 and the ratio of magnetic noise to the phase modulation signal. As shown in FIG. 8, the magnetic noise increases in proportion to the increase in rotational speed of the rotor 204. FIG.

Therefore, the gain adjustment circuit 141 adjusts the gain of the synthesizer 103 so as to decrease as the rotation speed of the rotor 204 increases. Here, gain adjustment circuit 141 adjusts the gain of synthesizer 103 so that the phase modulated signal output from synthesizer 103 exhibits a value as large as possible within a range that does not saturate (clip). As a result, even when the magnetic noise increases as the rotational speed of the rotor 204 increases, the phase modulated signal output from the combiner 103 does not saturate (clip), thereby suppressing the generation of harmonic components. be done.

Also, the gain adjustment circuit 141 adjusts the gain of the combiner 103 so that it increases as the rotation speed of the rotor decreases. As a result, when the magnetic noise decreases as the rotational speed of the rotor increases, the gain of the combiner 103 increases. That is, deterioration of the SN ratio is suppressed.

Specifically, the gain adjustment circuit 141 reads information about the gain of the combiner 103 corresponding to the rotational speed of the rotor 204 from the memory 142 and adjusts the gain of the combiner 103 to the gain read from the memory 142 . .

The memory 142 stores information (table) of the rotational speed of the rotor 204 and the corresponding gain of the combiner 103, as shown in FIG. 9, for example. This information is pre-measured or calculated. Here, all of the gain values stored in the memory 142 are set to values as large as possible within a range in which the phase-modulated signal output from the combiner 103 is not saturated (not clipped). If the information on the gain of the combiner 103 corresponding to the actual rotation speed of the rotor 204 is not stored in the memory 142, the gain set in the combiner 103 is obtained by using the information stored in the memory 142, for example. is calculated by linearly interpolating Note that the gain adjustment circuit 141 may adjust the gain of the synthesizer 103 using an arithmetic expression instead of using the table.

In this way, the control device 1 adjusts the gain of the synthesizer 103 based on the rotation speed of the rotor 204, thereby preventing deterioration of the SN ratio and generation of harmonic components, and removing magnetic noise. Angles can be detected with high accuracy.

(Calculation method of rotor rotation angle)
Next, a method for calculating the rotor rotation angle will be described with reference to FIG.

First, the carrier signal of the sine wave sinωt generated by the excitation circuit 301 is amplified by the amplification circuit 331 and then input to the excitation coil 206 .

A magnetic field is generated in the resolver 201 by being excited by a carrier signal of a sine wave sinωt supplied from the control device 1 to the exciting coil 206 . Here, the magnetic field generated in the resolver 201 fluctuates as the rotor 204 rotates. The detection coils 207 and 208 detect variations in the magnetic field generated in the resolver 201 and output them as two-phase resolver detection signals with a phase difference of 90°.

A differential amplifier circuit 105 amplifies the potential difference of the resolver detection signal (differential signal) detected by the detection coil 207 . Also, the differential amplifier circuit 106 amplifies the potential difference of the differential signal (resolver detection signal) detected by the detection coil 208 .

Here, the output signal Z1 of the differential amplifier circuit 105 and the output signal Z2 of the differential amplifier circuit 106 are represented by the following equations (1) and (2), respectively.

Z1=K·sin θm×sin ωt (1)
Z2=K·cos θm×sin ωt (2)

is the angular frequency of the carrier signal, t is the time, .theta.m is the rotation angle of the rotor of the resolver 201, and K is the amplitude component of the resolver detection signal.

The phase shifter 101 is, for example, an op-pass filter, and shifts the phase of the output signal Z1 of the differential amplifier circuit 105 by a shift amount φ1. Phase shifter 102 is, for example, an op-pass filter, and shifts the phase of output signal Z2 of differential amplifier circuit 106 by shift amount φ2.

For example, phase shifters 101 and 102 are designed to satisfy φ1-φ2=-90°. Specifically, for example, for carrier frequency fc=4.88 kHz, by designing the pole frequency of phase shifter 101 to be 1.99 kHz and the pole frequency of phase shifter 102 to be 11.8 kHz, φ1−φ2=−90° can be satisfied.

At this time, the output signal X1 of the phase shifter 101 and the output signal X2 of the phase shifter 102 are represented by the following equations (3) and (4), respectively.

X1=K·sinθm×sin(ωt−φ1) (3)
X2=K·cos θm×sin(ωt−φ2) (4)

Here, by substituting φ1=φ2−90° into the formula (3), X1 is expressed as the following formula (5).

X1=K・sinθm×sin(ωt+90°−φ2)
=K sin θm×cos(ωt−φ2) (5)

The synthesizer 103 synthesizes the output signal X1 of the phase shifter 101 and the output signal X2 of the phase shifter 102 and outputs the result. Here, the output signal Y of the synthesizer 103 is represented by the following equation (6) from equations (4) and (5).

Y=X1+X2
= K·sin θm×cos(ωt−φ2)+K・cos θm×sin(ωt−φ2)
=K sin(ωt-φ2+θm) (6)

For example, when φ1=−90° and φ2=0°, the output signal Y of the combiner 103 is represented by the following formula (7) from formula (6).

Y=K·sin(ωt+θm) (7)

That is, the synthesizer 103 generates a signal (phase modulated signal) in which the carrier signal is phase-modulated by the rotor rotation angle θm.

FIG. 10 is a diagram showing the relationship between carrier signals and phase modulation signals. In FIG. 10, the vertical axis indicates amplitude and the horizontal axis indicates time. As shown in FIG. 10, the carrier signal and the phase modulated signal have the same frequency but different phases. Therefore, it is possible to calculate the rotation angle of the rotor from the phase difference between the carrier signal and the phase modulated signal.

Specifically, it is preferable to detect the phase difference between the clock signal of the carrier frequency and the phase modulated signal shaped into a square wave, and calculate the rotation angle of the rotor from the detected phase difference.

FIG. 11 is a diagram showing the relationship between a carrier frequency clock signal and a shaped phase modulated signal. In FIG. 11, the vertical axis indicates amplitude and the horizontal axis indicates time. As shown in FIG. 11, there is a phase difference between the carrier frequency clock signal and the phase modulated signal after shaping according to the rotation angle of the rotor.

In the counter circuit 400, the CLK synchronizing circuit 403 detects the signal (shaped phase-modulated signal) obtained by phase-modulating the carrier frequency ω with the rotor rotation angle θm in synchronization with the reference clock signal. synchronizes the signal phase-modulated by the rotor rotation angle θm with the clock signal of the carrier frequency ω. A phase difference counter 404 counts the phase difference between the phase modulation signal and the carrier signal obtained by synchronous detection with the resolution of the reference frequency.

Microcomputer controller 500 and power circuit 600 control motor 202 based on the phase difference between the phase modulation signal and the carrier signal counted in counter circuit 400 .

According to the above-described method of calculating the rotor rotation angle, the phase shifter 101 having a pole with a frequency f1 lower than the carrier frequency fc shifts the phase of the output signal (first phase signal) of the differential amplifier circuit 105, and the carrier frequency Phase shifter 102 having a pole of frequency f2 higher than fc shifts the phase of the output signal (second phase signal) of differential amplifier circuit 106 . Then, the phase-shifted first phase signal and the phase-shifted second phase signal are synthesized. As a result, the rotor rotation angle can be detected with high accuracy, and an increase in circuit scale can be suppressed.

Also, the gain adjustment circuit 141 adjusts the gain of the synthesizer 103 based on the rotational speed of the motor 202 (in other words, the rotational speed of the rotor 204) calculated by the differentiation processor 505. FIG. Specifically, the gain adjustment circuit 141 adjusts the gain of the combiner 103 so as to decrease as the rotational speed of the rotor 204 increases. Here, gain adjustment circuit 141 adjusts the gain of synthesizer 103 so that the phase modulated signal output from synthesizer 103 exhibits a value as large as possible within a range that does not saturate (clip). As a result, even when the magnetic noise increases as the rotation speed of the rotor 204 increases, the phase modulated signal output from the combiner 103 does not stick to the saturation voltage (clip), so harmonic components are reduced. occurrence is suppressed.

Also, the gain adjustment circuit 141 adjusts the gain of the combiner 103 so that it increases as the rotation speed of the rotor decreases. As a result, when the magnetic noise decreases as the rotational speed of the rotor increases, the gain of the combiner 103 increases. That is, deterioration of the SN ratio is suppressed.

As described above, the control device 1 according to the present embodiment adjusts the gain of the combiner 103 based on the rotation speed of the rotor 204, thereby suppressing the magnetic noise while preventing deterioration of the SN ratio and generation of harmonic components. By removing it, the rotor rotation angle can be detected with high accuracy.

(Configuration example of a motor control system using a current detection resolver)
Although the resolver 201 is a transformer type resolver in the present embodiment, the resolver is not limited to this, and may be, for example, a current detection type resolver. A specific description will be given below with reference to FIG. 12 .

FIG. 12 is a diagram showing a modification of the motor control system shown in FIG. In the motor control system shown in FIG. 2, a transformer type resolver 201 was used. In contrast, the motor control system shown in FIG. 12 uses a current detection resolver 201a. The rest of the configuration of the motor control system shown in FIG. 12 is the same as that of the motor control system shown in FIG. 2, so description thereof is omitted.

The current detection resolver 201a includes a rotor 204, a stator 205, coils L1 to L4, and resistance elements R1 to R4. The rotor 204 is fixed to the rotating shaft 203 of the motor 202 and rotates as the motor 202 rotates. Stator 205 is provided to surround rotor 204 .

The coils L1 to L4 are arranged at positions of 0°, 90°, 180°, and 270°, respectively, about the rotating shaft 203 of the motor 202 in the stator 205 portion. The coils L1 to L4 are excited by a sinusoidal carrier signal supplied from the control device 1 to generate a magnetic field. Here, the magnetic fields generated in the coils L1 to L4 fluctuate as the rotor portion of the resolver 201 rotates. Therefore, the currents flowing through the coils L1 to L4 also fluctuate as the rotor portion of the resolver 201 rotates.

The resistance elements R1 to R4 respectively convert the currents flowing through the coils L1 to L4 into voltages and output them as resolver detection signals of 0°, 90°, 180° and 270° phases.

Specifically, the resistive element R1 is provided in series with the coil L1, and outputs the voltage at the node N1 between the resistive element R1 and the coil L1 as a 0° phase resolver detection signal. The resistance element R2 is provided in series with the coil L2, and outputs the voltage at the node N2 between the resistance element R2 and the coil L2 as a 90° phase resolver detection signal. The resistance element R3 outputs the voltage of the node N3 between the resistance element R3 and the coil L3 as a 180° phase resolver detection signal. The resistance element R4 is provided in series with the coil L4, and outputs the voltage at the node N4 between the resistance element R4 and the coil L4 as a 270° phase resolver detection signal.

The differential amplifier circuit 105 amplifies the potential difference between the resolver detection signals obtained by converting the currents flowing through the coils L1 and L3 into voltages using the resistance elements R1 and R3. Further, the differential amplifier circuit 106 amplifies the potential difference between the resolver detection signals obtained by converting the currents flowing through the coils L2 and L4 into voltages using the resistance elements R2 and R4. Therefore, for example, the differential amplifier circuit 105 outputs a sine wave signal, while the differential amplifier circuit 106 outputs a cosine wave signal. The phase difference between the output signals of the differential amplifier circuits 105 and 106 is ideally 90°, but may be substantially 90° in the range of 88° to 92°, for example.

In this manner, the control device 1 can achieve the same effect as when the resolver 201 is of the current detection type and of the transformer type.

Further, in the present embodiment, the case where the gain of the combiner 103 is adjusted by adjusting the resistance values of the input resistors R325 and R326 provided in the combiner 103 has been described as an example, but the present invention is not limited to this. do not have. A modification of the synthesizer 103 will be described below with reference to FIGS. 13 and 14. FIG.

(First modification of synthesizer 103)
FIG. 13 is a diagram showing a first modification of the control device 1 as a control device 1a. Compared to the control device 1 , the control device 1 a comprises a combiner 103 a which is a first modification of the combiner 103 . In the combiner 103a, the resistance value of the feedback resistor R328 can be adjusted by the gain adjustment circuit 141 instead of the input resistors R325 and R326. Since the specific configuration of the feedback resistor R328 is the same as the configuration shown in FIGS. 4 and 5, its description is omitted.

For example, the resistance value of the feedback resistor R328 is adjusted to decrease as the rotation speed of the rotor 204 increases. As a result, the gain of combiner 103 is reduced. On the other hand, the resistance value of the feedback resistor R328 is adjusted to increase as the rotation speed of the rotor 204 decreases. As a result, the gain of combiner 103 is increased.

The synthesizer 103a can implement functions similar to those of the synthesizer 103. FIG.

(Second modification of synthesizer 103)
FIG. 14 is a diagram showing a second modification of the control device 1 as a control device 1b. The control device 1b, compared to the control device 1, comprises a combiner 103b, which is a second modification of the combiner 103. FIG. In the combiner 103b, the gain adjustment circuit 141 can adjust the resistance value of the feedback resistor R328 in addition to the input resistors R325 and R326.

For example, as the rotation speed of the rotor 204 increases, the resistance values of the input resistors R325 and R326 are adjusted to increase, and the resistance value of the feedback resistor R328 is adjusted to decrease. As a result, the gain of combiner 103 is reduced. On the other hand, as the rotation speed of the rotor 204 decreases, the resistance values of the input resistors R325 and R326 are adjusted to decrease, and the resistance value of the feedback resistor R328 is adjusted to increase. As a result, the gain of combiner 103 is increased.

The synthesizer 103b can realize a function equivalent to that of the synthesizer 103. FIG. Furthermore, the synthesizer 103b can adjust the resistance values of the input resistor and the feedback resistor, so that the gain can be adjusted more finely.

<Embodiment 2>
FIG. 15 is a diagram showing a configuration example of a motor control system equipped with the control device 2 according to the second embodiment. Compared to the control device 1, the control device 2 further includes a correction circuit 143, an AD converter 144, a DA converter 145, resistance elements R146 and R148, and a capacitor C147. In the example of FIG. 15, the correction circuit 143 is provided in the microcomputer controller 500, the AD converter 144 and the DA converter 145 are provided in the counter circuit 400, and the resistance elements R146 and R148 and the capacitor C147 are provided in the analog circuit 300. is provided in

(Description of angle error component of resolver detection signal)
Before describing the details of the control device 2, first, the angular error component included in the resolver detection signal will be described.

FIG. 16 is a diagram showing an example of the carrier signal and the first and second phase signals (the respective output signals of the differential amplifier circuits 105 and 106) when there is no angular error component. In FIG. 16, the vertical axis indicates amplitude and the horizontal axis indicates time.

As shown in FIG. 16 , when there is no angular error component, the first phase signal is a sinusoidal phase-modulated signal in which the carrier signal is phase-modulated by the rotor rotation angle of the resolver 201 . The second phase signal is a cosine wave phase-modulated signal obtained by phase-modulating the carrier signal with the rotor rotation angle of the resolver 201 . Here, the envelopes of the first and second phase signals both form noiseless waveforms.

On the other hand, for example, if there is a winding variation in the resolver 201, the fixed-phase carrier signal remains as an angular error component (residual component) in the phase modulated signal. FIG. 17 is a diagram showing an example of the carrier signal and the first and second phase signals when there is an angular error component. In FIG. 17, the vertical axis indicates amplitude and the horizontal axis indicates time.

As shown in FIG. 17 , when there is an angular error component, the first phase signal is a sinusoidal signal obtained by phase-modulating the carrier signal containing the angular error component with the rotor rotation angle of the resolver 201 . The second phase signal is a cosine wave signal obtained by phase-modulating the carrier signal including the angle error component with the rotor rotation angle of the resolver 201 . That is, when there is an angular error component, the first and second phase signals include the angular error component phase-modulated by the rotor rotation angle.

Here, when there is an angle error component, the output signal (first phase signal) Z1 of the differential amplifier circuit 105 and the output signal (second phase signal) Z2 of the differential amplifier circuit 106 are obtained by the following equations. (8) and (9).

Z1=K·(α+sinθm)×sinωt (8)
Z2=K·(β+cos θm)×sin ωt (9)

is the angular frequency of the carrier signal, t is the time, .theta.m is the rotation angle of the rotor of the resolver 201, K is the amplitude component of the resolver detection signal, and α and .beta.

At this time, the output signal X1 of the phase shifter 101 and the output signal X2 of the phase shifter 102 are represented by the following equations (10) and (11), respectively.

X1=K·(α+sinθm)×sin(ωt−φ1) (10)
X2=K·(β+cosθm)×sin(ωt−φ2) (11)

Here, by substituting φ1=φ2−90° into the formula (10), X1 is expressed as the following formula (12).

X1=K・(α+sinθm)×sin(ωt+90°−φ2)
=K·(α+sinθm)×cos(ωt−φ2) (12)

The synthesizer 103 synthesizes the output signal X1 of the phase shifter 101 and the output signal X2 of the phase shifter 102 and outputs the result. Here, the output signal Y of the synthesizer 103 is represented by the following equation (13) from equations (11) and (12).

・・・（１３） Y=X1+X2
=K・(α+sinθm)×cos(ωt−φ2)+K・(β+cosθm)×sin(ωt−φ2)
=

(13)

For example, when φ1=−90° and φ2=0°, the output signal Y of the combiner 103 is represented by the following formula (14) from formula (13).

・・・（１４）

(14)

The second term on the right side of Equation (13) corresponds to the angular error component. A correction circuit 143, which will be described later, corrects the phase modulation signal so that the value of the second term on the right side of Equation (13) approaches zero.

Returning to FIG. 15, the description continues.
The AD converter 144 converts the phase modulated signal output from the synthesizer 103 into a digital signal and outputs the digital signal. A correction circuit 143 generates a correction signal whose phase and amplitude can be changed arbitrarily. The DA converter 145 converts the digital correction signal into an analog signal.

The resistance elements R146 and R148 are provided in series between the output terminal of the DA converter 145 and the inverting input terminal of the operational amplifier OP327. The capacitor C147 is provided between the node between the resistance elements R146 and R148 and the ground. Here, a low-pass filter is configured by the resistance element R146 and the capacitor C147.

The correction signal converted to analog by the DA converter 145 propagates through a low-pass filter composed of a resistance element R146 and a capacitor, and a resistance element R148, and is fed back to the inverting input terminal of the operational amplifier OP327.

Here, the correction circuit 143 arbitrarily adjusts the amplitude and phase of the correction signal so that the fluctuation width (amplitude) of the envelope of the phase modulated signal digitized by the AD converter 144 is minimized. That is, the correction circuit 143 generates an amplitude and phase correction signal that cancels out the angular error component contained in the phase modulation signal.

Other configurations and operations of the control device 2 are the same as those of the control device 1, so description thereof will be omitted.

Accordingly, the control device 2 uses the correction circuit 143 to generate an amplitude and phase correction signal that cancels out the angular error component contained in the phase modulated signal, and superimposes it on the phase modulated signal. As a result, the control device 2 can suppress the angle error component included in the phase modulated signal, so that the detection accuracy of the rotor rotation angle can be further improved.

The resistance element R148 is configured so that the resistance value thereof can be adjusted by the gain adjustment circuit 141 together with the input resistors R325 and R326. As a result, restrictions on gain adjustment are relaxed when the resistance value of the resistance element R148 is fixed.

In this embodiment, the control device 2 to which the correction of the phase modulated signal by the correction circuit 143 is applied to the control device 1 has been described, but the present invention is not limited to this. The correction of the phase modulated signal by the correction circuit 143 may be applied to the control devices 1a and 1b, for example.

<Embodiment 3>
FIG. 18 is a diagram showing a configuration example of a motor control system equipped with the control device 3 according to the third embodiment. The control device 3 further has a function of automatically calculating the gain of the combiner 103 according to the rotation speed of the rotor 204, compared with the case of the control device 1. FIG.

Specifically, the control device 3 further includes a monitor circuit 149 . The monitor circuit 149 is, for example, an AD conversion circuit, and monitors the output signal (phase modulation signal) of the synthesizer 103 when the rotor 204 is rotated at a predetermined rotational speed. The gain adjustment circuit 141 monitors the output signal of the combiner 103 by the monitor circuit 149 and searches for the gain of the combiner 103 so that the output signal of the combiner 103 exhibits a value as large as possible within a range in which the output signal of the combiner 103 does not clip. Then, the gain adjustment circuit 141 stores in the memory 142 the result of the search, that is, information on the rotational speed of the rotor 204 and the corresponding gain of the synthesizer 103 . By repeating such operations, the gain adjustment circuit 141 acquires a plurality of pieces of information on the rotational speed of the rotor 204 and the corresponding gain of the synthesizer 103 and stores them in the memory 142 .

FIG. 19 is a flowchart showing a method for acquiring information on the rotation speed of the rotor 204 and the corresponding gain of the synthesizer 103 by the control device 3 .

First, a threshold voltage VT is set for the output signal of the synthesizer 103 (step S101). The threshold voltage VT is set to the maximum voltage value that is allowed as the output signal of the synthesizer 103 . Specifically, the threshold voltage VT is set to a value slightly lower than the saturation voltage. For example, if the saturation voltage (limit voltage) is 4.5V, the threshold voltage VT is set to 4V. For example, the monitor circuit 149 outputs a monitor result of "1" when the output voltage of the combiner 103 is higher than the threshold voltage VT, and outputs "0" when the output voltage of the combiner 103 is below the threshold voltage VT. ” monitor result.

After that, the initial value of the gain of the synthesizer 103 is set (step S102). For the initial value of the gain, for example, the minimum value of the gain that the synthesizer 103 can take is set.

After that, the rotor 204 is rotated at a predetermined rotational speed (step S103). Here, for example, the rotor 204 is rotated at a rotation speed of 0 [r/min]. The monitor circuit 149 monitors the output signal of the synthesizer 103 at this time (step S104).

For example, when the monitor result of “0” is output from the monitor circuit 149 (NO in step S105), the output signal of the combiner 103 is equal to or lower than the threshold voltage VT. is increased by one step (step S106). It should be noted that the smaller the amount of change in the gain of the synthesizer 103 at this time, the more accurately the gain information can be obtained. Thereafter, monitor circuit 149 monitors the output signal of synthesizer 103 again (step S104).

On the other hand, when the monitor result of "1" is output from the monitor circuit 149 (YES in step S105), the output signal of the synthesizer 103 is higher than the threshold voltage VT. (or set at that time) is stored in the memory 142 together with the corresponding rotational speed information of the rotor 204 (step S107). If the monitor circuit 149 continues to output a monitor result of “0” even if the gain of the synthesizer 103 is maximized, the gain adjustment circuit 141 adjusts the maximum value of the gain to the rotational speed of the corresponding rotor 204. (YES in step S105→step S107).

After that, if acquisition of all the gain information of the multiple combiners 103 corresponding to the multiple rotor rotation speeds to be searched has not been completed (NO in step S108), the rotation speed of the rotor 204 is The rotation speed is switched to another rotation speed (for example, 500 [r/min]) that has not been selected yet (step S109). After that, the processes of steps S103 to S109 are repeated.

After that, when acquisition of all the gain information of the plurality of combiners 103 corresponding to the plurality of rotor rotational speeds to be searched is completed (YES in step S108), the controller 3 controls the rotation of the rotor 204. Acquisition of velocity and corresponding combiner 103 gain information ends.

As described above, the control device 3 according to the present embodiment can achieve effects equivalent to those of the control devices 1, 2, etc., and even if the identity of the resolver is unknown, the synthesized signal corresponding to the rotation speed of the rotor 204 The gain of the unit 103 can be automatically calculated.

As described above, the control apparatus according to the first to third embodiments adjusts the gain of the combiner 103 based on the rotational speed of the rotor 204, thereby preventing deterioration of the SN ratio and generation of harmonic components. Magnetic noise can be removed and the rotor rotation angle can be detected with high accuracy.

In each of the above embodiments, the counter circuit 400 and the microcomputer controller 500 can be implemented by hardware such as ASIC (Application Specific Integrated Circuit) or software. Also, part of the processing may be implemented by software, and the rest may be implemented by hardware. When implemented in software, a computer system having one or more CPUs such as microprocessors may be caused to execute programs relating to the processing of the functional blocks.

Also, the programs described above can be stored and supplied to computers using various types of non-transitory computer readable media. Non-transitory computer-readable media include various types of tangible storage media. Examples of non-transitory computer-readable media include magnetic recording media (e.g., flexible discs, magnetic tapes, hard disk drives), magneto-optical recording media (e.g., magneto-optical discs), CD-ROMs (Read Only Memory), CD-Rs, CD-R/W, semiconductor memory (eg, mask ROM, PROM (Programmable ROM), EPROM (Erasable PROM), flash ROM, RAM (Random Access Memory)). The program may also be supplied to the computer on various types of transitory computer readable medium. Examples of transitory computer-readable media include electrical signals, optical signals, and electromagnetic waves. Transitory computer-readable media can deliver the program to the computer via wired channels, such as wires and optical fibers, or wireless channels.

The invention made by the present inventor has been specifically described above based on the embodiments, but the present invention is not limited to the embodiments already described, and various modifications can be made without departing from the scope of the invention. It goes without saying that this is possible.

For example, neither the carrier frequency nor the amount of phase shift are limited to the numerical values in the above embodiments.

Further, for example, in the control device according to the first to fourth embodiments and the motor control system including the same, the conductivity type (p-type or n-type) of the semiconductor substrate, the semiconductor layer, the diffusion layer (diffusion region), etc. An inverted configuration may also be used. Therefore, when one conductivity type of the n-type and the p-type is defined as the first conductivity type and the other conductivity type is defined as the second conductivity type, the first conductivity type is the p-type, and the second conductivity type is the second conductivity type. It can be n-type, or conversely, the first conductivity type can be n-type and the second conductivity type can be p-type.

1 control device 1a, 1b control device 2 control device 3 control device 101 phase shifter 102 phase shifter 103 combiner 105 differential amplifier circuit 106 differential amplifier circuit 141 gain adjustment circuit 142 memory 143 correction circuit 144 AD converter 145 DA converter 149 monitor circuit 201 resolver 201a current detection resolver 202 motor 203 rotating shaft 204 rotor 205 stator 206 excitation coil 207 detection coil 208 detection coil 300 analog circuit 301 excitation circuit 329 bandpass filter 331 amplifier circuit 400 counter circuit 401 reference CLK circuit 402 Excitation CLK circuit 403 CLK synchronization circuit 404 Phase difference counter 500 Microcomputer controller 501 Position calculator 502 Serial communication device 503 Subtractor 504 Position gain calculator 505 Differential processor 506 Subtractor 507 Velocity gain calculator 508 Torque calculator 509 Three-phase Converter 510 Multiplier 511 Multiplier 512 Multiplier 600 Power circuit C147 Capacitor C705 Capacitor D305 Diode D306 Diode L1 to L4 Coil OP302 Operational amplifier OP315 Operational amplifier OP320 Operational amplifier OP327 Operational amplifier OP330 Operational amplifier OP701 Operational amplifier R1 to R_1 Resistance element R_1 to R_1 Resistance element Elements R146, R148 Resistance elements R311 to R314 Resistance elements R316 to R319 Resistance elements R325 Input resistance R325a Input resistance R325b Input resistance R326 Input resistance R328 Feedback resistance R702 Resistance element R703 Resistance element R704 Resistance elements SW1_1 to SW1_m Switch element SW2_1 Switch element TR303 TR304 transistor

Claims (17)

a first phase shifter for shifting the phase of the first phase signal among the first and second phase signals with different phases detected from the resolver excited by the carrier signal;
a second phase shifter for shifting the phase of the second phase signal;
By synthesizing the first phase signal phase-shifted by the first phase shifter and the second phase signal phase-shifted by the second phase shifter, the carrier signal rotates the rotor of the resolver. a combiner that outputs a phase-modulated signal modulated by angle;
a bandpass filter that removes magnetic noise contained in the phase modulated signal;
a gain adjustment circuit that adjusts the gain of the combiner based on the rotation speed of the rotor;
with
The gain adjustment circuit is configured to adjust the gain of the synthesizer so that the phase-modulated signal output from the synthesizer exhibits a maximum value within a predetermined range that does not saturate.
Control device. The gain adjustment circuit is configured to decrease the gain of the combiner as the rotation speed of the rotor increases, and increase the gain of the combiner as the rotation speed of the rotor decreases.
A control device according to claim 1 . The combiner is
a first input resistor configured to be adjustable in resistance value by the gain adjustment circuit, one end of which is supplied with the output signal of the first phase shifter;
a second input resistor configured to be adjustable in resistance value by the gain adjustment circuit, one end of which is supplied with the output signal of the second phase shifter;
an operational amplifier for amplifying a combined signal of signals output from the other ends of the first and second input resistors;
a feedback resistor that feeds back the output signal of the operational amplifier to the input of the operational amplifier;
having
A control device according to claim 1 . The combiner is
a first input resistor, one end of which is supplied with the output signal of the first phase shifter;
a second input resistor, one end of which is supplied with the output signal of the second phase shifter;
an operational amplifier for amplifying a combined signal of signals output from the other ends of the first and second input resistors;
a feedback resistor whose resistance value can be adjusted by the gain adjustment circuit and feeds back the output signal of the operational amplifier to the input of the operational amplifier;
having
A control device according to claim 1 . The combiner is
a first input resistor configured to be adjustable in resistance value by the gain adjustment circuit, one end of which is supplied with the output signal of the first phase shifter;
a second input resistor configured to be adjustable in resistance value by the gain adjustment circuit, one end of which is supplied with the output signal of the second phase shifter;
an operational amplifier for amplifying a combined signal of signals output from the other ends of the first and second input resistors;
a feedback resistor whose resistance value can be adjusted by the gain adjustment circuit and feeds back the output signal of the operational amplifier to the input of the operational amplifier;
has a
A control device according to claim 1 . further comprising a memory storing information of the rotation speed of the rotor and the corresponding gain of the combiner;
A control device according to claim 1 . further comprising a monitor circuit for monitoring the output signal of the combiner;
The gain adjustment circuit is configured to acquire information on the gain of the combiner corresponding to the rotation speed of the rotor using the monitor result of the monitor circuit.
A control device according to claim 1 . further comprising a memory storing information about the rotational speed of the rotor and the corresponding gain of the combiner obtained by the gain adjustment circuit;
A control device according to claim 7 . a correction circuit that generates a correction signal;
a resistive element through which the correction signal propagates;
The combiner is configured to output the phase-modulated signal by combining the correction signal supplied through the resistive element in addition to the first phase signal and the second phase signal,
The correction circuit is configured to generate the correction signal having an amplitude and phase that minimizes the fluctuation width of the envelope of the phase modulated signal,
The resistance element is configured so that the resistance value can be adjusted by the gain adjustment circuit,
A control device according to claim 1 . The phase difference between the first and second phase signals is approximately 90°,
The first and second phase shifters are configured so that the difference in the amount of phase shift by each is approximately 90°.
A control device according to claim 1 . the first phase shifter is configured to have a pole at a first frequency lower than the frequency of the carrier signal;
the second phase shifter is configured to have a pole at a second frequency higher than the frequency of the carrier signal;
A control device according to claim 1 . The first and second phase shifters are configured to set the frequency of the carrier signal to fc, the first frequency lower than the frequency of the carrier signal to f1, and the second frequency higher than the frequency of the carrier signal to f2. is configured to satisfy f1 = fc / n and f2 = fc × n, where n is a positive real number of
Control device according to claim 11 . A controller that detects the rotation angle of the rotor of the resolver based on the phase difference between the phase modulation signal and the carrier signal and controls the motor based on the detection result,
A control device according to claim 1 . a motor;
a resolver having a rotor attached to the rotating shaft of the motor;
14. The control device according to claim 13 , which controls the motor based on the rotation angle of the resolver;
A motor control system with Shifting the phase of the first phase signal by a first phase shifter among first and second phase signals with different phases detected from a resolver excited by a carrier signal,
shifting the phase of the second phase signal by a second phase shifter;
By combining the first phase signal phase-shifted by the first phase shifter and the second phase signal phase-shifted by the second phase shifter using a combiner, the carrier signal is the outputting a phase modulated signal modulated by the rotation angle of the rotor of the resolver;
removing magnetic noise contained in the phase-modulated signal;
adjusting the gain of the combiner based on the rotational speed of the rotor;
An error correction method for a control device,
adjusting the gain of the synthesizer so that the phase modulated signal output from the synthesizer exhibits a maximum value within a predetermined range without saturation;
Error correction method for controller. The gain of the combiner is decreased as the rotation speed of the rotor increases, and the gain of the combiner is increased as the rotation speed of the rotor decreases.
16. The error correction method for a control device according to claim 15 . monitoring the output signal of the combiner;
Acquiring gain information of the combiner corresponding to the rotation speed of the rotor using the monitored result,
Adjusting the gain of the combiner so as to correspond to the rotational speed of the rotor using the obtained result;
16. The error correction method for a control device according to claim 15 .

Images