JP6599024B2 - Power factor compensation power supply device and LED lighting device - Google Patents

Description

  The present invention relates to a power factor compensation power supply apparatus and an LED lighting apparatus that have a power factor correction (PFC) function and convert AC power into DC power.

  Conventionally, a power factor correction circuit comprising a DC / DC converter connected at the subsequent stage of a full-wave rectifier circuit is detected, and phase information of the input voltage after full-wave rectification is detected with high accuracy. A switching power supply device has been proposed in which the ON width of the switching elements constituting the DC / DC converter is widened to prevent the power factor from being lowered when it is low (see, for example, Patent Document 1 below).

JP2013-243798A

  In the conventional device of Patent Document 1 described above, an input filter is provided for the purpose of suppressing switching noise generated by switching on / off of the switching element to the power system on the input side. Due to the influence of the above, there is a problem that a phase difference which becomes a leading phase with respect to the input voltage occurs in the input current, and the power factor deteriorates.

  The present invention has been made to solve the above-described problem, and is a power factor compensating power source capable of achieving a high power factor by controlling the phase difference between the input current and the input voltage to be zero. An object is to provide a device and an LED lighting device.

A power factor compensation power supply device according to the present invention includes a power supply main circuit unit and a power supply control unit that controls the power supply main circuit unit,
The power supply main circuit section includes a full-wave rectifier circuit that full-wave rectifies an AC voltage of an AC power supply, an inductance element, and a switching element, and an output voltage that targets an input voltage obtained by the full-wave rectifier circuit. An input filter configured to include an input capacitor for suppressing outflow of a noise component due to the switching element to the AC power supply, an input voltage detection unit that detects the input voltage, and the output An output voltage detection unit for detecting the voltage,
The power supply control unit adjusts the phase of the input voltage from the signal detected by the input voltage detection unit with respect to a reference on-time of the switching element determined based on the signal detected by the output voltage detection unit. The output voltage is controlled to a desired voltage by turning on and off the switching element using the on-time of the switching element that is detected and corrected by multiplying the correction signal generated based on the phase. However, power factor compensation control is performed on the input current from the AC power source.

The LED lighting device according to the present invention includes a power supply main circuit unit and a power supply control unit that controls the power supply main circuit unit, and an LED module is connected to an output side of the power supply main circuit unit,
The power supply main circuit section includes a full-wave rectifier circuit that full-wave rectifies an AC voltage of an AC power supply, an inductance element, and a switching element, and an output current that targets an input current obtained by the full-wave rectifier circuit. An input filter configured to include an input capacitor for suppressing the outflow of noise components from the switching element to the AC power supply, and an input for detecting an input voltage obtained by the full-wave rectifier circuit A voltage detection unit, and an output current detection unit for detecting the output current,
The power supply control unit adjusts the phase of the input voltage from the signal detected by the input voltage detection unit with respect to the reference on-time of the switching element determined based on the signal detected by the output current detection unit. The output current is controlled to a desired current by turning on and off the switching element using the on-time of the switching element that is detected and corrected by multiplying the correction signal generated based on the phase. On the other hand, power factor compensation control is performed on the input current from the AC power source.

  In addition, the LED lighting device according to the present invention includes the power factor compensation power supply device having the above-described configuration, and an LED module is connected to the output side of the power supply main circuit unit, and between the converter and the LED module. Is provided with an LED current adjusting circuit for adjusting the current flowing through the LED module to a target current.

  According to the present invention, even when an input capacitor is provided in order to suppress switching noise, the phase information of the input voltage is detected, and the ON width of the switching element is adjusted as needed with the correction amount corresponding to the detected input voltage phase. Correction is performed to thereby bring the phase difference between the input current and the input voltage close to zero, so that a high power factor can be achieved.

It is a block diagram which shows the structure of the power factor compensation power supply apparatus in Embodiment 1 of this invention. It is explanatory drawing regarding operation | movement of the power supply control part in Embodiment 1 of this invention. It is explanatory drawing regarding operation | movement of the input voltage phase detection part in Embodiment 1 of this invention. It is explanatory drawing regarding operation | movement of the switch control part in Embodiment 1 of this invention. It is explanatory drawing of the correction amount calculation process of the switch control part in Embodiment 1 of this invention. It is a figure which shows the structural example of the input coil in Embodiment 1 of this invention. It is a wave form diagram which shows the simulation result of the reference example which does not apply this invention. It is a wave form diagram which shows the simulation result of Embodiment 1 of this invention. It is a wave form diagram which shows the state of the alternating current input current which has produced the phase advance and the phase delay with respect to the ideal alternating current input current in the case of the same phase as alternating current input voltage. In Embodiment 2 of this invention, it is explanatory drawing regarding the corrected amount about the case where an alternating current input has a phase advance with respect to an alternating current input voltage, and a case where there exists a phase lag. It is explanatory drawing which shows the change tendency of the correction amount with respect to the input voltage phase for introduction of the approximate expression which calculates correction amount in Embodiment 2 of this invention. It is explanatory drawing which shows the tendency for introduction of the approximate expression for calculating the corrected amount in Embodiment 2 of this invention in a table format. It is a wave form diagram which shows the simulation result of the reference example which does not apply this invention. It is a wave form diagram which shows the simulation result of Embodiment 2 of this invention. It is a wave form diagram which shows the simulation result of Embodiment 1 of this invention. It is a wave form diagram which shows the simulation result of the reference example which does not apply this invention. It is a wave form diagram which shows the simulation result of Embodiment 2 of this invention. It is a wave form diagram which shows the simulation result of Embodiment 1 of this invention. It is a figure for demonstrating the modification of the correction amount determination method in Embodiment 2 of this invention. It is a figure for demonstrating the modification of the correction amount determination method in Embodiment 2 of this invention. It is a figure for demonstrating the modification of the correction amount determination method in Embodiment 2 of this invention. It is a block diagram which shows the structure of the LED lighting apparatus in Embodiment 3 of this invention. It is a block diagram which shows the modification of a structure of the LED lighting apparatus in Embodiment 3 of this invention. It is explanatory drawing of the method of determining the phase reference regarding the pulsating current voltage obtained with a full wave rectifier circuit in Embodiment 1-3 of this invention.

Embodiment 1 FIG.
1 is a block diagram showing a configuration of a power factor compensating power supply apparatus according to Embodiment 1 of the present invention.

A power factor compensation power supply device 1 according to Embodiment 1 of the present invention includes a power supply main circuit unit 2 and a power supply control unit 3.
The power supply main circuit unit 2 is mainly composed of a full-wave rectifier circuit 14, an input filter 21, a DC / DC converter (hereinafter simply referred to as a converter) 4, and an output capacitor 15.
The full-wave rectifier circuit 14 is configured with a diode bridge to obtain a full-wave rectified voltage | vac | (hereinafter referred to as a pulsating voltage) from the AC input voltage vac supplied from the AC power supply 5.
The input filter 21 is for suppressing conduction noise generated by switching to the AC power supply 5 by a switching element 7 described later, and includes an input coil 20 and input capacitors 6 and 6a.
The converter 4 adjusts the pulsating voltage | vac | that has been full-wave rectified by the full-wave rectifier circuit 14 to a target DC output voltage vo.
The output capacitor 15 is for smoothing the pulsation of the output voltage of the converter 4 to obtain a DC output voltage vo.
A load 8 is connected to the DC power output side of the power supply main circuit unit 2.

Further, the power main circuit unit 2 is provided with a zero current detection unit 16, an input voltage detection unit 10, and an output voltage detection unit 9.
The input voltage detection unit 10 detects the magnitude of the pulsating voltage | vac | as an input voltage detection value vinsen which is an input signal. Although not shown here, for example, two or more divided voltages connected in series It consists of resistance.

  The output voltage detection unit 9 detects the magnitude of the DC output voltage vo as an output voltage detection value vosen. Although not shown here, for example, two or more voltage dividing resistors connected in series are used. Consists of. The contents of current detection by the zero current detection unit 16 will be described later.

  The converter 4 is constituted by a boost chopper circuit here. That is, in the converter 4, for example, a connection point between one end of the full-wave rectifier circuit 14 and one end of the input capacitor 6 is connected to one end of the reactor 18 that is an inductance element, and the other end of the reactor 18 is connected to the anode of the diode 17. The cathode of the diode 17 is connected to a connection point between one end of the output capacitor 15 and one end of the load 8. The connection point between the reactor 18 and the diode 17 is connected to one end of the switching element 7 (drain terminal in the case of an FET element), and the other end (source terminal in the case of FET element) of the switching element 7 is connected to the output capacitor 15. It is connected to a connection point between the end and the other end of the load 8.

  As the switching element 7, an FET (Field Effect Transistor) element, an IGBT (Insulated Gate Bipolar Transistor) element, or the like driven by an element drive signal Vg generated by the power supply control unit 3 is applied. Further, the above diode 17 can be changed to a switching element 17 such as an FET element or an IGBT element, and a synchronous rectification system can be employed in which the switching elements 7 and 17 are operated with reverse logic.

  The zero current detection unit 16 is for detecting the current iL flowing through the reactor 18 in order to determine the turn-on timing of the switching element 7. For example, a current detection resistor is provided between both ends of the current detection resistor. Is detected as a voltage conversion value iLsen corresponding to the reactor current iL.

  The zero current detection unit 16 is not limited to the method of detecting on the low potential side as shown in FIG. 1, but a voltage conversion corresponding to the reactor current iL on the high potential side by connecting a current detection resistor in series with the reactor 18. A method of detecting the value iLsen can also be adopted. In addition, an auxiliary winding having a reverse polarity may be provided in the reactor 18 without using a current detection resistor, and a voltage obtained from the auxiliary winding may be detected as the voltage conversion value iLsen.

  The power supply control unit 3 includes an input voltage phase detection unit 12, an output voltage control unit 11, and a switch control unit 13.

  Here, the input voltage phase detector 12 determines the phase binphase of the pulsating voltage | vac | (hereinafter also referred to as θ) from the input voltage detection value vinsen. This phase θ (= binphase) is hereinafter referred to as an input voltage phase θ.

  The output voltage control unit 11 outputs the detected output voltage at the timing when the input voltage phase detection unit 12 detects 0 and π corresponding to the bottom of the input voltage phase θ (= binphase) of the pulsating voltage | vac |. A difference is obtained from vosen and a preset target voltage value voref, and a control calculation is performed to determine an on-time (hereinafter referred to as a reference on-time) ton1 as a reference of the switching element 7.

  The switch control unit 13 determines a correction amount Kphase (which will be described in detail later) according to the input voltage phase θ (= vinphase), and drives the switching element 7 by multiplying the reference on time ton1 and the correction amount Kphase. The on-time ton2 of the element drive signal Vg to be determined is determined. Further, the switch control unit 13 determines the ON timing of the element drive signal Vg according to the voltage conversion value iLsen corresponding to the reactor current iL detected by the zero current detection unit 16.

  In the first embodiment, the input voltage phase θ is expressed with reference to the bottom of the pulsating voltage | vac | as shown in FIG. That is, since the pulsating voltage | vac | has a frequency twice that of the AC input voltage vac, one period is expressed by 0 to 180 degrees (0 to π).

  The power supply control unit 3 may be a general digital control circuit (including a circuit using software having the same function) that does not use an IC, and some of the components are digital control circuits. Further, an analog control circuit that does not use a digital control circuit may be used. In the first embodiment, a configuration in the case of a digital control circuit using a microcomputer will be described.

  Next, the processing operations of the output voltage control unit 11, the input voltage phase detection unit 12, and the switch control unit 13 constituting the power supply control unit 3 will be described in more detail.

First, how the output voltage control unit 11 determines the reference on time ton1 will be described.
The output voltage control unit 11 is the frequency of the pulsating voltage | vac | after full-wave rectification when the input voltage phase θ (= vinphase) is 0, π, that is, when the commercial frequency is 50 Hz (cycle 20 ms). The reference on-time ton1 is determined at an interval of 100 Hz (period 10 ms).

  The reason is that when the converter 4 is a step-up chopper circuit, the change ΔiL of the reactor current iL flowing through the reactor 18 having the capacitance L in the on period ton of the switching element 7 is pulsed as shown in the following equation (1). Since the relationship is proportional to the current voltage | vac |, as shown in FIG. 2, during one period of the pulsating voltage | vac |, the reference on-time ton1 of the switching element 7 is always kept constant and the current critical operation is performed. To do. For example, in FIG. 2, the reference on-time is set to ton1a for one period Ta of the pulsating voltage | vac |, and the reference on-time is set to ton1b for the next one period Tb. Thereby, the power factor improvement control can be performed by utilizing the property that the average current iLmean per unit time of the reactor current iL can be controlled in the same phase as the pulsating voltage | vac |.

  ΔiL = ton / L × | vac | (1)

  Note that the bottom 0 and π of the input voltage phase θ of the pulsating voltage | vac | are detected by the input voltage phase detection unit 12, and the method of the detection timing will be described later.

  The output voltage control unit 11 calculates the difference between the output voltage detection value vosen and the target voltage value voref, and determines the reference on-time ton1 for the switching element 7 so that the difference becomes zero. Specifically, the reference on-time ton1 is determined by classical control such as proportional-integral (PI) control and proportional-integral-derivative (PID) control, or modern control such as H∞ (H-infinity) control.

  Here, in order to perform the power factor improvement control, the output voltage control unit 11 is described as performing the control calculation of the reference on-time ton1 at the timing when the input voltage phase θ (= binphase) is 0 and π. However, the present invention is not limited to this method. For example, the control calculation may be performed every time the switching element 7 is turned on. In this case, the high-frequency component of the output voltage detection value vosen is sufficiently attenuated by adjusting an RC filter (not shown) provided in the output voltage detection unit 9, or a control coefficient (gain) used for the control calculation is sufficient. The effect equivalent to the method mentioned above can be acquired by making it small.

  On the other hand, the input voltage phase detector 12 detects the phase (0 to π) of the pulsating voltage | vac | from the input voltage detection value vinsen. Specifically, for example, as shown by the mark ● in FIG. 3, the input voltage detection value vinsen is sampled at a constant cycle by an AD converter (not shown), and the latest sampling value is set to x (n), one sampling before X (n-1), and x (n-2) ... before sampling, x (n-2)> x (n-1) <x (n) The x (n) point of the timing becomes the timing when the bottom of the pulsating voltage | vac | is detected.

  Then, the bottom detection timing is set to “zero”, the count-up process is performed for each sampling of the AD converter, etc., and the sampling frequency fsamp, the commercial frequency fcom of the AC power supply 5 and the count-up number n are used as follows. The input voltage phase θ (= vinphase) is calculated from the equation (2).

  θ = vinphase = 2 × fcom × n × π / fsamp (2)

  In the above-described detection method of the input voltage phase θ (= vinphase), a pulsating voltage bottom detection delay corresponding to a maximum of two sampling periods occurs. In order to reduce the influence of this pulsating voltage bottom detection delay, for example, the input voltage phase θ is determined using the following equation (3) based on the assumption of one cycle delay that is an average delay value: Good.

  θ = vinphase = 2 × fcom × (n + 1) × π / fsamp (3)

  Then, the switch control unit 13, which is a characteristic part of this embodiment, performs the reference on-time ton1 according to the correction amount adjustment equation shown in the following equation (8) according to the input voltage phase θ calculated by the above-described method. A correction amount Kphase as a correction signal for is determined. Then, the on-time ton2 of the element drive signal Vg for the switching element 7 is calculated as needed by the correction amount Kphase with respect to the reference on-time ton1 obtained by the output voltage control unit 11 using the following equation (4). Is finally determined.

  ton2 = ton1 × Kphase (4)

  Next, how the switch control unit 13 determines the timing for turning on / off the switching element 7 based on the voltage conversion value iLsen obtained by the zero current detection unit 16 will be described.

  In response to the element drive signal Vg, the switching element 7 continues to be on for the on time ton2 determined by the above-described equation (4), and then turns off the element drive signal Vg. As a result, the switching element 7 is turned off and the current iL flowing through the reactor 18 gradually decreases, and accordingly, the voltage conversion value iLsen detected by the zero current detection unit 16 also decreases.

  Then, as shown in FIG. 4A, at the timing when the voltage conversion value iLsen falls below a threshold voltage iLth preset in a comparator (not shown), the element drive signal Vg is switched on again, and the equation (4) The on state is maintained for the determined on time ton2. By controlling the ON / OFF timing of the switching element 7 by the switch control unit 13 in this way, the output voltage vo is controlled to a desired value, and not only the efficiency by the critical operation but also the power factor by the correction calculation is improved. Can be achieved.

The comparison of the voltage conversion value iLsen with the threshold voltage iLth uses a comparator here. However, the present invention is not limited to this. For example, the voltage conversion value iLsen is used as a drive signal input terminal (in the case of FET, a gate terminal). ) And the on-threshold voltage of the switching element 7 is configured as a substitute for iLth, so that the timing when the voltage conversion value iLsen falls below iLth can be detected.
In this case, the threshold voltage iLth can be adjusted so as to perform a critical operation in consideration of an element delay of a comparator or the like used for comparison of the capacitance or voltage level included in the zero current detection unit 16 in advance. preferable.

  4A shows an example in which the voltage conversion value iLsen is obtained by configuring the zero current detection unit 16 with a current detection resistor as shown in FIG. In the case where an auxiliary winding having a reverse polarity is provided in the reactor 18 as described above without using the current detection resistor and the voltage obtained from the auxiliary winding is obtained as the voltage conversion value iLsen, FIG. The waveform shown in FIG. Even in this case, the on-switching timing of the element drive signal Vg for the switching element 7 can be determined by the same method from the comparison between the voltage conversion value iLsen and the threshold voltage iLth by the comparator.

  Further, as shown in FIG. 2, the reference on-time ton1 is determined at an interval of one cycle of the pulsating voltage | vac |. However, a correction operation is performed by the equation (4) to turn on the on-time ton2 of the element driving signal Vg. It is preferable that the timing of determining is determined every time the element drive signal Vg is output, that is, in the output cycle of the element drive signal Vg.

  Here, in the technique described in the conventional patent document 1, in contrast to the problem of causing the power factor to decrease due to the leading phase of the input current iac by the input capacitors 6 and 6a, The advance phase of the AC input current iac with respect to the AC input voltage vac by 6a is estimated, and the on-time of the switching element 7 is controlled so that the current flowing through the converter 4 is corrected (delayed) by the advance phase using the correction amount Kphase. It is characterized by power factor improvement.

  Hereinafter, a method for determining an adjustment formula [formula (8) described later] for calculating the correction amount for determining the correction amount Kphase from the input voltage phase θ (= vinphase) in the switch control unit 13 will be described. .

  FIG. 5A shows an equivalent circuit of the input filter 21 and the converter 4. Here, the current flowing through the converter 4 is expressed by a current source I * (hereinafter, * indicates a vector), the full-wave rectifier circuit 14 is omitted for simplification, and the input filter 21 is The constituting input coil 20 is a reactor L, and the input capacitors 6 and 6a are represented as a combined capacitance C.

The input coil 20 shown in FIG. 5A is a hybrid choke coil that simultaneously reduces common mode noise and normal mode noise. However, the input coil 20 is not limited to this configuration.
FIG. 6A is an equivalent circuit diagram of the input coil 20 that is the hybrid choke coil shown in FIG.
FIG. 6B is a circuit diagram showing a configuration of an input coil 20a having a configuration different from that of the input coil 20 shown in FIG.
FIG. 6C is a circuit diagram illustrating a configuration of an input coil 20b having a configuration different from that of the input coil 20 illustrated in FIG.
The input coil 20 is not limited to the hybrid choke coil shown in the input coil 20 in FIG. 6A, and for example, a normal mode choke coil may be used as shown in the input coil 20a in FIG. 6B. Alternatively, a common mode choke coil may be used as shown in the input coil 20b of FIG.

  First, due to the influence of the capacitor C constituting the input filter 21, the AC input current iac * has a phase difference that is a leading phase with respect to the AC input voltage vac *. FIG. 5B shows a vector diagram in the case where no control for canceling is performed.

  Assuming that the current flowing through the converter 4 is I * = 0, the current Ic * flowing through the capacitor C of the input filter 21 is obtained by the following equation (5).

  Since the LC resonance frequency of the input filter 21 is sufficiently higher than the frequency of the AC power supply 5, the denominator of the equation (5) becomes “positive”, and the current Ic * flows through the capacitor C. From this, it can be seen that the AC input current iac is advanced by the phase difference Φ with respect to the AC input voltage vac.

  Therefore, as shown in FIG. 5C, for the current I * flowing through the converter 4, the current I * = Ip * −Ic * (provided that the AC input current iac is in phase with the AC input voltage vac). Ip * indicates the effective current), and the phase of the current I * flowing through the converter 4 is corrected (delayed) by Φ, thereby controlling the AC input current iac * only for the effective current Ip * component. To.

  Here, if the load 8 for the converter 4 is represented by a resistor R and a load voltage vo, and the loss in the power supply circuit is ignored, the effective current Ip * can be obtained by the following equation (6) from the law of conservation of energy.

  From the above equations (5) and (6), in order to make the AC input current iac in phase with the AC input voltage vac, the phase difference Φ to be corrected by the current I * flowing through the converter 4 is ).

Thus, the correction amount Kphase with respect to the reference on-time ton1 of the switching element 7 is determined using the phase difference Φ calculated by the equation (7). That is, using the input voltage phase θ (= binphase) detected by the input voltage phase detector 12, as shown in the following equation (8), an ideal current waveform (√2 · iac is a power factor “1”). The ratio between sin θ) and the correction current waveform (√2 · iac · sin (θ−Φ)) calculated using the equation (7) is set as a correction amount Kphase.
That is, the correction current waveform according to the phase difference Φ between the input current iac and the AC input voltage Vac by the input capacitors 6 and 6a and the ideal current waveform with the phase difference Φ set to zero are used. A correction amount Kphase is obtained by dividing the ideal current waveform from the correction current waveform.
Therefore, when the AC input current iac is in phase with the AC input voltage vac, that is, when the power factor is “1”, the correction amount Kphase = 1.
The input voltage phase θ (= binphase) is obtained from the above-described equation (2).

Kphase
= (√2 · iac · sin (θ−Φ)) / (√2 · iac · sinθ)
= Sin (θ−Φ) / sinθ (8)

  Then, the on-time ton2 of the element drive signal Vg for the switching element 7 is finally determined by performing a correction operation on the basis of the reference on-time ton1 according to the correction amount Kphase as necessary according to the above-described equation (4).

  As described above, the reference on-time ton1 is determined by an interval of one cycle of the pulsating voltage | vac |, but the correction amount Kphase is a function of the input voltage phase θ (= vinphase), and therefore, the element drive The determination of the on time ton2 of the signal Vg is performed every time the element drive signal Vg is output within one cycle of the pulsating voltage | vac |, that is, in the output cycle of the element drive signal Vg.

  Actually, the effect of this embodiment was confirmed by simulation using a circuit simulation of Myway Plus. The simulation conditions are AC input voltage vac = 200 V (50 Hz: frequency of pulsating voltage | vac | is 100 Hz), output voltage vo = 330 V, load 8 resistance R = 2000Ω (power 54.45 W), input of input filter 21 The reactance L of the coil 20 is 7 mH, and the combined capacitance C of the input capacitors 6 and 6 a of the input filter 21 is 0.42 uF. When the equation (7) is used, the phase difference Φ of the AC input current iac with respect to the AC input voltage vac is Φ = 10.98 [degree], and the simulation is performed with Φ = 10.98 [degree] in the equation (8). went.

  FIG. 7 shows a simulation result of a reference example to which this embodiment is not applied, and FIG. 8 shows a simulation result to which this embodiment is applied. Here, in both FIGS. 7 and 8, the output voltage detection value vosen, the target voltage value voref, the correction amount Kphase corresponding to the input voltage phase θ (= binphase), the reference on time ton1 and the correction amount Kphase are sequentially shown from the top. The ON time ton2 of the element drive signal Vg, which is a multiplication value, and the waveforms of the AC input voltage vac and the AC input current iac are shown.

  In FIG. 7, the correction amount Kphase = 1 is fixed for convenience of comparison with FIG. In FIG. 8, the correction amount Kphase is calculated, and the result of the formula (8) diverges to ± infinity when the input voltage phase θ (= binphase) is around 0 ° and 180 °, which makes control difficult. Here, the lower limit is set to “0” and the upper limit is set to “2”. 7 and 8, the waveforms of the AC input voltage vac and the AC input current iac at the bottom stage are for comparing the phase difference between them, and the AC input current iac is a value multiplied by 600 for easy comparison. Is displayed.

  In FIG. 7 as a reference example, for comparison with FIG. 8, the correction amount Kphase is always “1” for convenience, so that the on-time ton2 of the element drive signal Vg outputs the reference on-time ton1 as it is. On the other hand, in FIG. 8 showing the control method of the present embodiment, the on-time ton2 of the element drive signal Vg is compared to the reference on-time ton1 so as to correct the phase difference Φ between the AC input voltage vac and the AC input current iac. Has changed greatly.

  7 and FIG. 8, even if the waveform at the lowermost stage is confirmed, the lead phase can be confirmed in the AC input current iac with respect to the AC input voltage vac in FIG. 7 showing the reference example. 8, it can be confirmed that the phase difference Φ between the AC input voltage vac and the AC input current iac is substantially zero. Further, when the power factor is measured by circuit simulation, the power factor = 0.985 in the case of FIG. 7 showing the reference example, whereas the power factor = 0.985 in the case of FIG. 8 to which the present embodiment is applied. It was improved to 0.997, and this confirmed the effectiveness of the present embodiment for improving the power factor.

  In addition, although the method of calculating | requiring phase difference (PHI) by calculation using (7) Formula in calculation of correction amount Kphase was shown here, the phase detection means of alternating current input current iac was further added to the structure of FIG. The phase difference Φ may be obtained from the phase of both the measured AC input voltage vac and the AC input current iac, and the phase difference Φ may be obtained in advance by simulation or experiment.

  As described above, according to the first embodiment, the power supply control unit 3 calculates the reference on-time ton1 from the output voltage detection value vosen and the target voltage value voref, and responds to the input voltage phase θ (= vinphase). The on / off control of the switching element 7 using the on time ton2 obtained by multiplying the reference correction time Kphase by the reference on time ton1 allows the AC input to the AC input voltage vac by the input capacitors 6 and 6a constituting the input filter 21. Since the influence of the phase advance of the current iac can be suppressed, the AC input current iac and the AC input voltage vac can have the same phase and the same waveform, and the power factor can be further improved as compared with the conventional case.

Embodiment 2. FIG.
The second embodiment of the present invention differs from the first embodiment in the method of determining the correction amount Kphase according to the input voltage phase θ (= vinphase). That is, in the first embodiment, the correction amount Kphase is determined based on the equation (8) using the input voltage phase θ (= vinphase) obtained from the input voltage detection value vinsen. However, the equation (8) for determining the correction amount Kphase includes sin calculation and division, and it is difficult to realize the equation (8) by a CPU such as a microcomputer.

  Therefore, in the second embodiment, the switch control unit 13 can approximate the calculation of the equation (8) to determine the correction amount Kphase, and is intended to facilitate the mounting of a microcomputer or the like. It is said.

  Therefore, first consider formulating the equation (8). When the input voltage phase θ is displayed in degrees (the period of the pulsating voltage | vac | is 0 to 180 degrees), in FIG. 9A, the power factor is “1” in the solid line, that is, in the same phase as the AC input voltage. Shows the waveform of the ideal AC input current iacref, and the broken line shows the waveform of the AC input current iaclead in a state advanced by 10 degrees with respect to the ideal AC input current iacref due to the influence of the input capacitors 6 and 6a.

  In FIG. 9B, a solid line indicates a power factor “1”, that is, a waveform of an ideal AC input current iacref in the case of the same phase as the AC input voltage, and a broken line indicates an ideal current waveform iacref. The waveform of the AC input current iaclag delayed by 10 degrees is shown.

  FIG. 10A shows the result of calculating the correction amount Kphase by dividing the broken line AC input current iaclead in FIG. 9A by the ideal AC input current iacref in FIG. 9A based on the equation (8). Show. Further, in FIG. 10B, based on the equation (8), the correction amount Kphase was obtained by dividing the broken line AC input current iaclag in FIG. 9B by the solid line ideal AC input current iacref. Results are shown.

  10 (a) and 10 (b), in principle, the result of equation (8) diverges to ± infinity when the phase is around 0 ° and 180 °, and control becomes difficult. 8) The lower limit is set to “0” and the upper limit is set to “2” for the calculation result of equation (8).

  When the AC input current iac is advanced with respect to the AC input voltage vac, that is, when the denominator of the equation (5) is “positive”, the correction calculation is performed so as to delay the advanced phase. As shown in (a), the correction amount Kphase increases as the phase θ increases.

On the other hand, when the AC input current iac is delayed with respect to the AC input voltage vac, that is, when the denominator of the equation (5) is “negative”, the correction is made to advance the delayed phase, contrary to the above. Since the calculation is performed, the correction amount Kphase decreases as the phase θ increases as shown in FIG.
In the following description, it is assumed that the AC input current iac is in phase with the AC input voltage vac in consideration of the influence of the input capacitors 6 and 6a (that is, the state of FIG. 9A). Describe.

  Consider a case where the phase difference Φ of the AC input current iac with respect to the AC input voltage vac changes due to output power or the like. When the phase difference Φ of the AC input current iac with respect to the AC input voltage vac is changed to “5 degrees”, “10 degrees”, “15 degrees”, “20 degrees”, and “25 degrees”, respectively, The result of obtaining the correction amount Kphase with respect to the input voltage phase θ based on the graph is shown by the solid line in FIG.

  Here, as a feature of the second embodiment, the correction amount Kphase when the correction amount Kphase indicated by each solid line is approximated to a simple linear function is indicated by a broken line in FIG.

  The correction amount Kphase is a sawtooth signal having a period of 180 degrees by the respective linear functions shown by the broken lines in FIG. 11, and the expression (8), which requires processing time, is expressed as the following expression (9): It can be confirmed that it can be approximated to a linear function expression in which the correction amount Kphase simply increases in accordance with the input voltage phase θ.

  Kphase = A · θ + B (Limiter: 0 to 2) (9)

Referring to FIG. 11, the slope A and intercept B of the linear function equation represented by equation (9) can be set under the following conditions.
(I) The slope A is set to a “positive” value.
(Ii) As the phase difference Φ of the AC input current iac with respect to the AC input voltage vac increases, the change in the correction amount Kphase with respect to the change in the input voltage phase θ is increased, that is, the slope A is set to “large”.
(Iii) As a method of determining the intercept B, according to FIG. 11, the intercept B is set so that the correction amount Kphase is “1” when the input voltage phase θ is 90 degrees. That is, the intercept B is determined so that B = 1−A × (90 degrees).

Here, with respect to the above (ii), the condition that the phase difference Φ of the AC input current iac with respect to the AC input voltage vac is increased will be considered from the equation (7).
The constants used in equation (7) are: AC input voltage vac = 200 V (50 Hz: frequency of pulsating voltage is 100 Hz), output voltage vo = 400 V, load R = 2938Ω (power 54.45 W), input filter 21 Based on the reactor L = 7 mH of the input coil 20 and the combined capacitance C = 0.42 uF of the input capacitors 6, 6 a of the input filter 21, the output power Po to the load 8 of the power supply main circuit unit 2, the AC input voltage vac, AC FIG. 12 shows the calculation result of the phase difference Φ of the AC input current iac with respect to the AC input voltage vac when the input voltage frequency f is changed as a parameter.

  The phase difference Φ increases as the output power Po decreases from FIG. 12A, the phase difference Φ increases as the AC input voltage vac increases from FIG. 12B, and the AC input voltage frequency f from FIG. The larger the value, the larger the phase difference Φ.

  As described above, the phase difference Φ increases according to changes such that the output power Po to the load 8 decreases, the AC input voltage vac increases, and the AC input voltage frequency f increases. Accordingly, the input voltage phase is increased accordingly. A change in the correction amount Kphase with respect to the change in θ is set to be large, that is, the slope A of the equation (9) is set to be large.

  Note that the correction amount Kphase may be determined by reading the values of the slope A and the intercept B determined in advance for each combination of various parameters from the table memory, and the correction amount Kphase calculated for each combination of various parameters in advance. The value may be read from the table memory.

  As another method, for example, a means capable of detecting the power factor and the AC input current iac is added to the configuration of FIG. 1, and the pulsating voltage | vac | is improved so that the power factor is improved or the AC input current distortion is reduced. These values A and B can be adjusted so as to further improve the power factor by performing control to change the slope A and the intercept B for each cycle.

  In the second embodiment, as in the case of the first embodiment, the effect was confirmed by simulation using the circuit simulation of Myway Plus. The simulation conditions are AC input voltage vac = 242V (50 Hz: frequency of pulsating voltage | vac | is 100 Hz), output voltage vo = 390 V to load 8, reactance L of input coil 20 of input filter 21 = 7 mH, input filter The combined capacitance C of the 21 input capacitors 6 and 6a is 0.42 uF. Further, the output power Po to the load 8 was confirmed in two ways: (i) Po = 50 W (load resistance R = 3042Ω) and (ii) Po = 5 W (load resistance R = 30 kΩ).

(I) When Output Power Po = 50 W FIG. 13 shows a simulation result of a reference example to which the present invention is not applied, FIG. 14 shows a simulation result of the second embodiment, and FIG. 15 shows a simulation result of the first embodiment. 13, 14, and 15, the upper stage shows the correction amount Kphase corresponding to the input voltage phase θ, and the lower stage shows the waveforms of the AC input voltage vac and the AC input current iac.

  In FIG. 13, the correction amount Kphase = 1 is fixed for convenience of comparison with FIGS. 14 and 15. In FIG. 14, the inclination of the correction amount Kphase is determined so that the correction amount Kphase = 1 when the input voltage phase θ is 90 degrees (15-degree advance state in FIG. 11). Further, in FIG. 15, the correction amount Kphase is set such that the lower limit is “0” and the upper limit is “2”. 13, 14, and 15, the waveforms of the lowest AC input voltage vac and the AC input current iac are for comparing the phase difference between the two, and the AC input voltage vin is easy to compare. Indicates a value multiplied by 1/1000.

  When the power factor is confirmed by circuit simulation, it is 0.971 in FIG. 13, 0.995 in FIG. 14, and 0.995 in FIG. 15, and by using the approximation formula (9) of the second embodiment, It can be confirmed that there is a power factor improvement effect larger than that of the reference example, and that an effect substantially equivalent to the case of the expression (8) shown in the first embodiment can be obtained.

(Ii) When Output Power Po = 5 W FIG. 16 shows a simulation result of a reference example to which the present invention is not applied, FIG. 17 shows a simulation result of the second embodiment, and FIG. 18 shows a simulation result of the first embodiment. 16, 17, and 18, the upper stage shows the correction amount Kphase corresponding to the input voltage phase θ, and the lower stage shows the waveforms of the AC input voltage vac and the AC input current iac.

  In FIG. 16, the correction amount Kphase = 1 is fixed for convenience of comparison with FIGS. 17 and 18. In FIG. 17, the inclination of the correction amount Kphase is determined so that the correction amount Kphase = 1 when the input voltage phase θ is 90 degrees (a state of 25 degree advance in FIG. 11). Further, in FIG. 18, the correction amount Kphase has a lower limit set to “0” and an upper limit set to “2”. 16, 17, and 18, the waveforms of the lowest AC input voltage vac and the AC input current iac are for comparing the phase difference between them, and the AC input voltage vin is easy to compare. Indicates a value multiplied by 1/1000.

  When the power factor is confirmed by circuit simulation, the power factor is 0.673 in FIG. 16, 0.783 in FIG. 17, and 0.819 in FIG. 18. By using the approximation formula (9) in the second embodiment, It can be confirmed that there is a power factor improvement effect larger than that of the example, and that an effect of a level equivalent to that of the expression (8) shown in the first embodiment can be obtained.

  As described above, as can be seen from the simulation results (FIGS. 14 and 17) when the output power Po is 50 W and 5 W, a simple linear function formula based on the formula (9) of the second embodiment is used. It is understood that the power factor can be improved even if it is.

  Here, the correction amount Kphase is approximated to a simple linear function equation of the input voltage phase θ shown in the equation (9), so that the correction amount Kphase is generated as a sawtooth signal having a cycle of 180 degrees. However, the present invention is not limited to this, and the calculation of the correction amount Kphase can be determined as another function. This will be described next.

  Now, with the configuration of FIG. 1, the input voltage phase θ (= binphase) detected by the input voltage phase detector 12 is slightly shifted (delayed) from the phase of the actual pulsating voltage | vac |. .

  That is, FIG. 19 shows a simulation waveform when the input voltage phase θ detected by the input voltage phase detector 12 is delayed by 18 degrees with respect to the actual phase of the pulsating voltage | vac |. In FIG. 19, the upper part shows the relationship between the pulsating voltage | vac | and the correction amount Kphase (the pulsating voltage | vac | is displayed at a magnification of 1/200 in order to make the detection phase shift easy to see. ), And the lower part shows the relationship between the AC input voltage vac and the AC input current iac (in order to make the waveforms easier to compare, the AC input voltage vac is displayed at 1/1000 times). As can be seen from FIG. 19, the detected input voltage phase θ is delayed by 18 degrees with respect to the phase of the actual pulsating voltage | vac |, so that the AC input current iac is near 0 degrees of the pulsating voltage | vac |. It can be confirmed that the waveform is distorted. This deteriorates the power factor.

  In order to deal with this phenomenon, for example, as shown in the upper part of FIG. 20, the correction amount Kphase is not a sawtooth signal, but a period of 180 degrees inclined to the left and right with respect to the peak in accordance with the change of the input voltage phase θ. As shown in the lower part of FIG. 20, the waveform distortion of the AC input current iac near 0 degrees of the pulsating voltage | vac | can be suppressed, and the input voltage phase detection can be performed. It becomes possible to suppress the power factor deterioration due to the phase delay when the input voltage phase θ is detected by the unit 12.

  Alternatively, as shown in FIG. 21, in a period in which a phase delay occurs when the input voltage phase detector 12 detects the input voltage phase θ, the correction amount Kphase = 0 is held, and thereafter, a period of θ = 180 degrees. Then, it is possible to suppress the power factor deterioration due to the detection phase delay by setting the correction amount Kphase to a function that makes a sawtooth signal.

  As described above, the method of determining the correction amount Kphase according to the input voltage phase θ is not limited to a sawtooth signal, but can be given by a function that is relatively easy to perform arithmetic processing such as a triangular wave as shown in FIGS. it can.

  As described above, according to the second embodiment, the power supply controller 3 calculates the reference on-time ton1 from the output voltage detection value vosen and the target voltage value voref, and according to the input voltage phase θ (= vinphase). Thus, the reference on-time ton1 is multiplied by a correction amount Kphase determined by a function that is relatively easy to calculate, such as a sawtooth waveform obtained by equation (9) or a triangular waveform as shown in FIG. By controlling on / off of the switching element 7 using the obtained on time ton2, it is possible to suppress the influence of the phase advance of the AC input current iac on the AC input voltage vac by the input capacitors 6 and 6a constituting the input filter 21. Even if a digital control circuit using a relatively inexpensive microcomputer is used, the AC input current iac and the AC input voltage vac are in phase, Can be a waveform, it is possible to improve the power factor.

Embodiment 3 FIG.
FIG. 22 is a block diagram showing the configuration of the LED lighting device according to the third embodiment, and the same reference numerals are given to the components corresponding to those in FIG.

  The feature of the LED lighting device 27 of the third embodiment is the case where the load 8 is an LED module 22 in which a plurality of LEDs 23 are connected in cascade to the configuration shown in FIG. For this reason, the difference from the first embodiment is that the power supply control unit 3 includes an output current detection unit 24 instead of the output voltage detection unit 9 and an output current control unit 26 instead of the output voltage control unit 11. It is changed so that the target current value ioreef is set. Other configurations are the same as those shown in FIG.

  For example, the output current detection unit 24 includes a current detection resistor (not shown), and detects a potential difference generated between both ends of the current detection resistor as a voltage conversion value iosen corresponding to the output current flowing through the LED module 22. In addition, the connection configuration of the LED modules 22 is here that all the LEDs 23 are connected in series, but a plurality of LEDs 23 may be connected in series or in series-parallel. Furthermore, although the light source is the LED 23 here, it is not limited to this and can be changed to an organic EL or a laser diode.

  Here, the LED 23 is usually suitable for current control because of its VI characteristics, and therefore, in FIG. 21, the control is changed from output voltage control to output current control. With this configuration, the current flowing through the LED module 22 can be controlled by performing the same control as in the first and second embodiments. In addition, when a dimming function for adjusting the amount of light is installed, the dimming function can be realized if the target current value ioref inputted from the outside is variable.

  When the LED module 22 is used as a load, the output voltage vo does not change even when the current flowing through the LED module 22 is changed by the dimming function. In the first and second embodiments, the output power Po is one parameter for determining the correction amount Kphase. Ideally, it is necessary to calculate the power by detecting both the output voltage vo and the output current io. .

  However, in the third embodiment, the load 8 becomes the LED module 22 as shown in FIG. 22, so that the output current io is detected as the voltage conversion value iosen by the output current detector 24 without detecting the output voltage vo. By doing so, the correction amount Kphase can be adjusted. As a result, the advantages of reducing the number of pins of the microcomputer, shortening the processing time, reducing the wiring, and reducing the size and cost are obtained.

  Furthermore, since the dimming signal is normally input to the microcomputer from the outside to perform dimming control, the optimum correction amount Kphase after dimming can be obtained in advance even when the dimming signal is varied. It is possible to perform power factor correction control with a fast response speed against fluctuations.

  By the way, the LED lighting device 27 having the configuration shown in FIG. 22 converts the AC input into the DC output by the single-stage converter 4, but in this way, the commercial frequency ripple of the AC power source 5 is removed on the output side. There is a concern that the LED module 22 is visually observed as flickering of light.

  Therefore, by applying a two-stage converter, it is possible to simultaneously realize power factor improvement control on the input side and current stabilization control on the output side. Thus, FIG. 23 shows a configuration in the case of using a two-stage converter.

  23, in addition to the load 8 being the LED module 22, the LED lighting device 27 having the configuration shown in FIG. Is provided with an LED current adjustment circuit 25 (a detailed control configuration is omitted) as an output current adjustment DC / DC converter, and an output current for detecting a voltage conversion value iosen corresponding to the current flowing through the LED module 22 The detection unit 24 is provided, and the power supply control unit 3 is provided with an output current control unit 26 in addition to the output voltage control unit 11.

  In this case, the output current control unit 26 uses the LED current adjustment circuit 25 so that the output current io becomes constant according to the target current value ioreef inputted from the outside and the voltage conversion value iosen detected by the output current detection unit 24. At the same time, the output voltage control unit 11 sets the target voltage value voref to a value that is equal to or higher than the voltage of the LED module 22, and controls the converter 4 in the same manner as described above. . In the case of the configuration shown in FIG. 23, since the output voltage vo is constant regardless of the dimming degree, the output voltage of the converter 4 is generally controlled to a constant value regardless of the dimming degree. Advantages similar to those of the configuration of FIG. 22 can be obtained.

  As described above, according to the third embodiment, when the load 8 is the LED module 22, the output current detection unit 24 is provided in the power supply main circuit unit 2 to detect the output current io as the voltage conversion value iosen. In addition, the power supply control unit 3 is provided with an output current control unit 26 to calculate the reference on-time ton1 from the output current detection value iosen and the target current value ioref, and to make a correction according to the input voltage phase θ (= vinphase). The switching element 7 is on / off controlled using an on-time ton2 obtained by multiplying the amount Kphase by a reference on-time ton1. As a result, the influence of the phase advance of the AC input current iac on the AC input voltage vac by the input capacitors 6 and 6a can be corrected, so that the AC input current iac and the AC input voltage vac can have the same phase and waveform, thereby improving the power factor can do.

  In the first to third embodiments, in order to reduce the size of the power factor compensation power supply device 1 and the LED lighting device 27, all or a part of the devices are mounted on one integrated circuit, and one package is provided. IC may be included in For example, the power supply control unit 3 is preferably housed in a single control IC package.

  In addition, the present invention is not limited to the configuration of each of the above first to third embodiments, and can be freely combined with each of the first to third embodiments within the scope not departing from the gist of the present invention. The configurations of the first to third embodiments can be modified or omitted as appropriate.

Claims (14)

A power source main circuit unit, and a power source control unit for controlling the power source main circuit unit,
The power supply main circuit section has a full-wave rectification circuit for full-wave rectification of the AC voltage of the AC power supply, an inductance element and a switching element, and converts the input voltage obtained by the full-wave rectification circuit to a target output voltage. Converter, an input filter configured to include an input capacitor for suppressing the outflow of noise components due to the switching element to the AC power supply, an input voltage detector for detecting the input voltage, and the output voltage. An output voltage detection unit to detect,
The power supply control unit generates a reference on-time of the switching element determined based on the output voltage detected by the output voltage detection unit based on the phase of the input voltage detected from the input voltage. Correcting with a signal to obtain the on-time of the switching element, and using the corrected on-time of the switching element to control the switching element on and off,
Power factor compensation power supply. The power controller is
The switching element is detected by detecting the phase of the input voltage from the input signal detected by the input voltage detector and multiplying the reference on-time by the correction signal generated based on the phase of the input voltage. Is what you get on time,
Using the corrected on-time of the switching element, the switching element is turned on and off to control the output voltage to a desired voltage, and power factor compensation control is performed on the input current from the AC power supply.
The power factor compensation power supply device according to claim 1. The power control unit is based on the phase of the input voltage.
Using the correction current waveform according to the phase difference between the input current from the AC power supply and the input voltage by the input capacitor and the ideal current waveform with the phase difference being zero, The correction signal is obtained by dividing the ideal current waveform.
The power factor compensation power supply device according to claim 1 or 2. The power supply control unit includes a sawtooth function that changes linearly according to the phase of the input voltage, and a triangular wave function that tilts left and right with respect to a peak according to the phase of the input voltage. Generate a correction signal,
The power factor compensation power supply device according to claim 1 or 2. The power supply control unit determines the function so that the value of the correction signal is “1” when the phase of the input voltage is 90 degrees.
The power factor compensation power supply device according to claim 4. When the correction signal is a sawtooth function, the power supply control unit is configured such that when the phase information of the input voltage is smaller than 90 degrees, the correction signal is smaller than “1” and larger than 90 degrees. Determine the function so that the correction signal is greater than “1”;
The power factor compensation power supply device according to claim 5. The power supply control unit sets one of a slope of the sawtooth function and a slope of the triangular wave function when the output power of the power supply main circuit section is small.
The power factor compensation power supply device according to claim 4 or 5. The power supply control unit sets one of the slope of the sawtooth function and the slope of the triangular function when the input voltage is large.
The power factor compensation power supply device according to claim 4 or 5. The power supply control unit sets one of a slope of the sawtooth function and a slope of the triangular wave function when the frequency of the input voltage is large.
The power factor compensation power supply device according to claim 4 or 5. The power supply control unit sets upper and lower limiters for the correction signal.
The power factor compensation power supply device according to any one of claims 3 to 9. A power source main circuit unit and a power source control unit for controlling the power source main circuit unit, and an LED module is connected to the output side of the power source main circuit unit;
The power supply main circuit section has a full-wave rectifier circuit for full-wave rectification of the AC voltage of the AC power supply, an inductance element and a switching element, and converts the input current obtained by the full-wave rectifier circuit into a target output current. An input filter configured to include an input capacitor for suppressing outflow of noise components from the switching element to the AC power supply, and input voltage detection for detecting an input voltage obtained by the full-wave rectifier circuit And an output current detection unit for detecting the output current,
The power supply control unit generates a reference on-time of the switching element determined based on the output current detected by the output current detection unit based on the phase of the input voltage detected from the input voltage Correcting with a signal to obtain the on-time of the switching element, and using the corrected on-time of the switching element to control the switching element on and off,
LED lighting device. The power controller is
The switching element is detected by detecting the phase of the input voltage from the input signal detected by the input voltage detector and multiplying the reference on-time by the correction signal generated based on the phase of the input voltage. Is what you get on time,
Using the corrected on-time of the switching element, the switching element is turned on and off, thereby controlling the output current to a desired current and controlling the input current from the AC power source for power factor compensation.
The LED lighting device according to claim 11. A power factor compensation power supply device according to any one of claims 1 to 10, wherein an LED module is connected to an output side of the power supply main circuit unit, and the converter and the LED module are connected to each other. An LED current adjustment circuit that adjusts the current flowing through the LED module to a target current is provided in between.
LED lighting device. The power supply control unit includes a sawtooth function that changes linearly according to the phase of the input voltage, and a triangular wave function that tilts left and right with respect to a peak according to the phase of the input voltage. When generating the correction signal, when the dimming signal that is a command value of the current flowing through the LED module is large, one of the slope of the sawtooth function and the slope of the triangular wave function is set to be large.
The LED lighting device according to any one of claims 11 to 13.

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