JP6129549B2 - Transmitter - Google Patents

Description

  The present invention relates to a transmitter, and more particularly to a transmitter including a Cartesian Loop type negative feedback amplifier that performs nonlinear distortion compensation.

  A Cartesian loop negative feedback amplifier is a nonlinear distortion of a power amplifier that performs power amplification of a linear modulation signal such as QPSK (Quadrature Phase Shift Keying) modulation, 16-value QAM (Quadrature Amplitude Modulation), OFDM (Orthogonal Frequency Division Multiplexing), etc. Compensates and is mainly used in transmitters that perform wireless communications. In such a negative feedback amplifier, if the phase adjustment of the carrier wave signal input to the forward circuit or the feedback circuit is inappropriate, the negative feedback amplifier may become unstable and oscillation may occur. Oscillation caused by the negative feedback amplifier adversely affects other communication systems.

  As a technique related to the phase control of the carrier wave signal in the negative feedback amplifier, there are Patent Document 1 and Patent Document 2 below. In the negative feedback amplifier circuit of Patent Document 1, a baseband signal is orthogonally modulated by an orthogonal modulation circuit, and a quadrature demodulated signal obtained by orthogonally demodulating the signal amplified by the amplifier circuit by an orthogonal demodulator circuit is subtracted from the baseband signal. In the negative feedback amplifier circuit, the phase detection circuit inputs the quadrature demodulated signal and the baseband signal, compares the phases, and outputs the averaged phase difference time as phase control data to the phase control circuit. By performing phase control of the feedback signal based on the control data, a Cartesian loop negative feedback amplifier circuit that compensates for two-dimensional distortion included in the quadrature modulation signal is realized. In the transmitter using the non-linear distortion compensation circuit of the negative feedback method of Patent Document 2, the phase of the input baseband signal before adding the feedback signal and the phase of the signal before adding the feedback signal and before quadrature modulation By controlling the phase of the signal input by the quadrature demodulator that demodulates the feedback signal based on the compared information, the loop phase characteristics are kept stable and the occurrence of loop oscillation and spurious are suppressed and stable. A negative feedback amplifier of Cartesian loop that operates in the same manner is realized.

JP-A-7-58555 JP 2006-54907 A

  Thus, the negative feedback amplifier circuit of the Cartesian loop of Patent Document 1 and the negative feedback amplifier of the Cartesian Loop of Patent Document 2 detect the phases of the baseband signal and the feedback signal, and control the phase of the feedback signal. The negative feedback amplifier is operated stably. However, an offset occurs between the orthogonal component of the baseband signal and the orthogonal component of the feedback signal before being input to the comparator due to product manufacturing variations and temperature changes. Due to such an offset in the orthogonal component of the baseband signal and the orthogonal component of the feedback signal, the comparator erroneously detects the phase shift, and thus there is a problem that the offset must be adjusted.

  First, a method in which the conventional phase controller 111B detects the phase shift amount will be described with reference to FIGS. FIG. 5 is a diagram showing a configuration of a conventional phase controller 111B. FIG. 6 is a time chart showing the detection procedure of the phase shift in the phase controller 111B. The configuration of the conventional negative feedback amplifier is the same except that the phase controller 111A is replaced with the phase controller 111B in the negative feedback amplifier 10 of the first embodiment shown in FIG. The description of the configuration of the negative feedback amplifier 10 shown in FIG. 1 has been made in the first embodiment, and is omitted here.

  The conventional phase controller 111B includes an exclusive OR 501, a sample timing generation unit 502, a sampling function unit 503, a phase shift detection timing generation unit 504, a counter 505, a down edge detection unit 506, a phase shift direction detection unit 507, and The phase control information calculation unit 508 is configured.

When the Q-component signal is input from the comparator 110a and the q-component signal is input from the comparator 110b, the exclusive OR 501 performs an exclusive OR (EOR) operation between the Q component and the q component. Thus, the phase shift amount is calculated, and this phase shift amount is output to the sampling function unit 503 as a pulse signal.
The sample timing generation unit 502 generates a signal (hereinafter referred to as a sample timing signal) that is a timing for sampling the pulse signal output by the exclusive OR 501 to the sampling function unit 503 and outputs the signal to the sampling function unit 503. . The sample timing signal is generated so as to be a signal having a sufficiently fast period with respect to the period of the training signal for detecting the phase shift.
When the sampling function unit 503 receives the pulse signal from the exclusive OR 501 and the sample timing signal from the sample timing generation unit 502, the sampling function unit 503 samples the width of the High section of the pulse signal as the phase shift amount at the timing of the sample timing signal. A signal that counts the amount of phase shift in sampling (hereinafter referred to as phase shift amount count signal)
Is output to the counter 505.
The phase shift detection timing generation unit 504 generates an effective section of the training signal as a timing signal in order to measure the amount of phase shift, a signal serving as a start reference timing of the detection section (hereinafter referred to as a start timing signal), and an end reference timing. A signal (hereinafter referred to as a stop timing signal) is generated and output to the counter 505.
When the start timing signal is input from the phase shift detection timing generation unit 504, the counter 505 clears the count value to 0 and starts counting, and when the phase shift amount count signal is input from the sampling function unit 503, the counter 505 updates the count value and outputs the phase shift. When a stop timing signal is input from the detection timing generation unit 504, the count is stopped.
When the Q component signal is input from the comparator 110a, the down edge detection unit 506 detects the down edge timing at which the signal changes from High to Low, and indicates a signal indicating the down edge timing (hereinafter referred to as a down edge timing signal). ) And output to the phase shift direction detection unit 507.
When the phase shift direction detection unit 507 receives the q component signal from the comparator 110b and the down edge timing signal from the down edge detection unit 506, the phase shift direction detection unit 507 holds the state of the q component signal and the next input q component signal. By comparing the held q component signal with the down edge timing, it is detected whether the two signals are in the phase lag direction or the phase shift direction, and this phase shift direction is phase controlled. The information is output to the information calculation unit 508.
When the phase shift amount is input from the counter 505 and the phase shift direction is input from the phase shift direction detection unit 507, the phase control information calculation unit 508 generates phase control information from the phase shift amount and the phase shift direction and outputs the phase control information to the phase shifter 114. .

  Next, a method for detecting a phase shift in the phase controller 111B will be described with reference to FIG. FIG. 6A is a time chart showing a procedure for the phase controller 111B to detect a phase shift when the Q component signal input by the comparator 110a and the q component signal input by the comparator 110b are not offset. It is a chart. FIG. 6B is a time chart showing a procedure in which the phase controller 111B detects a phase shift when the Q component signal input by the comparator 110a and the q component signal input by the comparator 110b are offset. It is.

  First, the procedure for detecting the phase shift of the phase controller 111B when the Q component and the q component are not offset as shown in FIG. 6A will be described below. The Q component sine wave 601 is a sine waveform of the Q component of the baseband signal input from the baseband signal generator 101 by the comparator 110a. The Q component rectangular wave 602 is a rectangular waveform output from the comparator 110a when the Q component sine wave 601 and the reference voltage Vs are input to the comparator 110a. The q component sine wave 603 is a sine waveform of the component q of the feedback signal input from the direct demodulator 109 by the comparator 110b. The q component rectangular wave 604 is a rectangular waveform output from the comparator 110b by inputting the q component sine wave 603 and the reference voltage Vs to the comparator 110b. The phase shift amount detection rectangular wave 605 is a waveform obtained by calculating an exclusive OR of the Q component rectangular wave 602 and the q component rectangular wave 604. As shown in FIG. 6A, when the q-component sine wave 603 is delayed in phase with respect to the Q-component sine wave 601, the q-component rectangular wave 604 is also a rectangular wave whose phase is delayed with respect to the Q-component rectangular wave 602. Become. For this reason, the phase shift amount can be detected as the phase shift amount detection rectangular wave 605 by calculating the exclusive OR 501 of the Q component rectangular wave 602 and the q component rectangular wave 604.

  Further, the detection of the phase shift direction is performed by the phase shift direction detection unit 507. As shown in FIG. 6A, the phase shift direction detection unit 507 outputs the down edge (high-to-low change point) of the Q component rectangular wave 602 output from the down edge detection unit 506 and the comparator 110b. By confirming the state (High / Low) of the q-component rectangular wave 604, the phase advance / delay shift direction is detected. As shown in FIG. 6A, when the high state is detected, q indicates that the phase is delayed with respect to Q. Conversely, when the low state is detected, q indicates that the phase is advanced with respect to Q. Become. Further, the phase shift direction detection method does not necessarily need to be detected at the down edge of the Q component rectangular wave 602 output from the comparator 110a, but is compared at the up edge of the Q component rectangular wave 602 output from the comparator 110a. The state (High / Low) of the q component rectangular wave 604 output from the comparator 110b is confirmed, or is output from the comparator 110a at the up / down edges of the q component rectangular wave 604 output from the comparator 110b. It can also be detected by confirming the state (High / Low) of the Q component rectangular wave 602.

  Next, a detection example of the amount of phase shift when the phase shift detection section in the 4 KHz sine wave as the training signal is set to a half cycle or more and one cycle or less of the sine wave from the baseband signal generator 101 will be described. When the 4 KHz sine wave is shifted by 1 degree (π / 180 radians), the output of the exclusive OR 501 is “(Q component of the output of the comparator 110a) EOR (q component of the output of the comparator 110b) of FIG. ) "To become a pulse having a high level of 694 nsec width. Further, since this pulse is obtained twice in the phase shift detection section, the sum total is 1388 nsec (= 694 nsec × 2). In order to sample the pulse signal having a width of 694 nsec, a clock of 1.44 MHz (≈1 / (694 nsec)) or more is required. For example, when sampling with a sufficiently fast 10 MHz clock with respect to a 694 nsec width pulse, one clock is 100 nsec (10 MHz (≈ 1 / (100 nsec)). To count 694 nsec width pulses, 6.9 6 (a) “(Q component of the output of the comparator 110a) EOR (q component of the output of the comparator 110b)”, the high pulse section is generated twice in the phase shift detection section. Therefore, when it is accumulated, 13.8 counts (= 6.9 counts × 2) is obtained, so when the count value is 13.8 counts (actually 14 counts) or more, the phase is 1 degree. The phase control information is generated based on the phase shift information of the phase shift amount for one degree, and is output to the phase shifter 114.

  Next, a procedure for detecting the phase shift by the phase controller 111B in the case of the offset shown in FIG. 6B will be described below. The Q ′ component sine wave 611 is a sine waveform of the Q ′ component of the baseband signal offset to the upper side inputted from the baseband signal generator 101 by the comparator 110a. The Q ′ component rectangular wave 612 is a rectangular waveform output from the comparator 110 a by inputting the Q ′ component sine wave 611 and the reference voltage Vs to the comparator 110 a. The q ′ component sine wave 613 is a sine waveform of the component q of the feedback signal offset to the lower side inputted from the direct demodulator 109 by the comparator 110 b. The q ′ component rectangular wave 614 is a rectangular waveform output from the comparator 110 b by inputting the q ′ component sine wave 613 and the reference voltage Vs to the comparator 110 b. The phase shift amount detection rectangular wave 615 is a waveform obtained by calculating an exclusive OR of the Q ′ component rectangular wave 612 and the q ′ component rectangular wave 614. As shown in FIG. 6B, when a Q ′ component sine wave 611 offset upward with respect to the reference voltage Vs is input to the comparator 110a, a Q ′ component rectangular wave 612 output from the comparator 110a. Becomes a pulse signal having a longer High period than the Q component sine wave 601. Further, when the q ′ component signal 613 offset to the lower side with respect to the reference voltage Vs is input to the comparator 110 b, the q ′ component rectangular wave 614 output from the comparator 110 b is compared with the q component rectangular wave 604. Thus, the high period becomes a short pulse signal.

  In addition, in the pulse signal representing the phase shift amount obtained from “(Q ′ component of comparator 110a output) EOR (q ′ component of comparator 110b output)”, the phase shift amount detection of the pulse signal output for the first time is performed. The rectangular wave 615 is wider than the phase shift amount detection rectangular wave 605 of FIG. 6A, and the phase shift amount detection rectangular wave 615 of the pulse signal output for the second time is the phase of FIG. 6A. The width is narrower than the deviation amount detection rectangular wave 605. When this pulse signal is accumulated in the phase shift detection section, a difference occurs in the phase shift amount between FIG. 6 (a) and FIG. 6 (b). Further, since the state of the q ′ component sine wave 613 output from the comparator 110b at the down edge of the Q ′ component sine wave 611 is Low, the phase shift direction detection unit 507 performs the q ′ component with respect to the Q ′ component. Incorrect phase shift direction is detected such that the phase is advanced. For this reason, there is a problem that correct phase correction cannot be performed, and in order to perform correct phase correction, it is necessary to adjust peripheral devices so that offsets of signals input to the comparators 110a and 110b are at appropriate levels. There was a problem.

  This invention is made | formed in view of such a condition, and it aims at providing the transmitter which can solve the said subject.

The transmitter of the present invention, transmission inputs the in-phase and quadrature components of the baseband signal, the baseband signal to quadrature modulation by a carrier signal, to have the amplifier for amplifying the quadrature modulated linear modulation signal to a predetermined power level The amplifier is a Cartesian loop negative feedback amplifier that performs nonlinear distortion compensation, and is branched by a directional coupler that extracts a part of the output signal of the transmitter as a feedback signal, and the directional coupler. A quadrature demodulator that quadrature demodulates the feedback signal into a feedback in-phase component and a feedback quadrature component by the carrier signal, a phase controller that detects a phase shift amount and a phase shift direction of the baseband signal and the feedback signal, and A phase shifter for controlling the phase of the carrier signal input to the quadrature demodulator according to the detected phase shift and phase shift direction, The control unit includes baseband signal center point timing calculating means for calculating a center point timing of the in-phase component or quadrature component of the baseband signal, and a feedback signal center for calculating a center point timing of the in-phase component or quadrature component of the feedback signal. A point timing calculating means, and calculating a difference between a center point timing of the baseband signal and a center point timing of the feedback signal, and based on the difference, a phase shift of an in-phase component or a quadrature component of the baseband signal and the feedback signal And phase control information calculating means for calculating the quantity and the phase shift direction.
The baseband signal center point timing calculation means of the transmitter of the present invention includes a first comparator that inputs the baseband signal and a reference voltage and converts the baseband signal into a baseband signal rectangular waveform, and the baseband signal rectangular shape. First center point timing calculating means for calculating a center point timing from the timing of the up edge and the down edge of the waveform, the center point timing of the baseband signal rectangular waveform as the center point timing of the baseband signal, The feedback signal center point timing calculation means includes a second comparator that inputs the feedback signal and a reference voltage and converts the feedback signal into a feedback signal rectangular waveform, and calculates the center from the timing of the up edge and the down edge of the feedback signal rectangular waveform. A second center point timing calculating means for calculating a point timing, and a center point of the feedback signal rectangular waveform It is characterized in that the centering point timing of the feedback signal timing.
Also, the first comparator of the transmitter of the present invention inputs the in-phase component of the baseband signal and the second comparator inputs the in-phase component of the feedback signal, or The first comparator inputs the quadrature component of the baseband signal, and the second comparator inputs the quadrature component of the feedback signal.
The transmitter of the present invention, transmission inputs the in-phase and quadrature components of the baseband signal, the baseband signal to quadrature modulation by a carrier signal, to have the amplifier for amplifying the quadrature modulated linear modulation signal to a predetermined power level The amplifier is a Cartesian loop negative feedback amplifier that performs nonlinear distortion compensation, and is branched by a directional coupler that extracts a part of the output signal of the transmitter as a feedback signal, and the directional coupler. A phase shift amount between a quadrature demodulator that quadrature-demodulates the feedback signal into a feedback in-phase component and a feedback quadrature component by the carrier wave signal, the baseband signal, and an addition signal obtained by adding the baseband signal and the feedback signal And a phase controller that detects a phase shift direction, and the carrier that is input to the quadrature demodulator based on the detected phase shift and the phase shift direction A phase shifter for controlling the phase of the signal, the phase controller comprising: a baseband signal center point timing calculating means for calculating a center point timing of the in-phase component or quadrature component of the baseband signal; Adding signal center point timing calculating means for calculating the center point timing of the component or orthogonal component, and calculating a difference between the center point timing of the baseband signal and the center point timing of the addition signal, and from the difference, the baseband And a phase control information calculating means for calculating a phase shift amount and a phase shift direction of the in-phase component or the quadrature component of the signal and the addition signal.

  Since the transmitter of the present invention can detect the correct phase shift amount and phase shift direction even if the signal input to the comparator is offset, there is no need to adjust the offset due to product manufacturing variations and temperature changes. Thus, erroneous detection of phase shift by the comparator can be prevented, and the negative feedback amplifier can be operated stably.

It is a figure which shows the structure of the negative feedback amplifier of the transmitting apparatus which concerns on the 1st Embodiment of this invention. It is a figure which shows the structure of the phase controller which concerns on the 1st Embodiment of this invention. It is a timing chart which shows the detection procedure of the phase shift in the phase controller which concerns on the 1st Embodiment of this invention. It is a figure which shows the structure of the negative feedback amplifier of the transmitting apparatus which concerns on the 2nd Embodiment of this invention. It is a figure which shows the structural example of the phase controller which concerns on the conventional embodiment. It is a timing chart which shows the detection procedure of the phase shift in the phase controller which concerns on the conventional embodiment.

  Hereinafter, a first embodiment for carrying out the present invention (hereinafter referred to as “embodiment 1”) will be described with reference to the drawings. In order to stabilize the negative feedback loop in the negative feedback amplifier of the transmitter, the phase difference is set to 0 when a phase difference occurs between the orthogonal component of the input signal (baseband signal) and the orthogonal component of the feedback signal. As described above, the phase is adjusted by the phase controller. The negative feedback amplifier of the transmitter according to the first embodiment can adjust the phase by the phase controller even when the quadrature component of the input signal and the quadrature component of the feedback signal are offset.

  The configuration of the negative feedback amplifier 10 of the transmitter shown in FIG. 1 is used as the negative feedback amplifier 10 of the transmitter (not shown) of the digital radio apparatus using the Cartesian loop negative feedback linearizer system in the first embodiment. I will explain. The negative feedback amplifier 10 includes a baseband signal generator 101, an adder 102a, an adder 102b, a quadrature modulator 103, a bandpass filter (BPF) 104, a power amplifier (PA) 105, an antenna 106, a directional coupler 107, It comprises an attenuator (ATT) 108, a quadrature demodulator 109, a comparator 110a, a comparator 110b, a phase controller 111A, a reference signal generator 112, a PLL frequency synthesizer 113, and a phase shifter 114.

The baseband signal generator 101 outputs the in-phase component (hereinafter referred to as I component) of the baseband signal to the adder 102a, and the quadrature component (hereinafter referred to as Q component) of the baseband signal as the adder 102b and the comparator 110a. Output to.
The adder 102a adds the I component input from the baseband signal generator 101 and the in-phase component (hereinafter referred to as i component) of the feedback signal input from the quadrature demodulator 109 (i component is the subtraction input side of the adder 102a). “I component-i component” input to and added to the quadrature modulator 103 is output to the quadrature modulator 103.
The adder 102b adds the Q component input from the baseband signal generator 101 and the quadrature component (hereinafter referred to as q component) of the feedback signal input from the quadrature demodulator 109 (the q component is the subtraction input side of the adder 102b). “Q component−q component” input to and added to the quadrature modulator 103 is output to the quadrature modulator 103.

The reference signal generator 112 generates a reference frequency signal and outputs the reference signal to the PLL frequency synthesizer 113.
When the reference signal is input from the reference signal generator 112, the PLL frequency synthesizer 113 generates a carrier wave signal based on the reference signal and outputs it to the quadrature modulator 103 and the phase shifter 114.
The quadrature modulator 103 receives “I component-i component” from the adder 102 a, “Q component-q component” from the adder 102 b, and receives a carrier wave signal from the PLL frequency synthesizer 113. ”,“ Q component−q component ”, and a signal converted to a desired frequency by the carrier wave signal is output to a band pass filter (BPF) 104.
When a signal converted to a desired frequency is input from the quadrature modulator 103, the bandpass filter (BPF) 104 removes unnecessary spurious components from the signal and outputs the signal to the power amplifier (PA) 105.
The power amplifier 105 receives the signal from the bandpass filter (BPF) 104, amplifies the signal to a specified output level, and outputs the signal as a high frequency modulation signal, and outputs the high frequency modulation signal to the antenna 106.
When the antenna 106 receives the high frequency modulation signal, the antenna 106 transmits the high frequency modulation signal as a radio wave from the power amplifier 105 via the directional coupler 107.

When the phase shifter 114 receives the carrier signal from the PLL frequency synthesizer 113 and the phase control information from the phase controller 111A, the phase shifter 114 outputs the carrier signal whose phase is controlled by the phase control information to the quadrature demodulator 109.
The directional coupler 107 is provided in a high-frequency modulated signal path from the power amplifier 105 to the antenna 106 in order to realize a function as a negative feedback linearizer by a Cartesian loop. When a part of the high frequency modulation signal is input from the power amplifier 105, the directional coupler 107 outputs it to an attenuator (ATT) 108. A signal output from the directional coupler 107 to the attenuator (ATT) 108 is a feedback signal.
When the attenuator (ATT) 108 receives the feedback signal from the directional coupler 107, the attenuator (ATT) 108 adjusts the power level of the feedback signal to an appropriate value and inputs it to the quadrature demodulator 109.
The quadrature demodulator 109 receives the carrier signal whose phase is controlled from the phase shifter 114, and receives the feedback signal whose power level is adjusted to an appropriate value from the attenuator (ATT) 108, and the i component of the feedback signal And q component is taken out. The quadrature demodulator 109 outputs the i component of the feedback signal to the subtraction input side of the adder 102a, and outputs the q component of the feedback signal to the subtraction input side of the adder 102b and the comparator 110b.

When the Q component of the baseband signal is input from the baseband signal generator 101, the comparator 110a converts the Q component of the baseband signal into a rectangular wave signal based on the reference voltage Vs, and the Q component rectangular wave signal. Is output to the phase controller 111A.
The comparator 110b receives the q component of the feedback signal from the quadrature demodulator 109, converts the q component of the feedback signal into a rectangular wave signal based on the reference voltage Vs, and converts the q component rectangular wave signal to the phase controller. To 111A.
The phase controller 111A detects the phase difference between the Q component rectangular wave and the q component rectangular wave when the Q component rectangular wave signal is input from the comparator 110a and the q component rectangular wave signal is input from the comparator 110b. Thus, phase control information is generated and output to the phase shifter 114. The configuration of the phase controller 111A will be described later.

  The negative feedback amplifier 10 of the transmitter having such a configuration determines in advance a non-modulated signal (hereinafter referred to as a training signal) necessary for the baseband signal generator 101 to generate phase control information by the phase controller 111A. It is output at the timing of the transmitted section (hereinafter referred to as training section).

  Next, a method for detecting the phase shift amount by the phase controller 111A according to the first embodiment will be described with reference to FIGS. FIG. 2 is a diagram illustrating a configuration of the phase controller 111A according to the first embodiment. FIG. 3 is a time chart showing the detection procedure of the phase shift in the phase controller 111A.

  The phase controller 111A according to the first embodiment includes a phase shift detection timing generation unit 201, an up-edge detection unit 202a, a down-edge detection unit 203a, an up-edge detection unit 202b, a down-edge detection unit 203b, a pulse center point detection unit 204a, a pulse The center point detection unit 204b and the phase control information calculation unit 205 are configured.

  The phase shift detection timing generation unit 201 generates an effective section of a training signal for measuring the amount of phase shift as a timing signal, generates a start timing signal of the detection section, an up edge detection unit 202a, a down edge detection unit 203a, It outputs to the up edge detection part 202b and the down edge detection part 203b.

When the up-edge detection unit 202a receives the Q component signal from the comparator 110a and the start timing signal from the phase shift detection timing generation unit 201, the up-edge timing at which the Q component signal switches from Low to High using the start timing signal as a reference timing. Is detected. Specifically, the counter is cleared to 0 by the start timing signal input from the phase shift detection timing generation unit 201, and the counter is updated by 1 at the sampling timing. Further, the Q component signal is sampled, and the counter is stopped at the timing when the signal state is switched from Low to High. The count value thus counted is output as up-edge detection information to the pulse center point detection unit 204a.
When the down edge detection unit 203a receives the Q component signal from the comparator 110a and the start timing signal from the phase shift detection timing generation unit 201, the down edge timing at which the Q component signal switches from High to Low using the start timing signal as a reference timing. Is detected. Specifically, the counter is cleared to 0 by the start timing signal input from the phase shift detection timing generation unit 201, and the counter is updated by 1 at the sampling timing. Also, the Q component signal is sampled, and the counter is stopped at the timing when the signal state is switched from LHigh to Low. The count value thus counted is output to the pulse center point detection unit 204a as down edge detection information.

When the up-edge detection unit 202b receives the q-component signal from the comparator 110b and the start-timing signal from the phase shift detection timing generation unit 201, the up-edge timing at which the q-component signal switches from Low to High using the start timing signal as a reference timing. Is detected. Specifically, the counter is cleared to 0 by the start timing signal input from the phase shift detection timing generation unit 201, and the counter is updated by 1 at the sampling timing. Further, the q component signal is sampled, and the counter is stopped at the timing when the signal state is switched from Low to High. The count value thus counted is output as up-edge detection information to the pulse center point detection unit 204b.
When the down edge detection unit 203b receives the q component signal from the comparator 110b and the start timing signal from the phase shift detection timing generation unit 201, the down edge timing at which the q component signal switches from High to Low using the start timing signal as a reference timing. Is detected. Specifically, the counter is cleared to 0 by the start timing signal input from the phase shift detection timing generation unit 201, and the counter is updated by 1 at the sampling timing. Also, the Q component signal is sampled, and the counter is stopped at the timing when the signal state is switched from LHigh to Low. The count value thus counted is output to the pulse center point detection unit 204b as down edge detection information.

When the pulse center point detector 204a receives the up edge timing information of the Q component signal from the up edge detector 202a and the down edge timing information of the Q component signal from the down edge detector 202b, the pulse center point detector 204a receives the pulse signal in the Q component signal. The timing of the center point of the High section is calculated and output to the phase control information calculation unit 205.
When the pulse center point detection unit 204b receives the up edge timing information of the q component signal from the up edge detection unit 202b and the down edge timing information of the q component signal from the down edge detection unit 202b, the pulse center point detection unit 204b receives the pulse signal in the q component signal. The timing of the center point of the High section is calculated and output to the phase information calculation unit 205.
When the phase information calculation unit 205 receives the center point timing information of the Q component signal from the pulse center point detection unit 204a and the center point timing information of the q component signal from the pulse center point detection unit 204b, the phase information calculation unit 205 The phase control information is set by the center point timing information of the q component signal, and is output to the phase shifter 114.

  Next, a method for detecting a phase shift in the phase controller 111A will be described with reference to FIG. FIG. 3A is a time chart showing a procedure for the phase controller 111A to detect a phase shift when the Q component signal input by the comparator 110a and the q component signal input by the comparator 110b are not offset. It is a chart. FIG. 3B is a time chart showing a procedure in which the phase controller 111A detects a phase shift when the Q component signal input by the comparator 110a and the q component signal input by the comparator 110b are offset. It is.

  First, the procedure for detecting the phase shift by the phase controller 111A when the offset is not shown in FIG. 3A will be described below. The Q component sine wave 301 is a sine waveform of the Q component of the baseband signal input from the baseband signal generator 101 by the comparator 110a. The Q component rectangular wave 302 is a rectangular waveform output from the comparator 110a by inputting the Q component sine wave 301 and the reference voltage Vs to the comparator 110a. The q component sine wave 303 is a sine waveform of the component q of the feedback signal input from the direct demodulator 109 by the comparator 110b. The q component rectangular wave 304 is a rectangular waveform output from the comparator 110b by inputting the q component sine wave 603 and the reference voltage Vs to the comparator 110b. The phase shift amount detection rectangular wave 304 is a waveform obtained by calculating an exclusive OR of the Q component rectangular wave 302 and the q component rectangular wave 304. As shown in FIG. 3A, when the phase of the q-component sine wave 303 is delayed with respect to the Q-component sine wave 301, the q-component rectangular wave 304 is also a rectangular wave with a phase delayed with respect to the Q-component rectangular wave 302. Become. Therefore, the phase shift amount is detected from the phase difference between the center 305 of the Q component rectangular wave 302 and the center 306 of the phase shift detection rectangular wave 304.

  Next, the procedure by which the phase controller 111A detects the phase shift amount will be specifically described with reference to FIG. Within the phase shift detection interval, the up edge detection unit 202a detects the timing (t1) of the up edge (Low → High) of the Q component sine wave 301 with reference to the head (t0) of the detection interval, and the down edge detection unit 203a detects the timing (t2) of the down edge (High → Low) of the Q component sine wave 301. Similarly, the up-edge detection unit 202b detects the timing (t3) of the up-edge (Low → High) of the q-component sine wave 303, and the down-edge detection unit 203b detects the down-edge (High → Low) of the q-component sine wave 303. The timing (t4) is detected. In addition, the pulse center point detection unit 204a determines the center point timing of the high interval (pulse signal) from the timing (t1) of the up edge of the Q component sine wave 301 and the timing (t2) of the down edge of the Q component sine wave 301. Q_center (= t1 + (t2−t1) / 2) is calculated. Similarly, the pulse center point detection unit 204b calculates the center point of the high interval (pulse signal) from the timing (t3) of the up edge of the q component sine wave 303 and the timing (t4) of the down edge of the q component sine wave 303. Timing q_center (= t3 + (t4-t3) / 2) is calculated. The phase control information calculation unit 205 calculates q_center−Q_center (= (t4 + t3−t2−t1) / 2), which is a phase shift amount, from the calculated Q_center and q_center, and the phase advance / Detect the direction of delay.

Next, a detection example of the amount of phase shift when the phase shift detection section in the 4 KHz sine wave as the training signal is set to a half cycle or more and one cycle or less of the sine wave from the baseband signal generator 101 will be described. In order to calculate the amount of deviation, the counter is operated while sampling the pulse signals output from the comparators 110a and 110b with a 10 MHz clock. When a 4 KHz sine wave is sampled with a 10 MHz clock, the counter counts 2500 in one cycle.
For example,
Start of phase shift detection section: t0 = 0
Q-component up-edge timing of comparator 110a output: t1 = 417
Q component down edge timing of comparator 110a output: t2 = 1667
Q-component up-edge timing of comparator 110b output: t3 = 625
Q component down edge timing of comparator 110b output: t4 = 1875
Q_center = t1 + (t2-t1) / 2 = 417 + (1667-417) / 2
= 1042
q_center = t3 + (t4-t3) / 2 = 625 + (1875-625) / 2
= 1250
q_center-Q_center = 1250-1042 = 208
By
Phase shift amount: 360 × 208/2500 = 30 degrees Phase shift direction: a direction in which the q component is delayed with respect to the Q component is detected, the phase shift amount is “30 degrees”, and the phase shift direction is “q with respect to the Q component. The phase control information is generated based on the phase shift information in the direction in which the component is delayed, and is output to the phase shifter 114.

  Next, a procedure for detecting the phase shift by the phase controller 111A in the case of the offset shown in FIG. 3B will be described below. The Q ′ component sine wave 311 is a sine waveform of the Q ′ component of the baseband signal offset to the upper side inputted from the baseband signal generator 101 by the comparator 110 a. The Q ′ component rectangular wave 312 is a rectangular waveform output from the comparator 110 a by inputting the Q ′ component sine wave 311 and the reference voltage Vs to the comparator 110 a. The q ′ component sine wave 313 is a sine waveform of the feedback signal component q ′ input to the comparator 110 b from the direct demodulator 109 and offset downward. The q ′ component rectangular wave 314 is a rectangular waveform output from the comparator 110 b by inputting the q ′ component sine wave 313 and the reference voltage Vs to the comparator 110 b. As shown in FIG. 3B, when the phase of the q ′ component sine wave 313 is delayed with respect to the Q ′ component sine wave 311, the phase of the q ′ component rectangular wave 314 is also delayed with respect to the Q ′ component rectangular wave 312. It becomes a square wave. Therefore, the phase shift amount is detected from the phase difference between the center 315 of the Q ′ component rectangular wave 312 and the center 316 of the phase shift detection rectangular wave 314.

  Next, the procedure by which the phase controller 111A detects the phase shift amount will be specifically described with reference to FIG. Within the phase shift detection interval, the up-edge detector 202a detects the timing (t1 ′) of the up-edge (Low → High) of the Q ′ component sine wave 311 with reference to the start (t0 ′) of the detection interval, and the down The edge detection unit 203a detects the timing (t2 ′) of the down edge (High → Low) of the Q ′ component sine wave 311. Similarly, the up-edge detection unit 202b detects the timing (t3 ′) of the up-edge (Low → High) of the q ′ component sine wave 313, and the down-edge detection unit 203b detects the down-edge (High) of the q ′ component sine wave 313. → Low) timing (t4 ′) is detected. The pulse center point detection unit 204a determines the center of the high interval (pulse signal) from the timing (t1 ′) of the up edge of the Q ′ component sine wave 311 and the timing (t2 ′) of the down edge of the Q ′ component sine wave 311. The point timing Q_center (= t1 ′ + (t2′−t1 ′) / 2) is calculated. Similarly, the pulse center point detection unit 204b determines the high interval (pulse signal) from the up edge timing (t3 ′) of the q ′ component sine wave 313 and the down edge timing (t4 ′) of the q ′ component sine wave 313. ) Center point timing q_center (= t3 ′ + (t4′−t3 ′) / 2). The phase control information calculation unit 205 calculates the phase from the calculated Q_center and q_center, q_center−Q_center (= (t4 ′ + t3′−t2′−t1 ′) / 2), which is a phase shift amount, and the sign of q_center−Q_center. Detect the direction of advance / delay.

Next, a detection example of the amount of phase shift when the phase shift detection section in the 4 KHz sine wave as the training signal is set to a half cycle or more and one cycle or less of the sine wave from the baseband signal generator 101 will be described. In order to calculate the amount of deviation, the counter is operated while sampling the pulse signals output from the comparators 110a and 110b with a 10 MHz clock. When a 4 KHz sine wave is sampled with a 10 MHz clock, the counter counts 2500 in one cycle.
For example,
Start of phase shift detection section: t0 ′ = 0
Q ′ component up-edge timing of comparator 110a output: t1 ′ = 313
Q ′ component down edge timing of comparator 110a output: t2 ′ = 1771
Q ′ component up-edge timing of comparator 110b output: t3 ′ = 833
Q ′ component down edge timing of comparator 110b output: t4 ′ = 1667
Q′_center = t1 ′ + (t2′−t1 ′) / 2 = 313 + (1771-313) / 2
= 1042
q′_center = t3 ′ + (t4′−t3 ′) / 2 = 833 + (1667−833) / 2
= 1250
q'_center-Q'_center = 1250-1042 = 208
By
Phase shift amount: 360 × 208/2500 = 30 degrees Phase shift direction: a direction in which the q ′ component is delayed with respect to the Q ′ component is detected, the phase shift amount is “30 degrees”, and the phase shift direction is “Q ′ component. On the other hand, phase control information is generated based on phase shift information indicating that the q ′ component is delayed.

  As shown in FIG. 3, when the Q ′ component is offset upward with respect to the reference voltage Vs and inputted to the comparator 110a, the Q ′ component output from the comparator 110a of FIG. The pulse signal has a long High period compared to the Q component output from the comparator 110a in (a). However, when the center point timing Q′_center of the High section (pulse signal) is calculated, the same timing as Q_center calculated in FIG. When the q ′ component is offset downward with respect to the reference voltage Vs and input to the comparator 110b, the comparator 110b in FIG. 3B outputs the output q ′ component in the comparison in FIG. 3A. Compared to the q component output from the device 110b, the pulse signal has a shorter High period. However, when the center point timing q′_center of the High section (pulse signal) is calculated, the same timing as q_center calculated in FIG. Thus, even if the Q component signal input to the comparator 110a is an offset signal with respect to the reference voltage Vs, or the q component signal input to the comparator 110b is relative to the reference voltage Vs. Even for an offset signal, it is possible to detect the exact phase shift amount and the phase shift direction by detecting the center point timing of the High pulse signal and calculating the phase shift.

  In the first embodiment, in order to easily explain the phase shift between the Q component and the q component, or between the Q ′ component and the q ′ component, the phase difference is greatly illustrated, and an example in which the phase shift amount is 30 degrees will be described. did. However, in practice, when a phase shift greater than a predetermined threshold is detected, a fixed amount of phase is corrected each time. For example, when phase correction is performed when a 4 KHz sine wave is shifted by 1 degree (π / 180 radians), if the 4 KHz sine wave is sampled with a 10 MHz clock, the counter becomes 2500 counts in one cycle. In the case of the degree deviation, q_center−Q_center = 2500/360 = 6.9 counts. Therefore, when the count value is 6.9 or more (actually 7 counts) or more, it is determined that the phase has changed by one degree or more, and the phase control information is obtained from the phase deviation information of the phase deviation amount for one degree. Is generated and output to the phase shifter 114.

  Hereinafter, a second embodiment (hereinafter referred to as “second embodiment”) for carrying out the present invention will be described with reference to the drawings. In the second embodiment, the negative feedback amplifier 20 of the transmitter detects a phase shift between the quadrature component of the input signal (baseband signal) and the quadrature component obtained by adding the quadrature component of the input signal and the quadrature component of the feedback signal. The phase controller can also adjust the phase of the offset input signal orthogonal component and the feedback signal orthogonal component.

  The negative feedback circuit 20 of the transmitter of the digital radio apparatus using the Cartesian negative feedback linearizer in Embodiment 2 will be described using the configuration of the negative feedback amplifier 20 of the transmitter shown in FIG. Similarly to the negative feedback amplifier 10 of the first embodiment, the negative feedback circuit 20 includes a baseband signal generator 101, an adder 102a, an adder 102b, a quadrature modulator 103, a bandpass filter (BPF) 104, a power amplifier (PA). ) 105, antenna 106, directional coupler 107, attenuator (ATT) 108, quadrature demodulator 109, comparator 110a, comparator 110b, phase controller 111A, reference signal generator 112, PLL frequency synthesizer 113, and phase Device 114.

  However, in the second embodiment, the comparator 110b does not input the q component signal of the feedback signal from the quadrature demodulator 109 as in the first embodiment, but adds the Q component and the q component from the output of the adder 102b. The comparator 110b compares the signal obtained by adding the Q component and the q component with the reference voltage Vs. With this configuration, the phase controller 111A detects a phase shift between the input quadrature component and the component obtained by adding the input quadrature component and the feedback quadrature component, and adjusts the phase even in the offset input signal. It can be performed.

  In the first embodiment, the phase shift between the input quadrature component Q and the feedback quadrature component q is detected. However, the phase shift between the input common-mode component I and the feedback common-mode component i can also be detected. . In the second embodiment, the input quadrature component Q, and the phase shift of the signal Q-q obtained by adding the input quadrature component Q and the feedback quadrature component q are detected. However, the input in-phase component I and the input in-phase component I It is also possible to adopt a configuration for detecting a phase shift of the signal I-i obtained by adding I and the feedback in-phase component i.

In summary, the present invention has the following characteristics.
(1): The transmitter of the present invention inputs in-phase and quadrature components of a baseband signal, quadrature modulates the baseband signal with a carrier signal, amplifies the quadrature-modulated signal to a predetermined power level, and transmits it. In the transmitter, a directional coupler that extracts a part of the output signal of the transmitter as a feedback signal, and the feedback signal branched by the directional coupler is quadrature demodulated into a feedback in-phase component and a feedback quadrature component by the carrier wave signal A quadrature demodulator, a phase controller for detecting a phase shift amount and a phase shift direction of the baseband signal and the feedback signal, and the detected phase shift and the phase shift direction are input to the quadrature demodulator. A phase shifter for controlling a phase of the carrier signal, and the phase controller calculates a central point timing of an in-phase component or a quadrature component of the baseband signal. Subband signal center point timing calculation means, feedback signal center point timing calculation means for calculating center point timing of the in-phase component or quadrature component of the feedback signal, center point timing of the baseband signal, and center point timing of the feedback signal And a phase control information calculating means for calculating a phase shift amount and a phase shift direction of the in-phase component or the quadrature component of the baseband signal and the feedback signal.
(2): The baseband signal center point timing calculation means of the transmitter of (1) according to the present invention includes a first comparator that inputs the baseband signal and a reference voltage and converts them into a baseband signal rectangular waveform. First base point timing calculation means for calculating a center point timing from the timing of the up edge and the down edge of the baseband signal rectangular waveform, and the center point timing of the baseband signal rectangular waveform is determined as the baseband signal. The feedback signal center point timing calculation means inputs a feedback signal and a reference voltage and converts it into a feedback signal rectangular waveform, and an up edge and a down edge of the feedback signal rectangular waveform. A second center point timing calculating means for calculating a center point timing from the edge timing, and the feedback signal rectangular wave Is characterized in that the center point timing to the center point timing of the feedback signal.
(3): In the transmitter of the present invention of (2), the first comparator inputs the in-phase component of the baseband signal to the first comparator, and the second comparator supplies the feedback. The in-phase component of the signal is input, or the first comparator inputs the quadrature component of the baseband signal, and the second comparator inputs the quadrature component of the feedback signal. It is a feature.
(4): The transmitter of the present invention receives the in-phase component and the quadrature component of the baseband signal, quadrature modulates the baseband signal with the carrier signal, and amplifies the quadrature-modulated signal to a predetermined power level for transmission. In the transmitter, a directional coupler that extracts a part of the output signal of the transmitter as a feedback signal, and the feedback signal branched by the directional coupler is quadrature demodulated into a feedback in-phase component and a feedback quadrature component by the carrier wave signal An orthogonal demodulator, a phase controller that detects a phase shift amount and a phase shift direction of the baseband signal, and an addition signal obtained by adding the baseband signal and the feedback signal, and the detected phase shift A phase shifter for controlling a phase of the carrier signal input to the quadrature demodulator according to a phase shift direction, and the phase controller includes an in-phase component of the baseband signal. Or baseband signal center point timing calculating means for calculating the center point timing of the quadrature component, sum signal center point timing calculating means for calculating the center point timing of the in-phase component or quadrature component of the sum signal, and the baseband signal Phase control information calculation means for calculating a phase shift amount and a phase shift direction of an in-phase component or a quadrature component of the baseband signal and the addition signal from a center point timing and a center point timing of the addition signal. It is a feature.
(5): The baseband signal center point timing calculation means of the transmitter of the present invention of (4) includes: a first comparator that inputs the baseband signal and a reference voltage and converts them into a baseband signal rectangular waveform; First base point timing calculation means for calculating a center point timing from the timing of the up edge and the down edge of the baseband signal rectangular waveform, and the center point timing of the baseband signal rectangular waveform is determined as the baseband signal. The sum signal center point timing calculation means inputs a sum signal and a reference voltage and converts the sum signal into a sum signal rectangular waveform, and up and down edges of the sum signal rectangular waveform. A second center point timing calculating means for calculating center point timing from edge timing, and the added signal rectangular wave Is characterized in that the center point timing to the center point timing of the addition signal.
(6): The first comparator of the transmitter of the present invention of (5) receives the in-phase component of the baseband signal, and the second comparator and the in-phase component of the baseband signal. The sum of the feedback signal and the in-phase component is used as the addition signal, and the in-phase component is input, or the first comparator inputs the quadrature component of the baseband signal, and The second comparator is characterized in that a sum of the quadrature component of the baseband signal and the quadrature component of the feedback signal is used as the sum signal, and the quadrature component is input.

  As described above, the present invention is correct even from a signal offset to the comparator by detecting the center point timing of the pulsed comparator output signal in the transmitter having a negative feedback circuit. Since it is possible to detect the phase shift amount and the phase shift direction, it is not necessary to detect erroneous phase shift detection or to adjust the offset of the signal input to the comparator, and the negative feedback amplifier can be operated stably. it can. Furthermore, it is possible to adjust fluctuations in characteristics due to temperature changes and to adjust variations in characteristics from product to product.

  Although the present invention has been described with specific embodiments, it is needless to say that the above-described embodiments are examples of the present invention and are not limited to these embodiments.

  The present invention can be applied to a device equipped with a negative feedback amplifier.

10 .... Negative feedback circuit 20 .... Negative feedback circuit 101 ... Baseband signal generator 102a ... Adder 102b ...・ Adder 103... Quadrature modulator 104... Bandpass filter (BPF)
105... Power amplifier (PA)
106 ... Antenna 107 ... Directional coupler 108 ... Attenuator (ATT)
109... Quadrature demodulator 110 a... Comparator 110 b... Comparator 111 A... Phase controller 111 B. Reference signal generator 113 PLL frequency synthesizer 114 Phase shifter 201 Phase shift detection timing generator 202a ... ... Up-edge detector 202b ... Up-edge detector 203a ... Down-edge detector 203b ... Down-edge detector 204a ... Pulse center inspection Output unit 204b... Pulse center point detection unit 205... Phase information calculation unit 301... Q component sine wave 302. 303 ······ q component sine wave 304 ··· Q component rectangular wave 305... Q component rectangular wave center 306... Q component rectangular wave center 311... Q ′ component sine wave 312. ... Q 'component rectangular wave 313 ... q' component sine wave 314 ... q 'component rectangular wave 315 ......... Q' component rectangular wave Center 316... Q ′ component square wave center 501... Exclusive OR 502... Sample timing generation unit 503. 504... Phase shift detection timing generator 505... Counter 506... Down edge detector 507. ······································································· Q component sine wave Wave 603 ······ q component sine wave 604 ······ q component rectangular wave 605 ······· phase shift amount detection rectangular wave 611 ······· Q 'component Sine wave 612 ... Q 'component rectangular wave 613 ... q' component sine wave 614 ... q 'component rectangular wave 615 ... phase Deviation detection square wave

Claims (4)

In transmitter inputs the in-phase and quadrature components of the baseband signal, the baseband signal to quadrature modulation by a carrier signal, to have the amplifier for amplifying the quadrature modulated linear modulation signal to a predetermined power level,
The amplifier is a Cartesian loop negative feedback amplifier that performs nonlinear distortion compensation,
A directional coupler that extracts a part of the output signal of the transmitter as a feedback signal;
A quadrature demodulator that quadrature demodulates the feedback signal branched by the directional coupler into a feedback in-phase component and a feedback quadrature component by the carrier wave signal;
A phase controller that detects a phase shift amount and a phase shift direction of the baseband signal and the feedback signal;
A phase shifter that controls the phase of the carrier signal input to the quadrature demodulator according to the detected phase shift and phase shift direction;
The phase controller is
Baseband signal center point timing calculating means for calculating center point timing of the in-phase component or quadrature component of the baseband signal;
Feedback signal center point timing calculating means for calculating the center point timing of the in-phase component or the quadrature component of the feedback signal;
The difference between the center point timing of the baseband signal and the center point timing of the feedback signal is calculated, and from the difference , the phase shift amount and phase shift direction of the in-phase component or the quadrature component of the base band signal and the feedback signal are calculated. Phase control information calculating means for calculating;
A transmitter comprising: The baseband signal center point timing calculation means includes:
A first comparator that inputs the baseband signal and a reference voltage to convert the baseband signal into a baseband signal rectangular waveform;
First center point timing calculating means for calculating center point timing from the timing of the up edge and the down edge of the baseband signal rectangular waveform;
The center point timing of the baseband signal rectangular waveform is the center point timing of the baseband signal,
The feedback signal center point timing calculation means includes:
A second comparator that inputs the feedback signal and a reference voltage to convert it into a feedback signal rectangular waveform;
A second center point timing calculating means for calculating a center point timing from the timing of the up edge and the down edge of the feedback signal rectangular waveform;
The transmitter according to claim 1, wherein a center point timing of the feedback signal rectangular waveform is a center point timing of the feedback signal. The first comparator inputs the in-phase component of the baseband signal and the second comparator inputs the in-phase component of the feedback signal;
The transmission according to claim 2, wherein the first comparator inputs the quadrature component of the baseband signal, and the second comparator inputs the quadrature component of the feedback signal. Machine. In transmitter inputs the in-phase and quadrature components of the baseband signal, the baseband signal to quadrature modulation by a carrier signal, to have the amplifier for amplifying the quadrature modulated linear modulation signal to a predetermined power level,
The amplifier is a Cartesian loop negative feedback amplifier that performs nonlinear distortion compensation,
A directional coupler that extracts a part of the output signal of the transmitter as a feedback signal;
A quadrature demodulator that quadrature demodulates the feedback signal branched by the directional coupler into a feedback in-phase component and a feedback quadrature component by the carrier wave signal;
A phase controller that detects a phase shift amount and a phase shift direction between the baseband signal and an addition signal obtained by adding the baseband signal and the feedback signal;
A phase shifter that controls the phase of the carrier signal input to the quadrature demodulator according to the detected phase shift and phase shift direction;
The phase controller is
Baseband signal center point timing calculating means for calculating center point timing of the in-phase component or quadrature component of the baseband signal;
An addition signal center point timing calculating means for calculating a center point timing of an in-phase component or a quadrature component of the addition signal;
The difference between the center point timing of the baseband signal and the center point timing of the sum signal is calculated, and the phase shift amount and phase shift direction of the in-phase component or the quadrature component of the base band signal and the sum signal are calculated from the difference. Phase control information calculating means for calculating;
A transmitter comprising:

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