JP6083310B2 - Non-contact power feeding apparatus and control method thereof - Google Patents

Description

  The present invention relates to a non-contact power feeding device and a control method thereof.

  Conventionally, a non-contact power supply apparatus that charges a load such as a battery in a non-contact manner is known. Such a non-contact power feeding device includes a power supply side coil and a charging side coil, and is configured to transmit and charge high-frequency power by electromagnetic induction. The non-contact power supply apparatus is set so that the primary capacitor on the power transmission side resonates at the same resonance frequency as the inductance component on the secondary side including the load, and the power factor is set to “1”.

  In addition, as a non-contact power feeding device, a device that controls the frequency of high-frequency power according to the resistance component of a load has been proposed in order to perform efficient power transmission. In this non-contact power supply device, even if the resistance component of the load changes, the power factor of the supplied power can be set to “1” in order to control the frequency of the high frequency power (see Patent Document 1).

JP 2002-272134 A

  However, the contactless power supply device described in Patent Document 1 has a problem in that the intensity of the radiated electric field increases because the magnetic field generated from the coil increases. Therefore, when a structural member that reduces the magnetic field to suppress the radiated electric field intensity is provided, there is a problem that the apparatus itself is enlarged although the radiated electric field intensity can be suppressed.

  The present invention has been made in order to solve such a conventional problem, and an object of the present invention is to provide a non-contact power feeding apparatus capable of preventing an increase in size while suppressing a radiated electric field intensity and a control thereof. It is to provide a method.

  In the present invention, when the magnetic pole direction of the first coil at the time of power transmission / reception is opposite to the magnetic pole direction of the second coil, the first coil and the second coil are lower than the resonance frequency of the power receiving circuit. The inverter is driven by selecting a frequency within a range where the combined ampere turn with the coil is smaller than the combined ampere turn at the resonance frequency of the power receiving side circuit.

  The combined ampere turn is a concept including a combined current obtained by removing the concept of the number of turns from the combined ampere turn.

  According to the present invention, since the inverter is driven by selecting a frequency within the above range, the combined ampere turn can be reduced by selecting the frequency, and it is not necessary to provide a structural member, and the radiation electric field strength can be reduced. Therefore, enlargement can be prevented while suppressing the radiation electric field intensity.

It is a lineblock diagram showing the non-contact electric supply device concerning a 1st embodiment. It is the schematic which shows the magnetic field which generate | occur | produces when a 1st coil and a 2nd coil are disk type coils. It is the schematic which shows the magnetic field which generate | occur | produces when a 1st coil and a 2nd coil are solenoid type coils. It is a figure which shows the correlation with the drive frequency of an inverter when load electric power is made constant, and the synthetic | combination ampere turn in a disk type coil. It is a figure which shows the correlation with the drive frequency of an inverter when load electric power is made constant, and the synthetic | combination ampere turn in a solenoid type coil. It is a figure which shows the mode of the selection of the drive frequency by the drive frequency selection part shown in FIG. It is a correlation diagram which shows a frequency and the output phase of the inverter. It is a wave form diagram which shows the magnetic flux by the electric current which flows into a coil, (a) shows a 1st example, (b) shows a 2nd example, (c) has shown the 3rd example. It is a flowchart which shows the control method of the non-contact electric power feeder which concerns on 1st Embodiment. It is a block diagram which shows the non-contact electric power feeder which concerns on 2nd Embodiment. It is a figure explaining the adjustment method by the duty ratio which concerns on 2nd Embodiment, (a) shows before adjustment, (b) has shown after adjustment. It is a flowchart which shows the control method of the non-contact electric power feeder which concerns on 2nd Embodiment.

  Preferred embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a configuration diagram illustrating a contactless power supply device according to the first embodiment. As shown in FIG. 1, the non-contact power feeding device 1 is used for charging a load 24 mounted on a vehicle, for example, and includes a power feeding side circuit 10 and a power receiving side circuit 20. When the non-contact power feeding device 1 is used for charging a vehicle battery (an example of the load 24), the power feeding side circuit 10 is provided on the ground side, and the power receiving side circuit 20 is provided on the vehicle side.

  The power feeding side circuit 10 transmits high frequency power to the power receiving side circuit 20, and includes a commercial power source 11, a rectifying unit 12, a power factor correction boosting unit 13, an inverter 14, and a first capacitor C1. And a first coil L1.

The commercial power supply 11 is a 50 Hz or 60 Hz AC power supply. The rectifying unit 12 rectifies the AC voltage 12 from the commercial power supply 11. The power factor improving step-up unit 13 is a step-up type PFC circuit. By switching the switching element S ₀ , generation of harmonic current due to the alternating current rectified by the rectifying unit 12 is suppressed, and the power factor is increased to “ 1 ".

The inverter 14 includes a smoothing capacitor and switching elements S _{1 to} S ₄ , and generates high-frequency AC power by switching the switching elements S _{1 to} S ₄ . Here, the generated high-frequency current is supplied to a resonance circuit including the first capacitor C1 and the first coil L1. In addition, although the 1st capacitor | condenser C1 and the 1st coil L1 are connected in series, you may connect not only in this but in parallel.

  The power receiving side circuit 20 receives high frequency power from the power feeding side circuit 10 and includes a second coil L2, a second capacitor C2, a rectifying unit 21, a filter unit 22, a relay 23, and a load 24. And.

The second coil L2 receives high-frequency power by electromagnetic induction with the first coil L1, and constitutes a resonance circuit together with the second capacitor C2. In addition, although the 2nd capacitor | condenser C2 and the 2nd coil L2 are connected in series, you may connect not only in this but in parallel.

  The rectifier 21 is a rectifier circuit that rectifies high-frequency power from the resonance circuit into direct current. The filter 22 is configured by a smoothing capacitor that suppresses voltage fluctuation. The relay 23 is configured by a relay switch that can be switched on and off. The load 24 is a battery and is to be charged.

  Furthermore, as shown in FIG. 1, the power receiving circuit 20 includes a controller 25. The controller 25 includes a load power detection unit (power detection unit) 25a, a load control unit (command value calculation unit) 25b, and a coupling coefficient detection unit 25c. The load power detector 25a detects the power supplied to the load 24 to be charged. The load control unit 25b calculates a command value of power supplied to the load 24.

  The coupling coefficient detector 25c detects a coupling coefficient representing the degree of coupling between the first coil L1 and the second coil L2. The coupling coefficient detection unit 25c can be obtained from the induced voltage on the power receiving side circuit 20 generated when, for example, the relay 23 is opened and current is supplied only to the power feeding side circuit 10. It is not limited to.

  Note that the controller 25 may be equipped with the above functions in one control unit, or may be distributed and installed in a plurality of control units. For example, a battery control unit that controls a battery as the load 24 includes a load power detection unit 25a and a load control unit 25b, and controls wireless communication between the power supply side circuit 10 and the power reception side circuit 20 or a battery control unit. You may comprise so that the coupling coefficient detection part 25c may be provided in the non-contact electric power feeding control unit which communicates with.

  Further, the power supply side circuit 10 includes a power control unit (drive means) 15. The power control unit 15 drives the inverter 14 to generate high-frequency power to be supplied to the first coil L1, and includes a duty generation unit 15a, a drive frequency selection unit 15b, and a pulse generation unit 15c. I have.

  The duty generation unit 15a generates a duty ratio such that the current power Pout detected by the load power detection unit 25a becomes the command value Pref of the power detected by the load control unit 25b, for example, PI control Is done. The drive frequency selection unit 15 b generates a drive frequency for the inverter 14.

The pulse generation unit 15c generates a pulse having the frequency generated by the drive frequency selection unit 15b and the duty ratio generated by the duty generation unit 15a. The switching elements S _{1 to} S ₄ of the inverter 14 are driven by this pulse.

  In particular, in the non-contact power feeding device 1 according to the first embodiment, the first and second coils L1 and L2 are solenoid coils, and the drive frequency selection unit 15b has a frequency higher than the resonance frequency of the power receiving side circuit 20. A frequency is selected that is low and within a range in which the combined ampere turn of the first coil L1 and the second coil L2 is smaller than the combined ampere turn at the resonance frequency of the power receiving side circuit 20. Thereby, in 1st Embodiment, enlargement can be prevented, suppressing radiation electric field strength. Hereinafter, this point will be described in detail.

FIG. 2 is a schematic diagram showing a magnetic field generated when the first coil L1 and the second coil L2 are disk-type coils. In FIG. 2, the first coil L1 and the second coil L2
The winding direction is the same. In FIG. 2, the load 24 is indicated as a resistance _RL for convenience.

First, the disk type coil is one in which the axial directions of the coils L1 and L2 are the vehicle vertical direction as shown in FIG. In this case, the magnetic field is as follows during power transmission / reception. That is, when applying an AC voltage V1 with respect to the power feeding side circuit 10, and the voltage e ₁ and the current I ₁ is generated with respect to the first coil L1. In the power receiving side circuit 20, a voltage e ₂ and a current I ₂ are generated in the second coil L 2 based on the equation e ₁ · I ₁ = e ₂ · I ₂ .

At this time, a magnetic flux φ1 is generated in the _first coil L1. As is apparent from the magnetic flux φ1, _one end of the first coil L1 (upper side in FIG. 2) corresponds to the N pole in terms of a magnet, and the other end (lower side in FIG. 2) of the first coil L1. Corresponds to the S pole in terms of a magnet.

Similarly, a magnetic flux φ2 is generated in the _second coil L2. As is apparent from the flux phi _2, one end of the second coil L2 corresponds to the N pole as referred by a magnet (upper side in FIG. 2), the other end of the second coil L2 (lower side in FIG. 2) Corresponds to the S pole in terms of a magnet.

For this reason, in the example shown in FIG. 2, the magnetic pole directions of the first coil L1 and the second coil L2 are the same. The total magnetic flux φ _Total in the example shown in FIG. 2 is φ _Total = φ ₁ + φ ₂ .

That is, as shown in FIG. 2, if the axial direction of the first coil L1 and the second coil L2 is in the vehicle vertical direction, the magnetic flux phi ₁ and the second generated in the first coil L1 the flux phi ₂ generated in the coil L2 becomes the same direction, the whole of the magnetic flux phi _total each of the magnetic flux phi _1, phi
₂ is added. In FIG. 2, φm is a combined magnetic flux of magnetic fluxes φ ₁ and φ ₂ .

FIG. 3 is a schematic diagram showing a magnetic field generated when the first coil L1 and the second coil L2 are solenoid coils. In FIG. 3, the winding directions of the first coil L1 and the second coil L2 are the same. In FIG. 3, the load 24 is indicated as a resistance _RL for convenience.

The solenoid type coil is one in which the axial direction of the coils L1, L2 is the vehicle plane direction as shown in FIG. In this case, the magnetic field is as follows during power transmission / reception. That is, when applying an AC voltage V1 with respect to the power feeding side circuit 10, and the voltage e ₁ and the current I ₁ is generated with respect to the first coil L1. In the power receiving side circuit 20, a voltage e ₂ and a current I ₂ are generated in the second coil L 2 based on the equation e ₁ · I ₁ = e ₂ · I ₂ .

At this time, a magnetic flux φ1 is generated in the _first coil L1. As is apparent from the magnetic flux φ1, _one end of the first coil L1 (right side in FIG. 3) corresponds to the N pole in terms of a magnet, and the other end (left side in FIG. 3) of the first coil L1. This corresponds to the S pole in terms of a magnet.

Similarly, a magnetic flux φ2 is generated in the _second coil L2. As is apparent from the flux phi _2, one end of the second coil L2 (3 right) corresponds to the S pole as referred in the magnet, the other end of the second coil L2 (the left side in FIG. 3) is This corresponds to the N pole in terms of a magnet.

For this reason, in the example shown in FIG. 3, the magnetic pole directions of the first coil L1 and the second coil L2 are opposite. Then, the total magnetic flux φ _Total in the example illustrated in FIG. 3 is φ _Total = φ ₁ −φ ₂ .

That is, as shown in FIG. 3, if the axial direction of the first coil L1 and the second coil L2 is in the vehicle plane direction, the magnetic flux phi ₁ and the second generated in the first coil L1 the direction is opposite to the magnetic flux phi ₂ generated in the coil L2, the total magnetic flux phi _total becomes minus the flux phi ₂ from the magnetic flux phi _1. In FIG. 3, φm is a combined magnetic flux of magnetic fluxes φ ₁ and φ ₂ .

なお、Ｎ_１は第１のコイルＬ１のターン数であり、Ｓ_１は第１のコイルＬ１の内側面積である。また、Ｎ_２は第２のコイルＬ２のターン数であり、Ｓ_２は第２のコイルＬ２の内側面積である。さらに、Ｘは第１のコイルＬ１と第２のコイルＬ２との距離である。 Next, the magnitude of the radio wave (radiated electric field strength) depends on the magnitude of the magnetic flux of the first and second coils L1, L2. Here, the magnitude B of the magnetic flux can be expressed by the following formula (1) in the disk type coil, and can be expressed by the following formula (2) in the solenoid type coil.

N ₁ is the number of turns of the first coil L1, and S ₁ is the inner area of the first coil L1. N ₂ is the number of turns of the second coil L2, and S ₂ is the inner area of the second coil L2. Furthermore, X is the distance between the first coil L1 and the second coil L2.

なお、Ｖ_２は負荷２４の電圧であり、Ｌ_２は第２のコイルＬ２のインダクタンスであり、ωはインバータ１４の駆動周波数である。また、Ｃ_２は第２コンデンサＣ２の容量であり、Ｍは結合係数に依存して変化する数である。 When the areas S ₁ and S ₂ are fixed, the magnetic flux B flows through the number of turns N ₁ and N ₂ of the first and second coils L 1 and L 2 and the first and second coils L 1 and L 2. It depends on the currents I ₁ and I ₂ . Here, the combined ampere turn AT of the first and second coils L1 and L2 determined from the numbers of turns N ₁ and N ₂ and the currents I ₁ and I ₂ is expressed by the following formula (3) in the disk type coil. In the solenoid type coil, it can be expressed by the following formula (4).

V ₂ is the voltage of the load 24, L ₂ is the inductance of the second coil L ₂ , and ω is the drive frequency of the inverter 14. Also, C ₂ is the capacitance of the second capacitor C2, M is a number that varies depending on the coupling coefficient.

FIG. 4 is a diagram showing the correlation between the drive frequency of the inverter 14 and the combined ampere turn AT in the disk type coil when the load power is constant. As is clear from the equations (1) and (3), the smaller the synthetic ampere turn AT, the smaller the magnetic flux B, and the radiated electric field intensity can be suppressed. As shown in FIG. 4, the present inventors have found that in the case of a disk-type coil, the combined ampere turn AT becomes small at a frequency higher than the resonance frequency ω <b> 1 of the power receiving side circuit 20, which is a known value. Thus, even if the load power is constant, the optimum combined ampere turn AT can be obtained depending on the frequency.

  FIG. 5 is a diagram showing the correlation between the drive frequency of the inverter 14 and the combined ampere turn AT in the solenoid type coil when the load power is constant. On the other hand, as shown in FIG. 5 and formulas (2) and (4), the inventors of the present invention synthesized the signal at a frequency lower than the resonance frequency ω1 of the power receiving side circuit 20, which is a known value, in the case of a solenoid type coil. We found that the ampere turn AT is smaller. Thus, even if the load power is constant, the optimum combined ampere turn AT can be obtained depending on the frequency.

  Therefore, since the contactless power supply device 1 according to the first embodiment is a solenoid type coil in which the axial direction of the coils L1 and L2 is the vehicle plane direction, the drive frequency selection unit 15b is based on the resonance frequency ω1 of the power receiving side circuit 20. A lower frequency is selected, and the inverter 14 is driven at this frequency. FIG. 6 is a diagram showing how the drive frequency is selected by the drive frequency selection unit 15b shown in FIG. As shown in FIG. 6, there are four frequencies ω2 to ω5 that satisfy the command value Pref. When the current frequency of the current power Pout is fref, the drive frequency selection unit 15b sequentially increases the frequency, and the frequency ω2, ω3 at which the current power Pout and the command value Pref shown in FIG. Select either one. Note that the frequencies ω4 and ω5 are not selected because the frequencies are higher than the resonance frequency ω1.

  Then, the pulse generation unit 15c generates a pulse corresponding to the frequency selected by the drive frequency selection unit 15b as described above, and drives the inverter 14.

Here, the number of turns N ₁ and N ₂ of the first and second coils L1 and L2 is also set in advance, so that the formula (4) is simplified as follows. The combined current A when the number of turns N ₁ and N _{2 of} the first and second coils L1 and L2 is the same can be expressed by the following equation (5).

As described above, when the number of turns N ₁ and N _{2 of} the first and second coils L1 and L2 is the same, the combined current A can be reduced as is apparent from the above formulas (2) and (5). The more the magnetic flux B becomes, the more the radiation electric field intensity can be suppressed. Further, the combined current A is the same as the combined ampere turn AT shown in FIG. 5, and the combined current A becomes small at a frequency lower than the resonance frequency ω1 of the power receiving side circuit 20, which is a known value. Therefore, when the number of turns N ₁ and N ₂ are the same, the drive frequency selection unit 15b according to the first embodiment selects a frequency lower than the resonance frequency ω1 of the power receiving side circuit 20 based on the combined current A. The inverter 14 may be driven at this frequency.

In the above and the following description, since the combined current A is obtained by removing the concept of the number of turns N ₁ and N _{2 from} the combined ampere turn AT, it is assumed that it is a kind of combined ampere turn AT, and the combined ampere turn AT It shall be included in the concept of

  Furthermore, as shown in FIG. 5, the inventors of the present invention, when the frequency is too low in the frequency range lower than the resonance frequency ω1 of the power receiving side circuit 20, causes the combined ampere turn AT to occur at the resonance frequency ω1 of the power receiving side circuit 20. It was found to be larger than the synthetic ampere turn AT1. It has also been found that the frequency that is higher than the combined ampere turn AT1 at the resonance frequency ω1 of the power receiving side circuit 20 varies depending on the coupling coefficient. For this reason, by selecting a frequency smaller than the combined ampere turn AT1 at the resonance frequency ω1 of the power receiving side circuit 20 based on the coupling coefficient, the radiation electric field strength can be more reliably reduced.

  Specifically, as shown in FIG. 5, when the coupling coefficient k is 0.3, the combined ampere turn AT tends to decrease when it becomes lower than the resonance frequency ω1, and becomes a minimum value at the frequency ω6. Further, when the frequency is lowered, the combined ampere turn AT becomes the same value as that at the resonance frequency ω1 at the frequency ω7. Thereafter, the synthetic ampere turn AT further increases as the frequency decreases.

  Therefore, in the first embodiment, the drive frequency selection unit 15b sets ω7 <drive frequency of the inverter 14 <ω1 when the engagement coefficient k is 0.3. Thereby, the non-contact electric power feeder 1 which concerns on 1st Embodiment is trying to reduce a radiation electric field intensity | strength more reliably.

  Further, in the first embodiment, the drive frequency selection unit 15b selects a frequency in which the phase of the output when the inverter 14 is driven is delayed from the frequencies within the above range. Thereby, switching loss in the inverter 14 is reduced. Details will be described below.

  In general, when an inverter is driven at a frequency higher than the resonance frequency of the resonance circuit, the output phase when the inverter is driven is delayed. On the other hand, when the inverter is driven at a frequency lower than the resonance frequency of the resonance circuit, the phase of the output when the inverter is driven is advanced. And when an output phase becomes a phase advance, the switching loss in an inverter will become large. For this reason, the inverter is normally driven at a frequency higher than the resonance frequency of the resonance circuit.

  However, since the contactless power supply device 1 according to the present embodiment includes a solenoid type coil, the inverter 14 is driven at a frequency lower than the resonance frequency ω1 on the power receiving circuit side 20, and the output phase of the inverter 14 is delayed. The frequency range is very narrow compared to disk type coils. Therefore, the drive frequency selection unit 15b is lower than the resonance frequency ω1 of the power receiving side circuit 20, and the combined ampere turn AT is smaller than the combined ampere turn AT1 at the resonance frequency ω1 of the power receiving side circuit 20. If the frequency is selected without considering whether the phase is in the fast phase region or the slow phase region, the selected frequency may correspond to the frequency in the fast phase region, and switching loss may increase.

  Therefore, in this embodiment, the drive frequency selection unit 15b is lower than the resonance frequency ω1 of the power receiving side circuit 20, and the combined ampere turn AT is smaller than the combined ampere turn AT1 at the resonance frequency ω1 of the power receiving side circuit 20. In selecting a frequency within a certain range, a frequency at which the output phase when the inverter 14 is driven is selected is selected.

FIG. 7 is a correlation diagram showing the frequency and the output phase of the inverter 14. As shown in FIG. 7, the slow phase region is wide in the region higher than the resonance frequency ω1 of the power receiving side circuit 20, but the slow phase region is narrow in the region lower than the resonance frequency ω1 of the power receiving side circuit 20. For this reason, the power control unit 15 stores data as shown in FIG. 7 in advance. Then, the drive frequency selection unit 15b selects a frequency at which the phase of the output is delayed when the inverter 14 is driven in a region lower than the resonance frequency ω1 of the power receiving circuit 20 based on the stored data.

As is apparent from FIG. 7, the output phase curve of the inverter 14 varies depending on the coupling coefficient. Therefore, the drive frequency selection unit 15b selects a frequency at which the phase of the output when the inverter 14 is driven is delayed based on the coupling coefficient. That is, if a coupling coefficient k = X _1, the frequency range of the delayed phase is the range below ω9 higher than Omega8. Further, when the coupling coefficient k = X ₂ (> X ₁ ), the frequency range serving as a slow phase is a range higher than ω10 and lower than ω9.

  Accordingly, the drive frequency selection unit 15b firstly has a coupling coefficient lower than the resonance frequency ω1 of the power receiving side circuit 20, and the combined ampere turn AT is higher than the combined ampere turn AT1 when the power receiving side circuit 20 has the resonance frequency ω1. Find the frequency range to be reduced. Next, the drive frequency selection unit 15b determines a frequency range that is delayed as described above from the coupling coefficient. Then, the drive frequency selection unit 15b selects frequencies belonging to these two ranges.

  Furthermore, the power control unit 15 increases the phase difference between the current of the first coil L1 and the current of the second coil L2 to be greater than 90 degrees in order to further suppress the radiation field intensity. More specifically, the power control unit 15 sets the phase difference between the current of the first coil L1 and the current of the second coil L2 to be larger than 90 degrees and smaller than 270 degrees.

FIGS. 8A and 8B are waveform diagrams showing magnetic fluxes generated by currents flowing through the coils L1 and L2, in which FIG. 8A shows a first example, FIG. 8B shows a second example, and FIG. 8C shows a third example. Is shown. As shown in FIG. 8A, when the current phase is close to the same phase, the magnetic field B1 generated by the first coil L1 and the magnetic field B2 generated by the second coil L2 are strengthened to increase the total magnetic flux φ _Total. End up. On the other hand, by making the phase difference larger than 90 degrees and smaller than 270 degrees, both magnetic fluxes B1 and B2 can be made to cancel each other as shown in FIG. The magnetic field intensity can be suppressed by reducing the magnetic flux φ _Total . The phase difference is
It can be set by adjusting the impedance or frequency on the power receiving side circuit 20 side.

  As shown in FIG. 8C, when the difference between the amplitudes is large, the effect of canceling the magnetic fluxes B1 and B2 with each other is small even with the above phase difference. For this reason, in the first embodiment, the power control unit 15 preferably adjusts the mutual amplitude. In this respect, the combined ampere turn AT is an evaluation index composed of the mutual amplitude difference and phase difference of the currents flowing through the coils L1 and L2, and if the frequency for reducing the combined ampere turn AT is selected, the amplitude and phase are individually determined. Even without adjusting, the amplitude and phase are optimized as a whole.

  Next, the control method of the non-contact electric power feeder 1 which concerns on 1st Embodiment is demonstrated. FIG. 9 is a flowchart illustrating a method for controlling the non-contact power feeding device 1 according to the first embodiment. As shown in FIG. 9, first, the power control unit 15 inputs the current power Pout detected by the load power detection unit 25a and the command value Pref calculated by the load control unit 25b (S1).

  Next, the power control unit 15 inputs the coupling coefficient k detected by the coupling coefficient detection unit 25c (S2). Thereafter, the duty generation unit 15a of the power control unit 15 determines the duty ratio from the current power Pout detected by the load power detection unit 25a and the command value Pref calculated by the load control unit 25b (S3).

Next, the drive frequency selection unit 15b of the power control unit 15 selects the drive frequency of the inverter 14 (S4). At this time, as shown in the above theory, the drive frequency selection unit 15b is lower than the resonance frequency ω1 of the power receiving side circuit 20 and the combined ampere turn AT is the combined ampere at the resonance frequency ω1 of the power receiving side circuit 20. A frequency is selected that is smaller than the turn AT1 and whose output phase is delayed when the inverter 14 is driven. More specifically, the drive frequency selection unit 15b determines that the coupling coefficient k input in step S2 is lower than the resonance frequency ω1 of the power receiving side circuit 20, and the combined ampere turn AT is the resonance frequency of the power receiving side circuit 20. A frequency range that is smaller than the combined ampere turn AT1 at ω1 is determined. For example, as shown in FIG. 5, when the coupling coefficient k is 0.3, the lower limit value is ω7 and the upper limit value is ω1. That is, the drive frequency selection unit 15b determines the frequency range as ω7 <drive frequency of the inverter 14 <ω1. Next, the drive frequency selection unit 15b determines a frequency range in which the phase of the output when the inverter 14 is driven is delayed from the above frequency range. For example, as shown in FIG. 7, when the coupling coefficient k is X ₁ (only in this paragraph, X ₁ = 0.3), the lower limit value is ω8 and the upper limit value is ω9. The drive frequency selection unit 15b sequentially increases the frequency from the higher one of the lower limit values ω7 and ω8, and Pref = Pout in a range where the frequency does not exceed the lower one of the resonance frequencies ω1 and ω9. Will be selected.

Then, the pulse generation unit 15c of the power control unit 15 generates a pulse from the frequency selected by the duty ratio and S4 determined in step S3, switching the switching element S ₁ to S ₄ by the pulse Thus, the inverter 14 is driven (S5). Thereby, as explained with reference to the above formulas (2), (4) and (5), the radiation electric field intensity can be suppressed.

  Thereafter, the contactless power supply device 1 determines whether or not charging is complete (S6). At this time, the contactless power supply device 1 determines that charging is complete when the remaining capacity of the battery as the load 24 exceeds a predetermined value.

  If it is determined that charging has not been completed (S6: NO), the process proceeds to step S1. On the other hand, when it is determined that the charging is completed (S6: YES), the process shown in FIG. 8 ends.

In this way, according to the contactless power supply device 1 and the control method thereof according to the first embodiment, the combined ampere turn AT can be reduced by selecting the frequency, and the radiation electric field strength can be reduced without using structural members as much as possible. Can do. Present inventors in detail, depending on the current magnitude of the wave flows through the first and the number of turns N ₁ of the second coil L1, _L2, N ₂ and their, the number of turns N _1, of N ₂ and the current It has been found that the combined ampere turn AT of the first and second coils L1, L2 determined from the size becomes smaller at a frequency lower than the resonance frequency ω1 of the power receiving side circuit 20 in the case of a solenoid type coil. Further, in the case of a solenoid coil, the inventors of the present invention combine the combined ampere turn AT at the resonance frequency ω1 of the power receiving side circuit 20 if the frequency is too low in the frequency range lower than the resonance frequency ω1 of the power receiving side circuit 20. I found it larger than the ampere-turn AT. For this reason, the combined ampere turn AT at the resonance frequency ω1 of the power receiving side circuit 20 is a frequency lower than the resonance frequency ω1 of the power receiving side circuit 20 and the combined ampere turn AT of the first coil L1 and the second coil L2. By selecting a frequency smaller than AT, the radiation electric field strength can be reduced. Further, the reduction of the radiation electric field intensity can be realized by selecting the frequency without using a structural member for reducing the magnetic field as much as possible. Therefore, enlargement can be prevented while suppressing the radiation electric field intensity.

Furthermore, since the frequency in which the phase of the output when the inverter 14 is driven is selected from the frequencies within the above range, the switching loss in the inverter 14 can also be reduced. That is, at frequencies within the above range, the frequency range in which the output phase when driving the inverter 14 is delayed is narrow, and it is selected if it is not considered whether it is a fast phase region or a slow phase region. There is a possibility that the frequency corresponds to the frequency in the phase advance region and the switching loss becomes large. However, in the first embodiment, since the frequency range that becomes the slow phase region is determined and selected, the problem that the switching loss tends to increase in the case of the solenoid type coil can be solved.

  Moreover, since the phase difference between the current of the first coil L1 and the current of the second coil L2 is greater than 90 degrees, the generated magnetic fluxes cancel each other, and the radiation field strength can be further reduced.

  Next, a second embodiment of the present invention will be described. The non-contact power feeding device and the control method thereof according to the second embodiment are the same as those of the first embodiment, but the configuration and processing contents are partially different. Hereinafter, differences from the first embodiment will be described.

  FIG. 10 is a configuration diagram illustrating the contactless power supply device 2 according to the second embodiment. As shown in FIG. 10, in the non-contact power feeding device 2 according to the second embodiment, the drive frequency selection unit 15b selects a frequency based on the coupling coefficient k from the coupling coefficient detection unit 25c. Specifically, the drive frequency selection unit 15b stores a map indicating the correlation between the coupling coefficient k and the frequency as shown in FIG. 9, and when the coupling coefficient k from the coupling coefficient detection unit 25c is input, the map is displayed. Refer to to determine the frequency.

  Here, the frequency selected by the drive frequency selection unit 15b is a frequency that minimizes the combined ampere turn AT shown in FIG. Therefore, the radiation electric field intensity can be suppressed to the maximum. Here, as shown in FIG. 4, the selected frequency is lower than the resonance frequency ω1 of the power receiving side circuit 20, and the combined ampere turn AT is higher than the combined ampere turn AT1 at the resonance frequency ω1 of the power receiving side circuit 20. This is a frequency range where the frequency becomes smaller, and a frequency range where the phase of the output when the inverter 14 is driven is delayed, and within this range, the synthesized ampere turn AT is a minimum frequency. It is more desirable. This is because switching loss can be reduced. In the following description, it is assumed that the frequency selected by the drive frequency selection unit 15b is a frequency at which the combined ampere turn AT shown in FIG. 5 is minimum, but the phase of the output when the inverter 14 is driven is Needless to say, it is more desirable to consider the range of frequencies that are delayed.

式（６）によっても合成アンペアターンＡＴが最小となる周波数を決定することができるからである。また、ターン数Ｎ_１，Ｎ_２が同じである場合には、式（７）により合成電流Ａが最小となる周波数を決定することができるからである。 Further, the drive frequency selection unit 15b may store the following formula (6) or formula (7) instead of the map.

This is because the frequency that minimizes the combined ampere-turn AT can also be determined by equation (6). Further, when the turn numbers N ₁ and N ₂ are the same, the frequency at which the combined current A is minimized can be determined by the equation (7).

Here, in the second embodiment, since the frequency is uniquely determined by the coupling coefficient k, there is a possibility that the control of Pout = Pref may be hindered. Therefore, in the second embodiment, Pout = Pref is realized by controlling the duty ratio.

  FIGS. 11A and 11B are diagrams for explaining an adjustment method using a duty ratio according to the second embodiment, in which FIG. 11A shows before adjustment, and FIG. 11B shows after adjustment. As shown in FIG. 11A, when the frequency at which the combined ampere turn AT is minimum is ω6, Pout> Pref. Therefore, the duty generation unit 15a decreases the duty ratio. As a result, as shown in FIG. 11B, the output power is shifted downward in the figure, and Pout = Pref is realized at the frequency ω6.

  In FIG. 11, the case where the duty ratio is reduced has been described as an example. However, when Pout <Pref is satisfied at the frequency at which the combined ampere turn AT is minimum, the duty ratio is increased.

Furthermore, in the second embodiment, Pout = Pref is realized at a frequency at which the combined ampere turn AT is minimized by controlling the duty ratio. However, the present invention is not limited to this, and the duty ratio of the switching element S ₀ is changed. The boost ratio of the power factor correction booster 13 may be changed by calculation. This is also because Pout = Pref can be realized at a frequency at which the combined ampere turn AT is minimized, similarly to the control of the duty ratio.

  Next, a method for controlling the non-contact power feeding device 2 according to the second embodiment will be described. FIG. 12 is a flowchart illustrating a control method of the non-contact power feeding device 1 according to the second embodiment. In addition, the same code | symbol is attached | subjected to the process same as the process shown in FIG. 9, and description is abbreviate | omitted.

  After inputting the coupling coefficient k (after S2), the drive frequency selection unit 15b selects a frequency that minimizes the combined ampere turn AT from the map or Expression (6) and Expression (7) (S7). Next, the duty control unit 15a selects the duty ratio (or step-up ratio) so that Pout = Pref at the frequency selected in step S7 (S8).

  Thereafter, processing similar to steps S5 and S6 shown in FIG. 9 is executed, and the processing shown in FIG. 12 ends.

  In this way, according to the non-contact power feeding device 2 and the control method thereof according to the second embodiment, as in the first embodiment, an increase in size can be prevented while suppressing the radiated electric field strength, and the magnetic flux can be reduced. By canceling out, the radiation electric field strength can be further reduced.

  In addition, since the frequency that minimizes the combined ampere turn AT is selected, the radiation electric field strength can be further reduced. Further, when the current power Pout is not equal to the command value Pref due to selection of the minimum frequency, the current power Pout = the command value Pref is adjusted by adjusting at least one of the duty ratio and the step-up ratio. It can be. Therefore, the radiation electric field strength can be further reduced without affecting the charging power.

  In addition, the frequency range is lower than the resonance frequency ω1 of the power receiving side circuit 20 and the combined ampere turn AT is lower than the combined ampere turn AT1 at the resonance frequency ω1 of the power receiving side circuit 20, and the inverter 14 When the frequency range where the phase of the output when driven is slow and the frequency that minimizes the combined ampere turn AT is selected within these ranges, switching is performed while further reducing the radiation field strength. Loss can be reduced.

  As mentioned above, although this invention was demonstrated based on embodiment, this invention is not limited to the said embodiment, You may add a change in the range which does not deviate from the meaning of this invention, and combines each embodiment. Also good.

  For example, the contactless power feeding devices 1 and 2 according to the above-described embodiment are not limited to the illustrated circuit configuration, and various modifications are possible, for example, the power feeding side circuit 10 includes an insulating transformer.

  Further, in the above embodiment, the contactless power feeding devices 1 and 2 include the coupling coefficient detection unit 25c, but instead of this, data of the coupling coefficient k may be stored in advance. For example, when the non-contact power feeding devices 1 and 2 are used exclusively for a specific passenger car and the distance between the power feeding side circuit 10 and the power receiving side circuit 20 is already known, the power control unit 15 sends the data of the coupling coefficient k. You may make it memorize | store and select a frequency based on this coupling coefficient k.

  Further, when the non-contact power feeding devices 1 and 2 are used for charging the vehicle battery, since the distance between the power feeding side circuit 10 and the power receiving side circuit 20 based on the vehicle height can be predicted, the data of the coupling coefficient k is stored on the vehicle side. In addition, it may be configured to transmit to the power control unit 15 during charging.

DESCRIPTION OF SYMBOLS 1, 2 ... Non-contact electric power feeder 10 ... Electric power feeding side circuit 11 ... Commercial power supply 12 ... Rectification part 13 ... Power factor improvement pressure | voltage rise part 14 ... Inverter 15 ... Electric power control part (drive means)
DESCRIPTION OF SYMBOLS 20 ... Power-receiving side circuit 21 ... Rectification part 22 ... Filter part 23 ... Relay 24 ... Load 25 ... Battery controller 25a ... Load electric power detection part (electric power detection means)
25b ... Load control unit (command value calculation means)
25c ... Coupling coefficient detector C1 ... first capacitor L1 ... first coil C2 ... second capacitor L2 ... second coil

Claims (5)

Power is transmitted or received in a non-contact manner between the power supply side circuit having the first coil and the power reception side circuit having the second coil by at least magnetic coupling, and at the time of power transmission and reception, the first coil In the non-contact power feeding device in which the magnetic pole direction is opposite to the magnetic pole direction of the second coil,
The power supply side circuit includes drive means for driving an inverter to generate high frequency power to be supplied to the first coil,
The driving means is lower than a resonance frequency of the power receiving side circuit, and a combined ampere turn of the first coil and the second coil is higher than a combined ampere turn at the resonance frequency of the power receiving side circuit. A non-contact power feeding apparatus that drives the inverter by selecting a frequency within a smaller range. The non-contact power feeding apparatus according to claim 1, wherein the driving unit selects a frequency in which a phase of an output when the inverter is driven is delayed from frequencies within the range. The power receiving side circuit includes power detection means for detecting power supplied to a load to be charged, and command value calculation means for calculating a command value of power supplied to the load,
The drive means selects a frequency at which a combined ampere turn of the first coil and the second coil is minimized, and the power detected by the power detection means is calculated by the command value calculation means. 3. The contactless power supply device according to claim 1, wherein at least one ratio of a duty ratio and a step-up ratio is determined so as to match a command value of electric power. The contactless power supply device according to any one of claims 1 to 3, wherein the phase difference between the current of the first coil and the current of the second coil is greater than 90 degrees. Power is transmitted or received in a non-contact manner between the power supply side circuit having the first coil and the power reception side circuit having the second coil by at least magnetic coupling, and at the time of power transmission and reception, the first coil In the control method of the non-contact power feeding device in which the magnetic pole direction is opposite to the magnetic pole direction of the second coil,
A driving step of driving an inverter to generate high-frequency power to be supplied to the first coil;
In the driving step, the combined ampere turn of the first coil and the second coil is lower than the resonance frequency of the power receiving side circuit than the combined ampere turn at the resonance frequency of the power receiving side circuit. A method for controlling a non-contact power feeding device, wherein the inverter is driven by selecting a frequency within a smaller range.

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