JP5221111B2 - A power transmission coil, a power transmission device, a power reception coil, a power reception device, and a power transmission device. - Google Patents

Description

The present invention relates to a power transmission device in which a power transmission unit and a power reception unit can be separated, and in particular, a power transmission coil configured by winding a foil-like conductor in a spiral shape, a power transmission device equipped with the power transmission coil, and a mutual induction effect from the power transmission coil The present invention relates to a power receiving coil that receives power by the power receiving device, a power receiving device equipped with the power receiving coil, and a power transmission device including a power transmitting device and a power receiving device .

For example, as a power transmission device that transmits power, a configuration in which the power transmission coil 1 and the power reception coil 2 are separable as shown in FIG. 124 of the present application is known. That is, the power transmission coil 1 and the power reception coil 2 are arranged to face each other. When an alternating current is passed through the power transmission coil 1 via the capacitor 13 from the power transmission unit 3 including the AC power source 3b which is a power conversion means for converting DC power from the DC power source 12 into AC power, the power receiving coil 2 is caused by mutual induction. An electromotive force is induced in the current, and a current due to the electromotive force flows through the load RL of the power receiving unit 4 to perform power transmission. The power transmission coil 1 is configured by winding a foil-like conductor in a spiral shape. When an alternating current is passed through the power transmission coil 1, the electromotive force is induced in the power reception coil 2 due to the mutual inductive action because the power transmission coil and the power reception coil constitute a transformer. A theoretical formula for obtaining the primary impedance is described in Japanese Patent Laid-Open No. 11-274899 (Patent Document 8).

  However, a power transmission device that uses a coil using a foil conductor (hereinafter referred to as “foil conductor coil”) and can transmit a large amount of power with high efficiency has not yet been implemented. Although there are prior literatures on power transmission devices that can use a coil using a conductive wire to transmit large power with high efficiency, they have not yet been put into practice.

  From the shape of the conductor, it is considered that the skin effect of the foil conductor coil is reduced compared to the conductive wire. However, in order to reduce the DC resistance of the foil conductor coil, the conductor width must be increased in order to increase the conductor cross-sectional area, and the area of the entire coil increases. Therefore, it seems that the effect of the power transmission device that can transmit a large amount of power with high efficiency using the foil conductor coil is not yet clearly understood while having the advantage of being thin.

  Regarding a power transmission device using a foil conductor coil, Japanese Utility Model Publication No. 51-12929 (Patent Document 1) is one of the oldest patent documents in this field. Patent Document 1 describes a coil in which a foil-like conductor is spirally formed on a printed circuit board as a coil to be provided on the power receiving side. Hereinafter, prior arts will be listed for each element constituting a power transmission device using a foil conductor coil.

  First, in the present application, the terminology used is different depending on the cited document, so the terminology will be described first. A portion including the power transmission unit 3 and the power transmission coil 1 in FIG. 124 of the present application is expressed as a power transmission side, a power feeding side, a primary side, an input side, and the like. Indicated as primary coil. In addition, the part including the power receiving unit 4 and the power receiving coil 2 in FIG. 124 of the present application is referred to as a power receiving side, a secondary side, an output side, and the like, and the power receiving coil 2 is a power receiving coil, a power receiving coil, a secondary coil, 2 Indicated as secondary coil. Next, a resistance component existing in series in an equivalent circuit of a coil or a capacitor is generally expressed as ESR, and is called Equivalent Series Resistance in Japanese. In this application, reference is made to “Effective Series Resistance” because the frequency characteristics of ESR are referred to. Furthermore, a capacitor which is a capacitive element is usually written as a capacitor in Japanese. In the present application, the notation is unified for all capacitors, including cited references. And the means to convert the direct-current power contained in the power transmission part of an electric power transmission apparatus into alternating current is described with alternating current power. Further, a circuit in which a coil and a capacitor are connected in series is referred to as an LC series circuit, an LC series resonance circuit, or simply a series circuit. The notation “opposite” refers to the case where the power transmission coil 1 and the power reception coil 2 are arranged in a mutually inducible state. That is, both coils are inductively coupled. Therefore, the notation of facing is synonymous with the notation that both coils are inductively coupled.

  There are many patent documents regarding coils configured by winding a foil-like conductor in a spiral shape and conventional techniques for improving the performance of the foil conductor coil.

  For example, an example of a coil using a foil conductor is described in Japanese Patent Laid-Open No. 7-37728 (Patent Document 2). In the coil described in Patent Document 2, the width of the foil-like conductor wound spirally on the insulator is wider at the outer peripheral portion than the inner peripheral portion, and the resistance value per turn and the copper loss are the same. Thus, the heat generation of the entire coil surface is intended to be uniform.

  On the other hand, as a basic characteristic of the coil, it is known that the effective series resistance of the coil increases as the frequency increases due to skin effect, eddy current loss, and the like. As a countermeasure, the same applicant as Patent Document 2 has filed Japanese Patent Application Laid-Open No. 5-243060 (Patent Document 3). In Patent Document 3, a main winding having a low DC resistance and a sub winding having a low AC resistance are wound together, and the main winding and the sub winding are connected so as not to lose the inductance, thereby increasing the frequency. Describes a method for avoiding an increase in the effective series resistance of the coil. Patent Document 3 describes a relationship between the effective series resistance of the coil and the frequency.

  In addition, in the power transmission device configured as shown in FIG. 124 of the present application, the capacitor 13 connected in series to the power transmission coil 1 is equipped for the purpose of improving the power factor. With respect to the invention of a single capacitor having good characteristics, although many patents have been filed, there are hardly any references referring to the configuration and characteristics of a capacitor suitable for improving the power factor of the power transmission device described above. Japanese Patent Application Laid-Open No. 2002-353050 (Patent Document 4) describes the use of a film capacitor having a low dielectric loss tangent.

  Patent Document 4 describes the use of a diamagnetic metal as a method for preventing performance deterioration when a metal body comes close to an air-core coil in which a foil-like conductor is wound in a flat spiral shape.

  However, in order to improve the power factor using a capacitor, it is important to define the circuit configuration and characteristics of the AC power supply. However, the circuit configuration and characteristics of the AC power supply suitable for use in the power transmission device are mentioned. There is almost no literature.

  Regarding the AC power supply used in the power transmission device, Japanese Patent Application Laid-Open No. 2002-49428 (Patent Document 5) describes an outline of the impedance conversion unit. In addition, Japanese Patent Application Laid-Open No. 11-224822 (Patent Document 6) describes a method in which an AC power source using a DC-AC converter circuit generates a waveform close to a stepped sine wave. It is.

The AC power supply and the capacitor are referred to only in the above-mentioned patent documents, but there is almost no prior document characterized by the foil conductor coil in the power transmission device using the foil conductor coil. The above-mentioned Patent Document 1 also does not mention the specific configuration and characteristics of the foil conductor coil. As prior art closer to the present, especially table 20 05-525705 (Patent Document 7), a power transmission apparatus using a foil conductor coil as "power transmission coil" is disclosed.

However, although it is the gist of the present invention, the power that can be transmitted and the power transmission efficiency are low, the power receiving device equipped with the coil having a configuration that can be inductively coupled with the foil conductor coil from the power transmission device of the power transmission device using the foil conductor coil. There is no prior document that can perform long-distance power transmission in the section. Although there are prior arts that perform long-distance power transmission to a coil built in an IC card, all use electromagnetic waves of several tens of MHz or more (for example, JP-A-10-32248). However, these are not capable of transmitting power at a distance of 50 cm or more at a frequency of 1 MHz or less as in the present invention. Therefore, the description of the prior art documents that describe the above-described effects is omitted.
Japanese Utility Model Publication No. 51-12929 (claims, FIG. 3) Japanese Unexamined Patent Publication No. 7-37728 (paragraph numbers 0012 to 0014, FIG. 1) JP-A-5-243060 (paragraph numbers 0010 to 0011, 0014, claims 1, 2 and 9) JP 2002-353050 A (paragraph numbers 0016, 0017, 0019 to 0022, FIG. 4) JP 2002-49428 A (Claim 1, paragraph numbers 0006, 0025, FIG. 1) JP-A-11-224822 (Claim 1, paragraph numbers 0002 to 0004,0025) JP 2005-525705 A (Claim 1, paragraph numbers 0018, 0043, 0063, 0120, FIG. 2a, FIG. 6a, FIG. 6c) JP-A-11-27489 (paragraph numbers, 0018 to 0021, 4 to 6 formulas)

  Hereinafter, the problems of the conventional technology will be clarified for each component that improves the conventional technology described above.

  As a power transmission device using a foil conductor coil, there has been a possibility of practical application from the technical level at that time, such as Patent Document 1. At the time of 1974 (Showa 49) when Patent Document 1 was filed, vacuum tubes were still used for communication equipment, telephone exchanges, etc., and there were no high-speed semiconductor elements. An embodiment of a power transmission device driven at a frequency of is described.

  However, the power transmission coil and the power reception coil, which are power transmission means, constitute a transformer. Although the primary and secondary coils are inseparable, it is clear that transformers designed for commercial power supplies of 50-60 Hz cannot be used at any frequency, for example 5 Hz or 10 kHz. That is, the transformer which is a power transmission means has a lower limit and an upper limit of usable frequencies. The notation of several tens of kHz does not specify a frequency range that can be used as a power transmission coil. As described above, there is no conventional technique that considers the frequency range that can be used as the power transmission coil, not limited to Patent Document 1.

  Next, the coil described in Patent Document 2 is a coil in which a foil-like conductor is wound in a flat spiral shape, and the resistance value per turn is increased by making the conductor width of the outer peripheral part wider than that of the inner peripheral part. Are intended to be the same, and the copper loss per turn is the same.

  Patent Document 2 describes an embodiment in which a foil conductor coil is equipped with a magnetic material plate. However, Claim 1 of Patent Document 2 includes a case where the magnetic material plate is not equipped. However, the claims are later amended and limited only to the case where a magnetic material plate is provided (Japanese Patent No. 3359099). When considering the case of an air core with reference to Patent Document 2, when the copper foil width of the outer peripheral portion is widened, the conductor volume of one turn is larger in the outer peripheral portion than in the inner peripheral portion, and the conductor The surface area also increases. As a result, it is obvious that the heat dissipation state of the outer peripheral portion is improved. If the copper loss (Joule loss) for one turn is constant, the temperature rise of the outer peripheral portion is lower than that of the inner peripheral portion. Therefore, at least in the case of an air core, the effect as described in Patent Document 2 cannot be expected.

  And from paragraph number 0015 of patent document 2, the copper loss described in patent document 2 is based on DC resistance calculated from the electrical conductivity of metallic copper. Further, paragraph number 0018 of Patent Document 2 specifies “static magnetic field calculation program” and “DC inductance”. Therefore, the inductance and effective series resistance described in Patent Document 2 are not actually measured values based on AC. Even with the calculated value, the decrease rate of the resistance value is 12% (decrease to 88%). The inductance is reduced to 98% and it is described that there is almost no change.

  However, referring to FIG. 9 of Patent Document 3, although the coil configuration is different, the effective series resistance at 200 kHz is twice the DC resistance even in a single-layer coil. In addition, although the coil configuration is also different, referring to FIG. 2 of Patent Document 3, even the frequency characteristics of the effective series resistance of the coil improved as an example, at 100 kHz compared to about 4 mΩ at 100 Hz. About 15 mΩ, which is about 4 times. Thus, even if the improvement rate of 12% described in paragraph No. 0019 of Patent Document 2 is a reduction rate of effective series resistance in a high frequency region, it is described in the example of Patent Document 2. In the coil, there is almost no effect of reducing the power loss due to the effective series resistance in the high frequency region.

  In addition, the specific configuration of the coil described in Patent Document 2 does not always provide the same effect when the outer diameter, the number of turns, and the like of the coil change. In addition, it is clear from FIG. 9 of Patent Document 3 that the relationship between the effective series resistance of the coil and the frequency changes depending on the configuration of the coil (for example, the number of winding layers). Furthermore, the effective series resistance and the rate of change of the inductance are merely described as ratios, and the actual values are unknown. Regardless of supplementing the actual measured value of the effect with the calculated value, if the effect is not shown as the actual measured value, it cannot be said that the effect has been proved.

  Moreover, the applicants of Patent Document 2 and Patent Document 3 are the same. A characteristic actually required as a foil conductor coil for power transmission is a frequency characteristic of an effective series resistance described in Patent Document 3. Even if the DC resistance is reduced, a foil conductor coil suitable for power transmission cannot be realized. Further, paragraph numbers 0029 to 0030 of Patent Document 2 describe “skin depth”, “eddy current”, and factors that increase the effective series resistance of the foil conductor coil. Nevertheless, Patent Document 2 does not describe anything about an increase in effective series resistance due to an increase in frequency.

  Similar to Patent Document 2, in a coil configured by winding a foil conductor, there are many prior documents in which the width of the foil conductor is different between the inner peripheral portion and the outer peripheral portion. However, in these prior documents, there is almost no description regarding inductance, which is an important characteristic of the coil. When the actual measurement value is not shown as in Patent Document 2, it cannot be determined that the Q of the coil is not lowered. It cannot be said that a coil with good performance could be realized with a configuration rule that lowers the Q of the coil.

  Unlike a coil composed of a conductive wire, in a foil conductor coil, the circumference of the conductor is longer than the conductor cross-sectional area, so that the influence of the skin effect can be reduced. However, since the foil thickness is thin, there is a problem that the direct current resistance increases. For this reason, the foil width is widened to prevent an increase in DC resistance. However, when the foil width is widened, the number of windings is reduced and the inductance is reduced. For this reason, the distance between the foil conductors is made as narrow as possible, and the number of turns is increased to ensure the inductance. However, when the distance between the foil conductors is narrowed, almost all the magnetic flux generated by the alternating current flowing through the foil conductors passes through the adjacent foil conductors. This increases the effective series resistance of the coil due to eddy current loss. Eddy current loss is proportional to the conductor volume. Therefore, although the foil thickness is thin, the influence of eddy current loss increases because the foil width is wide. There is no prior art that clarifies these issues and clearly indicates the solution means and effects.

  As described above, the effect of the foil conductor coil is not clarified, and this is a first problem in which a power transmission device with good power transmission performance cannot be realized.

  Furthermore, in patent document 2, the magnetic material is equipped on both surfaces of the foil conductor coil. However, the coil having such a configuration cannot be used for power transmission because the magnetic flux is shielded by the magnetic material. As a result of further trial by the inventor of the present application, when the magnetic material is provided on both surfaces of the foil conductor coil, it has been confirmed that although the inductance increases, the effective series resistance also significantly increases. As will be described later, what is important in the present invention is to avoid power loss due to effective series resistance. The same applies to a transformer as a power transmission means, and the transformer described in Patent Document 2 cannot be said to have an appropriate configuration. Further, paragraph numbers 0029 to 0030 of Patent Document 2 describe that the magnetic material has a laminated structure. However, as long as the inventors of the present application have verified, the function and effect of the magnetic material of the present invention having a laminated structure is different from the function and effect described in Patent Document 2. Details thereof will be described later.

  In Patent Document 4, paragraph numbers 0019 to 0021 describe that the inductance and effective series resistance of the coil fluctuate when a metal body is close to the air-core coil. In the power transmission device, when the inductance of the coil and the effective series resistance fluctuate, power cannot be sent stably. In particular, when a power factor improving capacitor is equipped, the frequency at which the reactance of the power transmission coil is zero or the frequency at which the impedance is minimized as viewed from the AC power source fluctuates.

  Paragraph Nos. 0019 to 0021 of Patent Document 4 include copper (Cu), zinc (Zn), lead (Pb), bismuth in order to prevent fluctuations in the characteristics of the air core coil when a metal body comes close to the air core coil. Although it is described that the diamagnetic metal of (Bi) is provided between the coils, it is assumed that the description that the diamagnetic metal is provided between the coils is an error that the diamagnetic metal is provided on the opposite side of the opposing surface of the coil. Moreover, although bismuth having an atomic number of 82, which is a heavy metal and not generally used, is described, it is generally known than lanthanum (La), which is a diamagnetic metal having an atomic number of 57 smaller than 82, or bismuth. No noble metal gold (Au), which is a diamagnetic metal having an atomic number of 79, is described.

  Moreover, Patent Document 4 does not describe any performance comparison by installing a diamagnetic metal and another metal (paramagnetic metal, ferromagnetic metal), and when the metal body is close to the air-core coil. There is no description of how the influence of the difference between the diamagnetic metal and the other metal is different, or the effect of avoiding the influence when the metal body actually approaches the air-core coil. According to the inventor's verification, in order to eliminate the decrease in inductance and the increase in effective series resistance, which are the effects when a metal body comes close to the air-core coil, the necessary factor as the metal plate equipped in the coil is the metal plate It has been confirmed by experiments that this is not due to the magnetic properties but merely due to the thickness of the metal plate, and details thereof will be described later with reference to experimental data.

  125 is an equivalent circuit diagram of the power transmission coil of the power transmission device configured as shown in FIG. 124, where Le is the reactance of the power transmission coil, Re is the effective series resistance of the power transmission coil, C1 is the capacitor for power factor improvement, Rc Represents the effective series resistance of the power factor correction capacitor.

  In paragraph No. 0017 of Patent Document 4, the dielectric loss tangent tan δ is small as a capacitor for canceling the residual reactance Le of the power transmission coil 1 and improving the power factor (described as being for resonance in Patent Document 4). It is described that a capacitor and a capacitor having a dielectric material such as polystyrene or polypropylene are used. However, a capacitor using polystyrene or polypropylene as a dielectric does not always have a smaller dielectric loss tangent tan δ at any frequency than a capacitor using other dielectrics.

  Furthermore, when the inventors of the present application made additional trials, some polystyrene capacitors could not be used due to heat generation. In addition, the polypropylene capacitor has different power transmission performance depending on the configuration, and a phenomenon in which the power transmission performance decreases depending on the capacitance and frequency is observed.

  FIG. 126 is a diagram illustrating an equivalent circuit of a capacitor. In FIG. 126, Xc represents the reactance of the capacitor, Rc represents the effective series resistance of the capacitor, and Lc represents the parasitic inductance of the capacitor.

  The dielectric loss tangent tan δ is defined as tan δ = 1 / Q (no unit) using the Q of the capacitor. The Q of the capacitor is represented by Q = Xc / Rc, where Xc is the reactance of the capacitor and Rc is the effective series resistance of the capacitor at the frequency at which the Q of the capacitor is measured. According to JIS regulations, the dielectric loss tangent tan δ of a capacitor is measured at 1 kHz or 1 MHz by capacitance. When the capacitance is C (F), the reactance Xc (Ω) of the capacitor is Xc = 1 / ωC (Ω), and the reactance Xc is an angular frequency ω (ω = 2πf, f is a frequency (Hz). ) And the capacitance C.

Using the symbols in FIG. 126, the dielectric loss tangent tan δ is tan δ = ωC · Rc = Rc · Xc. The capacitance C of the capacitor is usually 10 ⁻⁶ F or less. Therefore, at a low frequency such as 1 kHz, the value of ωC is a very small value of 10 ⁻⁶ or less. On the other hand, the effective series resistance Rc of the capacitor differs depending on the type of capacitor and the capacitance, but is usually 10Ω or less at 1 kHz. Therefore, normally, the value of the dielectric loss tangent tan δ of the capacitor is 0.001 or less, which is a JIS prescribed value described later.

  In general, a capacitor having a low effective series resistance Rc (Ω) in a high frequency region is a ceramic capacitor. Therefore, the ceramic capacitor has a high Q and a low dielectric loss tangent tan δ. The inventors of the present invention actually measured the Q of a multilayer ceramic capacitor having a capacitance of 0.1 μF at 1 MHz, where Q≈18, tan δ = 1 / 18≈0.055, and the tangent angle δ is δ≈ It was 3 degrees.

  Further, when the Q of a polypropylene capacitor having a capacitance of 0.1 μF, which is different from a polypropylene capacitor having a good power transmission performance described later, was measured at 1 MHz, Q≈5 and tan δ = 1 / 5≈0. 2. The tangent angle δ was δ≈11.3 degrees. That is, as long as the dielectric loss tangent tan δ in the high frequency region is compared, the ceramic capacitor has better characteristics than the polypropylene capacitor. However, as far as the inventor of the present application experimentally verified, the multilayer ceramic capacitor has a lower power transmission performance than the polypropylene capacitor.

  Patent Document 4 simply describes that “dielectric loss tangent is small”. If it conforms to the JIS regulations, as described above, an optimum capacitor for the power transmission device can be selected simply by measuring the dielectric loss tangent of the film capacitor at 1 kHz and comparing the performance. However, as will be described later, the optimum capacitor for the power transmission device cannot be selected only by the dielectric or tangent of the capacitor. That is, it is necessary to specify a specific value of dielectric loss tangent considering frequency characteristics, a specific capacitance, a relationship between both, and a characteristic factor other than the dielectric loss tangent.

  Therefore, the dielectric loss tangent tan δ of the capacitor used for improving the power factor of the power transmission device needs to be determined in consideration of the dielectric (capacitor type), capacitance, frequency characteristics, and the like. . The frequency characteristics of the dielectric loss tangent tan δ will be described later.

  Further, paragraph number 0018 and claim 3 of Patent Document 4 describe a method of connecting two capacitors to both ends of a coil. These are also described in FIG. 1 of Japanese Patent Application Laid-Open No. 2005-6396, FIG. 1 of Japanese Patent Application Laid-Open No. 2005-6459, FIG. 1 of Japanese Patent Application Laid-Open No. 2005-6460, and the like. However, although this circuit configuration has the effect of increasing the withstand voltage of the capacitor, there is no prior document that mentions the correlation between this circuit configuration and power transmission performance. Detailed operational effects of this circuit configuration will also be described later.

  That is, in the conventional technology, the configuration and characteristics of the capacitor used for power factor improvement of the power transmission device, the circuit configuration and the operation effect using the power factor improvement capacitor of the power transmission device are not clarified at all, Therefore, a power transmission device with good power transmission performance cannot be realized. In other words, it is not possible to select a capacitor with good performance for use in the high-frequency power circuit and to define its use conditions. This is the second problem in this field.

  125, as described above, in the series resonance circuit, the voltage of the AC power supply is Vs (V), the resonance frequency is fr (Hz), the effective series resistance of the power transmission coil is Rw (Ω), and the effective series of the capacitors. When the resistance is Rc (Ω), a large current Ir = Vs / (Rw + Rc) (A) flows through the capacitor at the resonance point. However, Patent Document 4 completely describes the current that can be passed through the capacitor. can not see.

  Further, Patent Document 4 describes that the resonance frequency of the power receiving unit alone is set lower than the resonance frequency of the power transmission unit alone. The effect of the present invention is different from that of Patent Document 4, and the power transmission unit of the power transmission device of the present invention is a simple series resonance circuit. Therefore, when Qr of the LC series circuit composed of the power transmission coil and the power factor correction capacitor is Qr, a voltage Qr times the AC power supply voltage is applied to the capacitor. Therefore, performance specifications regarding the withstand voltage and the withstand current of the capacitor are extremely important. Details thereof will be described later.

  In addition, FIG. 2 of Patent Document 4 does not have an AC power source configuration that allows current to flow in both directions, and cannot drive a circuit in which a coil and a capacitor are connected in series. Therefore, the circuit of FIG. .

  As described above, the conventional technology can realize a circuit having an optimum characteristic for driving a series circuit in which a power factor improving capacitor having an optimum characteristic for improving power transmission performance and a power transmission coil are combined. Not yet another challenge in this area of technology.

  Next, FIG. 1 of Patent Document 5 describes a circuit including only an inductor and a capacitor as an impedance converter. However, this circuit is known as a π-L type circuit used for impedance conversion of the output stage of a wireless transmitter (for example, Linear Amplifier Handbook, Haruo Yoneda, P48, CQ Publisher, July 15, 1972). ,reference). In this example, when the AC power source is viewed from the power transmission coil, the impedance of the power transmission coil and the impedance of the AC power source are merely matched. Passive elements such as inductors and capacitors are used for impedance conversion. These passive elements function as impedance conversion filters, but it is known that loss occurs in a filter circuit using passive elements (for example, Linear Amplifier Handbook, Haruo Yoneda P43, CQ Publishing Co., Ltd.) , July 15, 1972). Therefore, when the AC voltage source is converted into an AC current source, which is the effect described in Patent Document 5, or vice versa, the occurrence of power loss is unavoidable in the LC filter unit.

  The immittance described in Patent Document 5 is a generic term for impedance, admittance, reactance, susceptance, resistance, conductance, and the like, and the immittance conversion described in Patent Document 5 is synonymous with impedance conversion.

  Patent Document 6 describes a circuit configuration example of an output portion of an AC power source that can cause a current to flow bidirectionally through the coil L. Paragraph No. 0003 of Patent Document 6 describes a circuit that can be used as a switching element of 20 KHz. It is described that the degree is a limit and it is difficult to perform PWM control. On the other hand, paragraph No. 0009, FIG. 1, and FIG. 6 of Patent Document 6 describe a system that is substantially PWM control. Moreover, when the circuit of FIG. 1 of Patent Document 6 is driven with a waveform like FIG. 6 of Patent Document 6, the zero point of the waveform shown in FIG. 6 of Patent Document 6 is high impedance, It is not zero potential.

  Further, from paragraph number 0014 of Patent Document 6 and FIG. 9, E1> E2. For example, in FIG. 10 of Patent Document 6, when Q2, Q3, Q6, and Q7 are ON, in FIG. 9 of Patent Document 6, E1 positive electrode → Q7 → D3 → E2 positive electrode → E2 negative electrode → D2 → Q6. → A short-circuit current flows from the positive electrode of E1 through E2 to the negative electrode of E1 through the path of E1 negative electrode. Therefore, the current from the DC power supplies E1 and E2 does not flow through the coil connected to the output, and the circuit of FIG. 9 in Patent Document 6 does not operate. FIGS. 10 and 12 of Patent Document 6 corresponding to FIG. 9 of Patent Document 6 are simulated by a computer on the assumption that FIG. 9 of Patent Document 6 operates even when viewed from the plotted graph. It seems that it is not actually measured value. Further, as described above, in the circuit of FIG. 9 of Patent Document 6, an output waveform as shown in FIG. 10 of Patent Document 6 cannot be obtained. Moreover, Patent Document 6 does not clearly show the characteristics of the coil drive circuit and does not mention the DC-AC conversion efficiency.

  As an example of a power transmission device using a foil conductor coil, paragraph number 0043 of Patent Document 7 describes using a coil in which a foil conductor is formed in a flat spiral shape. It is described that by devising the winding method of the foil conductor, the power required by the power receiving device can be transmitted no matter where the multiple power receiving coils smaller than the power transmitting coil are placed. (Patent Document 7, paragraph number 0018). However, referring to FIG. 6a of Patent Document 7, even if a wide foil-like conductor is formed in a portion where the gap between adjacent conductors is wide, the portion where the gap between adjacent conductors is narrow inevitably becomes the conductor width. Becomes narrower, the effective series resistance of the coil becomes higher.

  Moreover, FIG. 6c of patent document 7 is well-known as described in FIG. 2 of patent document 2, and is the same structure as FIG. 2 which patent document 7 uses as a prior art example. In paragraph Nos. 0063 and 0120 of Patent Document 7, as a function and effect, the magnetic field lines rotate, so that the receiving coil generates magnetic flux generated by the transmitting coil regardless of the angle of the direction of the receiving coil with respect to the transmitting coil. Although it can be supplemented, this requires at least two power transmission coils.

  In addition, if the coils are arranged so that the magnetic fields generated by the two coils are orthogonal to each other, and a current having a phase difference of 90 degrees is passed through each coil, the magnetic fields generated by the currents flowing through both coils will rotate. Is a known theory of electromagnetism, and is used, for example, as a drive coil for a magnetic bubble memory (see, for example, Japanese Patent Laid-Open No. 59-44713, page 2, lower right, lines 1 to 10). Therefore, in the configuration of the power transmission coil using one coil described in claim 1 of Patent Document 7, the power reception coil can capture the magnetic flux generated by the power transmission coil at any angle as described above. The effect cannot be achieved.

  As described above, the conventional technology that is not consistent with electromagnetic theory or circuit theory cannot apply the theory of mutual induction that transmits power between separable coils. The characteristics of the coil constituting the transmission device, the transformer combined with the coil, the capacitor, and the AC power supply deviate from the above theory. This divergence point and the problems caused by the divergence are not clear, and this is an issue in this field, and no solution has been made.

  In addition, various proposals have been presented for a coil configured by winding a foil-like conductor into a flat spiral, and a power transmission device using the coil, but there are various problems as described above. Therefore, it has not yet reached practical use.

  However, according to the inventor's verification, it is difficult to transmit large power with high efficiency, but unlike Patent Document 7, a power transmission coil using a foil-like conductor is wide in all directions, centering on the power transmission coil. Magnetic flux can be formed in the range. For this reason, it is possible to transmit power over a wide range at least for long-distance power transmission and centering on the power transmission coil.

  In addition, by using a coil configured by winding a foil-like conductor in a flat spiral shape as a power transmission coil, the power reception device can be placed in any position on the power transmission coil where the power reception coil smaller than the power transmission coil is placed. Necessary power can be transmitted. Alternatively, it has been confirmed that power can be transmitted regardless of the relative angle between the power transmission coil and the power reception coil. And the power transmission device with such performance can improve performance over the conventional power transmission device capable of long-distance transmission using electromagnetic waves of several tens of MHz. However, there is no prior art that mentions the above-described effects.

  The power transmission device having the above-mentioned performance is, for example, an automatic ticket gate system that can pass through a ticket gate machine with a non-contact IC card in a pocket, and other power receiving devices that do not incorporate a secondary battery such as interiors, games, and toys. There are various application uses such as long-distance power transmission to a power receiving unit incorporating a secondary battery or a small capacity secondary battery or an electric double layer capacitor.

  That is, in the conventional technology, a coil configured by winding a foil-like conductor into a flat spiral shape that is optimized in accordance with the required specification of the power transmission device cannot be realized. In addition, in the prior art, not only a power transmission coil formed of a foil-like conductor but also a power transmission device with good power transmission performance having an optimal peripheral circuit for driving the coil cannot be realized.

Therefore, first, a foil conductor coil with good power transmission performance, which is the first problem described above, must be realized and its operating conditions must be defined. Next, the second problem of defining the characteristics and configuration of the power factor improving capacitor of the power transmission device and selecting a capacitor suitable for improving the power factor of the power transmission device must be solved.
Accordingly, an object of the present invention is to provide a power transmission coil configured with a foil conductor capable of long-distance power transmission, a power transmission device equipped with a power transmission coil and an optimal capacitor, an optimal power reception coil and power reception device, a power transmission device and a power reception device, and To realize a power transmission device comprising:

The present invention is a power transmission coil used in a power transmission unit of a power transmission device in which a power transmission unit and a power reception unit are configured to be separable and transmit power from the power transmission unit to the power reception unit by mutual inductive action. This is an air-core coil that is formed into an air-core shape by winding a foil-like conductor in a spiral shape on an insulating material. The effective series resistance of the power transmission coil alone is Rw (Ω), and the change in the effective series resistance of the power transmission coil In order to detect this, the transformer coil is configured with the same coil as the power transmission coil facing the power transmission coil, and the effective series resistance of the power transmission coil when both ends of the same coil opposed to the power transmission coil are short-circuited. If Rs (Ω), the power transmission coil uses a predetermined foil conductor so that the maximum frequency f1 (Hz) satisfying the relationship Rs ≧ Rw is at least 200 kHz, Specified by the winding method The outer diameter and the inner diameter are configured.

A power transmission coil with good power transmission performance can be realized by using two coils formed by winding a foil-like conductor on an insulating material and selecting the coils by the measurement method as described above.

Specifically, when the thickness of the foil-shaped conductor is t and the width is H, at least the width H is 10 times the thickness t and the width H is 1 mm or more. The foil conductor is wound in a single layer spiral so that the insulating material and the width H of the foil conductor are in contact with each other, and at least the maximum outer dimension of the power transmission coil is 10 times the width H or more. The minimum outer shape is 10 cm or more, and the number of windings of the foil-shaped conductor is 4 or more.

Specifically, the thickness t of the foil conductor is 0.3 mm or less and 10 μm or more, and a gap of 5 × t or more is provided between the conductors of adjacent foil conductors.

By configuring the power transmission coil as described above, a power transmission coil with good power transmission performance can be realized. Specifically, the influence of the skin effect can be reduced by defining the thickness and width of the foil conductor, and the direct current resistance can be reduced when the thickness t of the foil conductor has a predetermined thickness.

More specifically, the width of the gap provided between adjacent foil-shaped conductors in the outermost peripheral portion of the power transmission coil is w1, and the width of the gap provided between adjacent foil-shaped conductors in the innermost peripheral portion of the power transmission coil is w2. Then, w2> w1> 0, and the width of the gap increases from w1 as it goes from the outermost periphery to the inner periphery, and the width w2 of the gap is at least 5 × t or more.

In this example, since the magnetic flux density in the center is high and the magnetic flux density in the outer periphery is low, a blank is provided in the inner periphery, the inner periphery is loosely wound, and the outer periphery is densely wound. The magnetic flux density on the surface can be made uniform.

Preferably, at least one coupling wire is wound between the foil conductors, the coupling wire is electrically insulated from the foil conductor for power transmission of the air core coil, and electromagnetically the air core It is comprised so that it may be in the state equivalent to the foil-like conductor for power transmission of a coil, and a honey-couple transformer.

By providing such a coupling line, the coupling line can be used for signal transmission, detection of the operating state of the power transmission coil, self-excited oscillation, and the like.

Preferably, the power transmission coil uses a conductive wire as a part of a conductor forming the coil, or
Whether the width H of a part of the foil-shaped conductor forming the coil is different from other parts,
Alternatively, at least a part of the foil conductor forming the coil is composed of at least two electrically separated foil conductors.

The thicker the foil-shaped conductor, the more effective the magnetic flux generated by the foil-shaped conductor spreads in the direction perpendicular to the width H. By thickening the outer peripheral portion, the magnetic flux on the coil surface is made uniform, and The power transmission distance can be increased. By using two electrically separated foil-like conductors, the DC resistance of the coil can be lowered and the upward spreading of the magnetic flux can be suppressed.

Preferably, the area of the power receiving coil is ¼ or less of the area of the power transmitting coil, and the power receiving coil has a rectangular shape having a short side and a long side, and the power transmitting coil and the power receiving coil face each other. The number of foil conductors intersecting the short side wound in a direction perpendicular to the short side and the number of foil conductors intersecting the long side wound in the direction perpendicular to the long side The foil-like conductor is wound so that they are close to each other.

In this example, in the power transmission coil facing the power reception coil, the power required for the power reception coil is increased over the entire surface of the power transmission coil by increasing the number of foil conductors included in the magnetic flux supplement area of the power reception coil. Can send.

Preferably, in the power transmission coil, a part or all of the foil conductor forming the coil is wound at an angle with respect to the winding surface of the foil conductor.

In this example, by giving an angle to the winding surface of the foil conductor, the magnetic flux direction and magnetic flux intensity are changed, and the coil of the power transmission device having various performances such as long distance, non-direction, single direction, etc. realizable.

Preferably, at least one magnetic material plate is provided on the side opposite to the winding surface of the foil conductor of the insulating material.

By using a magnetic material sheet or a magnetic material plate, it is possible to reduce radio wave interference due to unnecessary radiation.

Preferably, when a metal plate is provided to shield the influence when a metal comes close to the power transmission coil of the previous invention and to prevent fluctuations in the characteristics of the power transmission coil, or when the power transmission coil is installed on a metal material In the case where at least one of the magnetic material plate and the insulating material is equipped on the opposite side of the foil-shaped conductor winding surface of the insulating material forming the air-core coil of the power transmission coil,
・ Include only one magnetic material plate,
Whether the predetermined thickness M is (10 mm) or less,
When the outer diameter of the power transmission coil is D (mm) and M is an insulating material including air, where M> D / 10,
The metal plate or the metal material is a metal plate made of a metal or an alloy having a paramagnetic or diamagnetic magnetic property with a thickness M (mm) of 0.1 mm or more,
・ Contain at least two or more magnetic plates
-One magnetic material plate and an insulating material having a predetermined thickness (10 mm) or less,
-When it is only an insulating material having a predetermined thickness (10 mm) or more,
The metal plate or metal material is a metal plate or metal material made of a metal or an alloy having an arbitrary magnetic property with a thickness M (mm) of 0.01 mm or more.

In this example, the inductance of the power transmission coil can be increased by equipping the power transmission coil with a magnetic material plate. In addition, radio wave interference due to unnecessary radiation can be reduced. By making the size of the magnetic material plate larger than the size of the power transmission coil, it is possible to further increase the inductance of the power transmission coil and reduce radio wave interference due to unnecessary radiation. Further, by using an insulating material containing air, it is possible to suppress an increase in the effective series resistance of the coil when the frequency is increased.

Furthermore, in order to prevent a change in the inductance of the power transmission coil, a lead wire taken out of the power transmission coil from the conductive wire end of the inner peripheral portion of the power transmission coil is fixed to the winding surface of the foil conductor, and the two of the two terminals of the power transmission coil are fixed. The lead wire from the inner periphery is fixedly formed on the back surface of the insulating material on which the foil conductor is wound using the foil conductor, and one of the two terminals of the power transmission coil If the metal plate is equipped on the back side of the insulating material on which the foil-like conductor is formed, the conductor wire end of the inner periphery of the power transmission coil is connected to the metal plate at the center of the coil. Let the plate be one of the two terminals of the power transmission coil.

An electric wire taken out from the center of the foil conductor coil having a large size changes the inductance as a part of the coil. The electric wire taken out from the end of the conductive wire at the center of the coil is subject to fluctuations in the inductance of the coil by fixing the positional relationship of the outgoing wire to the winding surface of the foil conductor coil by any of the above methods. Can be prevented.

Preferably, a metal frame larger than the outer dimension of the power transmission coil is included, and a part of the metal frame overlaps a part of the winding surface of the power transmission coil.

In this example, power can be transmitted not only to the upper surface of the power transmission coil but also to the power reception coil outside the metallic frame. Thus, the magnetic flux can be spread over a wide range that is greater than the area of the power transmission coil. The metal frame is preferably made of a ferromagnetic metal such as iron.

In another aspect of the present invention, there is provided a power transmission coil according to the previous invention, an AC power source comprising power conversion means for converting DC power into AC power, and a capacitor (13) for canceling the positive reactance of the power transmission coil. A power transmission device including at least a capacitor (13) connected in series between an output of an AC power source and a reference coil, and a series resonance circuit is configured by a single power transmission unit without facing another coil to the reference coil When the frequency (series resonance frequency) at which the reactance Xc of the capacitor and the reactance Xi of the power transmission coil become equal is fr (Hz) and the output frequency of the AC power supply is set to fr, When Q is Qr, the output voltage of the AC power source is boosted to Vt, the voltage across the capacitor is Vc, and Vt is boosted to Vc when the resonant current is flowing through the series resonant circuit. Up ratio H, H = Vc / Vt, and when the capacitor (13), which satisfies the H ≧ 0.9 × Qr, following condition.

Q and Qr of the series resonant circuit have a relationship of 1 / Qr = 1 / Qi + 1 / Qc, where Qi of the reference coil is Qi and Q of the capacitor is Qc. At the series resonance frequency, the reactance of the reference coil and the capacitor becomes zero, and a voltage Qr times the AC power supply voltage is generated at both ends of the reference coil and the capacitor. This is called the boosting effect of the series resonance circuit. A power transmission device with good performance can be realized by selecting a capacitor using this boosting action and installing the capacitor in the power transmission device.

In another example of the present invention, a capacitor (13) is connected in series between the power transmission coil of the previous invention, the output of the AC power source, and the reference coil, and the other coil is not opposed to the reference coil. When the series resonance circuit is configured as a single unit, the output voltage of the AC power supply is a square wave with a duty of 50%, the voltage of which changes between a zero potential and a predetermined potential, and the reactance Xc of the capacitor, If the frequency (series resonance frequency) at which the reactance Xi of the reference coil becomes equal is fr (Hz), and the effective series resistance of the coil is Ri (Ω) and the output impedance Zs (Ω) of the AC power supply at fr, then Zs < (Ri / 5) is satisfied, and the positive peak value and the negative peak value of the AC voltage waveform at both ends of the capacitor are symmetric with respect to zero potential due to the DC current blocking action of the capacitor. In an actual series resonance circuit, when the AC voltage waveform across the capacitor is shifted from the zero point and the output frequency of the AC power supply is set to fr, one capacitor is connected to the GND terminal of the AC power supply and the reference coil. Capacitor whose ratio of positive peak value Vp and negative peak value Vn with respect to zero potential at both ends of capacitors connected in series is S = Vp / Vn, which is the same as the capacitor (the type, configuration and capacitance are all the same) The ratio of the positive peak value Vp to the zero potential of the voltage across the capacitor connected in series with the other terminal of the AC power supply and the reference coil, and the negative peak value Vn is S1 = Vp / Vn, AC power supply A capacitor is connected in series between the GND terminal of the power supply and one terminal of the power transmission coil, and the same capacitor as the capacitor is connected between the other terminal of the AC power source and the other terminal of the power transmission coil. When the ratio of the positive peak value Vp to the zero potential of the voltage across the capacitor connected to the other terminal when connected in columns and the negative peak value Vn is Sn = Vn / Vp, the capacitor (13 )But,
1) The capacitance of the capacitor is C (μF), and the coefficient α is α = log (C / 0.001 μF) (where C ≧ 0.01 μF, α ≧ 1; C <0.01 μF, α = 1) When the predetermined coefficient B is B = 1 + 0.06α (no unit), whether the capacitor (13) satisfies the relationship S ≦ B,
2) Alternatively, the capacitor (13) has all the following conditions: Fa = (Sp-1) / (S-1) <0.75,
Fb = (Sp-1) / (S1-1) <0.75,
Fc = (Sn-1) / (S-1) <0.75,
Fd = (Sn-1) / (S1-1) <0.75,
If at least one of the conditions is satisfied, and the condition 2) is satisfied, a capacitor (between the GND terminal of the AC power supply and one terminal of the power transmission coil) is satisfied. 13) are connected in series, and a capacitor (13) is connected in series between the other terminal of the AC power supply and the other terminal of the power transmission coil.

In this example, the voltage waveform at both ends of the capacitor is symmetric with respect to the zero point due to the DC current blocking action of the capacitor, but in an actual series resonance circuit (power transmission circuit), the voltage waveform of the capacitor is shifted from the zero point. To do. A capacitor having good power transmission characteristics can be selected by utilizing the asymmetry of the voltage across the capacitor with respect to the zero point. In addition, capacitors with poor power transmission performance that cannot be excluded by the definition of dielectric loss tangent tan δ can be swept out of the specified value. Therefore, it is possible to select a capacitor with good power transmission performance more accurately. As a result, a power transmission device with good power transmission performance can be realized. Furthermore, when a capacitor is connected to both ends of the coil, the value of Vp / Vn approaches 1 and the power transmission performance can be improved.

In still another example of the present invention , a positive voltage is applied to one terminal A and a negative voltage is applied to the other terminal B of the power transmission coil of the previous invention and the two terminals of the capacitor (13). Dielectric absorption when Kp is positive, a positive voltage is applied to the other terminal B, and a positive voltage is applied to one terminal A is Kn, and Kr is the ratio of Kp and Kn, Kp ≧ Kp ≧ When Kn, Kr = Kp / Kn, and Kp <Kn, when Kr = Kn / Kp, the capacitor (13) satisfies the condition 1 ≦ Kr <1.5. .

Dielectric absorption is a direct current characteristic of a capacitor, and is specified by a nonpolar capacitor in JIS. The dielectric absorption Kp measured by applying a positive voltage to one of the two terminals of the nonpolar capacitor is different from the dielectric absorption Kn measured by applying a positive voltage to the other terminal. By defining the asymmetry of dielectric absorption of this capacitor, it is possible to select a capacitor with better power transmission performance more accurately. As a result, a power transmission device with good power transmission performance can be realized.

Preferably, at least one value of Kp or Kn is 1% or less.

If the capacitor satisfies the definition of dielectric asymmetry, the lower the dielectric absorption value, the better the performance. However, a capacitor that does not satisfy 1 ≦ Kr <1.5, such as Kp of 0.1% and Kn of 1%, does not perform well, but a capacitor that has good dielectric absorption characteristics does not have dielectric absorption. The numbers are often low.

In still another example of the present invention, when an integrating circuit including the power transmission coil of the previous invention and a capacitor (13) is configured, the integrating circuit positively outputs a reference pulse generating means and a constant current of a predetermined same value. A current source capable of flowing in opposite directions, an initializing means for setting the output voltage of the integrating circuit to zero, and a pulse number measuring means for measuring at least N = 1000 counts as a reference pulse for measuring the forward / reverse integration time Including,
1) When the initialization unit sets the output voltage of the integration circuit to zero, the pulse measurement value of the pulse number measurement unit is initialized to zero, and then a constant current Ip in the positive direction is applied to the capacitor for a predetermined time T ( S), and after the elapse of a predetermined time T, when a predetermined constant current In in the negative direction of | Ip | = | In | is passed through the capacitor, the output voltage of the integrating circuit becomes zero. Time is Tn, pulse count is N for a predetermined time T, pulse count is Tn for time Tn until the voltage across the capacitor becomes zero,
2) When the output voltage of the integration circuit is set to zero by the initialization means, the pulse measurement value of the pulse number measurement means is initialized to zero, and then a constant current In in the reverse direction is applied to the capacitor for a predetermined time T. When a predetermined constant current Ip in the positive direction of | Ip | = | In | is passed through the capacitor after a lapse of a predetermined time T, the time until the output of the integrating circuit becomes zero is Tp, When the pulse count for the predetermined time T is N and the pulse count for the time Tn until the voltage across the capacitor becomes zero is Np, the absolute value of the difference between Nn and Np, | Nn−Np | , 0.003 × N capacitors or less (13).

In this example, it is possible to select a capacitor with good power transmission performance in a short time compared to the dielectric absorption measurement method defined in JIS. This action is defined as the asymmetry of the integrating circuit by the capacitor.
Each of the above-described capacitor selection methods has a correlation with the power transmission performance, and a capacitor with good power transmission performance can be obtained by using one or more selection methods.

More preferably, at least one of the absolute value of the difference between N and Nn, | N−Nn |, the absolute value of the difference between N and Np, and | N−Np | is 0.004 × Capacitor (13) with N counts or less.

In addition to the asymmetry of the integration circuit using the capacitor described above, it is more preferable if the ratio between the charging time and the discharging time of the capacitor using the integration circuit constituted by the capacitor is specified and the measured value is smaller than the specified value.

Further, as the prescribed operating conditions of the power transmitting device, the effective series resistance in the electricity transmission coil alone, Rw (Omega), constitutes a transformer with a power receiving coil of the inductive coupling is configurable to face the power transmission coil and the power transmitting coil If the effective series resistance of the power transmission coil when the both ends of the power reception coil are short-circuited is Rs (Ω), and the power transmission coil is Rs ≧ Rw, the maximum frequency satisfying the relationship of f1 (Hz) The capacitance of the capacitor (13) is selected so that the output frequency fa (Hz) can be set to f1 or less, the power transmission coil is driven by the AC power source, and the power is transmitted from the power transmission unit to the power reception unit.

In this example, the power transmission coil determined by the power transmission coil and the power reception coil has a maximum frequency f1 that satisfies the relationship of Rs ≧ Rw, and the power transmission coil is driven by an AC power supply, thereby realizing a high-performance power transmission device. .

Preferably, when both ends of the power receiving coil facing the power transmission coil of the power transmission device of the previous invention are opened, the effective series resistance of the power transmission coil is Rn (Ω), and the power transmission coil has a relationship of Rs ≧ Rn ≧ Rw. If the satisfied maximum frequency is f2 (Hz), the frequency fd (Hz) for driving the power transmission coil is set to a frequency equal to or lower than f2 (Hz).

In this example, by satisfying Rs ≧ Rn ≧ Rw at the frequency at which power is transmitted, a coil having a smaller effective series resistance Rw (Ω) can be selected, and an optimum frequency range for power transmission can be defined. Further, both the single coil and the transformer with the coils facing each other approach the ideal theoretical characteristics, and the power transmission performance can be improved as compared with the conventional one.

Preferably, the thermal resistance is θi (° C./W), the allowable operating temperature of the power transmission coil is Tw (° C.), the ambient temperature of the place where the power transmission coil is installed is Ta (° C.), Assuming that the alternating current flowing through the power transmission coil is Ia (A), at fd (Hz), power transmission is performed so that the power transmission coil satisfies the relationship Rw ≦ (Tw−Ta) / (Ia2 × θi). The power is transmitted from the unit to the power receiving unit.

In this way, by defining the thermal conditions based on the effective series resistance Rw (Ω) and the alternating current Ia (A), a turn that determines the upper limit of the alternating current Ia of at least one coil or the effective series resistance Rw of one coil. The upper limit of the number and the frequency region where the effective series resistance Rw (Ω) is small can be defined.

Preferably, the output frequency fa of the AC power supply is set to a frequency (series resonance frequency) fr (Hz) at which the reactance Xc of the capacitor and the reactance Xi of the power transmission coil are equal, and the power transmission coil is driven by the AC power supply, fa (Hz ), Rw (Ω) is the effective series resistance of the power transmission coil alone, Rc (Ω) is the effective series resistance of the capacitor at fa (Hz), and Zs (Ω) is the output impedance of the AC power supply at fa (Hz). In addition, the AC power supply satisfies at least Zs ≦ (Rw + Rc) × 4.

In this example, power loss due to the output impedance of the AC power supply is avoided. The relationship between each effective series resistance and Zs is preferably Zs <Rc <Ri.

Another aspect of the present invention is a power receiving coil that receives power from the power transmitting device of the previous invention, wherein the power receiving coil is formed on an air core so as to have a magnetic flux supplement area Sn, and Sn is greater than an area of the power transmitting coil. Sn is approximately 2% or more of the area of the power transmission coil, the effective series resistance of the power transmission coil when the power transmission coil itself is Rw, the power reception coil is short-circuited, and is opposed to the power transmission coil. Is selected so that the frequency f1 that satisfies the relationship of Rs ≧ Rw is 1 MHz or more.

By measuring the frequency characteristics of the effective series resistance of the power transmission coil with the shorted power reception coil facing the power transmission coil, a power reception coil suitable for the power transmission coil is obtained, and as a transformer that is a power transmission means composed of both coils. The characteristics can be improved.

Still another aspect of the present invention is a power receiving device, which includes at least the power receiving coil of the previous invention and a load.

The power receiving coil of the present invention can appropriately supply the power received from the power transmitting coil to the load, and the power receiving device of the present invention can be realized if the power receiving coil and the load are included at least.

Preferably, the power receiving coil is selected such that the maximum outer dimension is smaller than the minimum outer dimension of the power transmission coil, and the power receiving coil is arbitrarily positioned with respect to the winding surface of the power transmission coil at an arbitrary position in the vicinity of the power transmission coil. If the voltage set above a certain level is required to operate the load connected to the receiving coil at an angle and in any direction, and the electromotive force of the coil alone is insufficient, the receiving coil The capacitance of the capacitor is selected so that the reactance of the two terminals in which the power receiving coil and the capacitor are connected in parallel, or the point where the impedance becomes a maximum value is close to the AC power supply frequency fs. When a certain current or more is required for the load connected to the coil to operate, and the induced current in the coil alone is insufficient. The capacitor is connected in series between the power receiving coil and the load, and the point where the reactance of the two terminals where the power receiving coil and the capacitor are connected in series is zero or the impedance is a minimum value is close to the AC power supply frequency fs. Thus, the capacitance of the capacitor is selected.

In the present invention, a power receiving device capable of transmitting power over a long distance to a load that requires a high operating voltage can be realized by connecting a capacitor in parallel to the power receiving coil. Alternatively, by connecting a capacitor in series between the power receiving coil and the load, a power receiving device capable of transmitting power over a long distance to a load that requires a high operating current can be realized.

Still another aspect of the present invention is a power transmission device, which includes the power transmission device and the power reception device of the previous invention.

By configuring the power transmission device by combining the power transmission device and the power reception device of the previous invention, even in a low frequency region of 1 MHz or less, the power reception coil having a smaller magnetic flux supplement area than the power transmission coil Thus, electric power can be transmitted even when the magnetic flux supplementing surface of the power receiving coil is at an arbitrary angle with respect to the winding surface of the power transmitting coil at a distance of about the outer diameter of the power transmitting coil.

  According to the present invention, by increasing the width H of the foil conductor, a power transmission coil having a larger area can be realized by the coil, and a magnetic flux is formed over a wide range on the upper surface of the power transmission coil. Long-distance power transmission to a place away from the vehicle becomes possible.

  Moreover, even if the winding surface of a power transmission coil and the winding surface of a receiving coil are not parallel, and it is arbitrary angles and positional relationship, electric power can be transmitted.

In addition, a newly devised coil selection method can realize a long-distance power transmission coil formed of foil conductors, which was selected by this transmission coil and the newly devised capacitor selection method. A power transmission device capable of long-distance power transmission can be realized by providing a capacitor and setting the output frequency of the AC power supply to a specified value. Furthermore, a power receiving coil suitable for long-distance transmission can be realized by the above-described coil selection method, and a power receiving device equipped with the power receiving coil and a power transmission device including the power receiving device and the power transmitting device can be realized.

  Furthermore, since electric power is transmitted between coils having a large area ratio, the coupling coefficient between the power transmitting coil and the power receiving coil is small. For this reason, since the power transmission coil is not affected by the power receiving side, the AC power source may be fixed at a frequency determined by the power transmission coil and the capacitor so that the reactance is zero, and feedback control, AC power source and power transmission coil Complicating means such as comparing the voltage phase, detecting the state of the power receiving side, and adjusting the AC voltage and frequency of the AC power supply is not required. Therefore, the power transmission device can be realized at low cost.

(Explanation of differences in power transmission characteristics depending on the configuration of the power transmission coil)
An example of a coil suitable for power transmission is a coil wound in a plane spiral shape. As a conductor constituting the coil, a conductor having a circular or square cross section is usually used. These are referred to as “conductor coils”. In the power transmission device, the power transmission coil and the power reception coil can be separated. Therefore, the inventor of the present application combined the various coils as shown below with the power transmission device shown in FIG. 124 and actually performed a power transmission test.
(1) Power transmission coil: Conductive coil Power receiving coil: Conductive coil (2) Power transmission coil: Foil conductor coil Power receiving coil: Conductive coil (3) Power transmission coil: Conductive coil Power receiving coil: Foil conductive coil (4) Power transmission coil: Foil conductive coil Power receiving coil: foil conductor coil As a result, in the combination of the conductive coil of (1) above, good power transmission performance is obtained. The inventor of the present application summarized the results and has filed an application as PCT2007 / JP2007 / 061012. On the other hand, when a foil conductor coil is used for at least one of the power transmission coil or the power reception coil, it has been found that the power transmission performance is lower than in the case (1).

  Conductive coils and foil conductor coils used for actual measurement have a large effective series resistance. Therefore, in order to reduce the negative overcurrent, an incandescent bulb of 100V and 40W was used as a load. In this case, the load current IL (A) is IL = 40 W / 100 V = 0.4 A. The load resistance value RL (Ω) is RL = 100V / 0.4A = 250Ω, which is sufficiently larger than the effective series resistance of each coil, which is several Ω. Therefore, at least the power loss due to the effective series resistance of the power receiving coil is sufficiently small as 40 × (7/250) ≈1 W compared to the power consumption of the load.

  However, although the foil conductor coil is structurally good in heat dissipation, it has been found that heat is generated and there is an upper limit to the power that can be transmitted. Thus, it can be seen that the power transmission performance of the foil conductor coil is inferior to that of the conductor coil unless the foil conductor coil is used at a large voltage and a small current. Further, the foil conductor coil has a larger outer diameter than the conductor coil. Therefore, the foil conductor coil is not suitable for transmitting a large amount of power with high efficiency with a limited area coil. However, when used at a high voltage and a small current, a large electric power can be transmitted even with a foil conductor coil. The effect of the foil conductor coil in this case is different from the effect of the conductive wire coil. Details thereof will be described later.

(Other effects of foil conductor coil)
The inventor of the present application tried to make the coil of the power transmission unit larger than the coil of the power reception unit as in Patent Document 7 described above. And in order to transmit electric power to the receiving coil to which the load of high resistance value was connected, various foil conductor coils were made from about A5 size to a square of 60 cm square. As a result, it has been found that the foil conductor coil for power transmission having a large size as described above can transmit power over a long distance to a power receiving coil smaller than the power transmission coil. Furthermore, it has been found that the power receiving coil can receive power even if the magnetic flux supplement surface of the power receiving coil is not parallel to the winding surface of the power transmitting coil. As described above, the power transmission unit configured by the foil conductor coil can form a magnetic flux in a wide range around the coil. This function and effect is completely different from the function and effect of generating magnetic flux substantially parallel to the coil surface near the coil surface as in Patent Document 7. The effects described above are not capable of transmitting large power with high efficiency. However, in the present invention, long-distance power transmission is possible even in a frequency region of 1 MHz or less due to mutual induction. Depending on the size and configuration of the power transmission coil, power transmission over a long distance of 50 cm or more is possible with the power transmission coil as the center. Conventionally, there has been no prior art referring to the function and effect of such a foil conductor coil.

  Hereinafter, the above-described power transmission device of the present invention will be described in detail.

(Description of configuration of power transmission device)
FIG. 1 is a block diagram of a power transmission device 100 according to an embodiment of the present invention. In FIG. 1, a power transmission device 100 includes a power transmission unit 3 that operates as a power transmission device and a power reception unit 4 that operates as a power reception device. The power transmission unit 3 includes a DC power source 12, a power transmission control circuit 3a, and a power transmission coil 1. The power receiving device 4 includes a power receiving coil 2, a power receiving control circuit 4a, and a load RL. The power transmission coil 1 and the power reception coil 2 are disposed to face each other.

  The power transmission unit 3 and the power reception unit 4 are configured to be separable. When the power transmission unit 3 and the power reception unit 4 are coupled, the power transmission coil 1 and the power reception coil 2 are arranged to face each other, so that the power transmission coil 1 and the power reception coil 2 function as a transformer.

  The power transmission control circuit 3a of the power transmission unit 3 includes at least an AC power source 3b that operates as power conversion means such as an inverter circuit that converts the DC power source 12 into AC power. A series circuit in which the power factor improving capacitor 13 is connected in series to the power transmission coil 1 is driven by an AC power source 3 b at a predetermined frequency or lower to transmit power to the power receiving unit 4. The power reception unit 4 receives the power transmitted from the power transmission coil 1 by the power reception coil 2. The power reception control circuit 4a supplies the received power to the load RL. The power reception control circuit 4a includes a rectifier circuit that converts AC power into DC power. When the load RL operates with AC power such as an incandescent lamp or LED, the power receiving control circuit 4a can be omitted and the load RL can be directly connected to the power receiving coil 2.

  The power transmission unit 3 of the power transmission device 100 illustrated in FIG. 1 is configured to drive a series circuit in which a power factor improving capacitor 13 is connected in series to the power transmission coil 1 by supplying AC power from an AC power source 3b. ing. The output frequency fa (Hz) of the AC power supply 3b is determined by the inductance L1 (H) of the power transmission coil 1 and the capacitance C1 (F) of the power factor correction capacitor 13, and the reactance of the series circuit becomes zero. It is set close to the frequency frx (Hz) or the frequency frz (Hz) at which the impedance of the series circuit is minimized.

  In addition, alternating current means what can send an electric current through the coil connected to the output terminal to the forward direction and a reverse direction. Hereinafter, the power conversion means for converting the DC power supply 12 into AC power is referred to as AC power supply 3b. The output frequency of the AC power supply 3b is expressed as fa (Hz). Further, the frequency at which the power transmission coil 1 is driven by the AC power supply 3b is denoted as fd (Hz). The frequency at which the power receiving coil 2 receives power is denoted as fj (Hz). In this case, naturally, fa = fd = fj (Hz). fa, fd, and fj are frequencies used for power transmission. Both fa and fd are the frequencies of the power transmission unit 3, and the operational effect is also only the difference between the drive unit and the driven unit. However, the operational effects of fd and fj are different and will be described below. In the embodiment described later, the frequency for transmitting power is defined as fs (Hz), but fs is also equal to fa, fd, and fj.

(Description of operation of power transmission device)
The opposing power transmission coil 1 and power reception coil 2 shown in FIG. 1 are air-core coils, of which the effective series resistance of the power transmission coil alone is Rw (Ω). The effective series resistance of the power transmission coil 1 when the power reception coil 2 facing the power transmission coil 1 is short-circuited is Rs (Ω). As will be described later, when the frequency is low, the relationship between Rw and Rs is Rs> Rw. On the other hand, when the frequency increases, the relationship between Rw and Rs is Rs <Rw. The frequency satisfying Rs <Rw varies depending on the coil. That is, there is an upper limit for the frequency satisfying the relationship of Rs ≧ Rw, and the upper limit value varies depending on the coil. As described above, this upper limit value serves as a criterion for selecting a coil with good power transmission performance, can define a frequency region in which the coil constituting the power transmission device can be used, and can realize a power transmission device with good power transmission performance. is there.

  Therefore, in the power transmission device 100 according to the embodiment of the present invention, when the power transmission coil 1 has the highest frequency satisfying Rs ≧ Rw as f1 (Hz), the AC power supply included in the power transmission unit 30 The output frequency fa (Hz) is set to a frequency region of f1 (Hz) or less, and power is transmitted to the power receiving unit 40. When fa (Hz) is set as described above, the power transmission coil 1 is driven at the frequency fd = fa (Hz). The frequency fj received by the power receiving coil 2 satisfies the condition of fj ≦ f1. That is, the power transmission coil 1 satisfies the condition of fd ≦ f1. As a matter of course, the power transmission coil 1 satisfies the relationship of Rs ≧ Rw at fd (Hz). Further, the frequency fj (Hz) at which the power receiving coil 2 receives power is conditional on being f1 (Hz) or less. That is, the power receiving coil satisfies the condition of fj ≦ f1. As a matter of course, the power receiving coil 2 satisfies the relationship of Rs ≧ Rw at fj (Hz).

  As described above, the notation that “the driving frequency fd (Hz) is set to f1 (Hz) or lower” of the power transmission coil indicates that the driving frequency fd (Hz) of the power transmission coil satisfies the condition “fd ≦ f1”. It is synonymous with “Yes”. The expression “satisfying the condition of fd ≦ f1” is synonymous with the expression “the power transmission coil satisfies the relationship of Rs ≧ Rw at fd (Hz)”. In addition, the notation that the power receiving coil receives the power “frequency fj (Hz) is equal to or less than f1 (Hz)” indicates that the frequency fj (Hz) at which the power receiving coil receives power is “fj”. ≦ satisfying f1 ”.

  The expression “satisfying the condition of fj ≦ f1” is synonymous with the expression “the power receiving coil satisfies the relationship of Rs ≧ Rw at fj (Hz)”. Since the frequency of the received power cannot be set in the power receiving unit 40, the frequency fj (Hz) at which the power receiving coil 2 receives power is defined, and f1 of the power receiving coil 1 determined by the power receiving coil 2 and the power transmitting coil 1 is fj. That is the condition. Hereinafter, the conditions that the power transmission apparatus should satisfy are defined by any of the above-described notations.

  Furthermore, let Rn (Ω) be the effective series resistance of the power transmission coil 1 when the power reception coil 2 facing the power transmission coil 1 is opened. A maximum frequency satisfying Rs ≧ Rn ≧ Rw is defined as f2 (Hz). As will be described later, f2 (Hz) is lower than f1 (Hz). The power transmission device 100 sets the output frequency fa (Hz) of the AC power supply 3b included in the power transmission control circuit 30a to a frequency region of f2 (Hz) or less, and transmits power to the power receiving unit 40. When fa (Hz) is set as described above, the power transmission coil 1 is driven at the frequency fd = fa (Hz). That is, the power transmission coil 1 satisfies the condition of fd ≦ f2. As a matter of course, the power transmission coil 1 satisfies the relationship of Rs ≧ Rn ≧ Rw at fd (Hz). At this time, the frequency fj (Hz) at which the power receiving coil 2 receives power is conditional on being f2 (Hz) or less. That is, the power receiving coil 2 satisfies the condition of fj ≦ f1. As a matter of course, the power receiving coil 2 satisfies the relationship of Rs ≧ Rn ≧ Rw at fj (Hz). Hereinafter, similarly to the relationship between f1 and fd or the relationship between f1 and fj described above, the relationship between f2 and fd or the relationship between f2 and fj is defined by any notation.

Furthermore, when the thermal resistance of the coil is θi (° C./W), the allowable operating temperature of the coil is Tw (° C.), the ambient temperature of the place where the coil is installed is Ta (° C.), and power is transmitted. When the alternating current flowing through the coil is Ia (A), the relationship of (Tw−Ta) ≧ θi (Rw × Ia ² ) is satisfied.

(Specific example of coil used in power transmission unit: Embodiment 1)
2 is a perspective view showing a foil conductor coil 10a which is an example of a power transmission coil used in the power transmission unit of the power transmission device according to the embodiment of the present invention, and FIG. 3 is a foil conductor coil shown in FIG. It is a figure which expands and shows the cross section along line AA of 10a.

  In FIG. 1, a foil conductor coil 10a is formed as a flat single-layered air core on a flat surface 22 of a plate-like member 20 made of a square-shaped insulating material in a counterclockwise direction from the outside toward the inside. It is formed in a square shape and a spiral shape. In addition, the blank part in which the foil conductor is not wound is provided in the center part of the foil conductor coil 10a.

  As shown in FIG. 2, the foil-shaped conductor 30 has a cross section that intersects the direction in which the current flows approximately rectangular, more preferably rectangular, and the length (thickness) of one side of the rectangle is t (mm). When the length (width) of one side intersecting one side is H (mm), at least the width H is 10 times or more the thickness t, and the width H is selected to be 1 mm or more. Each foil conductor 30 is in contact with the plate-like member 20 on one side indicating the width H, and the adjacent foil conductors 30 have an interval w of at least 0.1 mm or a width H of the foil conductor. In addition, there is an interval of w = H / 40 (mm) or more, and at least the minimum outer dimension d1 of the outer shape is configured to be 15 times or more of the width H. Furthermore, the foil conductor coil 10a satisfies the condition that the number of turns is 4 turns or more and the self-inductance is at least 2 μH or more.

  The shape of the cross section of the foil-shaped conductor 30 is rectangular, the ratio of the width H to the thickness t, and H / t being defined as 10 or more, the ratio of the surface area Sf of the foil-shaped conductor 30 to the volume Vf, Sf / This is to increase Vf and reduce the influence of the skin effect. Furthermore, the reason why the width H is selected to be 0.5 mm or more is to suppress an increase in the DC resistance of the coil. Preferably, the width H is selected to be 1 mm or more.

  As described above, in the foil conductor coil, the distance between adjacent foil conductors is reduced as much as possible to reduce the direct current resistance of the coil and ensure the inductance. However, the effective series resistance Rw of a single coil increases with an increase in frequency due to eddy current loss in the coil having such a configuration. Therefore, the above-described interval w is provided between adjacent foil conductors. Thereby, an increase in the effective series resistance of the foil conductor due to eddy current loss can be avoided.

  The minimum outer shape of the power transmission coil is 10 cm or more in order to generate magnetic flux over a wide range, and the thickness t of the foil conductor 30 is selected to be 0.3 mm or less, and between adjacent foil conductors. When the spacing is S, the width H of the foil conductor satisfies H ≦ w × 40 because the distance between the conductors existing in the magnetic flux path generated by the adjacent foil conductor 30 is shortened. This is to reduce eddy current loss. By adopting such a coil configuration, a large-area power transmission coil can be realized, and a magnetic flux is formed over a wide range on the top surface of the power transmission coil, so long-distance power transmission away from the power transmission coil surface. Is possible.

  As shown in FIG. 2, one end of the covered electric wire 40 is electrically connected to the inner peripheral end of the central portion of the foil conductor 30 by soldering or the like. One end of the covered electric wire 50 is electrically connected to the end portion by soldering or the like. The covered electric wire 40 is fixed on the plate-like member 20 with an adhesive or the like. However, you may fix the covered electric wire 40 on the plate-shaped member 20 with a metal fitting etc., without using an adhesive agent.

  When a large-area coil is realized with a small number of turns, the inductance of the wire itself drawn from the coil conductor and the mutual inductance due to the relative positional relationship between the coil conductor cannot be ignored, and the coil viewed from the end of the wire cannot be ignored. The self-inductance varies depending on the relative position between the wire and the coil. Therefore, in this embodiment, one end on the center side of the foil-like conductor 30 is connected to one end of the covered electric wire 40, and the covered electric wire 40 is fixed to the plate-like member 20. By configuring the coil in this way, the relative position between the covered electric wire 40 and the foil-shaped conductor 30 does not vary, so that the self-inductance of the coil can be made constant.

  Alternatively, an electric wire is taken out from the center of the coil to the back surface of the plate-like member 20 around which the foil-like conductor is wound, and on the back surface of the plate-like member 20 so as to be close to the winding end position of the outermost periphery of the coil. The foil-like conductor 30 may be attached, and the electric wire connected to the outside of the coil may be drawn from there.

  When the foil conductor coil 10a is used as a power transmission coil, the covered wires 40 and 50 are connected to an AC power source (not shown), but the inductance is adjusted by appropriately selecting the length of the covered wires 40 and 50. it can. This also applies to the case where a capacitor is connected in series to at least one of the covered wires 40 and 50 and connected to an AC power source. The point where the reactance of the coil and the capacitor is zero as viewed from the AC power source is as follows. , There is an advantage that it can be adjusted by the length of 50.

  The foil conductor coil 10a configured as described above can be used as the power transmission coil 1 of the power transmission device shown in FIG.

(Description of power receiving coil)
4 and 5 show examples of the power receiving coil provided on the power receiving side of the power transmission device. FIG. 4 shows the power receiving coil 20a wound in a flat plate spiral shape, and FIG. 5 shows the power receiving coil 20b wound in a cylindrical shape. A magnetic flux supplement surface Sm is formed so as to surround the surface of the power receiving coil 20a of FIG. In FIG. 5, the inner side of the winding surface of the power receiving coil 20b wound in a cylindrical shape is formed as the magnetic flux supplement surface Sn. Hereinafter, Sm and Sn represent a magnetic flux supplement surface and an area of a coil having an arbitrary configuration.

  By increasing the width H of the foil conductor 30 shown in FIG. 3, the foil conductor coil 10a can realize a power transmission coil Lp having a large area, and a magnetic flux is formed over a wide range on the upper surface of the power transmission coil Lp. Therefore, long-distance power transmission to a place away from the power transmission coil surface is possible, and the winding surface of the foil conductor coil 10a and the magnetic flux supplement surface Sm of the power reception coil of any configuration shown in FIGS. Even if Sn is not parallel, power can be transmitted to the power receiving coil.

  The reason why the minimum outer dimension d1 is selected to be 15 times the width H or more and the number of turns of the foil conductor 30 is selected to be 4 turns or more is that the foil conductor coil 10a forms a magnetic flux in a wide range.

(Description of characteristics related to power transmission coil)
The effective resistance of the foil conductor coil 10a alone is Rw (Ω), and one or more other coils having an arbitrary configuration that can be inductively coupled to the foil conductor coil 10a are connected to the foil conductor coil 10a shown in FIG. When the maximum frequency satisfying Rs (Ω) and Rs ≧ Rw is f1 (Hz) when the effective series resistance of the foil conductor coil 10a is short-circuited and close to all other adjacent coils, Another coil is selected so that f1 is 100 kHz or more, and the foil conductor coil 10a is driven at fd (Hz), which is a frequency of f1 (Hz) or less.

  Furthermore, the effective resistance of the foil conductor coil 10a alone is Rw (Ω), and one or more other coils having an arbitrary configuration are brought close to the foil conductor coil 10a shown in FIG. When the effective resistance of the foil conductor coil 10a when opened is Rn (Ω) and the maximum frequency satisfying Rs> Rn ≧ Rw is f2 (Hz), the foil conductor coil 10a is in a frequency region below f2. Is driven at fd (Hz), which is a frequency equal to or lower than f1 (Hz). Thereby, since the transformer which combined the foil conductor coil 10a and the other coil of arbitrary structures can be brought close to a theoretical ideal state, power transmission efficiency can be improved.

  Further, when another coil identical to the opened foil conductor coil 10a is opposed to the foil conductor coil 10a, the effective series resistance of the foil conductor coil 10a is Rn1 (Ω), and the foil conductor coil 10a has Rs1 ≧ Rn ≧ Assuming that the highest frequency satisfying the relationship of Rw is f2o (Hz), the foil conductor coil 10a satisfies the relationship of f2o ≧ 50 kHz. In other words, the foil conductor coil 10a satisfies the relationship of Rs1 ≧ Rn ≧ Rw at 50 kHz.

Furthermore, when the thermal resistance of the coil is θi (° C./W), the allowable operating temperature of the coil is Tw (° C.), the ambient temperature of the place where the coil is installed is Ta (° C.), and power is transmitted. (Tw−Ta) ≧ θi (Rw × Ia ² ) is satisfied, where Ia (A) is the alternating current flowing through the coil.

  Next, the highest frequency f1 (Hz) satisfying the relationship of Rs ≧ Rw, the satisfactory condition described above, and the highest frequency f2 (Hz) satisfying the relationship of Rs ≧ Rn ≧ Rw, (Tw−Ta) ≧ θi (Rw × Ia2) will be described. In addition, since this description has the same effect also in other embodiment, description is abbreviate | omitted in embodiment described below.

  First, the relationship of Rs ≧ Rw and Rs ≧ Rn ≧ Rw will be described. Here, since the circuit theory is cited, the power transmission coil is expressed as a primary coil, and the power reception coil is expressed as a secondary coil.

(Explanation of equivalent circuit)
FIG. 6 is a diagram illustrating an equivalent circuit of the transformer. FIG. 7 shows an equivalent circuit that clearly shows the effective series resistance of the air-core coil alone, and FIG. 8 shows the equivalent circuit of the transformer that clearly shows the effective series resistance R1 of the primary coil and the effective series resistance R2 of the secondary coil. It is a figure showing a circuit. FIG. 9 is a diagram showing an equivalent circuit of the transformer when the secondary coil is short-circuited in FIG. FIG. 10 is a diagram illustrating an equivalent circuit of the transformer when the load resistance RL is connected to the secondary coil in FIG.

  First, in order to obtain the theoretical relationship between Rw, Rn, and Rs, the impedance on the primary side of the transformer, Z1, is obtained. In FIG. 6, L1 is the inductance of the primary coil, L2 is the inductance of the secondary coil, M is the mutual inductance between the primary coil and the secondary coil, V1 is the voltage across the primary coil, and V2 is The voltage across the secondary coil (load resistance RL), I1 is the current flowing through the primary coil, I2 is the current flowing through the secondary coil, RL is the load resistance (pure resistance), and Z1 is the primary input impedance Represents.

In the equivalent circuit shown in FIG. 6, the following circuit equation is established, and the pure resistance component (effective resistance) and reactance component (inductance) of Z1 can be obtained by solving the following simultaneous equations. The circuit equation of FIG. 6 is described below. Note that j ² = −1, and ω is an angular frequency (ω = 2πf, f is a frequency (Hz)).

V1 = jωL1 · I1 + jωM · I2 (1)
V2 = jωM · I1 + jωL2 · I2 (2)
V2 = −RL · I2 (3)
What we want to find is Z1 = V1 / I1. Therefore, V2 and I2 may be eliminated from the above three simultaneous equations.

Substituting equation (3) of the above simultaneous equations into equation (2) and eliminating V2,
0 = jωM · I1 + (jωL2 + RL) I2
When the above equation is solved with respect to I2 and substituted into the above equation (1), and I2 is eliminated,
V1 = (jωL1 + ω ² M ² / (jωL2 + RL)) I1
Since Z1 = V1 / I1, from the above equation, Z1 is
Z1 = jωL1 + ω ² M ² / (jωL2 + RL).

Since an actual transformer has an effective resistance R1 on the primary coil and an effective resistance R2 on the secondary coil, considering the circuit of FIG. 9, if R1 is added to the primary coil with RL = R2,
Z1 = (R1 + jωL1 + ω ² M ² / (jωL2 + R2)
It becomes. In the above equation, ω ² M ² / (jωL2 + R2) is multiplied by (−jωL2 + R2) / (− jωL2 + R2) = 1.
Z1 = (R1 + R2 · ω ² M ² / (ω ² L2 ² + R2 ² ))
+ Jω (L1-L ² · ω ² M2 / (ω ² L2 ² + R2 ² )),
If A ² = ω ² M ² / (ω ² L2 ² + R2 ² ), then Z1 is
Z1 = (R1 + A ² R2) + jω (L1−A ² L2) (4)
It becomes. Since ω ² > 0, M ² ≧ 0, L2 ² > 0, and R2 ² > 0, obviously, A ² ≧ 0. That is, in FIG. 8, the input impedance Z1 of the primary coil is
Z1 = R1 + jωL1 (5)
As is clear from the comparison between the equations (5) and (4), as shown in FIG. 9, when the secondary coil of the transformer is short-circuited, the effective resistance R1 of the primary coil increases. It can be seen that the inductance L1 decreases. The above-mentioned is known in the above-mentioned “University Course Electrical Circuit (1)” written by Katsuro Ohno and Tetsuo Nishi, published by Ohmsha (August 20, 2001, first edition, June 30, 1968) Circuit theory.

  The above formulas (5) and (4) are basic formulas used to explain the conditions of Rs ≧ Rw, Rs ≧ Rn ≧ Rw, and to explain the relationship between Rw, Rn, and Rs.

  Next, a specific example of the foil conductor coil 10a shown in FIG. 2 will be described. There are some overlaps, but the definition of the symbol is clear. Rw is the effective resistance of the foil conductor coil 10a alone (R1 in FIG. 7), and Rn is the foil conductor coil when another air core coil faces the foil conductor coil 10a and the facing air core coil is open. The effective resistance of 10a (R1 in FIG. 8) and Rs are the effective resistance of the foil conductor coil 10a when another air core coil faces the foil conductor coil 10a and the facing air core coil is short-circuited (FIG. 9). R1) and kr are coupling coefficients between the coils approximately obtained from Rw and Rs.

  Further, when the inductance of the foil conductor coil 10a alone is Lw, and when the other air-core coil faces the foil conductor coil 10a and the opposed air-core coil is short-circuited, the inductance of the foil conductor coil 10a is Ls. , A coupling coefficient approximately obtained from Lw and Ls is expressed as ki. An approximate method for obtaining kr and ki will be described later.

  When L1 indicates the coil itself, L1 is a symbol, and when L1 is a numerical value of inductance, a unit is added as L1 (H). The same applies to resistors such as R1 and Rw. However, when Rs ≧ Rw, etc. are described with an equal sign or an inequality sign, when Rw is described before or after the description that Rw etc. is used in the calculation or when it is used for calculation , “Rw is 2Ω”, etc. When the unit and the unit are described immediately after Rw, etc., when it is clear that it is a numerical value in the explanation of the characteristic diagram, etc., the unit addition is omitted. ing.

  The highest frequency that satisfies the condition of Rs ≧ Rw is expressed as f1 (Hz), and the highest frequency that satisfies the condition of Rs ≧ Rn ≧ Rw is expressed as f2 (Hz). When f1 and f2 are simply written, f1 is the highest frequency (Hz) that satisfies the condition of Rs ≧ Rw, and f2 is the highest frequency (Hz) that satisfies the condition of Rs ≧ Rn ≧ Rw. Show.

  In the present invention, the coupling coefficient between the power transmission coil and the power reception coil is often low. Therefore, there are cases where Rs = Rw. For example, there is a case where the area of the power receiving coil is about 2% compared to the area of the power transmitting coil. Alternatively, there are cases where the distance between the power transmission coil and the power reception coil is long. Therefore, if the highest frequency satisfying the condition of Rs ≧ Rw is f1 (Hz), the fd (Hz), fa (Hz), and fj (Hz) are the same as f1 (Hz). Good. Therefore, the relationship between fd (Hz), fa (Hz), fj (Hz) and f1 (Hz), f2 (Hz) is not expressed as f1 “less than”, but is expressed as f1 “below”. .

First, the condition of Rs ≧ Rw will be described. As described above, the relationship of Rs> Rn = R1 = Rw is established between the transformer configured as shown in FIG. 8 and the transformer configured as shown in FIG. 9 in circuit theory. However, referring to a characteristic diagram showing the relationship between Rw, Rs, Rn and frequency, which will be described later, in the high frequency region, the relationship of Rs> Rn = R1 = Rw is not satisfied, and Rs <Rw, It has become. In the above-described equation (4), A ² is
A ² = ω ² M ² / (ω ² L2 ² + R2 ² ),
As described above, A ² ≧ 0. Therefore, the pure resistance component of the primary side impedance of the transformer shown in FIG. 8 and FIG.
Rs = (R1 + A ² R2) = (Rw + A ² R2) ≧ Rw,
When the secondary side is opened, Rn = R1 = Rw.

  As described above, if the circuit theory is followed, Rs ≧ Rw must be satisfied, but from the actual measurement result, the above inequality, Rs ≧ Rw, does not hold when in the high frequency region.

Again, if you look for the above ^{A 2,}
Since A ² = ω ² M ² / (ω ² L2 ² + R2 ² ) and Rs ≧ Rw does not hold because it is in the high frequency region, in A ² ω ² L2 ² >> R2 ² holds ,
A ² ≈ω ² M ² / ω ² L2 ² = M ² / L2 ² .

According to the known circuit theory, if the coupling coefficient between the primary coil and the secondary coil is k,
The relationship of M ² = k ² L1 · L2 is established. If the primary and secondary coils are the same,
Since L1 = L2 = Lw and R1 = R2 = Rw,
A ² = M ² / L ^{2 2} = k ² L1 · L2 / L2 ² = k ² Lw / L2.

In this manner, the square of approximately coupling coefficient from known Lw (H) and L2 (H), can be obtained ^{k 2.} For example, when the coil 10A is used on both the primary side and the secondary side, Lw = L2, Rw = R2,
Rs≈Rw + k ² Rw, k ² Rw = Rs−Rw, k ² = (Rs−Rw) / Rw (6)
Ls≈Lw−k ² Lw, k ² Lw = Lw−Ls, k ² = (Lw−Ls) / Lw (7)
As a result, the square of the coupling coefficient k can be approximately calculated, and the coupling coefficient k is determined by taking the square root of the right side of the above equations (6) and (7). Here, the coupling coefficient obtained by taking the square root of the right side of the equation (6) is denoted as kr, and the coupling coefficient obtained by taking the square root of the right side of the equation (7) is denoted by ki.

kr = √ ((Rs−Rw) / Rw),
ki = √ ((Lw−Ls) / Lw),
Kr is approximately obtained from Rw and Rs, and ki is approximately calculated from Lw and Ls. The above formulas (6) and (7) are for the case where the same coil is used for the primary side coil and the secondary side coil, and when the primary side coil and the secondary side coil are different,
Rs≈Rw + k ² R2, k ² R2 = Rs−Rw, k ² = (Rs−Rw) / R2 (8)
Ls≈Lw−k ² L2, k ² L2 = Lw−Ls, k ² = (Lw−Ls) / L2 (9)
kr = √ ((Rs−Rw) / R2),
ki = √ ((Lw−Ls) / L2),
As a result, the coupling coefficient can be obtained approximately. Note that R2 (Ω) is the effective series resistance of the power receiving coil 2 in FIG. 9, and L2 (H) is the self-inductance of the power receiving coil 2 in FIG.

Rw in the expression (6) and R2 in the expression (8) are both positive values. In the expressions (6) and (8), if Rs ≧ Rw is not satisfied, the expression (6), (8 ) The right side of the expression is negative. In this case, kr ² <0, and mathematically, the coupling coefficient kr becomes an imaginary number. Although the two coils are actually inductively coupled opposite to each other, the coupling coefficient cannot be imaginary, and there are imaginary coupling coefficients in electromagnetic induction and circuit theory. do not do.

  Rs−Rw ≧ 0, that is, Rs ≧ Rw is a condition that should be satisfied, and when the highest frequency that satisfies this condition is f1 (Hz), the power transmission coil 1 is in a frequency region below f1 (Hz). It must be used for power transmission. Therefore, it is preferable to select a coil having a high maximum frequency f1 (Hz) that satisfies the relationship of Rs ≧ Rw.

  However, the present application is directed to an invention relating to a power transmission device capable of transmitting power over a long distance from a power transmission coil having a large area to a power reception coil having a small area. Under certain limited conditions, a certain amount of power (approximately 30W) can be transmitted with a transmission efficiency of about 80% between foil conductor coils having an area ratio of about 1: 3 or less, or between foil conductor coils and conductor coils. it can. However, it is difficult to transmit high power with high efficiency between foil conductor coils of the same area. On the other hand, in a conducting wire coil, large power can be transmitted with high efficiency between the same coils. Therefore, the difference in action and effect between a coil obtained by winding a single conductor wire having a circular cross section in a plane spiral shape and a coil obtained by winding a foil-like conductor in a plane spiral shape will be described below.

(Specifications and characteristics of each coil)
Table 1 is a diagram showing the specifications of each coil in the data cited in order to show the operation and effect of the present invention. The coils 1A to 1C are conductive coils, and the coils 10A to 10F are foil conductor coils. The specifications of each coil of the coil 1J, which is a power receiving coil formed in (1), are described.

(Characteristics of coils 1A to 1C)
FIG. 11 is a diagram showing the relationship between Rw, Rs, Rn and frequency of a coil 1A in which a single conductor of 1 mm is closely wound with an outer diameter of 70 mm and 25 turns.
FIG. 12 is a diagram showing a relationship between Rw, Rs, Rn and frequency of a coil 1B in which a 0.6 mm single conductor wire is closely wound with an outer diameter of 70 mm for 40 turns. FIG. 13 is a diagram showing the relationship between Rw, Rs, Rn and frequency of a coil 1C in which a single conductor of 0.3 mm is closely wound with an outer diameter of 70 mm for 70 turns.

11 to 13, let Rw1 be the effective series resistance of each coil when the frequency is 10 kHz, and Rw2 the effective series resistance of each coil when the frequency is 1 MHz. The increase rate s1 of the effective series resistance is s1 = Rw2 / Rw1. s1 and f1 are
In the coil 1A, s1 = 3.8Ω / 0.08Ω = 47.5, f1≈70 kHz
In the coil 1B, s1 = 7.4Ω / 0.36Ω = 20.5, f1≈210 kHz
In the coil 1C, s1 = 14.4Ω / 1.7Ω = 8.47, f1 = 820 kHz
Multiplying s1 and f1,
In the coil 1A, s1 × f1 = 48.7 × 70 = 3325
In the coil 1B, s1 × f1 = 22.1 × 210 = 4313
In the coil 1C, s1 × f1 = 8.47 × 820 = 6945
When the conductor outer diameter is d (mm), the logarithm of d × 10 is taken and multiplied by the above s1 × f1,
In the coil 1A, 2681 × log (10) = 3325
In coil 1A, 4258 × log (6) = 4258 × 0.77 = 3359
In the coil 1A, 6945 × log (3) = 6945 × 0.47 = 3313
As shown, the single conductor shows almost perfect correlation between f1 and s1. Furthermore, when multiplying s1 × f1 obtained above by the wire diameter d (mm),
In the coil 1A, 2681 × 1 = 2681
In coil 1B, 4258 × 0.6 = 2555
In the coil 1C, 6945 × 0.3 = 2 083
As shown, the values are substantially the same.

  As described above, it can be seen that the coil formed of a single conductor in a plane spiral shape has a decrease in f1 (Hz) as the wire diameter d increases and the increase rate s1 of the effective series resistance depending on the frequency is large. Thus, in the coil formed in a plane spiral shape with a single conductor, there is a correlation between the wire diameters d, f1, and s1.

(Characteristics of coils 10A to 10E)
FIG. 14 shows a case in which two coils 10A, each of which is a six-turn winding of a copper foil having a foil thickness of 70 μm, with a foil width H = 10 mm and a gap w = 2 mm, are used on the same A4 size printed circuit board. It is a figure which shows the relationship between Rs and Rn and a frequency. FIG. 15 shows Rw, Rs, Rn in the case where two coils 10B wound with 8 turns of copper foil having a thickness of 70 μm, with a foil width H = 7 mm and a gap w = 2 mm are used on the same printed circuit board. It is a figure which shows the relationship of a frequency. In FIG. 14 and FIG. 15, f1 (Hz) which is the highest frequency satisfying the relationship of Rs ≧ Rw and f2 (Hz) which is the highest frequency satisfying Rs ≧ Rn ≧ Rw are specified. ing.

  In both the coil 10A shown in FIG. 14 and the coil 10B shown in FIG. 15, the maximum frequency f1 (Hz) that satisfies the relationship of Rs ≧ Rw is about 350 kHz, and the maximum frequency that satisfies the relationship of Rs ≧ Rn ≧ Rw. F2 (Hz) is about 130 kHz. That is, even if coils are formed with the same dimensions, no difference is observed in f1 (Hz) and f2 (Hz) due to the difference in foil width. Further, the ratio s1 between the effective series resistance Rw of each coil unit at 10 kHz and the effective series resistance Rw of each coil unit at 1 MHz will be examined. In the coil 10A, 0.33 / 0.11 = 3, and in the coil 10B, 0.57 / 0.19 = 3. The increase rate s1 of the effective series resistance of both coils is about 3 times, and no difference is seen between the two coils. On the other hand, s1 of the coil 1C is about 8. However, the maximum frequency f1 (Hz) that the coil 1C satisfies the relationship of Rs ≧ Rw is about 820 kHz, and the maximum frequency f2 (Hz) that satisfies the relationship of Rs ≧ Rn ≧ Rw is about 250 kHz. Both f1 (Hz) and f2 (Hz) are higher than the coils 10A and 10B. As described above, the maximum frequency f1 (Hz) satisfying Rs ≧ Rw and the maximum frequency f2 (Hz) satisfying the relationship of Rs ≧ Rn ≧ Rw are different between the conductor coil and the foil conductor coil. I understand that.

  In the conductor coil, it has been found that f1 (Hz), which is the highest frequency that satisfies the relationship of Rs ≧ Rw, is the upper limit of the frequency that can be used for power transmission. Further, by selecting the opposing coil, the f1 (Hz) can be increased, and the power transmission performance can be improved. Details thereof are described in PCT2007 / JP2007 / 061012. However, although there are differences between the opposing coils and the effects of the conductive wire coils, even if the coils 10A and 10B have f1 of around 350 kHz, they can actually be used at a frequency of 1 MHz or more. However, when power is transmitted between the two coils 10A and between the two coils 10B, about 350 kHz which is f1 (Hz) which is the highest frequency satisfying Rs ≧ Rw can be used for power transmission. The highest frequency.

  Next, the inventor of the present application uses two coils 10C in which the number of turns is the same as that of the coil 10A, only the foil width H is 7 mm, and the interval w between adjacent foil conductors is increased to 5 mm. , Rs, Rn frequency characteristics were measured.

  FIG. 16 is a diagram showing the relationship between Rw, Rs, Rn and frequency when two coils 10C are opposed to each other.

  According to FIG. 16, it can be seen that f1 (Hz), which is the highest frequency that satisfies the relationship of Rs ≧ Rw, has increased to about 3.2 MHz. Moreover, f2 (Hz) which is the highest frequency satisfying the relationship of Rs ≧ Rn ≧ Rw is increased to 1 MHz.

  Since f1 of the coil 10A is about 350 kHz and f2 is about 120 kHz, f1 rises about 9 times and f2 also rises about 9 times. Note that the inductance Lw (H) of the single coil is 7.1 μH for the coil 10A and about 7.7 μH for the coil 10C. Since the number of turns is the same, the coil 10A and the coil 10C have substantially the same inductance. However, the effective series resistance of the coil 10A is smaller due to the difference in foil thickness and foil width.

  From the above results, when two identical foil conductor coils are used, f1 (Hz) which is the highest frequency satisfying Rs ≧ Rw is determined only by the interval w between adjacent foil conductors, and the number of turns and foil width Seems to have no correlation. However, an appropriate foil conductor coil cannot be selected only by comparing f1 when the same coil is opposed. Actually, the coil 10A having a low effective series resistance although f1 is low has better power transmission performance than the coil 10C. This point will be described later. Next, consider the foil thickness.

  FIG. 17 shows the frequency of Rw, Rs, Rn and the frequency of the coil 10D when the coil 10C is opposed to the coil 10D in which the copper foil of the coil 10C is soldered and finished to a foil thickness of about 0.3 mm. It is a figure which shows a relationship. FIG. 18 is a diagram illustrating the relationship between Rw, Rs, and Rn and the frequency of the coil 10C when the coil 10D is opposed to the coil 10C. FIG. 19 is a diagram showing the relationship between the effective series resistance Rw of the coil 10A alone, the effective series resistance Rw of the coil 10C alone, the effective series resistance Rw of the coil 10D alone, and the frequency.

  First, referring to FIG. 19, it can be seen that the effective series resistance Rw of the coil alone is not lowered until the coil 10D is equivalent to the coil 10A, but is improved over the coil 10C. The difference in effective series resistance Rw between the coil 10A and the coil 10D is a difference in DC resistance. Solder has a large specific resistance. The coil 10A has a foil thickness of 70 μm. If a copper foil having a foil thickness of about 0.1 mm is used for the coil 10D, a DC resistance substantially equal to that of the coil 10A can be obtained.

  Next, referring to FIG. 16, as in FIG. 15, the maximum frequency f2 (Hz) that satisfies Rs ≧ Rn ≧ Rw exists around 1 MHz. The maximum frequency f1 (Hz) that satisfies Rs ≧ Rw is 2.3 MHz in FIG. 16 but is 4 MHz or more in FIG. 17 and is not shown. This is presumably because when the coil 10D is formed, the flatness of the substrate is broken due to heat during soldering, and the distance between adjacent faces is longer than that in FIG. This tendency is the same when the coil 10D is opposed to the coil 10C in FIG. From FIG. 16 to FIG. 18, in order to increase f1 (Hz) which is the highest frequency satisfying Rs ≧ Rw, and to decrease the effective series resistance Rw of a single coil, the foil width t is reduced by reducing the foil thickness t. And the interval w between adjacent foil conductors may be widened. That is, it can be seen that up to about one-tenth of the foil width H, the foil thickness t is increased and the interval w between adjacent foil conductors is increased. If this rule is satisfied, the foil thickness t can be increased to about 0.3 mm.

  Next, an embodiment of a foil conductor coil that can be used as a power receiving coil will be described.

  FIG. 20 is a diagram showing the relationship between Rw, Rs, Rn and frequency when two coils 10E wound with 40 turns with a foil width of 0.5 mm and a gap of 0.5 mm are used. FIG. 21 is a diagram showing the relationship between Rw, Rs, Rn and frequency when using two coils 10F wound for 15 turns with a foil width of 4 mm and a gap of 0.1 mm prepared as a comparative example.

  Referring to FIG. 20, the coil 10E satisfies the relationship of Rs ≧ Rw and Rs ≧ Rn ≧ Rw up to 4 MHz or more. It can be inferred that this is because the gap w is wider than the foil width H. Referring to FIG. 21, the maximum frequency f1 (Hz) at which the coil 10F satisfies the relationship Rs ≧ Rw is 150 kHz. The maximum frequency f2 (Hz) at which the coil 10F satisfies the relationship of Rs ≧ Rn ≧ Rw is 25 kHz. The power transmission device 100 of the present invention does not cause the power receiving coil 2 to be close to and opposed to the power transmitting coil 1, and the power receiving coil 2 may be separated from the power transmitting coil 1. Therefore, it is preferable to transmit power in a frequency region of 250 kHz or less because of radio interference. Therefore, it is necessary that the power transmission coil 1 has the highest frequency f1 (Hz) that satisfies the relationship of Rs ≧ Rw, and the upper limit for using the power transmission coil is at least 200 kHz. When two coils 10F are used, none of the coils can be used as a power transmission coil. In this case, the coil 10F is a comparative example.

  On the other hand, the characteristics when the coil 10F is opposed to another coil change considerably.

  FIG. 22 is a diagram showing a relationship between Rw, Rs, Rn and frequency of the power transmission coil 10F when the coil 10F is used as the power transmission coil and the coil 10E is used as the power reception coil.

  In FIG. 21, the maximum frequency f1 (Hz) at which the coil 10F satisfies the relationship Rs ≧ Rw is 130 kHz, but in FIG. 22, the maximum frequency f1 at which the coil 10F satisfies the relationship Rs ≧ Rw. (Hz) has risen to 2.3 MHz. Similarly, in FIG. 21, the maximum frequency f2 (Hz) at which the coil 10F satisfies the relationship of Rs ≧ Rn ≧ Rw is 25 kHz, but in FIG. 22, the coil 10F has Rs ≧ Rn ≧ Rw, The maximum frequency f2 (Hz) that satisfies the above relationship has increased to 1.2 MHz. Thus, in the foil conductor coil, f1 and f2 can be significantly increased by selecting an opposing coil. This effect is the same when the coil opposed to the conductor coil is a foil conductor coil, and f1 and f2 of the conductor coil can be significantly increased. An example of a combination of a conductive wire coil and a foil conductor coil will be described later.

  Thereafter, when the same coil is combined as the power transmission coil, the coil 10A having the lowest maximum frequency f1 (Hz) satisfying Rs ≧ Rw and the coil 10F that cannot be used as the power transmission coil when the same coil is combined. Let's see the characteristics by combining.

  FIG. 23 is a diagram illustrating a relationship between Rw, Rs, and Rn and the frequency of the coil 10A when the coil 10A is used as the power transmission coil and the coil 10F is used as the power reception coil. FIG. 24 is a diagram illustrating the relationship between Rw, Rs, and Rn and the frequency of the coil 10F when the coil 10A is used as the power transmission coil and the coil 10F is used as the power reception coil.

  In FIG. 23, the maximum frequency f1 (Hz) at which the coil 10A satisfies the relationship of Rs ≧ Rw is 1 MHz or more. Also in FIG. 24, the maximum frequency f1 (Hz) at which the coil 10F satisfies the relationship of Rs ≧ Rw is 1 MHz or more. Assuming that the frequency used for power transmission is 1 MHz, when the coil 10A and the coil 10F are combined, f1 of each coil is 1 MHz or more, which satisfies a predetermined condition. However, depending on the place where the coil 10F is installed on the coil 10A, f1 may be 1 MHz or less. However, when the short-circuited coil 10F is installed on the surface of the coil 10A, if there is a location where Rs of the coil 10A is larger than Rw, the predetermined condition is satisfied.

  Thus, a predetermined condition when the same coil is combined, for example, a power transmission coil where f1 does not satisfy 1 MHz, and a predetermined condition when the same coil is combined, such as a power receiving coil where f1 does not satisfy 1 MHz. By combining these, the power transmission coil satisfies a predetermined condition. The power receiving coil also satisfies a predetermined condition. This is presumed to be because the area ratio between the power transmission coil and the power reception coil is large. In the conductive wire coil, the maximum frequency f1 (Hz) satisfying the relationship of Rs ≧ Rw is 70 mm in diameter and the coil 1A having a diameter of 70 mm and the maximum frequency f1 (Hz) satisfying the relationship of Rs ≧ Rw is 200 kHz in diameter. When the coil 1C having a length of 70 mm is combined, the f1 of the coil 1A is 70 kHz, and the f1 of the coil 1C is not changed to 200 kHz. Thus, the foil conductor coil has different characteristics from the conductor coil.

  And in FIG. 25, the relationship between Rw, Rs, Rn and the frequency of the power transmission coil, the coil 10A when the solenoid coil 10J having a short length made of a single conductor of 0.3 mm is used as the power receiving coil. Show.

  23 to 25 are equivalent to the coil 10F shown in FIG. 22, and the maximum frequency f1 (Hz) at which the coil 10A satisfies the relationship of Rs ≧ Rw is significantly higher than that in FIG. I understand. Thus, by selecting the power receiving coil with reference to the power transmitting coil, the power transmitting coil can increase the maximum frequency f1 (Hz) that satisfies the relationship of Rs ≧ Rw. Specifically, f1 can be set to 200 kHz or more. In any figure, f2 exists between 2 MHz and 4 MHz, but f1 is 4 MHz or more.

  A single coil in which a foil-like conductor is wound in a plane spiral shape has good frequency characteristics of effective series resistance, and the coil has a very high Q, so that it can be used for power transmission. In particular, when a series resonant circuit is formed using a capacitor, an alternating current can be passed through the power transmission coil, and the large current can form a magnetic flux in a wide range around the power transmission coil. As a result, electric power can be transmitted to a coil having an arbitrary configuration that is separated from the power transmission coil.

(Relationship of Rs ≧ Rn ≧ Rw)
Next, Rs ≧ Rn ≧ Rw will be described. As described above, if the circuit theory is followed, the relationship of Rs> Rn = Rw must be satisfied, but as with the relationship of Rs ≧ Rw, when the frequency increases, Rn becomes larger than Rw. From FIG. 11 to FIG. That is, when the frequency increases above, Rn becomes larger than Rs. For example, referring to FIG. 22, the relationship between f1 and f2 is f1> f2. This relationship holds for all coils.

  In this example, as is apparent from FIG. 22, when the maximum frequency satisfying the condition of Rs ≧ Rn ≧ Rw is f2 (Hz), the coil 10A is used for power transmission at a frequency equal to or lower than f2. It is a condition.

  When the maximum frequency satisfying the relationship of Rs ≧ Rn ≧ Rw is set to f2 (Hz), the coil 10C is opposed to the coil 10C and the coil 10C by using the coil 10C for power transmission at a frequency equal to or lower than f2. Both (electromagnetically coupled) transformers approach the ideal ideal state and can improve power transmission performance.

  In FIG. 22, f1 and f2 are clearly shown. As f1 is higher, f2 that can be used in a theoretically ideal state is higher. Although the foil-like conductor coil of the present invention can form magnetic flux in a wide range, the higher the frequency for driving the coil, the wider the magnetic flux can be spread. Therefore, it is preferable to select a coil having high f1 and f2.

  As described above, the power transmission coil is driven at the maximum frequency f1 (Hz) or less satisfying the relationship of Rs ≧ Rw, and the power transmission coil is operated at the maximum frequency f2 (Hz) or less satisfying the relationship of Rs ≧ Rn ≧ Rw. The conventional technology cannot define the frequency at which the coil can be used, such as driving.

(Regarding the regulation of coil thermal conditions)
Next, the condition of (Tw−Ta) ≧ θi (Rw × Ia ² ) will be described.

As described above, the output frequency fs (Hz) of the AC power supply is set to a point (resonance point) where the reactance determined by the inductance of the power transmission coil alone and the capacitance of the capacitor becomes zero. That is, the current flowing through the power transmission coil is all consumed by the effective series resistance Rw of the power transmission coil at fs (Hz), and the power of (Rw × Ia ² ) W is converted into heat as Joule loss by Rw.

  In implementing this invention, the thermal resistance θi (° C./W) of the coil is determined by the coil structure and installation conditions. However, the power transmission coil using the foil-like conductor in the present invention has a large conductor area, and unlike the solenoid or spiral coil, the entire surface of the conductor is in contact with air.

  Normally, the higher the temperature, the more heat is dissipated to the surroundings. Therefore, it is necessary to solve the heat diffusion equation accurately. However, for coils with various structures, heat constants such as specific heat are taken into account. Since it is difficult to solve the thermal diffusion equation, the thermal resistance θi (° C./W) is simply obtained by the following method.

  First, an initial coil temperature T1 (° C.) is obtained at a place where the power transmission coil is installed. A DC constant current Id (A) is passed through the coil, the voltage Vd (V) across the coil is measured, and the power consumption of the coil is determined as Pd = Vd × Id (W). Since the resistance value of the metal wire increases as the temperature rises, the voltage Vd across the coil rises, so Vd is recorded with a pen recorder or the like to obtain an average value, or the Vd is monitored successively with an A / D converter or the like. It is desirable to take an average value. When the thermal equilibrium is reached, the coil temperature T2 (° C.) is measured. The thermal resistance θi (° C./W) is obtained as θi = (T2−T1) / Pd (° C./W). This measurement is preferably performed as an average value by measuring several times while changing the current value of Id.

The thus obtained thermal resistance θi (° C./W) has an effective series resistance Rw determined by the effective series resistance Rw (Ω) of the coil under actual use conditions and the current Ia (A) flowing through the coil. When the consumed power, Rw × Ia ² (W), is multiplied, the coil temperature rise value under actual use conditions, Tr (° C.) is obtained. Tr = θi × Rw × Ia ² (° C.) If the temperature at which the coil can be operated is Tw (° C.) and the ambient temperature of the place where the coil is installed is Ta (° C.), Tr = Tw−Ta. If (Tw−Ta) ≧ θi × Rw × Ia ² (° C.) is not satisfied, the usable temperature of the coil will be exceeded, making it difficult to implement the present invention.

  The inequality defines the condition of Rw or Ia. At the frequency fs at which power is transmitted, the effective series resistance Rw is a variable obtained by actual measurement of the power transmission coil alone, a variable obtained by actual measurement of the current Ia (A) flowing through the power transmission coil, and other Tw (° C. ), Ta (° C.), and θi (° C./W) are known constants. Therefore, if Rw (Ω) is obtained, the upper limit value of Ia (A) is defined, and conversely if Ia (A) is determined, the upper limit value of Rw is defined. Ra is the sum of the direct current resistance Rd (Ω) and the alternating current resistance Ra (Ω), and Rd and Rw can be directly measured. Therefore, by determining Ia (A), the effective value increases with the number of turns. The upper limit value of the series resistance Rw (Ω) can be defined, and the frequency range in which power can be transmitted can be defined from the relationship between the effective series resistance Rw (Ω) and the frequency.

  That is, both 1V × 10A and 10V × 1A have the same 10 W power, but the power loss due to the effective series resistance of the coil is 100 times that of 1A in the case of 10A. The power transmission efficiency in the power transmission coil cannot be improved unless the power loss due to the effective series resistance of the coil is specified in consideration of the current Ia (A) flowing through the coil, not the power.

  However, even if an alternating current having an effective value of about 3 A is passed through the coil 10A, which is the coil 10a of FIG. 1 according to an embodiment of the present invention, at a frequency of 1 MHz, the heat dissipation of the coil itself is very high due to the configuration of the coil. As it is good, almost no coil heat is observed.

  The configuration of the power receiving coil is not particularly limited, and a coil having an arbitrary configuration such as a solenoid, spiral, honeycomb, or meander shape can be used. The present invention performs long-distance transmission from the power transmission coil to the power reception coil, and since the power received by the power reception coil is small, it is not particularly necessary to define the configuration and characteristics of the coil. The power receiving coil shown in FIGS. 4 and 5 represents a magnetic flux supplement surface of the power receiving coil, and the angle between the magnetic flux supplement surface and the foil-like conductor winding surface of the power transmission coil shown in FIG. Even if it is an angle, there is an excellent effect that power can be transmitted, which cannot be realized by the conventional technology.

  As described above, the coil 10E and the coil 10F are an embodiment of the power receiving coil configured as shown in FIG. 3, and the coil 10J is an embodiment of the power receiving coil configured as shown in FIG. As described above, the frequency region in which the power transmission coil can be driven is determined by the configuration of the power reception coil.

(Difference in characteristics between conductive coil and foil conductor coil)
When two identical coils are used and the maximum frequency f1 (Hz) satisfying the condition of Rs ≧ Rw and the maximum frequency f2 (Hz) satisfying the condition of Rs ≧ Rn ≧ Rw are obtained. explain.

  In the conductor coil, f1 and f2 obtained when two identical coils are used do not change so much even when combined with other conductor coils. For example, f1 of the conductive wire coil 1A shown in FIG. 11 is about 67 kHz, and f2 is about 25 kHz. As with the coil 1A, there is a coil 1D having a diameter of 70 mm and f1 of 3.2 MHz and f2 of 780 kHz when two identical coils are used. Even if the coil 1D is opposed to the coil 1A, the f1 of the coil 1A is 110 kHz, the f2 is 88 kHz, and the f1 and f2 of the coil 1A do not rise so much. In the case of a conductor coil, once the coil configuration is determined, the frequency characteristics of Rw, Rs, and Rn are almost determined. In addition, when a coil having a low f1 measured using two identical coils is opposed to a coil having a high f1, the f1 of a coil having a high f1 becomes low. As long as the inventors measured using various coils, there is no coil having characteristics contrary to this correlation.

  On the other hand, in the foil conductor coil, even if f1 measured by the same coil is low, it has been confirmed that f1 is greatly increased by changing the opposing coil. The characteristics of this foil conductor coil are the same even when combined with a conductor coil. When the coil 1F, which is a foil conductor coil, is combined with the coil 1A, which is a conductive coil, as described above, f1 of the conductive coil and the coil 1A greatly increases.

  FIG. 26 is a diagram illustrating the relationship between Rw, Rs, and Rn and the frequency of the coil 1A when the coil 10F is opposed to the coil 1A.

Referring to FIG. 26, it can be seen that the maximum frequency f1 (Hz) at which the coil 1A satisfies the relationship of Rs ≧ Rw has increased to 1.25 MHz. However, the frequency characteristic of the effective series resistance Rw of the coil 1A alone is poor. Further, the DC resistance of the coil 10F is large. Therefore, due to the above-described definition of the thermal condition of (Tw−Ta) ≧ θi × Rw × Ia ² (° C.), it is difficult to transmit a large amount of power between the coil 10A and the coil 10J. In this case, the coil 10F is used as a power transmission side coil, and power of about 100 V × 0.2 A = 20 W is input except for the capacitor. When the power transmission frequency is about 100 kHz, the effective series resistance Rw of the coil 10A is about 0.5Ω. Under this condition, power of about 20 V × 0.9 A = 18 W can be extracted from the coil 10 </ b> A that is a power receiving side coil. Thus, when the foil conductor coil is used, high power cannot be transmitted with high efficiency unless the operating conditions are specified in detail. Conventionally, only the high power is transmitted with high efficiency, and as described above, the power loss due to the current flowing through the coil and the effective series resistance of the coil is not considered at all.

  Furthermore, in the foil conductor coil, when the same coil is used, it is assumed that there are a coil with f1 of 300 kHz and a coil with f1 of 200 kHz. It has been confirmed that when a coil having f1 of 300 kHz and a coil having f1 of 200 kHz are combined, f1 increases in any coil. For example, when the coil 10B is opposed to the coil 10A having both f1 of about 350 kHz, the f1 of the coil 10A rises to 500 kHz or more. When the coil 10C is opposed to the coil 10A, f1 of the coil 10A rises to 1 MHz or more.

  Therefore, when using a foil conductor coil, it is only necessary to satisfy the condition that “the same foil conductor coil is not used”. If this condition is satisfied, at least long-distance power can be transmitted from the large-area power transmission foil conductor coil, which is the gist of the present application, to the power receiving coil that can be inductively coupled to the power transmitting coil including the conductive wire coil. It becomes like this. Alternatively, high power can be transmitted with high efficiency under specific conditions.

  The foil conductor coil has better frequency characteristics of the effective series resistance Rw of the coil alone than the coil around which the conductive wire is wound. For example, from FIG. 16, which is a characteristic diagram of the coil 10C, a transformer in which two foil conductor coils having good frequency characteristics of Rw are opposed to each other has a higher coupling coefficient than the ratio of Rs and Rw. Further, the highest frequency f1 (Hz) satisfying the relationship of Rs ≧ Rw and the highest frequency f2 (Hz) satisfying the relationship of Rs ≧ Rn ≧ Rw are also high. However, when the inventor of the present application conducted a power transmission test using two coils 10C, good power transmission performance was not obtained despite the large coil area. Although it cannot be determined, this point seems to be one of the reasons why the practical application of transformers using foil conductor coils is difficult.

  The following is an example of actual implementation of the present invention. The coil 10B which is an example of the coil 10a in this embodiment is used, and the coil 10B is driven by a square wave having an effective value of 5 V at a frequency of 1 MHz through a capacitor. The drive current is about 1.5A. When the same coil is used, the LED connected to the coil 1A, which has f1 of about 70 kHz, can be lit at about 40 cm on the upper surface of the coil 10a. As a precaution, the relationship between Rw, Rs, Rn and frequency of the coil 10A and the coil 1A is confirmed.

  FIG. 27 is a diagram illustrating a relationship between Rw, Rs, and Rn of the coil 10A and the frequency when the coil 1A is opposed to the coil 10A. FIG. 28 is a diagram illustrating a relationship between Rw, Rs, and Rn of the coil 1A and the frequency when the coil 10A is opposed to the coil 1A.

Referring to FIG. 27, the maximum frequency f1 (Hz) at which the coil 10A satisfies the relationship of Rs ≧ Rw is increased to 4 MHz or more, and the maximum frequency f2 (at which the relationship of Rs ≧ Rn ≧ Rw is satisfied) Hz) is also rising to 1.2 MHz or higher. Referring to FIG. 28, the maximum frequency f1 (Hz) that the coil 1A satisfies the relationship of Rs ≧ Rw is increased to 2.8 MHz, and the maximum frequency f2 that satisfies the relationship of Rs ≧ Rn ≧ Rw. It can be seen that (Hz) rises to 700 kHz.

  Thus, when the foil conductor coil is used, the maximum frequency f1 (Hz) satisfying the relationship of Rs ≧ Rw and the maximum frequency f2 (Hz) satisfying the relationship of Rs ≧ Rn ≧ Rw can be increased. . In the same coil, even if the maximum frequency f1 (Hz) that satisfies the relationship of Rs ≧ Rw is 70 kHz, the coil 1A can be used as a power receiving coil at 1 MHz or more.

  Here, the grounds for defining f1 as 100 kHz will be described. The highest frequency that can be used for power transmission is 120 kHz in the lowest country (France). Actually, there are variations in the components, so if the power transmission frequency is set to 120 kHz, it may exceed 120 kHz. Therefore, it is defined that Rs> Rw is satisfied at a minimum of 100 kHz with an allowance of about 20%. For f2, since f2 <f1, 50 kHz, which is half the value of f1, is defined as a guide.

  However, a frequency exceeding 1 MHz is used in a space where radio waves are completely shielded, or when the public property is very high. For example, an automatic ticket gate at rush hour. Therefore, regardless of the above definition, f1 and f2 are preferably as high as possible.

(Regarding the frequency range in which the coil can be used)
The upper limit of the frequency at which the coil can be used for power transmission is the highest frequency f1 (Hz) that satisfies the relationship of Rs ≧ Rw, and the highest frequency f2 (Hz) that satisfies the relationship of Rs ≧ Rn ≧ Rw. Although it can obtain | require, the lower limit of the frequency which can use a coil is demonstrated. The performance as a coil is represented by Q. The Q of the coil varies depending on the frequency. When an alternating current is applied to a single coil, the frequency at which the phase difference θ between the current flowing through the coil and the voltage is 80 degrees or less is the lower limit frequency at which the coil can be used. FIG. 14 clearly shows the phase angle of the coil 10 </ b> A, and 10 kHz is a lower limit for using the coil 10 </ b> A.

  The relationship between Q and θ is tan θ = Q, and tan 80 degrees = 5.67. Therefore, it is sufficient that the coil Q has a frequency equal to or higher than about 5.5 or more. The Q of the coil becomes Q = ωL / Rw when the symbols described above are used. Therefore, by reducing Rw and securing the inductance, it can be used in a wide frequency range. At the lower limit frequency at which the coil can be used, Rw is almost the direct current resistance of the coil, so it is important to ensure the inductance. Even if the coil has a high f1 and can be used up to a high frequency, if the inductance is small, the frequency at which Q is 5.5 or less is also high. A method for reducing the DC resistance of the coil and securing the inductance will be described later. As described above, the present invention can realize a power transmission device with good power transmission performance by selecting a coil with good power transmission performance that could not be selected by the conventional technology and setting the frequency range in which the coil can be used. It has an excellent effect.

(Description of Coil of Other Embodiment: Embodiment 2)
FIG. 29 is a diagram showing a coil 10b of a power transmission device according to another embodiment of the present invention. The foil conductor coil 10a shown in FIG. 2 has a plate-like member 20 and a foil-like conductor 30 formed in a square shape, whereas the foil conductor coil 10b shown in FIG. 29 has a disk-like plate-like member 20a. The foil conductor 30a is formed in a circular spiral on the top. Conditions such as the thickness and width of the foil-like conductor 30a are the same as those described in FIGS.

  The foil conductor coil according to the present invention is not limited to the square foil conductor coil 10a shown in FIG. 2 or the circular foil conductor coil 10b shown in FIG. 29, but various shapes such as an elliptical shape and a polygonal shape. Can be formed. Moreover, in the coil 10a shown in FIG. 2, the center part of the square side which is AA cross section of FIG. 2 can be expanded or narrowed. The effect can change the magnetic flux direction and intensity | strength on the said coil surface similarly to embodiment mentioned later.

(Description of Umbrella Coil: Embodiment 3)
In FIG. 30, the plate-like member 20b is formed in an umbrella shape in cross section, and the foil-like conductor 30b is formed on the plate-like member 20b to constitute the foil conductor coil 10c. In FIG. 30, the angle formed by the two lines d2 and d3 in contact with the upper surface of the plate-like member 20b is preferably set between 180 degrees and 90 degrees. In order to make the plate-shaped member 20b an umbrella-shaped cross section, the whole can be formed into a conical shape or a pyramid shape.

(Description of other configuration of coil: Embodiment 4)
FIG. 31 is a diagram showing a coil 10d of a power transmission device according to still another embodiment of the present invention. In this embodiment, in order to sparsely wind the inner peripheral part side of the foil-like conductor 30b and to dense the outer peripheral part side, each interval becomes wider toward the inner peripheral part side and narrower toward the outer peripheral side. In this way, the foil-like conductor 30b is formed on the plate-like member 20. That is, the adjacent foil conductors 30b and 30b in the inner peripheral portion are wound with a relatively wide interval w1, whereas the adjacent foil conductors 30b and 30b in the outer peripheral portion are The coils are sequentially approached and wound with a small interval w2 on the outermost periphery. The interval w1 ≧ 0.5 mm is satisfied, and w2 ≧ w1 is satisfied. The interval w2 is selected to be twice or more the width H of the foil conductor.

  From the description in paragraph No. 0028 of Patent Document 2, the flat spiral coil has a high magnetic flux density in the central part and a low magnetic flux density in the outer peripheral part, so that a blank is provided in the inner peripheral part and the inner peripheral part is made sparse. By winding and winding the outer peripheral portion densely, the magnetic flux density on the coil surface can be made uniform.

  FIG. 32 shows a coil of a power transmission device according to another embodiment of the present invention, and is an enlarged view showing a cross section of a portion corresponding to line AA in FIG. The embodiment shown in FIG. 32 is configured such that the foil-like conductor 30 is perpendicular to the winding surface of the coil as it goes to the outer peripheral portion. 8e and 8f of Patent Document 7 disclose a method in which the long side of the copper foil is perpendicular to the winding surface as it goes to the outer peripheral portion. This makes it difficult to operate at high frequencies. This embodiment is characterized in that a dielectric (insulator) does not exist between adjacent or opposing foil conductors, and there is no resin or the like having a high dielectric constant on the upper surface of the foil conductor 30, and only an air layer is present. It is configured not to exist.

  In the example shown in FIG. 32A, the outer peripheral portion is inclined from the flat surface 22 of the plate-shaped member 21 to form the inclined surface 23, and the tip portion is adjacent to the inclined surface 23 with respect to the flat surface 23. A right-angle surface 24 that is a right angle is formed. Then, the foil conductors 30 are formed on the flat surface 22, the inclined surface 23, and the right-angled surface 24, respectively. Instead of forming the inclined surface 23, a right-angle surface 24 that rises at a right angle from the outermost periphery of the flat surface 23 may be formed.

  In the example shown in FIG. 32 (B), a spiral groove 26 is formed in the plate-like member 21, and among the opposing side walls of each groove 26, the same outer side walls facing the inner peripheral side are respectively the same as each other. An inclined surface 27 having an angle is formed, and a foil-like conductor 30 is formed on the inclined surface 27.

  In the example shown in FIG. 32C, a spiral groove 29 is formed in the plate-like member 21, and the outer peripheral side of the side walls that are substantially perpendicular to the bottom surfaces of the plurality of grooves 29 are opposed to each other. A foil conductor 30 is formed on the wall 31.

  In the example shown in FIG. 32 (D), the inclined surface 33 with an inclination angle that gradually rises from the flat surface 32 toward the outer peripheral portion, and the surface facing the inside of the groove 34 has a steep angle with respect to the flat surface 32. The inclined surface 35 and the side wall surface 37 whose surface facing the inside of the groove 36 at the outermost periphery is a right angle are formed on the plate member 21 so as to face the inner peripheral side. A foil-like conductor 30 is formed on each of the inclined surfaces 33 and 35 and the side wall surface 37.

  In the example shown in FIG. 32E, the angle of the inclined surface 40 on the center side is gentle and the inclination angle of the inclined surfaces 41 to 44 becomes steeper toward the outer periphery, and the side wall surface 45 whose outermost periphery is a right angle. It is formed so as to face the inner peripheral side. A foil-like conductor 30 is formed on each surface 41 to 45.

  33 is a diagram showing the direction and strength of the magnetic flux on the coil surface of FIG. 3, which is a cross-sectional view of the coil 10a shown in FIG. In FIG. 33, the direction of the magnetic flux is represented by an angle with respect to the flat surface 22, and the strength of the magnetic flux is represented by the length of a double arrow line. Further, FIG. 33 shows the load resistance of the flat plate spiral coil 20a of FIG. 4 to which the non-inductive load resistance is connected is perpendicularly opposed to the AA cross section of FIG. 3 at various angles. The angle perpendicular to the magnetic flux supplementing surface Sm where the voltage at both ends is maximum is taken as the direction of the magnetic flux, and the voltage across the load resistance is shown as the strength of the magnetic flux. As is clear from FIG. 33, a magnetic flux is generated in a direction close to the vertical direction with respect to the flat surface 22 in the central portion of the coil 10a, and in a direction close to parallel to the flat surface 22 in the outer peripheral portion of the coil 10a. It can be seen that magnetic flux is generated.

  FIG. 34 shows a coil of a power transmission device according to another embodiment of the present invention, and is an enlarged view showing one of the foil conductors present in the cross section of the portion corresponding to line AA in FIG. In FIG. 34A, a part of the foil-like conductor 30 is in contact with the flat part 22, and a part has an angle α with respect to the flat surface 22. In FIG. 34B, a part of the foil conductor is in contact with the flat portion 22, the central portion has an angle α with respect to the flat surface 22, and the tip portion has an angle β with respect to the flat surface 22. In this case, α> β. On the other hand, FIG. 34C shows a case where α <β. Further, FIG. 34D shows a case where β is larger than 180 degrees. FIG. 34E shows a case where the foil conductor 30 is not entirely in contact with the flat portion 22 and has a portion having an angle α with respect to the flat surface 22 and an angle β. Further, the foil-like conductor 30 may have a circular arc shape as shown in FIG. 34 (F), and the arc bending direction is as shown in FIG. 34 (G). It may be in the opposite direction. As shown in FIG. 34 (H), a part of the arc may be in contact with the flat portion 22 as in FIG. 34 (A), or the curve constituting the arc may be an ellipse, hyperbola, parabola, etc. It may be part of an arbitrary curve. As described above, the method of winding the foil conductor with an angle with respect to the flat surface 22 is not limited to that illustrated in FIG. 34, and there are various embodiments.

  FIG. 35 is a diagram showing the magnetic flux Φ generated by the current flowing through the conductor. From FIG. 35, the magnetic flux Φ generated by the current flowing through the conductor generates a circular magnetic flux in the conducting wire 51 having a circular cross section as shown in FIG. 35A, but as shown in FIG. When the cross section of the conductor wire 52 is rectangular, it is assumed that the conductor wire 52 has an oval or elliptical shape having a major axis in the direction perpendicular to the width H. As shown in FIG. As the ratio of H / t increases, the major axis of the ellipse seems to be longer. When the magnetic flux Φ generated by each foil conductor constituting a coil wound several turns is synthesized, it is considered that the direction and strength of the magnetic flux as shown in FIG. 33 are obtained. Therefore, a part or all of the foil-shaped conductor forming the coil 10a is wound around the flat surface 22 while changing the width thereof, or a part or all of the foil-shaped conductor is angled with respect to the flat surface 22. It is possible to change the magnetic flux intensity and direction of FIG. For example, with the configuration as shown in FIG. 32A, the magnetic flux intensity in the vertical direction of the flat surface 22 can be increased at the outer periphery of the coil, and the entire surface on the coil is perpendicular to the flat surface 22. The magnetic flux intensity in the direction can be made close to uniform. In the coil having the configuration as shown in FIG. 2, when the plane of the power receiving coil wound in a flat spiral is parallel to the flat surface 22, the power required by the power receiving coil is obtained at any part on the coil surface. Transmission is possible.

(Description of other configuration of coil: Embodiment 5)
In FIGS. 32A to 32E described above, the embodiment in which a part or all of the foil conductor is wound at an angle with respect to the winding surface of the foil conductor is shown. There are a wide variety of embodiments, such as the vertical positional relationship with the winding surface of the foil-like conductor wound at an angle. As shown in FIG. 34, a part or all of the foil conductor is wound at an angle with respect to the winding surface of the foil conductor, thereby changing the magnetic flux direction and magnetic flux intensity shown in FIG. A coil of a power transmission device having various performances such as non-direction and single direction can be realized.

  As described above, FIG. 33 shows the intensity and direction of the magnetic flux with respect to the line AA in FIG. 2, but the intensity and direction of the magnetic flux changes except for the line AA in FIG. Therefore, the angle between the flat surface 22 and the foil-like conductor 30 is determined by adopting various embodiments as shown in FIG. 34 with respect to the winding surface of FIG. realizable. This effect is the same in other embodiments described later, and when the shape of the coil is not circular, the embodiment shown in FIG. 34 can realize a power transmission coil having the required performance.

  36 is a view showing a power transmission coil 10e according to another embodiment of the present invention, and FIG. 37 is a cross-sectional view taken along line BB in FIG.

  A power transmission coil 10e shown in FIG. 36 is formed such that a portion indicated by a dotted line in the foil-like conductor 30 of the power transmission coil 10a shown in FIG. The power transmission coil 10a shown in FIG. 2 can spread magnetic flux over a wide range, and can transmit power over a long distance, parallel to the plane of the power reception coil wound in a flat spiral, the power transmission coil Lp, and the flat surface 22. Even if not, power can be transmitted.

  Therefore, as shown in FIG. 34, the required performance can be realized by winding the foil conductor so as to have an angle with the flat surface 22. The foil-like conductor that forms an angle with the flat surface 22 may be a part as shown in FIG. 36 instead of one turn. In addition to the cross-sectional view of FIG. 34, various embodiments exist. However, a winding method in which an insulator 30a having a high dielectric constant exists between the foil conductors 30 as shown in FIG. 38 is not preferable.

(Description of other configuration of coil: Embodiment 6)
FIG. 39 is a diagram showing a coil 10f of a power transmission device according to still another embodiment of the present invention.

  In FIG. 39, the width H of the part 55 of the foil conductor forming the coil 10f is wider than the width H of the other part 56 of the foil conductor. In Patent Document 2, the effect of reducing the DC resistance of a coil formed by winding a foil-like conductor is intended, and in order to make the resistance value of one turn the same, the conductive foil width H of the outer peripheral portion is increased. However, the thicker the conductive foil, the more effective the magnetic flux generated by the conductive foil spreads in the direction perpendicular to the width H. Therefore, by thickening the outer periphery, the magnetic flux on the coil surface is made uniform. In addition, the power transmission distance can be increased.

  Further, in FIG. 39, the DC resistance can be reduced by making the thickness t of the foil conductor of the other portion 56 thicker than the thickness t of the foil conductor of the portion 55 of the foil conductor.

  FIG. 40 is a diagram showing a coil 10g of a power transmission device according to still another embodiment of the present invention. 40, the foil conductor coil 10g has a plurality of strip-like foil-like conductors 61, 61,... Arranged in parallel on a plate-like member 60, and one end of the innermost circumferential foil-like conductor 61. One end of the conducting wire 62 is connected to the other end, the other end of the conducting wire 62 is connected to the other end of the adjacent foil-like conductor 61, and the conducting wire 62 is sequentially connected to the foil-like conductor 61 arranged in parallel to be wound in a spiral shape. It is a turn. The conditions such as the cross-sectional shape of the foil conductor 61 are the same as in the description of FIG.

  As the conducting wire 62, a conducting wire (vinyl wire) obtained by collecting a plurality of conducting wires and applying an insulation coating, or a twisted conducting wire (Litz wire) obtained by twisting a plurality of conducting wires provided with an insulating coating is used. An increase in the effective resistance of the coil can be suppressed. Further, by using a lead wire twisted with the vinyl wire or the litz wire as the lead wire 62, the magnetic flux can be canceled and the magnetic flux density on the coil surface can be made uniform. Further, as indicated by the double-headed arrow described in FIG. 33, a magnetic flux is formed in a direction perpendicular to the foil-like conductor and upward, so that the magnetic flux supplement surface Sm of the flat spiral power receiving coil 20a shown in FIG. 2 can transmit power when the relative position of the coil 10a shown in FIG. 2 is specific.

  FIG. 41 is a diagram showing a coil 10h of a power transmission device according to still another embodiment of the present invention. In this embodiment, a plurality of foil conductors 63 are formed in an inverted U shape, a plurality of conductors 64 are formed in a U shape, and the foil conductors 63 and the conductors 64 are combined to form a spiral shape. The foil conductor coil 10h is configured.

  FIG. 42 is a diagram showing a coil 10i of a power transmission device according to still another embodiment of the present invention. In this embodiment, the outer peripheral side of the foil conductor coil 10 a shown in FIG. 2 is constituted by a foil-like conductor 65, one end of a conductor 66 is connected to the inner circumference of the foil-like conductor 65, and the inner circumference is a conductor 66. A spiral foil conductor coil 10i is configured.

  In the embodiment of FIGS. 40, 41 and 42, the same effect as that of the embodiment of FIG. 39 can be obtained.

(Description of other configuration of coil: Embodiment 7)
FIG. 43 is a diagram showing a coil 10j of the power transmission device according to still another embodiment of the present invention. The coil 10j of this embodiment is a foil conductor 67 that divides the foil conductor 66 corresponding to the foil conductor 30 on the inner peripheral side of the foil conductor coil 10a shown in FIG. Formed. As described above, as the width H of the foil conductor is wider, the magnetic flux formed by the current flowing through the foil conductor spreads in the direction perpendicular to the width H. Place two or more foil-shaped conductors in parallel at locations where you want to reduce the DC resistance of the coil but want to suppress the spread of the magnetic flux upward, and connect them with solder 68 etc. at the corners and one turn at a time. To do. These have the same effects as the embodiment in which the width of a part of the foil-like conductor forming the coil of FIG. 39 is changed.

(Description of other configuration of coil: Embodiment 8)
The embodiment of FIGS. 39 to 43 described above has an effect of changing the direction in which the magnetic flux spreads and the intensity and direction of the magnetic flux on the coil surface, and thus any one of the conductors forming the power transmission coil is used. One embodiment may be used, and a plurality of embodiments may be used in combination.

(Description of other configuration of coil: Embodiment 9)
FIG. 44 is a diagram showing a coil 10k of a power transmission device according to still another embodiment of the present invention. In this embodiment, the foil conductors 30 of the foil conductor coil 10a shown in FIG. 2 are partially dense and sparse. That is, the foil conductors 30c and 30d are spirally formed on the insulating member 2 in the same manner as the foil conductor coil 10a shown in FIG. 2, but in FIG. 44, between the foil conductors 30c in the horizontal portion. The sparsely wound portion 71 is formed by widening the interval, and the beeswound portion 72 is configured by narrowing the interval between the foil-like conductors 30d in the vertical portion.

  That is, the power receiving coil that receives power from the coil 10k has short sides and long sides having different lengths. In the case where the area of the power receiving coil is approximately ¼ or less of the area of the power transmitting coil, the number of foil conductors intersecting the short side wound in the direction perpendicular to the short side of the power receiving coil, and the long side The foil conductors constituting the power transmission coil are wound so that the value of the number of the foil conductors intersecting with the long film wound in the perpendicular direction is close.

  The thus configured foil conductor coil 10k is used as a power transmission coil Lp, and a power receiving coil 73 indicated by a one-dot chain line is disposed opposite to the foil conductor coil 10k in the loosely wound portion 71 of FIG. Alternatively, a power receiving coil 74 indicated by a one-dot chain line is arranged opposite to the bead winding portion 72. The power receiving coil 73 is formed in a rectangular shape having a minimum outer dimension of M1 on the short side and a minimum outer dimension of M2 on the long side. In the case of this embodiment, the power receiving coil 74 can be obtained because the power receiving coil 74 has a larger number of foil-like conductors of the power transmitting coil that enters the winding surface cross-sectional area of the power receiving coil or the magnetic flux supplement surface than the power receiving coil 73. The received power is characterized by being almost uniform at any position on the surface of the power transmission coil.

  FIG. 45 is a diagram illustrating a correspondence relationship between the power transmission coil and the power reception coil in the embodiment illustrated in FIG. 44. Although the power transmission coil 10k shown in FIG. 44 is formed in a square shape, when the power transmission coil 75 is formed in a rectangular shape as shown in FIG. 45, the area is S1, and the short side of the rectangle is b1. . On the other hand, the power receiving coil 76 is rectangular and has a smaller area S2 than the power transmitting coil 75, and b1 is selected to be larger than b2 when the short side of the rectangle is b2. Yes. The power receiving coil 76 is disposed so as to face the portion of the power transmitting coil 75 having the largest number of foil conductors.

  That is, more specifically, as shown in FIG. 44, the power receiving coil 73 is opposed to the loosely wound portion 71 of the power transmitting coil 10k, and the power receiving coil 74 is opposed to the honeycomb wound portion 72, Each of them is arranged so as to face a portion where the foil conductor 30b is large.

With such a winding method, a blank portion in which the foil conductor is not wound is generated at the center of the power transmission coil 10k. Therefore, when the power receiving coil 73 is disposed at the center of the power transmitting coil 10k, the number of foil conductors of the power transmitting coil 10k disposed in the magnetic flux supplement surface of the power receiving coil 73 is reduced. However, the central portion of the power transmission coil 10k has a high magnetic flux density, and the power reception coil can output a sufficient electromotive force due to the mutual induction action. When the power receiving coil 73 moves from the central portion of the power transmitting coil 10k to the outer peripheral portion, the number of foil conductors of the power transmitting coil 10k located in the magnetic flux supplement surface Sm of the power receiving coil 73 increases. Thereby, the electromotive force of a receiving coil is securable.

  Thus, when the power receiving coils 73 and 74 have a shape other than a circle or a regular polygon, a magnetic flux generated by a plurality of portions facing the power receiving coils 73 and 74 in the foil conductor 30b of the power transmitting coil 10k. Are the same. Therefore, by increasing the number of foil-shaped conductors 30b per area in the power transmission coil 10k facing the power reception coils 73 and 74, necessary power is sent to the power reception coils 73 and 74 over the entire surface of the power transmission coil 10k. be able to.

  The power receiving device including the power receiving coil 73 and the power receiving coil 74 is an embodiment of the power receiving device corresponding to the power transmitting device including the foil conductor coil 10k.

  More specifically, for example, a cordless mouse that does not require a battery can be realized by mounting the power receiving coil 73 on the casing to form a cordless mouse and mounting the foil conductor coil 10k on the mouse pad. Since the number of foil conductors forming the foil conductor coil 10k included in the area of the power receiving coil 73 is increased regardless of the position of the foil conductor coil 10k, the cordless mouse operates. Can be ensured at any position of the foil conductor coil 10k.

  Usually, a mouse has a vertically long shape. Therefore, the shape of the power reception coil built in the mouse is also vertically long. Currently, a cordless mouse that transmits power from a power transmission coil built into a commercially available mouse pad to a power reception coil built into the mouse has a power transmission coil composed of a conductive wire that is wound around the end face of the mouse pad. Yes. Therefore, the shape of the mouse pad has to be vertically long. By adopting this method, the horizontal dimension of the mouse pad can be made longer than the vertical dimension.

  In addition, the mouse pad of the said system currently marketed cannot be used on a metal desk. The foil conductor coil of the present invention can also be used on a metal desk. Details thereof will be described later.

  When the power transmitting coil 10k and the power receiving coils 73 and 74 are the same size or foil-like conductors having similar areas, high efficiency and high power transmission is not impossible, but it has been experimentally confirmed that it is difficult. ing. However, when a coil composed of a foil-like conductor having an area of about 1/5 or less comes close to a large-area power transmission coil and the coupling coefficient increases, the power transmission side is affected.

(Description of other configuration of coil: Embodiment 10)
FIG. 46 is a diagram showing an example in which a metal frame is brought close to the coil of the power transmission device according to the embodiment of the present invention. A power transmission coil 75 is disposed on a part of the upper surface of the metal frame 80. 46A shows an example in which the coil 10a is arranged on the upper surface of the metal frame 80, and FIG. 46B shows the coil 10a in the metal frame 80 so that the coil 10a and the metal frame 80 are parallel to each other. An example of arrangement is shown. 46A, when the magnetic flux supplementing surface Sm of the flat plate spiral receiving coil 20a shown in FIG. 3 is parallel to the flat surface of the metal frame 80, Not only that, but at least inside the metal frame 80, power can be transmitted up to about 20 cm above the flat surface of the metal frame 80.

  Furthermore, as shown in FIG. 46 (B), when the magnetic flux supplement surface Sm of the power receiving coil 20a is on the same plane as the flat surface of the metal frame 80, power transmission is possible up to a distance of 1 m or more from the coil 10a. It becomes. Thus, the metal frame 80 has an effect of extending the power transmission distance beyond the area of the coil 10a. Specifically, the size of the metal frame 80 is set to 60 cm × 2 m using a B4 size coil 10a. In this case, on the same plane as the flat surface of the metal frame 80, electric power can be transmitted to the power receiving coil that is 2 m away from the power transmitting coil within the metal frame.

  Alternatively, by configuring the metal frame as shown in FIG. 46B, electric power can be transmitted to the magnetic flux supplement surface of the power receiving coil 20a up to a plane perpendicular to the flat surface of the coil 10a.

  As the material of the metal frame 80, it is preferable to use a ferromagnetic metal such as iron. Further, the metal frame 80 may have a flat plate shape, a square pipe shape, an L angle shape, or the like. For example, a frame equipped with a passage sensor used in current automatic ticket gates can be used as the metal frame 80. Within this automatic ticket gate, power of about 50 mW can be transmitted to the coil built in the IC card. In order to secure an electromotive force, it is necessary to increase the number of windings of the power receiving coil, but if a conductive wire having a diameter of about 30 μm is wound about 50 turns, there is no problem in volume.

  Thus, the transmission distance of the received power of about 50 mW can be extended to 1 m or more with the transmitted power of about 10 W. The power of about 50 mW is sufficient for operating an IC card or the like currently used. As described above, this embodiment has an extremely excellent effect that cannot be realized by the conventional technique.

(Coil configuration to ensure the inductance of the foil conductor coil)
FIG. 47 is a diagram showing the structure of the coil 10m for ensuring the inductance. In FIG. 47, at least one magnetic material plate 91 as a shielding plate is provided on one side facing the coil 90. Thus, by providing at least one magnetic material plate 91 on one surface side of the coil 90, the inductance of the carp 90 can be ensured. Such a configuration of the coil 10m is different from the case of providing a magnetic material plate on both surfaces of the foil conductor coil as in Patent Document 2, and the inductance increases only by about 20%. However, the effective series resistance of the foil conductor coil hardly increases. By providing a foil conductor coil having a wide foil width H and a wide gap w with a magnetic material plate as shown in FIG. 47, inductance can be ensured. Thereby, the Q of the coil can be secured at a low frequency, and the frequency range in which the coil can be used is expanded.

  As described above, in the coil 10m, in the coil 90, the effective series resistance of the power transmission coil alone in the air core state is Rwa (Ω), and the power reception coil facing the power transmission coil in the air core state is short-circuited. When the effective series resistance of the power transmission coil is Rsa (Ω), a power transmission coil that satisfies Rsa> Rwa is used at 100 kHz. For example, the coil 90 uses the coils 10a to 10k according to the embodiment of the present invention described above.

  Furthermore, when the coil 10m equipped with the magnetic material plate 91 is a power transmission coil, it is preferable that the coil 10m satisfies Rsa> Rwa at 100 kHz.

  Assuming that the maximum frequency at which the coil 10m satisfies Rsa> Rwa is f1 (Hz), the power transmission device 100 equipped with the coil 10m transmits power at a frequency of f1 (Hz) or less.

  When the coil 10m is the power transmission coil 1 of the power transmission device 100, the coil 10m is driven at a frequency fa (Hz) equal to or less than f1 (Hz) by the AC power source 3b illustrated in FIG.

  The output frequency fa (Hz) of the AC power supply 3b included in the power transmission unit 3 of the power transmission device 100 is set to a frequency of f1 (Hz) or less.

  When the power transmission unit 3 of the power transmission device 100 includes the coil 10m, the power transmission unit 3 including the coil 10m is a power transmission device of the power transmission device of the present invention.

(About other coil configurations to ensure inductance)
FIG. 48 is a diagram showing a coil 10 n in which an insulating plate 92 is provided between the coil 90 and the magnetic material plate 91. In FIG. 48, the insulating plate 92 can suppress an increase in the effective series resistance Rwa (Ω) of the coil 10n when the frequency is increased. Further, it is possible to prevent the Q of the coil 10n from being lowered when the frequency is increased.

  FIG. 49 is a diagram showing a coil 10p in which two layers of a magnetic material plate 911 and a magnetic material plate 912 are provided on one surface side of the coil 90. FIG. In FIG. 49, the coil 10p is provided with the magnetic material plate 911 and the magnetic material plate 912, so that the inductance of the coil 90 can be increased and the Q of the coil can be increased as compared with the configuration of the coil 10m.

  FIG. 50 is a diagram showing a coil 10q in which an insulating plate 92 having a thickness of I (mm) including air is provided between two magnetic material plates 911 and 912 constituting the coil 10p shown in FIG. In FIG. 50, the insulating plate 92 can suppress an increase in the effective series resistance Rwa (Ω) of the coil 10q when the frequency is increased. Further, it is possible to prevent the Q of the coil 10q from being lowered when the frequency is increased. Such a configuration increases the thickness of the coil but is suitable for use in a power transmission coil. In particular, the power transmission coil converts electrical energy into magnetic energy, and it is preferable to use the power transmission coil to ensure inductance. The thickness I (mm) of the insulating plate 92 is preferably half or more than the thickness of the magnetic material plate 911 or 912. The effect of the insulating plate 92 is as described above.

(Regarding the coil configuration that prevents fluctuations in coil characteristics when a metal object is in close proximity)
FIG. 51 is a diagram illustrating the structure of the coil 10r that prevents the fluctuation of the characteristics of the coil 90 when a metal body approaches the coil 90. FIG. In FIG. 51, a metal plate 95 has a thickness M (mm), and an insulating plate 92 is provided on one side of the coil 90 so as to face the coil 90 via a predetermined distance G (mm). The dimension of the metal plate 95 is equal to or greater than the dimension of the coil 90 and is disposed so as to face the entire surface of the coil 90. For the coil 90, for example, the coils 10a to 10k according to the embodiment of the present invention described above are used. The predetermined distance G (mm) is the same as the thickness of the insulating plate 92, and the predetermined distance G is selected to be 10 mm or 10% or more of the coil outer diameter D. Since the characteristic variation of the coil 90 is small, the longer the predetermined distance G (mm) is, the better. When the power transmission coil is equipped with the metal plate 95 as shown in FIG. 51, a coil capable of preventing characteristic fluctuations when another metal body is in proximity to the coil 90 can be realized.

(About the coil configuration that prevents the proximity effect of the metal body and increases the inductance)
FIG. 52 is a diagram showing the configuration of the coil 10s that prevents the proximity effect of the metal body and increases the inductance. 52, in FIG. 51, a magnetic material plate 91 is used instead of a predetermined distance G (mm) provided between the insulating plate 92 and the metal plate 95 disposed on the opposite side of the surface facing the coil 90. Is provided. For the coil 90, for example, the coils 10a to 10k according to the embodiment of the present invention described above are used. The metal plate 95 is a metal or alloy having diamagnetic or paramagnetic magnetic properties, and the thickness is selected to be 0.1 mm or more. The metal plate 95 has the same dimensions as the magnetic material plate 91.

  FIG. 53 is a diagram showing a configuration of a coil 10t that prevents the proximity effect of the metal body and increases the inductance according to another embodiment of the present invention. In FIG. 53, the insulating plate 92 is provided on the surface facing the coil 90, and the magnetic material plate 91 and the metal plate 95 are provided on the opposite side surface of the insulating plate 92 facing the coil 90. The effect of the insulating plate 92 is as described above. The metal plate 95 is a metal or alloy having diamagnetic or paramagnetic magnetic properties, and the thickness is selected to be 0.1 mm or more.

  FIG. 54 is a diagram showing a configuration of a coil 10u that prevents the proximity effect of the metal body and increases the inductance according to another embodiment of the present invention. In FIG. 54, the insulating plate 92 is provided between the magnetic material plate 91 and the metal plate 95. The effect of the insulating plate 92 is as described above. The metal plate 95 is a metal or alloy having diamagnetic or paramagnetic magnetic properties, and the thickness is selected to be 0.1 mm or more.

  FIG. 55 is a diagram showing a configuration of a coil 10v that prevents the proximity effect of the metal body and increases the inductance according to another embodiment of the present invention. In FIG. 55, a magnetic material plate 911 and a magnetic material plate 912 are provided on one side of the coil 90. By providing the two magnetic material plates 911 and 912, the inductance of the coil 90 can be increased and the Q of the coil 90 can be increased. Further, by providing the two magnetic material plates 911 and 912, fluctuations in coil characteristics due to the type and thickness of the metal plate 95 provided on the magnetic material plate 912 side can be reduced. Therefore, the metal plate 95 can have any magnetic property and thickness different from the coil 10r to the coil 10 described above. More preferably, the metal plate 95 is a metal or alloy having diamagnetic or paramagnetic magnetic properties and having a thickness of 0.01 mm or more. For the coil 90, for example, the coils 10a to 10k according to the embodiment of the present invention described above are used.

  FIG. 56 is a diagram showing a configuration of a coil 10w that prevents the proximity effect of the metal body and increases the inductance according to another embodiment of the present invention. In FIG. 56, a magnetic material plate 911 is provided on one side of the coil 90. An insulating plate 92 is provided between the magnetic material plate 911 and the magnetic material plate 912, and a metal plate 95 is provided on the magnetic material plate 912 side. The effect of the insulating plate 92 is as described above. The metal plate 95 has the same dimensions as the magnetic material plates 911 and 912. The thickness I (mm) of the insulating plate 92 is selected similarly to the coil 10q.

  FIG. 57 is a diagram showing a configuration of a coil 10x that prevents the proximity effect of the metal body and increases the inductance according to another embodiment of the present invention. In FIG. 57, the insulating plate 92 is provided between the magnetic material plate 912 and the metal plate 95. The effect of the insulating plate 92 is as described above. The metal plate 95 has the same dimensions as the magnetic material plate 912.

  FIG. 58 is a diagram showing a configuration of a coil 10y that prevents the proximity effect of the metal body and increases the inductance according to another embodiment of the present invention. In FIG. 58, the insulating plate 92 is provided on one side of the coil 90. The insulating plate 92 is provided with magnetic material plates 911 and 912 and a metal plate 95. The effect of the insulating plate 92 is as described above. The metal plate 95 has the same dimensions as the magnetic material plates 911 and 912.

  Even when three or more magnetic material plates are used, the place where the insulating material is provided is between the plurality of magnetic material plates 911 and 912, between the coil 90 and the magnetic material plate 91, as shown in FIG. There are various embodiments, such as between the plate 91 and the metal plate 95. The effect of the insulating plate 92 is as described above.

  As described above, in the coils 10m to 10y, the coil 90 is short-circuited to the coil 90 with the effective series resistance of the power transmission coil alone in the air-core state being Rwa (Ω) and the power-receiving coil facing the power transmission coil in the air-core state. When the effective series resistance of the power transmission coil is Rsa (Ω), a power transmission coil satisfying Rsa> Rwa is used at 100 kHz. For example, the coil 90 uses the coils 10a to 10k according to the embodiment of the present invention described above.

  Furthermore, when the coil 10y from the coil 10m equipped with either the magnetic material plate 91 or 911, 912 and the metal plate 95 is used as the power transmission coil, the coil 10m to the coil 10y satisfies Rsa> Rwa at 100 kHz. It is preferable.

  Assuming that the maximum frequency satisfying Rsa> Rwa from coil 10m to coil 10y is f1 (Hz), power transmission device 100 equipped with coil 1y from coil 1m transmits power at a frequency of f1 (Hz) or less. .

  When the coil 10m to the coil 10y are the power transmission coil 1 of the power transmission device 100, the coil 10m to the coil 10y are driven by the AC power source 3b shown in FIG. 1 at a frequency fa (Hz) equal to or less than f1 (Hz). The

  The output frequency fa (Hz) of the AC power supply 3b included in the power transmission unit 3 of the power transmission device 100 is set to a frequency of f1 (Hz) or less.

  Furthermore, when the coil 10m to the coil 10y are the power receiving coil 2 of the power transmission device 100, the frequency fj (Hz) at which the coil 10y receives power from the coil 10m is f1 (Hz) or less. .

  When the power transmission unit 3 of the power transmission device 100 includes the coil 10m to the coil 10y, the power transmission unit 3 including the coil 10m to the coil 10y becomes the power transmission device of the power transmission device according to the embodiment of the present invention.

  Detailed operational effects of the coils 10m to 10y, which are coils equipped with at least one of the metal plate 95 and the magnetic material plates 91, 911, and 912 described above, will be described in detail later.

(Characteristics of coil 10m to coil 10y)
FIG. 59 shows the relationship between the effective series resistance Rw (Ω) in each coil when the coil 10F shown in FIG. 21 is used as the coil 10m shown in FIG. 47 and the coil 10n shown in FIG. FIG.

  FIG. 60 is a diagram showing the relationship between the Q frequency of each coil when the coil 10F is used as each of the coils 10m and 10n.

  As is clear from FIG. 59, when the coil 10n having the coil 90 equipped with the insulating plate 92 and the magnetic material plate 91 and the coil 10m having the coil 90 equipped with the magnetic material plate 91 are compared, for example, the effective series resistance at 1 MHz. Rw (Ω) is lower in the coil 10n. As described above, by mounting the insulating plate 92 on the coil 90, the effective series resistance Rw (Ω) of the coil in the high frequency region can be lowered. This tendency is the same even when the coil 10q in which the coil 90 is equipped with the magnetic material plate 911, the insulating plate 92, and the magnetic material plate 912 is compared with the coil 10p in which the coil 90 is equipped with the magnetic material plates 911 and 912. .

  Further, as is apparent from FIG. 59, the coil 10n having the coil 90 equipped with the insulating plate 92 and the magnetic material plate 91 has a coil Q at 1 MHz having a Q higher than that of the coil 10m equipped with only the magnetic material plate 91, for example. high. As is apparent from FIGS. 59 and 60, even if the insulating plate 92 is provided in the coil 90, there is almost no difference in effective series resistance Rw (Ω) and Q at a frequency of 100 kHz. Thus, by providing the coil 90 with the insulating plate 92, the Q of the coil in the high frequency region can be increased. The effect of the insulating plate 92 is the same in the coil 10s to the coil 10y equipped with the magnetic material plate 91 or 911, 912 described above. Therefore, the description regarding the insulating plate 92 is omitted for the coils 10s to 10 and 10w to 10y.

(Explanation when various metal plates face the coil)
FIG. 61 is a characteristic diagram showing the effective series resistance Rw (Ω) of the single coil 10F and the inductance Lw (μH) of the single coil 10F when various metal plates are closely opposed using the coil 10F. .

  As shown in FIG. 61, the coil of the configuration (1) represents the effective series resistance Rw (Ω) and the inductance Lw (μH) of the coil 10F alone. The coil of the configuration (2) is left blank because it is used when a magnetic material plate described later is provided. The coil of the configuration (3) is in a state where an aluminum (Al) foil having a thickness of 12 μm is brought close to the coil 10F. The coil of the configuration (4) is in a state in which an aluminum plate having a thickness of 0.1 mm is brought close to the coil 10F. The coil of the configuration (5) is in a state in which an aluminum plate having a thickness of 0.5 mm is brought close to the coil 10F. The coil of the configuration (6) is in a state in which an aluminum plate having a thickness of 3 mm is brought close to the coil 10F. The coil having the configuration (7) is in a state in which a copper foil (Cu) having a thickness of 35 μm is brought close to the coil 10F. The coil of the configuration (8) is in a state in which a copper plate having a thickness of 0.1 mm is brought close to the coil 10F. The coil of the configuration (9) is in a state in which a copper plate having a thickness of 0.5 mm is brought close to the coil 10F. The coil of the configuration (10) is in a state in which an iron plate (Fe) having a thickness of 0.5 mm is brought close to the coil 10F. In FIG. 61, the effective series resistance Rw (Ω) and the inductance Lw (μH) at 100 kHz of the coils of the configurations (3) to (10) are compared with the characteristics of the coil 10F alone of the configuration (1). It is shown as a bar graph.

(Construction and characteristics of the coil excluding the influence of proximity to the metal plate)
FIG. 62 is a characteristic diagram showing the effective series resistance Rw (Ω) of the coil 10Fa alone and the inductance Lw (μH) of the coil 10Fa alone when various metal plates are brought close to the coil 10F with an interval of 10 mm. It is. In other configurations, the metal plate provided in the coil 10Fa is the same as described above. Coil 10Fa is an embodiment of coil 10r shown in FIG.

  FIG. 63 shows the effective series resistance Rw (Ω) of the coil 10Fb alone and the inductance Lw (μH) of the coil 10Fb alone when various metal plates are brought close to the coil 10F on which the single magnetic material plate 91 is provided. FIG. The coil of the configuration (2) is in a state in which one magnetic material plate 91 is provided in the coil 10Fb and no metal plate is brought close to the coil 10Fb. In other configurations, the metal plate provided in the coil 10F is the same as described above. Coil 10Fb is an embodiment of coil 10m shown in FIG.

  FIG. 64 shows the effective series resistance Rw (Ω) of the coil 10 Fc alone and the coil 10 F alone when various metal plates are brought close to the coil 10 Fc in which the two magnetic material plates 911 and 912 are provided on the coil 10 F. It is a characteristic view which shows inductance Lw ((micro | micron | mu) H). The coil of the configuration (2) is in a state where the coil 10Fc is equipped with two magnetic material plates 911 and 912 and the metal plates are not brought close to each other. In other configurations, the metal plate provided in the coil 10Fc is the same as that described above. The coil 10Fc is an embodiment of the coil 1v shown in FIG.

  61 to 64, the frequency characteristics of the effective series resistance Rw (Ω) of the coil 10F alone shown in FIG. 20 is referred to, and the measurement is performed by selecting 100 kHz where the effective series resistance Rw (Ω) is substantially equal to the DC resistance of the coil 10F. It is.

  First, the characteristic diagram shown in FIG. 61 will be examined. FIG. 61 shows that the effective series resistance Rw is about 1.2Ω and the inductance Lw is about 13 μH in the configuration of the coil 10F alone. In the coil 10F having the configuration (3) in which 12 μm aluminum foils are closely opposed to each other, the effective series resistance Rw is 3Ω or more, and the inductance Lw is reduced to about 4 μH. When the characteristic diagrams of the configuration (4) to the configuration (6) in which the aluminum plate of various thicknesses which are paramagnetic metals are opposed to the coil 10F are seen, when the aluminum thickness is 0.1 mm or more, the thickness is As the value increases, the effective series resistance Rw (Ω) of the coil decreases and the inductance Lw (μH) increases. This tendency is the same in the coils of configurations (8) and (9) in which copper, which is a diamagnetic metal, is opposed to the coil 10E. When the copper plate is thin, the effective series resistance Rw (Ω) of the coil decreases and the inductance Lw (μH) also decreases. When the thickness of the copper plate is about 0.5 mm, the effective series resistance Rw (Ω) of the coil has characteristics that are not significantly different from the 0.5 mm aluminum plate of the configuration (5). When a 0.5 mm iron plate, which is a ferromagnetic metal, is placed close to the coil 10F, the effective series resistance Rw (Ω) is the same as when the aluminum foil in the configuration (3) is placed close to the coil 10F. Is 2.5 times or more, and the inductance Lw is reduced to about 5.5 μH. That is, except for the ferromagnetic metal, the coil characteristics are not changed due to the magnetic properties of the metal as described in Patent Document 4, but the metal plate that is in close proximity to the coil wound in a plane air-core spiral shape. Coil characteristics vary depending on the thickness of the coil.

  In the coils of configurations (3) to (10) shown in FIG. 61, the effective series resistance Rw (Ω) is excessive and the inductance Lw (μH) is excessive compared to the air-core state. Therefore, the coils of configurations (3) to (10) shown in FIG. 61 cannot actually be used as coils of the power transmission device. FIG. 61 shows data to be compared with FIGS. 62 to 65 shown below. In FIG. 61, when the aluminum foil having the configuration (3), the copper foil having the configuration (7), and the iron plate having the configuration (10) are provided, the increase rate of the effective series resistance is large. However, in the conductive wire coil, the increase rate of the effective series resistance in the configuration (3), the configuration (7), and the configuration (10) is much larger than that in FIG. In the foil conductor coil, the increase rate of the effective series resistance in the configurations (3), (7), and (10) is small. These will be described later. The increase rate of the effective series resistance Rw (Ω) of the 35 μm-thick copper foil of the configuration (7) is higher than the increase rate of the effective series resistance Rw (Ω) of the aluminum foil of the configuration (3) having a thickness of 12 μm. The reason for the small number is probably due to the thickness.

  FIG. 62 shows the characteristics of the above-described coil 10Fa in which the various metal plates shown in FIG. 61 are opposed to the coil 10F via a 10 mm insulator as the coil 10r shown in FIG. As is clear from FIG. 61, in FIG. 62, the increase rate of the effective series resistance Rw (Ω) and the decrease rate of the inductance Lw (μH) are also small. It can be seen that the characteristics of the coil 10Fa are greatly improved by using an insulator of 10 mm. In particular, the value of the inductance Lw has decreased only to about 10 μH compared to about 13.7 μH in the air-core state. The value of Lw (μH) is almost the same in each configuration, and can be used for a power transmission device. However, when the 12 μm thick aluminum foil of the configuration (3), the 35 μm thick copper foil of the configuration (7), and the 0.5 mm iron plate as the ferromagnetic material of the configuration (10) face the coil 10F, the coil 10F Since the increase rate of the effective series resistance Rw (Ω) is large and power loss due to Rw (Ω) occurs, such a configuration is not suitable for use in a power transmission device.

  That is, referring to FIG. 62, a metal or alloy having means for providing a certain predetermined distance G (mm) between the coil and the metal and having a diamagnetic or paramagnetic magnetic property of 35 μm or more is provided as the metal plate. By using this, it is possible to eliminate the influence of a metal body close to the air-core coil. In the configurations other than the configurations (1) and (2) shown in FIG. 62, various metals including a ferromagnetic metal such as iron are brought close to the opposite side of the metal plate to the coil facing surface. As for the change in μH), the effective series resistance Rw (Ω) slightly changes in the configuration (3), but is not observed at all in the other configurations. In addition, there is no change in power transmission performance.

  In paragraph No. 0022 of Patent Document 4, it is described that a metal plate having a size smaller than that of the magnetic field type antenna (coil) may be used. However, in order to eliminate the influence of the proximity of the metal body of the coil formed by winding the conducting wire in a plane spiral shape, the above-mentioned predetermined distance G (mm) is provided, and the thickness is 0. It must be equipped with a metal or alloy plate of the same size as the coil that is 1 mm or more.

  Note that paragraph number 0022 of Patent Document 4 describes that the metal plate is divided. When the inventor of the present application divided a 0.1 mm copper foil and measured the characteristics at 100 kHz with the coil having the configuration (8) using the coil 10 Fa, Lw = 13.1 μH, Rw = 1, 31. Ω. This configuration has better characteristics than Lw = 12.5 μH and Rw = 1.33Ω when the copper plate is not divided. As described above, this is presumably because the eddy current loss that increases in proportion to the volume of the metal body decreases. This is also described in paragraph No. 0022 of Patent Document 4. However, in the configuration in which the 0.1 mm copper foil is divided and installed, when a 0.5 mm thick iron plate is brought close to the opposite surface of the copper plate coil, the Lw decreases to 12.3 μH and the Rw becomes 1. Increased to 38Ω. If the copper plate is not divided, the inductance Lw (μH) is smaller than the divided copper plate, but the effective series resistance Rw (Ω) is small. There was no change in both Lw (μH) and Rw (Ω) even when they were close to each other. Therefore, in an embodiment in which the metal plate described in paragraph No. 0022 of Patent Document 4 is divided and an embodiment in which the dimension of the metal plate is made smaller than the dimension of the coil, it is described in Paragraph No. 0022 of Patent Document 4. The effect of eliminating the fluctuation of the coil characteristics when the metal body is close to the coil is not expected.

  As described above, in the embodiment of the coil 10r shown in FIG. 51, by providing a means for making the distance between the coil and the metal plate constant between the coil and the metal plate, another metal body comes close to the back surface of the metal plate. However, fluctuations in the inductance Lw and effective series resistance Rw of the coil 10r can be eliminated. A coil 10r shown in FIG. 51 is suitable for a power transmission unit in which metal plates can be installed at regular intervals on the back surface of the coil. When the power transmission unit is placed on a steel desk, fluctuations in the inductance Lw and effective series resistance Rw of the coil 10r can be eliminated, and predetermined power transmission performance can be maintained. The predetermined distance G is 10 mm in the coil 10F, and good results are obtained. However, the predetermined distance G (mm) varies depending on the outer diameter D of the coil. Unlike the conductive coil, the foil conductor coil is not easily affected by the proximity of the metal plate. Since the outer diameter D of the coil 10F is 100 mm, the predetermined distance G (mm) is determined by, for example, G ≧ D / 50 = 2 mm with a margin. The foil conductor coil has less characteristic variation due to the proximity of the metal plate than the conductive coil. In the foil conductor coil, it is preferable to obtain the predetermined distance G based on the foil width H, the foil thickness t, and the like.

  The above-described commercially available cordless mouse uses a conductive coil. As described above, the conductive wire coil is easily affected by the metal body. Use a foil conductor coil and a mouse pad. By configuring the coils 10r to 10x as shown in FIGS. 51 to 57, it is possible to realize a mouse pad for power transmission that has no change in power transmission performance on a metal desk or a non-metal desk. . The foil conductor winding method shown in FIG. 44 is adopted. This makes it possible to transmit power necessary for the mouse to operate regardless of where the mouse is placed on the mouse pad, and to realize a mouse pad that can be used on a metal desk or a non-metal desk.

  Next, consider FIG. 63, which is a characteristic diagram of the embodiment of the coil 10s shown in FIG. The configuration (2) in FIG. 63 shows the effective series resistance Rw (Ω) and the inductance Lw (μH) of the coil 10Eb alone in which a magnetic material plate having a thickness of 0.3 mm is attached to the coil 10F. In the coil 10Fb alone, the inductance Lw (μH) is increased and the effective series resistance Rw (Ω) is hardly changed compared to the coil 10F alone in the air-core state. The effective series resistance Rw (Ω) and the inductance Lw (μH) of each coil of the configuration (3) to the configuration (10) in which the same metal plate as that of FIGS. 61 and 62 is provided on the opposite surface of the coil 10Fb are shown. ing. In the configurations (3) to (10), the value of the inductance Lw is almost the same. However, as is apparent from FIG. 63, as in FIG. 62, a configuration (3) equipped with a 12 μm thick aluminum foil, a configuration (7) equipped with a 35 μm thick copper foil, 0 The coil of the configuration (10) equipped with a steel plate having a thickness of 0.5 mm has a high effective series resistance Rw (Ω).

  Furthermore, the characteristics of each component of the coil 10u having the configuration shown in FIG. 54, which is another embodiment of the present invention, will be examined with reference to FIG. The coil 10u having the configuration shown in FIG. 54 is equipped with two magnetic material plates 911 and 912. The two magnetic material plates 911 and 912 are preferably stacked with an insulating layer. Referring to FIG. 64, the characteristics of the coil 10Fc alone in which the coil 10FF is provided with two magnetic material plates are shown in the configuration (2), and the inductance Lw is about 20 μH, which is about 20 μH compared to about 13 μH in the air-core state. 1.6 times. Compared to FIG. 63, in all of the configurations (3) to (10), the inductance Lw exceeds 18 μH. Further, in all of the configurations (3) to (10), the value of the inductance Lw (μH) is substantially the same. Compared with FIG. 63, the 12 μm-thick aluminum foil of the configuration (3), the 35 μm-thick copper foil of the configuration (7), and the 0.5 mm iron plate as the ferromagnetic material of the configuration (10) are used as the coil 10Fc. Even when facing each other, there is a feature that there is almost no change in the effective series resistance Rw (Ω). That is, by mounting two magnetic material plates on top of each other, the coil is not affected by the magnetic properties or thickness of the metal provided on the opposite side of the coil facing surface of the magnetic material plate. With the configuration like the coil 1u having the configuration shown in FIG. 54, the above-described proximity effect of the metal body can be eliminated with a thin metal such as an aluminum foil.

  In addition, as the magnetic material plate, a thickness of 0.01 mm to 1.5 mm and various configurations such as a magnetic material powder solidified with a binder, an amorphous type, and a ferrite type were tested. The characteristics due to the difference in the metal plates were the same as those in FIGS. 57 to 59 described above. Further, in the configuration of the coil 1m and the coil 1v, when the increase in inductance was large, the increase in effective series resistance was also large. At 100 kHz, the increase rate of the inductance and the increase rate of the effective series resistance were almost equal for any magnetic material plate. As will be described later, these actual measurement results indicate that this configuration rule has generality.

  In the embodiment in which the power receiving coil 73 is equipped to form a cordless mouse and the foil conductor coil 10k is a mouse pad, the mouse pad is insulated by configuring the foil conductor coil 10k as shown in FIGS. The power transmission performance can be sent to the cordless mouse without changing the power transmission performance regardless of whether it is installed on a desk or metal desk.

(Explanation that the power transmission coil configuration rules are general)
As described above, a coil with good performance cannot be realized simply by defining a specific configuration of the coil. However, the specific configuration of the coil in the present embodiment exhibits the same effects regardless of the coil type, winding method, outer diameter, and the like. That is, the specific configuration of the coil in the present embodiment has an effect of suppressing fluctuations in coil characteristics when a metal body is close to the back surface of the coil regardless of the wire type, winding method, outer diameter, and the like of the coil. An example is shown below.

  FIG. 65 is a characteristic diagram in which the effective series resistance Rw (Ω) and the inductance Lw (μH) are measured for the coil 10F shown in FIG. 20 in the same configuration as the coil 10r shown in FIG. . In FIG. 65, the interval between the coil 10F and the metal plate is set to 5 mm and measured.

  FIG. 66 is a characteristic diagram in which the effective series resistance Rw (Ω) and the inductance Lw (μH) are measured for the coil 10E shown in FIG. 21 in the same configuration as the coil 10r shown in FIG. 51, as in FIG. . In FIG. 66, the interval between the coil 10E and the metal plate is set to 10 mm and measured.

  When comparing FIG. 62, FIG. 65, and FIG. 66, in the coil 10r having the configuration shown in FIG. 51, the metal plate 95 is configured using an aluminum foil as an example (3), and is configured using a copper foil as an example (7). ), It can be seen that the effective series resistance Rw (Ω) of each of the coils of the example (10) configured using the iron plate is increased as compared with the coils of other configurations.

  This tendency is the same for both the coil 10E and the coil 10F in the configuration of the coil 10s shown in FIG. 52 and the configuration of the coil 10u shown in FIG. That is, when only a predetermined distance G (mm) is provided between the coil and the metal plate, when one magnetic material plate is provided between the coil 90 and the metal plate 95, the metal plate 95 is configured using an aluminum foil. In the example (3), the example (7) configured using the copper foil, and the example (10) configured using the iron plate, the effective series resistance Rw (Ω) is higher than that of the coil having the other configuration. To increase. When two or more magnetic plates are provided between the coil 90 and the metal plate 95, there is almost no increase in the effective series resistance Rw (Ω) as shown in FIG. 63 depending on the type and thickness of the metal plate 95.

  The effect of the coil 10r having the configuration shown in FIG. 51 is to prevent fluctuations in coil characteristics when a metal body is close to the opposite side of the coil facing surface, as described in Patent Document 4. The operational effects of the coil 10s having the configuration shown in FIG. 52 and the coil 10v having the configuration shown in FIG. 55 also prevent fluctuations in coil characteristics when a metal body is close to the opposite side of the opposing surface of the coil. As another effect of the coil 10s having the configuration shown in FIG. 52 and the coil 10v having the configuration shown in FIG. In order to ensure the inductance, the configuration of the coil 10v shown in FIG. 55 is preferable. As described above, the coil 10v is hardly affected by the material and thickness of the metal plate. Therefore, even if it is the coil 10p shown in FIG. 49, the proximity | contact effect of a metal body can be excluded.

  In FIGS. 62 to 64, data of the coil 10t, the coil 10u, the coil 10v, the coil 10w, the coil 10x, and the coil 10y that are formed of the metal plate, the magnetic material plate, and the insulating plate are omitted. This is because the effect of the metal plate 95 is the same as shown in FIGS. As shown in FIGS. 59 and 60, the coil having such a configuration has an effect of suppressing the increase of the effective series resistance Rw (Ω) in the high frequency region and increasing the Q. In actual measurement, in the high frequency region, the effective series resistance Rw (Ω) of each coil described above decreases compared to the coils 10s and 10v, and the Q of each coil increases compared to the coils 10s and 10v. Has been confirmed to do.

  Note that the coil 10Ea, which is an example of the coil 10r shown in FIG. 51, has a maximum frequency f1 satisfying Rs> Rw of 10 MHz or more and a maximum frequency f2 satisfying Rs> Rn ≧ Rw of 2 MHz or more. Since there is no influence of the magnetic material plate and the effects other than the prevention of the change of the coil characteristics due to the proximity of the metal body are the same as those of the coils 1A to 10E, the description about the interval between the opposing coils is omitted.

  Needless to say, the coil 90 from the coil 10m to the coil 10y has an effective series resistance Rw (Ω) of the power transmission coil alone in the air core state, and a power receiving coil facing the power transmission coil in the air core state is short-circuited. When the effective series resistance of the power transmission coil is Rs (Ω), a power transmission coil that satisfies Rs> Rw at 100 kHz is used. For example, as the coil 90, the coils 10A to 10D according to the embodiment of the present invention described above are used.

  Furthermore, when the coil 10s equipped with the magnetic material plate 91 or 511, 912 is used as the power transmission coil, it is preferable that the coil 10s to the coil 10y satisfy Rs> Rw at 100 kHz.

  Assuming that the maximum frequency satisfying Rs ≧ Rw from coil 10m to coil 10y is f1 (Hz), power transmission device 100 equipped with coil 10m to coil 10y transmits power at a frequency of f1 (Hz) or less. .

  When the coil 10m to the coil 10y are the power transmission coil 1 of the power transmission apparatus 100, the coil 10m to the coil 10y are fd (Hz) having a frequency equal to or lower than f1 (Hz) by the AC power source 3b illustrated in FIG. Driven.

  The output frequency fa (Hz) of the AC power supply 3b included in the power transmission unit 3 of the power transmission device 100 is set to a frequency of f1 (Hz) or less.

  Furthermore, when the coil 10s and the coil 10v of the embodiment described above are used as a power transmission coil, the same coil is used as a power reception coil, the inductance of a single power transmission coil is Lwa (H), and both coils are inductively coupled. Assuming that the inductance of the power transmission coil when short-circuited is Lsa (H), the coil 10s and the coil 10v satisfy Lwa> Lsa at 100 kHz and have a frequency of f1 (Hz) or less. It is preferable that Lwa> Lsa is satisfied at the frequency used for power transmission.

  The maximum frequency f2a (Hz) satisfying Rna (Ω) and Rsa> Rna ≧ Rwa as the effective series resistance of each coil when the coil 10y to the coil 10y described above are used as the power transmission coil and the opposite power receiving coil is opened. ), The power transmission coil is preferably used in a frequency region of f2a (Hz) or less.

  When the coil 10s and the coil 10v described above are used as a power transmission coil and the inductance of one coil when the opposing power receiving coil is opened is Ln, the relationship of at least Ln> Lw> Ls is 100 kHz. Satisfactory, and in the frequency region below f1 (Hz), the coil 10s and the coil 10v satisfy the relationship Ln> Lw> Ls at the frequency fa (Hz) used for power transmission. Is preferred.

  More preferably, when the coil 10m to the coil 10y have a maximum frequency f2 (Hz) satisfying Rs> Rn ≧ Rw, the coil 10m to the coil 10y are used in a frequency region of f2 (Hz) or less, and f2 ( Hz), the following relationship is satisfied: Ln> Lw> Ls.

  The above-mentioned definition of the thermal condition can be satisfied by obtaining the thermal resistance θi by the same method as that described above.

  When the power transmission unit 3 of the power transmission device 100 includes the coil 10m to the coil 10y, the power transmission unit 3 including the coil 10m to the coil 10y becomes the power transmission device of the power transmission device of the present invention.

(Explanation when connecting the metal plate to the coil center wire)
67 is a diagram in which a metal plate mounted on a coil is used as a wire to be taken out from the inner periphery of the coil in the coil 10r shown in FIG. 51 to the coil 10y shown in FIG.

  In the coil 90 having the configuration of the coil 10m to the coil 10y, the conducting wire taken out from the center of the coil 90 is thicker by the thickness of the conducting wire. 67, a conductive wire through hole is provided at the center of the insulating plate 92 of the coil 10r shown in FIG. 51, and the conductive wire 901 at the inner periphery of the coil is connected to the metal plate 95. An end 902 of the outer peripheral portion of the coil 90 is used as one terminal, and an end 951 of the metal plate 95 is used as the other terminal. As a method for connecting the conductive wire 901 and the metal plate 95, various methods such as soldering and welding can be used. This configuration is applicable not only to an insulating plate but also to a magnetic material plate.

(Example in which a coupling wire is provided in a coil)
FIG. 68 is a diagram of the coil when a thin foil-like conductor is wound between the foil conductors of the coils shown in FIGS. 47 to 58 and the winding line is taken out as a coupling wire 30d. In this example, the foil conductor 30a is wound in the circular spiral shape shown in FIG. 29. However, the foil conductor 30 is wound in the square shape shown in FIG. Also good.

  68 can be used to detect the operating state of the coil. Alternatively, it can be used for signal transmission. In addition, self-excited oscillation can be performed by applying positive feedback using an inverting amplifier. In particular, the power transmission unit of the power transmission device according to the present invention is a simple series resonance circuit, and by using a self-excited oscillation circuit, the frequency at which the power transmission coil is automatically driven is adjusted to an optimum value.

  The coupling line 30d is in a substantially tightly coupled state with the power transmission coil 1. The winding ratio is 1: 1. Therefore, an AC voltage having the same amplitude and phase as that of the power transmission coil 1 appears in the coupling line 30d. This coupling line 30d is a signal transmission function between a power transmission unit and a power reception unit in other embodiments of the present invention described later, detection when a metal body is close to the power transmission coil, and when a power reception coil to which a load is connected is close Can be used for discrimination.

  When the embodiment of FIG. 68 described above is applied, the foil conductors in the inner peripheral portion are collectively connected to the metal plate 95, and the foil conductor taken out from the outer peripheral portion is divided into the power transmission and the coupling line 30a. Alternatively, the coupling wire 30a may be taken out without being connected to the common line and may be a four-terminal coil. In this case, since the power transmission coil and the coupling line are insulated, the degree of freedom in circuit configuration increases when used for signal detection, self-excited oscillation, and the like.

(Example of coil for power transmission)
A coil 10m shown in FIG. 47 to a coil 10y shown in FIG. 58 are also examples of the power transmission coil used in the power transmission device described above.

(Power transmission coil drive conditions)
In addition, even if it is an air-core coil or equipped with a magnetic material plate, the above-described f1 (Hz) and f2 (Hz) are increased so that power is transmitted at f1 (Hz) or lower than f2 (Hz). A power transmission device with good performance cannot be realized unless the driving conditions of the coil for driving the coil are defined.

(Embodiment 15)
Preferably, a receiving coil having a foil conductor wound in a spiral shape is provided on the back surface of the insulating material, and Sc is a total area of the foil conductors on the winding surface of the power transmission coil and the power receiving coil, and a gap between the winding surfaces of the power transmission coil and the power receiving coil If Si is the total area, satisfying Si> Sc, the effective series resistance of the power transmission coil when the power reception coil is opened is Rn (Ω), and the effective series of the power transmission coil when the power reception coil is short-circuited When the maximum frequency satisfying Rs (Ω) and Rs ≧ Rn is f3 (Hz), it is used in the frequency region below f3.

  This configuration can also be used as a transformer in which the power transmission coil and the power reception coil cannot be separated. In this configuration, the configuration using the magnetic material described above can be applied. When used as a transformer, magnetic material plates are provided on both sides of the foil conductor coil. As described above, it is preferable that an increase in effective series resistance can be reduced by providing an insulating plate between the foil conductor coil and the magnetic material plate.

(Explanation regarding metals used in the present invention)
In the embodiment of the present invention, the material of the conductor forming the conducting wire is not particularly limited, but all the coils described in this embodiment use copper as the conductor. Although copper having a small specific resistance is preferably used as the conductor, other metals or alloys having a small specific resistance can also be used as the conductor.

  In addition to diamagnetism, paramagnetism, and ferromagnetism, the magnetic properties of metals include antiferromagnetism. However, attention is focused on the magnetic property that increases the effective series resistance Rw (Ω) of the coil. The metal plate mounted on the coil may be anything other than a metal or alloy that is simply attracted to the permanent magnet.

  The inventor of the present application made measurements using various metals such as titanium, brass, and stainless steel. The 0.1 mm thick titanium plate showed the same characteristic variation as the 0.1 mm thick aluminum plate. A brass plate with a thickness of 0.5 mm showed the same characteristic variation as a copper plate with a thickness of 0.5 mm. The stainless steel plate adsorbed on the 0.5 mm thick permanent magnet caused characteristic deterioration more than the 0.5 mm thick iron plate. Thus, the metal plate may be selected depending on whether it is attracted to the permanent magnet or not.

(Measurement device used for measuring coil characteristics)
The effective series resistance and inductance of each coil described above were measured up to 1 MHz using an Agilent LCR meter, and 4284A, 1 to 10 MHz were measured using a Hewlett Packard LCR meter, 4275A. . In addition, since the measurement of 1 to 10 MHz can be performed only at each point of 1, 2, 4, and 10 MHz, for example, when Rs ≧ Rw is satisfied at 4 MHz and Rs> Rw is not satisfied at 10 MHz. Estimates the maximum frequency f1 (Hz) satisfying Rs ≧ Rw by interpolation.

  FIG. 69 is a diagram illustrating a circuit configuration that is an example of an embodiment of a power transmission device according to the present invention. The power transmission device 100d includes a power transmission unit 3d for transmitting power and a power reception unit 4d for receiving power transmitted from the power transmission unit 3d.

  69 includes a power transmission coil Lp, a capacitor Cp, an AC power supply V, a power receiving coil Lr, a capacitor Cm, and a load RL. The power transmission device 100d illustrated in FIG. 69 includes an AC power supply V and a power transmission coil Lp. A capacitor Cp is connected in series between them. The power receiving coil Lr receives power from the power transmitting coil Lp in an arbitrary shape and supplies the power to the load RL. The capacitor Cp is connected in series with the power transmission coil Lp in order to cancel the reactance of the power transmission coil Lp. The output frequency fa (Hz) of the AC power supply V is set close to the point where the reactance determined by the power transmission coil Lp and the capacitor Cp becomes zero or the point where the impedance becomes minimum.

  In the power receiving unit 4d shown in FIG. 69, the capacitor Cm is connected in parallel to the power receiving coil Lr. The frequency at which the susceptance determined by the power receiving coil Lr and the capacitor Cm is zero or the frequency at which the impedance is maximized is set to be the same as the output frequency of the AC power supply V. Thereby, a voltage larger than that when the capacitor Cr is not provided at both ends of the power receiving coil Lr is generated by the parallel resonance action, so that the voltage required by the load can be obtained on the power receiving side.

  The power receiving unit 4d illustrated in FIG. 69 can be used when a voltage is required rather than a current to operate the load RL. In particular, the present invention is applicable to transmitting power over a long distance to a power receiving side device equipped with a load having a high forward voltage and a low operating current, such as a blue LED.

  FIG. 70 is a diagram illustrating a circuit configuration which is an example of another embodiment of the power transmission device according to the present invention. The power transmission device 100e includes a power transmission unit 3e for transmitting power and a power reception unit 4e for receiving power transmitted from the power transmission unit 3e.

  The power transmission device 100e shown in FIG. 70 has a circuit configuration that is used when a current rather than a voltage is required to operate the load RL. The capacitor Cn is connected in series to the power receiving coil Lr, and the frequency at which the reactance determined by the power receiving coil Lr and the capacitor Cn becomes zero or the frequency at which the impedance is minimized is set to be the same as the output frequency of the AC power supply V. Due to the resonance action, the impedances at both ends of the series circuit of the power receiving coil Lr and the capacitor Cn are reduced, and a larger current can be supplied to the load RL than when the capacitor Cn is not provided. Therefore, the current required by the load RL can be obtained on the power receiving side. The power receiving unit 4e shown in FIG. 70 can be applied to transmit power over a long distance to a power receiving side device equipped with a load that requires a starting current even when the operating voltage of the motor or the like is low.

  In the embodiment shown in FIGS. 69 and 70, it is not necessary to use the capacitors Cm and Cn provided on the power receiving side that are used for the power transmission side capacitor Cp defined in the present invention, and the load RL is in an open state. As long as it is not short-circuited, a generally used capacitor may be used. In the embodiment of FIG. 69, when the load RL is in an open state, a high voltage is generated across the capacitor Cm, which may exceed the operable voltage of the capacitor Cm. In FIG. This is because a high voltage is generated at both ends of the capacitor Cn and may exceed the operable voltage of the capacitor Cn.

  71 is a diagram illustrating a circuit configuration on the power receiving side of FIG. 69 or 70. In addition, since the power transmission coil Lp in which the capacitor Cp is connected in series is driven by the AC power source V, the power receiving side circuit of FIG. 69 includes diodes D21 to D24 whose ON time is about several nS as shown in FIG. It is preferable to equip a rectifier circuit with four bridges. 69 and 70, the power transmitting coil Lp and the power receiving coil Lr are close to each other, and the magnetic flux supplement surface Sn of the cylindrical coil 20b illustrated in FIG. 5 and the flat surface 22 are parallel to each other. Even when the coupling coefficient is high, it is preferable that the power receiving side circuit is equipped with a rectifier circuit having four diodes D21 to D24 with ON times on the order of several nS as bridges, and provided with a smoothing capacitor Cd. Dz is a voltage limiting Zener diode for preventing damage to the capacitor Cd, the diodes D21 to D24, the load RL and the like due to an excessive voltage.

  The coil 10a as an example of the embodiment shown in FIG. 2 described above can be used as a power receiving coil for a load that requires a high voltage, and the coil 10b shown in FIG. 29 receives a power for a load that requires a current. Can be used for coils. However, it is necessary to select a receiving coil with an appropriate configuration depending on the load conditions.

(Embodiment of power transmission device)
With reference to FIG. 69, the AC power source, the power transmission coil using the foil conductor, and the power transmission units 3d and 3e equipped with the capacitor to the power reception units 4d and 4e with a transmission distance of several tens of cm or more, A power transmission device capable of transmitting power can be realized.

  72 is a diagram showing an example of an AC power supply 3b included in the power transmission control circuit 3a shown in FIG. In FIG. 72, AC power supply 3b includes a control circuit 3c and switching elements Q1 and Q2. The control circuit 3c is supplied with a DC voltage Vd (V) from the DC power supply 12, and the control circuit 3c alternately applies control signals to the gates of the switching elements Q1 and Q2. A DC voltage Vd (V) is supplied to the drain of the switching element Q1, and the source of the switching element Q1 and the drain of the switching element Q2 are commonly connected to one electrode of the capacitor C1. The source of the switching element Q2 is connected to GND (ground).

  The control circuit 3c alternately turns on the switching elements Q1 and Q2, and supplies a square wave that is a non-sinusoidal wave to the power transmission coil 1 via the capacitor C1. This square wave is a waveform having a duty of 50% and an amplitude Vd (V), the level of which changes between Vd (V) and GND. Since the capacitor C1 and the power transmission coil 1 constitute an LC series resonance circuit such as the equivalent circuit of FIG. 125 including the power reception unit not shown in FIG. 1, the capacitor C1 and the power transmission coil 1 resonate based on the square wave signal. Then, a sine wave alternating current with an amplitude of VL (V) is transmitted from the power transmission coil 1 to the power reception coil 2 shown in FIG. 1 and supplied to the load RL. As will be described later, the amplitude VL (V) is boosted several to several tens of times higher than the amplitude Vd (V).

(Description of LC series resonance circuit)
FIG. 73 is a circuit diagram for measuring the characteristics of the LC series resonance circuit. In FIG. 73, a reference coil Ls, a measurement capacitor Cx, and a resistor R2 are connected between the AC power output terminal of the AC power supply 3b and GND. An oscilloscope 85 is connected to both ends of the measurement capacitor Cx, and the voltage across the capacitor Cx is measured. R2 is a resistance of about 0.1Ω and measures the AC current flowing through the LC series resonance circuit by measuring the voltage across R2, and the AC power supply 3b has a reactance Xc of the capacitor Cx to be evaluated and a reference coil The output frequency is set so that reactance Xi of Ls alone is equal.

  74 is an equivalent circuit diagram including a pure resistance component (effective series resistance) of each element constituting the circuit for measuring the characteristics of the LC series resonance circuit of FIG. In FIG. 74, Rc (Ω) is the effective series resistance of the capacitor C, Ri is the effective series resistance of the power transmission coil 1, and Rm (Ω) is the effective series resistance Ri of the coil 1 and the effective series resistance Rc of the capacitor C. The added value (Ri + Rc) (Ω), and the output impedance Zs (Ω) indicates the output impedance of the AC power supply 3b.

(Description of capacitor)
First, the inventor of the present application uses the same AC power source 3b, which is a component other than the capacitor, and the power transmission coil 1, the power reception coil 2, and the load RL in the circuit of FIG. I tried the test. As a result, it has been found that even when capacitors having the same capacitance are used, the power transmission performance differs depending on the capacitor dielectric and configuration. Moreover, even if it was the capacitor comprised by the same dielectric material, it discovered that electric power transmission performance differed with the structure of a capacitor. Furthermore, even if the power transmission coil 1 is changed and the configuration is exactly the same as the dielectric of the capacitor, it has been found that the power transmission performance varies depending on the capacitance.

  Therefore, the present inventor prepared 17 types of capacitors C1a to C1s of 0.01 μF, and first measured the characteristics of the capacitors with an LCR meter. C1a and C1b are polystyrene (PS), C1c and C1g are polypropylene (PP), C1d is polycarbonate (PC), C1e and C1f are polyphenylene sulfide (PPS), C1j and C1q are polyethylene (PE), C1h, C1m, C1r, C1s is made of ceramic (CE), C1i, C1k, C1n, and C1p are made of polyethylene terephthalate (PET) as dielectrics.

  Table 2 shows the characteristics of these capacitors rearranged in descending order of the effective series resistance Rc at 100 kHz. In addition, about each capacitor shown in Table 2, about 5 each is measured, an average value is calculated | required, and the characteristic value of the capacitor close | similar to an average value is described. However, capacitors that do not serve as basic data, such as capacitors whose characteristics cannot be measured, capacitors whose measurement values are not reproducible, and capacitors whose power transmission performance is extremely poor, are excluded from Table 2. Further, the above-described polystyrene capacitors that cannot be used due to heat generation and polypropylene capacitors that have extremely poor power transmission performance are also excluded from Table 2.

In Table 2, C represents the capacitance of the capacitor, and its unit is nF . Rc represents the effective series resistance of the capacitor at each frequency, Xc represents the reactance of the capacitor at each frequency, and the unit is Ω. Q represents the Q of the capacitor at each frequency and is unitless. tan δ represents the dielectric loss tangent of the capacitor at each frequency and is unitless.

  In addition, although not described in Table 2, the power transmission performance is also measured for a capacitor using polyethylene naphthalate (PEN) as a dielectric. The power transmission performance of the polyethylene naphthalate capacitor will be described later.

  Regarding the frequency characteristics of these capacitors, usable frequency regions will be described later with examples. Using Table 2 as basic data, the characteristics of the capacitor will be discussed. First, the characteristics of the capacitor related to the embodiment of the present invention defined in JIS are listed in Table 3.

(Principle of capacitor characteristic measurement)
First, the principle of measuring capacitor characteristics will be described. FIG. 75 is an example of a principle circuit diagram of a measurement circuit that measures the capacitance of a capacitor, and is a principle diagram showing an example of a capacitance measurement method in an LCR meter.

  The LCR meter having the circuit configuration of FIG. 75 includes an AC constant current source 86 and an AC voltmeter 87. In FIG. 75, Rz is a contact resistance when a probe or the like is brought into electrical contact with the terminal of the capacitor C. The capacitance of the capacitor C is measured by flowing an AC constant current I (A) from the AC constant current source 86 to the capacitor C and measuring the voltage V (V) across the capacitor C with an AC voltmeter 87. A so-called four-terminal measurement method is used so that the voltage V (V) across the capacitor C can be accurately measured. This is to eliminate the influence of the contact resistance Rz (Ω) shown in FIG. 75 when measuring a minute current or minute voltage.

  Although not shown, the LCR meter shown in FIG. 75 includes means for detecting a phase difference θ between the phase of the AC constant current source 86 and the voltage phase measured by the AC voltmeter 87. Therefore, the LCR meter shown in FIG. 75 can measure complex impedance.

Since the reactance Xc (Ω) of the capacitor is impedance Z (Ω),
Xc (Ω) = Z (Ω) = V (V) / I (A). When the angular frequency for measuring the capacitance C (F) of the capacitor C is ω, the capacitance C (F) is obtained as C (F) = Xc (Ω) / ω. Actually, Z is a complex impedance, and the phase of the current I (A) is advanced by θ (0 ° ≦ θ ≦ 90 °) from the phase of the voltage V (V). Therefore, the reactance Xc (Ω) of the capacitor is obtained as Xc (Ω) = Z · sin θ (Ω). The effective series resistance Rc (Ω) of the capacitor is obtained as Rc (Ω) = Z · cos θ (Ω). Therefore, C (F) = Xc (Ω) / ω = Z (Ω) · sin θ / ω. As apparent from the above equation, the Q of the capacitor is Q = Xc (Ω) / Rc (Ω), so that Q = (Z · sin θ / Z · cos θ) = tan θ. Q is a function of only the phase difference, θ, between the current I and the voltage V, regardless of the numerical values of the voltage V and the current I. The phase difference θ is obtained as θ = tan−1 (Q).

(Explanation of time change of capacitance)
FIG. 76 shows the fluctuation of the capacitance for 5 minutes from the measurement start time when the capacitance of the polyethylene terephthalate capacitor C1n and the ceramic capacitor C1r having poor power transmission performance is measured by the LCR meter shown in FIG. It is a graph to show.

  When the inventor of the present application measured the capacitance of a capacitor having poor power transmission performance with the LCR meter shown in FIG. 75, the capacitance display was not stable, and it took several minutes until a constant display was obtained. It cost. Also in the LC series circuit of FIGS. 72 and 73, an alternating current Ic (A) flows through the capacitor. Accordingly, it takes a considerable time from the moment when Ic flows until the capacitance of the capacitor is stabilized. This indicates that in the circuits of FIGS. 72 and 73, the frequency at which the reactance becomes zero varies with time. That is, when the polyethylene terephthalate capacitor C1n and the ceramic capacitor C1r shown in Table 2 are used in the circuits of FIGS. 72 and 73, the circuit cannot be stably operated when the current Ic (A) fluctuates. If this is not actually confirmed by the circuits shown in FIGS. 72 and 73, the capacitance is measured with a general LCR meter having a resolution of several digits, and the time until the capacitance value is stabilized is seen. It is easy to understand.

  From the actual measurement results, a capacitor that is not stable in about 5 seconds or less has poor power transmission performance. That is, it is necessary that the measured numerical value after 5 seconds from the start of measurement does not vary by ± 0.1% or more. Alternatively, it is necessary that the variation rate of capacitance is 1% or less in a measurement time of 5 minutes, and the variation rate of capacitance is 0.2% or less in a measurement time of 1 minute from the start of measurement.

  In the LCR meter having the circuit configuration of FIG. 75, as described above, a so-called four-terminal measurement method is used. Even if the capacitance is measured using such a minute current or minute voltage, the capacitance cannot be measured stably. Therefore, as shown in FIG. 73, a capacitor whose measured capacitance is not stable is not stable even in the circuits of FIGS. 69 and 70, power transmission performance is poor, and power transmission performance varies. To do.

  It is generally known that the temperature characteristics of a ferroelectric substance, for example, some ceramic capacitors, are very poor, and if the temperature is about 70 ° C., the capacitance may be about half. Such a capacitor having the characteristic that the capacitance greatly varies depending on the temperature cannot be used in the present invention. However, even a capacitor using ceramic as a dielectric has a good capacitance temperature characteristic. As a guide, the capacitance at 25 ° C. is used as a reference, and the temperature change of the capacitance between 0 ° C. and 85 ° C. is ± 5% or less. .

(Explanation of boost ratio)
Next, the operation of the basic circuit constituting the series resonant circuit shown in FIG. 74 will be described. In FIG. 74, the current I (A) flowing through the circuit becomes the maximum value Ir (A) at the frequency ω where ωL (Ω) = (1 / ωC) (Ω). This is a state where the capacitor C1 and the residual inductance Le are both short-circuited in FIG.

  Henceforth, when expressing with a resonance frequency or a resonance point, it shall refer to the frequency used as (omega) L (ohm) = (1 / omegaC) (ohm). That is, in the series resonance circuit, the reactance Xi (Ω) of the coil and the reactance Xc (Ω) of the capacitor are frequencies fr where Xi (Ω) = Xc (Ω). The inventor of the present application used the same AC power source 3b and the power transmission coil 1, prepared various capacitors, and measured the maximum value Ir (A). In FIG. 74, the power transmission coil 1 configured in a plane air-core shape is used. The output impedance Zs of the AC power supply 3b at 200 kHz is 0.2Ω. The open output voltage Vo (V) of the AC power supply 3b is fixed at an effective value of 1 V (2 Vp-p). The output impedance Zs (Ω) of the AC power supply 3b is the output voltage of the AC power supply 3b when the peak value of the open output voltage of the AC power supply 3b is Vo (V) and a non-inductive resistance of 2Ω is connected to the output of the AC power supply 3b. The peak value of V is determined as Vt (V), and Zs (Ω) = 2 (Ω) × (Vo−Vt) (V) / Vo (V).

Similarly, the effective series resistance Ri of the power transmission coil 1 alone at 200 kHz is 1.35Ω, the inductance is about 60 μH, and the reactance Xi is 80Ω. Therefore, when the effective series resistance of the capacitor is Rc (Ω), the maximum value Ir (A) of the current I (A) flowing through the circuit is
Ir (A) = Vo (V) / (Zs (Ω) + Rw (Ω) + Rc (Ω)). And the both-ends voltage Vi (V) of the power transmission coil 1 becomes Vi (V) = Xi (Ω) × Ir (A). That is, if the output impedance Zs (Ω) of the AC power supply 3b is sufficiently low and the effective series resistance Rc (Ω) of the capacitor satisfies Rc (Ω) << Ri (Ω), the power transmission coil 1 If the single Q is Qi, both ends of the power transmission coil 1 are
A voltage of Vi (V) = Vo (V) × Qi is generated. This Qi is called a boost ratio H. Actually, considering the Q and Qc of the capacitor, the Q and Qr of the series resonant circuit are
1 / Qr = 1 / Qc + 1 / Qi. When Qc >> Qi is not established, the step-up ratio H is H = Qr. As described above, at the frequency where ωL1 (Ω) = 1 / ωC1 (Ω), the reactance Xi (Ω) of the coil and the reactance Xc (Ω) of the capacitor C are equal. Therefore, the voltage Vc (V) across the capacitor becomes equal to Vi (V). The inventor of the present application uses various capacitors having a nominal value of 0.01 μF shown in Table 2, and sets the voltage Vc (V) between both ends of the capacitor and the boost ratio H and H = Vc (V) / Vo (V) described above. I measured it with an oscilloscope 85.

  FIG. 77 is a graph with the effective series resistance Rc of each capacitor at 200 kHz shown in Table 2 as the X axis and the step-up ratio H and the drive circuit current Id (mA) as the Y axis. The black dots on the characteristics shown in FIG. 77 indicate the actual measurement value H of the boost ratio and the drive circuit current ID in each capacitor. In FIG. 77, Ht is a theoretical value of the step-up ratio calculated from Qr. In addition, a black point is a measured value of each capacitor, and an intermediate point is interpolated by a graph.

  FIG. 78 is a circuit diagram in which a resistor R3 is added in series to the circuit of FIG. The action of R3 shown in FIG. 78 will be described later. The output impedance Zs (Ω) of the AC power supply 3b is as extremely low as 0.2Ω. However, in this embodiment, the effective series resistance Ri (Ω) of the coil and the effective series resistance Rc (Ω) of the capacitor are 2Ω or less, and Zs cannot be ignored. Therefore, the output voltage of the AC power source connected to the LC series circuit is set to Vt (V) so that Zs need not be considered. Vt (V) is smaller than Vo (V), but if Vt (V) is measured, the current flowing through the LC series circuit can be calculated regardless of Zs (Ω). Hereinafter, Vt (V) and Vo (V) will be properly used according to the embodiment.

From FIG. 77, it can be seen that the boost ratio H differs depending on the capacitor. At the series resonance point, the reactance component becomes zero and only the effective series resistance Rm (Ω) which is a pure resistance component. Thereby, the voltage Vi (V) generated at both ends of the coil is
Vi (V) = Ir (A) × Xi (Ω).

The voltage Vc (V) generated across the capacitor is
Vc (V) = Ir (A) × Xc (Ω).

  Since Xi (Ω) = Xc (Ω) at the series resonance point, Vi (V) = Vc (V). Since Vi (V) and Vc (V) are both 180 degrees out of phase and have the same amplitude, the voltage between point A and point B in FIG. 78 is theoretically zero. That is, the point A and the point B are in the same state as a short circuit. Actually, since there is an effective series resistance Rc (Ω) of the capacitor and an effective series resistance Ri (Ω) of the coil, a sine wave residual voltage is generated between the points A and B.

Current Ir (A) = Vo (V) / (Zs (Ω) + Ri (Ω) + Rc (Ω)),
Substituting Zs = 0.2Ω and Ri = 1.35Ω,
Zs + Ri + Rc = Rc + 1.55Ω, and if Rc≈0Ω, the circuit current Ir (A) at the resonance frequency is theoretically Ir = 1V / 1.55Ω = 666 mA.

  In the circuit of FIG. 73, R2 is a resistance for AC current measurement of 0.1Ω, and a polystyrene capacitor C1a having an effective series resistance Rc (Ω) of Table 2 of 0.01Ω is used, as shown in FIG. The maximum value of the voltage across R2 was 83 mV. Therefore, a current of peak value, 83 mV / 0.1Ω = 830 mA flows in the circuit of FIG. That is, an effective current of 830 / √2 = 586 mA is flowing, and a result almost as expected is obtained. The difference between the theoretical value 666 mA and the actually measured value 586 mA is presumed to be because the AC current measuring resistor R2 is added in series to the circuit.

The effective series resistance Rm (Ω) in the circuit of FIG. 74 is Rm = Ri + Rc = 1.55Ω. At the resonance point, the power Pr consumed by the effective series resistance Rm is
Pr = 0.586 (A) ² × 1.55 (Ω) = 0.52 W. The output voltage of the DC power supply 12 is 2 V, the output current is 0.26 A, and the output power is 2 V × 0.26 A = 0.52 W, which is a result that almost agrees with the theory.

However, some capacitors in Table 2 are 60p
Ir (A) = Vo (V) / (Zs (Ω) + Ri (Ω) + Rc (Ω)), and
The relationship Ir (A) = Vt (V) / (Ri (Ω) + Rc (Ω)) = Vt (V) / (Rm (Ω)) is not satisfied. Further, the boost ratio H does not satisfy the theoretical value of the boost ratio H obtained from the relationship 1 / Qr = 1 / Qc + 1 / Qi, where Q is Q of the series resonance circuit. That is, H ≠ Qr.

For example, the ceramic capacitor C1r in Table 3 has an effective series resistance Rc of 1.57Ω at 200 kHz. Therefore, Zs + Rw + Rc is
Zs + Rw + Rc = 0.2 + 1.35 + 1.57 = 3.12Ω.

In FIG. 74, the circuit current It at the theoretical resonance point is an effective value,
It = 1V / 3.12Ω = 320 mA.

Therefore, the theoretical power consumption Pt, Pt = I2R (W) in FIG.
Pt = I ² R = 0.32A × 0.32A × 3.12Ω = 0.32W.

On the other hand, the actual measurement current of the drive circuit of FIG. 69 is 63 mA, and the power input to the circuit of FIG.
2V × 0.063A = 0.126W. That is, theoretically, the circuit of FIG. 70 that should consume 0.32 W consumes only 0.126 W in actual measurement.

In addition, when the circuit current I in FIG.
I = √ (P / R) = √ (0.126 W / 3.12Ω) = 200 mA.

  The theoretical value It is 320 mA, but the actual measurement value I is 200 mA.

The voltage Vc across the capacitor calculated from the theoretical value is an effective value,
Vc = 75.4Ω × 0.32A = 23.17V.

  The theoretical boost ratio Ht is Ht = 23.17 / 1 = 23.17, but the actually measured boost ratio H is only about 10.8. The step-up ratios Ht and H represent the capacitor voltage.

Summarizing the above results, the ratio between theoretical and measured values is
In current, 200 mA / 320 mA = 0.625
In voltage, 10.8 / 23.17 = 0.466
In power, 126mW / 320mW = 0.394
It becomes. Both values are smaller than the theoretical values. However, as will be described later, it may be larger than the theoretical value. This is presumed that the capacitance of the capacitor is not 0.01 μF, and from Table 2, there is a maximum deviation of about ± 10%, and the resonance frequency is shifted from 200 kHz. Using the series resonance circuit of FIG. 70, the performance of the capacitor can be determined in this way.

  In FIG. 77, the theoretical boost ratio Ht = Qr is plotted. In FIG. 77, the ceramic capacitor C1h deviates from the theoretical value calculated from the effective series resistance Rc. This is a capacitor that does not match the theoretical value described above with reference to the ceramic capacitor C1r, and does not satisfy the following rules. This will be described later.

From the above experimental results, the theoretical step-up ratio Ht and the ratio of the actually measured step-up ratio H, H / Ht,
If the condition of H / Ht> 0.9 is satisfied, predetermined power transmission performance described later can be ensured.

Referring to FIG. 77, there is a capacitor where H / Ht> 1. As described above, the capacitance of these capacitors is smaller than 0.01 μF, and the resonance frequency is increased. Since the coils are the same, the reactance Xi (Ω) of the coils is increased. As a result, the reactance Xi (Ω) of the capacitor also increases. The drive circuit current IDr (A) at the resonance point is determined only by the effective series resistance Ri (Ω) of the coil and the effective series resistance Rc (Ω) of the capacitor. Therefore, since the drive circuit current IDr (A) is the same, the voltage across the capacitor,
It is assumed that Vc (V) = Xc (Ω) × IDr (A) increases.

Here, if It (A) = Vt (V) / (Ri (Ω) + Rc (Ω)), the ratio of the theoretical circuit current It to the actually measured circuit current Ir, Ir / It is
If the condition of Ir (A) / It (A)> 0.9 is satisfied, predetermined power transmission performance described later can be ensured. This is because, at the series resonance point, when the output voltage of the AC power supply 3b is Vt (V) and the voltage across the capacitor is Vc (V), the condition of Vc (V) / Vt (V)> 0.9 is satisfied. It is equivalent to being satisfied. As described above, H = Vc (V) / Vt (V). Therefore, the condition of Vc (V) / Vt (V)> 0.9 is equivalent to H> 0.9 × Qr, where H is the actually measured boost ratio and Qr is the Q of the LC series circuit. . Depending on the difference in resonance frequency, H> Ht may be satisfied. However, since normally Ht> H, it is not necessary to specify the upper limit of H in particular. From the actual measurement result of FIG. 77, when H> Ht, H was about 1.2 times Ht. Note that Ht is defined because the symbol of the boost ratio H is defined, but Ht = Qr. This is the same as the theory described above.

The ratio of theoretical circuit power Pt to measured circuit power P, P / Pt,
If the condition of 0.8 <P / Pt is satisfied, predetermined power transmission performance described later can be ensured.

  The above-described correlation between H / Ht, I / It, P / Pt, and power transmission performance will be described later with examples. Next, 1 / Qr = 1 / Qc + 1 / Qi is calculated.

From Table 2, Xc = 72.4Ω at 200 kHz of the ceramic capacitor C1r, and Xc (Ω) = Xi (Ω) at the resonance point.
Qi = Xi / Ri = 72.4Ω / 1.35Ω = 53.62
Qc = Xc / Rc = 72.4Ω / 1.57Ω = 46.11
1 / Qi + 1 / Qc = 1 / 53.62 + 1 / 46.11 = 0.04
Qr = 1 / 0.041 = 24.8.
Although there is a slight difference from the theoretical boost ratio Ht = 23.17 obtained above, the theoretical boost ratio Ht obtained above takes into account the output impedance Zs of the AC power supply, and the output impedance Zs. , Ri + Rc = 1.35Ω + 1.57Ω = 2.92Ω, Ir = Vt (V) / (Ri (Ω) + Rc (Ω)) = 1V / 2.92Ω = 0.342A,
Vc = 72.4Ω × 0.342A = 24.8V, and Ht = Qr. As described above, the actual boost ratio H is 10.8, and the capacitor C1r has a theoretical correlation,
The relationship 1 / Qr = 1 / Qc + 1 / Qi is not satisfied.

  As described above, regarding the characteristics of the capacitor used in the circuit configuration of FIGS. 1 and 72 in which an alternating current flows in the capacitor itself, the difference between the theoretical value and the experimental value has been completely studied. There is no prior literature.

  From FIG. 77 and the calculation, it was found that if there is a capacitor whose boost ratio H matches the theoretical value, the boost ratio H differs from the theoretical value Ht depending on the capacitor. Therefore, the inventor of the present application measured the voltage across the capacitor with the oscilloscope 85 again in FIG.

(Explanation of Examples of Capacitor Performance Evaluation Method and Performance Evaluation Device)
79 is an AC voltage waveform across the polypropylene capacitor C1c of Table 3 at the resonance point of FIG. FIG. 80 is an AC voltage waveform across the ceramic capacitor C1r shown in Table 2. FIG. 81 is an AC voltage waveform across the ceramic capacitor C1r when R3 is about 30Ω in the circuit of FIG.

  79 and 80 are obtained by measuring the voltage across each capacitor with reference to GND in FIG. 73 with R2 in FIG. 73 removed. The GND of the oscilloscope 85 must be connected to the GND of the AC power supply 3b as shown in FIG. This is because when a positive peak value Vp and a negative peak value Vn with respect to GND, which will be described later, are measured and a ratio of Vp and Vn, Vp / Vn, is obtained, Vp / Vn> 1. It is. When reverse connection is established, Vp / Vn <1, which is not within the numerical value specification. Also, if the ground of the oscilloscope 85 is not connected to a low impedance point, the observed waveform becomes unstable.

  As can be seen from a comparison between FIG. 79 and FIG. 80, in FIG. 79, the voltage waveform across the polypropylene capacitor C <b> 1 c is almost symmetrical with respect to the zero point (GND in FIG. 73). On the other hand, in FIG. 81, the voltage waveform at both ends of the ceramic capacitor C1r is shifted in the plus direction with respect to the zero point, and has a positive and negative asymmetric waveform. That is, a positive DC component is included in the voltage waveform across the ceramic capacitor C1r. Such a phenomenon becomes more prominent when a non-inductive resistor R3 of about 30Ω is inserted in series in the circuit of FIG. 78 described above. FIG. 81 shows a waveform that is largely shifted in the positive direction with respect to the zero point. It has been experimentally confirmed that when the voltage across the polypropylene capacitor C1c is measured under the same conditions as in FIG. 81, there is very little shift in the positive direction.

  FIG. 82 is a graph of the capacitors shown in Table 2 with the effective series resistance Rc at 200 kHz as the X axis and the shift ratio S from the sine wave as the Y axis. As shown in FIG. 82, there is no correlation between the effective series resistance Rc of each capacitor indicated by a black dot and the shift ratio S from the sine wave.

  FIG. 83 is a graph in which the shift ratio from the zero point of each capacitor shown in Table 2 is taken as the X axis, and the Y axis is taken as the power transmission performance. FIG. 84 is a graph of the capacitors shown in Table 2 with the effective series resistance Rc at 200 kHz as the X axis and the Y axis as the power transmission performance. FIG. 85 is a graph of each capacitor shown in Table 2 with the dielectric loss tangent tan δ as the X axis and the Y axis as the power transmission performance.

  83 to 85, the power transmission performance is indicated by secondary side power P2 (W) and transmission efficiency η. The secondary power P2 is the maximum power P2 (W) that can be transmitted to the secondary side with the AC voltage on the power receiving side being constant, the power transmission coil 1, the power receiving coil 2, the relative position of both coils, and the load resistance value being the same. Indicates. The transmission efficiency η is the effective value of the AC voltage obtained by monitoring the both ends of the DC power Pd (W) input to the power transmission side and the load resistance with the oscilloscope 85 and calculating from the pp voltage Vp (V) at both ends of the load resistance. Ve (V), Ve = Vp / 2√2 (V), the ratio η of the load power Ps (W) calculated from the above, η = PsW / PdW. The secondary power P2 is 3.25 W or more and the power transmission efficiency η is 80% or more as a reference, and the X-axis condition that satisfies this criterion is defined. In the following description, power transmission performance means secondary power P2 and transmission efficiency η.

  The above criteria is because the power loss of the power transmission device occurs due to Zs, Rc, and Ri in FIG. 74, and the power loss of 1 W is the upper limit for practical use in power transmission around 4 W. When calculated from this condition, the power transmission efficiency is 75%. Moreover, although the magnitude | sizes of a coil differ, the efficiency which converts the electrical energy of an induction heater (electromagnetic cooking device) into heat energy is about 85%. Therefore, 80% transmission efficiency is defined as an intermediate value between 75% and 85%. The maximum power that can be transmitted to the secondary side is about 4.1 W. Therefore, from the 80% transmission efficiency, the secondary power lower limit is defined as 4.1 × 0.8≈3.25 W.

  When the time average value of the output voltage of the AC power supply 3b is not zero and includes a DC component, such as a square wave of zero potential and plus potential, both ends of the capacitor in the circuits of FIGS. 73 and 78. The voltage Vc (V) is a sine wave shifted to the plus side with respect to the zero potential. Further, in the circuit of FIG. 78, when R3 is a value larger than Xc = 80Ω, for example, about 100Ω, the Q of the series resonant circuit decreases, the waveform approaches a triangular wave, and the minimum value of the voltage waveform across the capacitor Becomes higher than zero potential.

  In practice, in FIG. 73, it is difficult to make the effective series resistance Ri (Ω) of the coil 1 and the output impedance Zs (Ω) of the AC power supply zero. Therefore, the reactance Xc (Ω) of the capacitor or the reactance Xi (Ω) of the coil 1, the effective series resistance Ri (Ω) of the coil 1, and the output impedance Zs (Ω) of the AC power source are measured in advance. Observe the waveform.

  Compared with the voltage Vc (V) applied to the capacitor, the current Ic (A) flowing through the capacitor is advanced by 90 degrees. Multiplying the instantaneous values of Vc (V) and Ic (A) and integrating for one period, the product of Vc and Ic becomes zero. That is, the capacitor which is a reactive element does not consume AC power. It is assumed that a DC voltage Vj as shown in FIG. 81 is superimposed on the current applied to the capacitor. In this case, multiply the instantaneous value of Vc and Ic and integrate for one period.

Vc = Vj + Vm · sinφ (V), Ic = cosφ (A),
Vc × Ic = (Vj + Vm · sinφ) × cosφ (W)
= Vj · cos φ + Vm · sin φ · cos φ (W)
= Vj · cosφ + Vm · (1/2) sin2φ (W)
Mathematically, if φ is an independent variable and Vj · cos φ and Vm · sin 2φ are definitely integrated from 0 to 2π, both become zero. That is, even if a DC component is superimposed on an AC voltage applied to the capacitor, the capacitor does not consume power. This is the effect that the capacitor blocks the direct current and allows only the alternating current to pass.

  Therefore, in FIG. 78, the both-ends voltage of resistance R3 through which a direct current flows and the both-ends voltage of power transmission coil 1 are substantially symmetrical with respect to the zero point (GND in FIG. 70). On the other hand, the voltage across the capacitor C is shifted with respect to the zero point as shown in FIGS.

Assuming the above circuit theory, the shift ratio S is obtained as follows with respect to the zero point of the AC voltage waveform Vc across the capacitor. First, the sum of all effective series resistances in the series circuit is Rr, the reactance of the capacitor at the resonance frequency is Xcr (Ω), the phase angle is θ (degrees), and the pp value of the voltage across the capacitor is Vp (V). Assuming that the shift value from the zero point is Vm (V), as described above,
tan ⁻¹ (Xcr / Rr) = θ (degrees),
Vm = Vcp × (½) cos θ (V).

In the above equation, reactance at 200 kHz, Xc = Xi = 80Ω, effective series resistance, Ri = 1.35Ω, output impedance of AC power supply, Zs = 0.2Ω, sum of all effective series resistances in the series circuit, Substituting Rr = 1.55Ω,
tan ⁻¹ (Xcr / Rr) = tan ⁻¹ (80 / 1.55) = 88.89 degrees Vm = Vcp × (½) cos 88.89 degrees = (Vcp / 2) · 0.0193
Vm = Vcp × 0.00968.

Therefore, when an ideal capacitor is used, assuming that the absolute value of the positive peak value of the capacitor is Vp and the absolute value of the negative peak value is Vn,
Vp = (Vp−Vn) × 1.000968 (V)
Vn = (Vp−Vn) × 0.9903 (V)
Is the theoretical value. The ratio of Vp and Vn, Vp / Vn is
Vp / Vn = 1.00968 / 0.9903 = 1.0195.

  As a precaution, the above shift value is calculated using an effective value instead of pp.

Vp = ((Vp−Vn) / 2√2) × 1.000968 (V)
Vn = ((Vp−Vn) / 2√2) × 0.9903 (V)
((Vp−Vn) / 2√2) is a constant. Therefore, the ratio of Vp (V) to Vn (V), Vp / Vn (no unit), remains the same as Vp / Vn = 1.0195.

  When the effective series resistance Rc (Ω) of the capacitor satisfies Rc << Rr, the ratio of Vp to Vn, Vp / Vn, is Vp / Vn <1, taking into account significant figures and measurement errors. .02 should be satisfied. In Table 2 described above, only some capacitors such as C1c and C1f satisfy Vp / Vn <1.02.

On the other hand, when Rc> 0.1Ω, the effect of Rc comes out. From the data in Table 3, with a slight margin as the average value of Rc, Rc = 1.5Ω, Rr = 3Ω Recalculating with the above formula,
tan ⁻¹ (80/3) = 87.85 degrees Vcp × (1/2) cos 87.85 = Vcp · 0.0187
Vp = (Vp− (Vn)) × 1.0187 (V)
Vn = (Vp− (Vn)) × 0.9812 (V)
The ratio of Vp and Vn, Vp / Vn is
Vp / Vn = 1.0187 / 0.9812 = 1.0381.

  In this way, in consideration of the effective resistance of the capacitor and taking into account significant figures and measurement errors in Table 2 described above, 1 <Vp / Vn <1.04. However, a capacitor that satisfies this condition is also C1e. , C1a and C1b only. That is, it cannot be considered that this asymmetry with respect to the zero potential is caused only by the effective series resistance Rc of the capacitor. It must be considered that the waveform asymmetry with respect to zero potential is caused by factors other than Rc. For example, Lc shown in FIG. 126 can be considered. Therefore, in FIG. 73, a capacitor that satisfies the power transmission performance definition that P2> 3.25 W and η> 80% from the condition that the shift ratio S is S <1.06 is selected. I do not get.

  When the integration circuit Gi is connected to both ends of the capacitor Cx shown in FIG. 73, the output voltage Vg of the integration circuit Gi is Vg = Vp−Vn and is proportional to the shift ratio S. In order to measure the shift ratio S with the oscilloscope 85, there is a limit to the measurement accuracy. However, the amplitude values Vp (V) and Vn (V) of the voltage waveform of the capacitor can be measured with the oscilloscope 85 almost accurately. If the shift ratio S is S = Vg / Vn + 1 = (Vp−Vn + Vn) / Vn = Vp / Vn, a more accurate shift ratio S is obtained. Alternatively, positive and negative peak hold circuits may be provided at both ends of the capacitor to obtain Vp and Vn.

  The oscilloscope 85 shown in FIG. 73 constitutes a means for measuring the positive peak value Vp and the negative peak value Vn. That is, each measuring means described above describes the evaluation apparatus for a capacitor for a high-frequency power circuit according to the present invention.

  It goes without saying that the higher the secondary power, the better the transmission efficiency, regardless of the above rules. When the polypropylene capacitor C1c is used, the power transmission side power in FIG. 70 is 4.5 W, and the secondary side AC power is 4.1 W. The sum of the output impedance Zs and the effective series resistance Ri of the coil is 1.55Ω, and the power transmission side voltage is an effective value of 1V. Therefore, the Joule loss of P = 1V2 / 1.55Ω = 0.65W is generated by the output impedance Zs and the effective series resistance Ri of the coil. That is, the calculated transmission efficiency is 100% or more. The surplus of 0.25 W appears because the waveform on the secondary side is not a perfect sine wave, and the pp voltage value divided by 2√2 does not give an accurate effective value. Is done. That is, when the polypropylene capacitor C1c is used, the transmission efficiency of the power transmission system (between the power receiving and receiving coil 2) is extremely 100% excluding the power loss due to the output impedance Zs of the AC power supply 3b and the effective series resistance Rc of the capacitor. It seems close.

  As is clear from FIG. 83, as the capacitor having asymmetrical positive / negative, that is, Vp / Vn, as shown in FIG. 80 becomes larger than 1, the power transmission performance deteriorates. On the other hand, a capacitor with Vp / Vn close to 1 has good power transmission performance.

  In FIG. 83, it can be seen that as Vp / Vn becomes larger than 1, the curve of the power transmission characteristic monotonously decreases. Referring to FIG. 80, when the circuit of FIG. 73 is driven with a square wave of zero potential and positive potential, the sine wave voltage waveform of the capacitor has a shift ratio S of 1.06 or less from the zero point. It will be necessary. However, the specified value, 1.06, was measured at 200 kHz using a 0.01 μF capacitor as described above. Under this condition, the value of Vp / Vn when using the capacitor C1c and the reference coil is 1.02. However, when the same capacitance as the capacitor C1c having a capacitance of 0.47 μF and another reference coil were used and measured at 50 kHz, Vp = 22.8V, Vn = 20.7V, Vp / Vn = 1. 1

  Depending on the reference coil and frequency, the present inventor has conducted various experiments. As a result, when the capacitance is C (μF) and the coefficient γ is γ = log (C / 0.001), It was found that the specified value can be approximated as 1 + 0.06 × γ. As an example, at 0.47 μF, γ = log (0.47 / 0.001) = log (470) = 2.67. The specified value is 1 + 0.06 × γ = 1 + 0.162 = 1.162.

  When the same polyethylene terephthalate capacitor as the 0.1 μF capacitor C1i is used at 100 kHz, Vp = 16.5V, Vn = 15.1V, and Vp / Vn = 1.09. γ becomes γ = log (0.1 / 0.001) = log (100) = 2. Therefore, the specified value is 1 + 0.06 × α = 1 + 0.12 = 1.12. When the capacitance is 0.01 μF, γ = log (0.01 / 0.001) = log (10) = 1. Therefore, the specified value is 1 + 0.06 × γ = 1 + 0.06 = 1.06. When the capacitance is 0.01 μF or less, the specified value is 1.06.

  Therefore, when the predetermined coefficient is B and the capacitance of the capacitor is C (μF), when C (μF) <0.01, B = 1.06. When C (μF)> 0.01, the predetermined coefficient B is B = 1 + 0.06 × log (C / 0.001).

  The value of Vp / Vn described above also varies depending on the reference coil and the frequency of power transmission. Therefore, the predetermined coefficient B is obtained from the capacitance of the capacitor by the above-described calculation formula. Then, using the power transmission coil 1 that is actually used, Vp and Vn of the voltage across the capacitor are measured at the frequency used for power transmission. Under this condition, when the predetermined count is B, the capacitor only needs to satisfy the condition of Vp / Vn ≦ B.

  On the other hand, in FIG. 87, as described above, as the effective series resistance Rc (Ω) of the capacitor increases, the effective series resistance Rc increases and decreases around 1.4Ω after the power transmission characteristic curve decreases. Such monotonic singularities are seen. Also in FIG. 88, as the dielectric loss tangent tan δ of the capacitor increases, the curve of the power transmission characteristic does not monotonously decrease and a singular point is seen. However, in both FIG. 84 and FIG. 85, it is possible to define the characteristics from the correlation between the characteristics as a capacitor and the power transmission performance. In FIG. 84, when the effective series resistance Rc of the capacitor is 1.55Ω or less, the above-described power transmission performance standard is satisfied. The above 1.55Ω is obtained by adding the effective series resistance Ri = 1.35Ω of the power transmission coil 1 alone and the output impedance Zs = 0.2Ω of the AC power supply 3b.

  However, FIG. 84 shows data at a capacitance of 0.01 μF and a frequency of 200 kHz. Therefore, it is preferable that the reactance considering the frequency and the capacitance is defined by the dielectric loss tangent tan δ in the equation. As far as FIG. 82 is referred to, the relationship between the power transmission performance and the dielectric loss tangent is not monotonously decreased as the X value increases, as shown in the graph of FIG. At a predetermined minimum frequency, for example, 100 kHz, 200 kHz, etc., the dielectric loss tangent tan δ of the capacitor is 2% or less, which is a minimum satisfactory condition. However, there is a singular point that does not satisfy the above-described reference values, such as a secondary power of 3.25 W or more and a transmission efficiency of 80% or more in a region where the dielectric loss tangent tan δ of the capacitor is 2% or less. Therefore, the dielectric loss tangent tan δ of the capacitor is more preferably 1% or 0.5% or less.

  FIG. 86 is a graph showing the frequency characteristics of the effective series resistance Rc of a capacitor of the same type as the ceramic capacitor C1r having poor power transmission performance but having a different capacitance.

  FIG. 87 is a graph showing the frequency characteristic of the dielectric loss tangent tan δ of a capacitor of the same type as the ceramic capacitor C1r having poor power transmission performance but having a different capacitance.

  FIG. 88 is a graph showing the frequency characteristics of the effective series resistance Rc of a capacitor of the same type as the polypropylene capacitor C1c with good power transmission performance and a different capacitance.

  FIG. 89 is a graph showing the frequency characteristic of the dielectric loss tangent tan δ of a capacitor of the same type as the polypropylene capacitor C1c with good power transmission performance and a different capacitance.

  As far as FIG. 87 and FIG. 89 are referred to, it can be seen that the frequency characteristic of the dielectric loss tangent tan δ differs depending on the dielectric and capacitance of the capacitor. As shown in FIG. 87, the ceramic capacitor C1r using a dielectric having a high relative dielectric constant has a small dielectric loss tangent tan δ of a capacitor having a large capacitance in the low frequency region. However, in the high frequency region, the dielectric loss tangent tan δ of the capacitor having a large capacitance is large. That is, in a capacitor using a high dielectric, the increase rate of the dielectric loss tangent tan δ accompanying the increase in frequency is smaller as the capacitance is smaller. On the other hand, referring to FIG. 89, the polypropylene capacitor C1c using a dielectric having a low relative dielectric constant has a large dielectric loss tangent tan δ of a capacitor having a large capacitance in almost all frequency regions. 87 and 89 coincide with the actual measurement results described above. That is, it is necessary to select the type of capacitor according to the capacitance and the operating frequency of the power transmission device.

  Referring to FIG. 87, in the ceramic capacitor C1r made of a dielectric material having a high relative dielectric constant, only a capacitor having a capacitance of 0.47 μF at 200 kHz has a dielectric loss tangent tan δ of 1% or less. On the other hand, referring to FIG. 89, the polypropylene capacitor C1c formed of a dielectric material having a low relative dielectric constant has a capacitance of 0.01 μF to 0.47 μF and a dielectric loss tangent tan δ of 1% or less at 200 kHz. It has become.

  As mentioned at the outset, to improve the power factor is to cancel the positive reactance. The LC series circuit has a function of returning a non-sinusoidal waveform such as a square wave, a triangular wave, and a sawtooth wave to a sine wave. For that purpose, the Q of the LC circuit must be high, and as a rule, the capacitor needs a Q of at least 10 or more. As shown in Table 2, it is necessary to select a capacitor satisfying the characteristics as described above for a capacitor having a Q of several tens, regardless of a capacitor having a Q exceeding 1000 or 10,000 at 100 kHz.

  In FIG. 84 where the effective series resistance Rc of the capacitor is the X axis and the power transmission performance is the Y axis, and in FIG. 85 where the dielectric loss tangent tan δ of the capacitor is the X axis and the power transmission performance is the Y axis, the capacitor C1h is in the specified region. Inside. On the other hand, in the graph of FIG. 83, all the capacitors C1h are outside the specified region. Thus, in the present invention, it is possible to exclude capacitors having poor power transmission performance that cannot be defined only by the effective series resistance Rc of the capacitor and the dielectric loss tangent tan δ of the capacitor. According to such a rule, a capacitor having good power transmission performance is selected to configure the power transmission device. As a result, there is an extremely excellent effect that a power transmission device with good power transmission performance can be realized.

  88, when the frequency is in a low frequency region of about 20 kHz, the effective series resistance Rc of the capacitor is several Ω to several tens Ω, and the power loss due to the effective series resistance Rc increases. Therefore, it is preferable that the lower limit of the frequency at which a capacitor having a capacitance of 0.01 μF can be used for power transmission is about 20 kHz. However, when the capacitor has a capacitance of 0.1 μF or more, the effective series resistance Rc is reduced to a value that causes almost no power loss even at 10 kHz. However, the presence of the effective series resistance Rc (Ω) deteriorates the power transmission performance. Therefore, when the effective series resistance of the power transmission coil 1 is Ri (Ω), power transmission is performed at a minimum frequency that satisfies at least Ri (Ω)> Rc (Ω) as the effective series resistance Rc (Ω) of the capacitor. Is preferred. The effective series resistance Rc (Ω) of the capacitor described above relates to the definition of the lowest frequency at which the capacitor can be used. The maximum frequency at which the capacitor can be used will be described later by showing the frequency characteristics of the capacitor impedance.

(Explanation of dielectric absorption)
Next, as one of factors having a correlation with the power transmission performance, the dielectric absorption of the capacitor can be considered. Dielectric absorption is a direct current characteristic of the capacitor. There are various theories about the cause of the dielectric absorption. As one theory, it is considered that the dielectric polarization occurs while the DC voltage Vw (V) is applied to both terminals of the capacitor for a long time. A DC voltage is no longer applied to the capacitor, and both terminals of the capacitor are short-circuited to discharge the charge accumulated in the capacitor. When the capacitor is opened after discharging, an open circuit voltage Vb (V) is generated across the capacitor. The ratio of Vw (V) to the capacitor Vb (V) is the dielectric absorption K, and K = Vb (V) / Vw (V). The smaller the K, the better the capacitor characteristics. The dielectric absorption K is always a positive value smaller than 1 and is a unitless numerical value.

(Explanation when one capacitor is installed at each end of the coil)
In paragraph 0018 of Patent Document 4, it is described that a capacitor having twice the capacitance necessary for the resonance frequency is provided at each end of the coil. As described above, other patent documents have similar descriptions. In the previous invention, it has already been described that when a reference coil is used to measure the voltages Vp and Vn across a capacitor, the performance of the capacitor can be judged by the shift ratio S from the zero point of Vp and Vn. The inventor of the present application pays attention to this point, and Vp, Vn, S = Vp / Vn, both when the reference coil is equipped with one capacitor and when two capacitors are provided at both ends of the reference coil. And I measured the power transmission performance.

  In FIG. 73, the source (Vd-terminal) of the switching element Q2 is connected to the GND of the oscilloscope 85, and the connection point between the reference coil Ls and the capacitor Cx is connected to the signal input terminal of the oscilloscope 85. As described above, when measuring Vp and Vn, R2 is short-circuited. By connecting in this way, Vp> Vn and S = Vp / Vn> 1.

  FIG. 90 is a connection diagram for measuring the voltage across the capacitor with an oscilloscope when the capacitor is connected to the VOUTH side. FIG. 91 is a connection diagram in the case where each capacitor voltage is measured with an oscilloscope when one capacitor is connected to each end of the power transmission coil 1.

  The output of the AC power supply 3b shown in FIGS. 90 and 91 is indicated by terminals VOUTH and VOUTL having low impedance. A series circuit of the capacitor Cx and the power transmission coil 1 is connected to terminals VOUTH and VOUTL of the AC power supply 3b shown in FIG. The GND of the oscilloscope 85 is connected to VOUTH or VOUTL of the AC power supply 3b. A voltage input IN of the oscilloscope 85 is connected to a connection point between the capacitor Cx and the power transmission coil 1.

  In FIG. 91, a series circuit of a capacitor Cx1, a power transmission coil 1, and a capacitor Cx2 is connected to terminals VOUTH and VOUTL of the AC power supply 3b. When the capacitor voltages of the two capacitors Cx1 and Cx2 are simultaneously measured by the same oscilloscope 85a and 85b, the outputs OUTH and OUTL of the AC power supply 3b are short-circuited via the GND of the oscilloscopes 85a and 85b. It is necessary to measure Vp and Vn separately.

  In order to measure the voltage across the capacitor Cx, it is necessary to connect the GND of the oscilloscopes 85a and 85b to the reference voltage point. In FIG. 73, the capacitor Cx and the coil Ls are interchanged, and one terminal of the capacitor Cx is connected to the drain (+ terminal of Vd) of the switching element Q1. One terminal of the reference coil Ls is connected to the source of the switching element Q2 (the negative terminal of Vd). An LC series resonance circuit is configured by connecting the other terminal of the capacitor Cx and the other terminal of the coil Ls. The drain of the switching element Q1 (+ terminal of Vd) is connected to the GND of the oscilloscope 85, and the connection point of the reference coil Ls and the capacitor Cx is connected to the signal input terminal of the oscilloscope 85. In this way, when the oscilloscope 85 measures the voltage across Cx of the LC series resonance circuit in which the capacitor Cx and the coil Ls are interchanged in FIG. 73, Vn> VP. Therefore, the shift ratio is S1, and S1 = Vn / Vp> 1.

  When two capacitors Cx1 and Cx2 are provided at both ends of the reference coil Ls, in order to measure Vp and Vn, the connection method of the oscilloscope 85 may be changed as described above. When two capacitors Cx1 and Cx2 are provided at both ends of the reference coil Ls and the connection method of the oscilloscope 85 is changed as described above, Vp <Vn, so that Sn = Vn / Vp> 1 and Sn Define When two capacitors are provided at both ends of the reference coil and the connection method of the oscilloscope 85 is the same as that in FIG. 73, Sp = Vp / Vn. S = Vp / Vn and S1 = Vn / Vp are the shift ratios when one capacitor is installed in the reference coil.

  The inventor of the present application connected two capacitors shown in Table 2 in parallel to create two composite capacitors having a capacitance of 0.02 μF. Vp, Vn, when using one synthetic capacitor having a capacitance of 0.01 μF with this synthetic capacitor connected in series and when connecting a synthetic capacitor with a capacitance of 0.02 μF to both ends of the reference coil S, S1, Sp, and Sn were measured. The effective series resistance Rc (Ω) of each capacitor described in Table 2 is half of Rc (Ω) in a synthetic capacitor having a capacitance of 0.01 μF. However, Rc (Ω) is equal to Rc (Ω) described in Table 2 in the case of a synthetic capacitor having a capacitance of 0.01 μF in which two synthetic capacitors having a capacitance of 0.02 μF are connected in series. When Vp, Vn, S, Sp, and Sn are measured under such conditions, the conditions are the same as when one single capacitor is used.

As a result of measurement, Sp and Sn were almost equal in any capacitor. Also, S and S1 were almost equal. However, Sp was often smaller than S. As a general experimental result, if the value of (Sp-1) / (S-1) is 0.75 or less, the power transmission performance of the capacitor is one at each end of the coil than when one capacitor is installed in the coil. It was better to equip. For example, if S = 1.08 and Sp = 1.05,
(Sp-1) / (S-1) = 0.05 / 0.08 = 0.625 <0.75, which satisfies the conditions.

If S = 1.1 and Sp = 1.08,
(Sp-1) / (S-1) = 0.08 / 0.1 = 0.8> 0.75, which does not satisfy the condition. If (Sp-1) / (S-1) = Fa,
Fb = (Sp-1) / (S1-1)
Fc = (Sn-1) / (S-1)
Fd = (Sn-1) / (S1-1)
The equivalent result can be obtained. Any of Fa to Fd may be 0.75 or less. This sorting method is a sorting method that does not depend on the reference coil, capacitance, frequency, or the like as compared with the sorting method in which the value of S = Vp / Vn is simply measured and is equal to or less than the predetermined numerical value B.

  It is presumed that the connection of one capacitor to each end of the coil is due to the effect that the value of Vp / Vn is close to 1. As a cause of this, there is a possibility that the effective series resistance Re (Ω) varies in the equivalent circuit of FIG. 125 due to the value of the alternating current flowing through the capacitor C1. Or as mentioned above, the possibility of the influence of the equivalent series inductance Lc of a capacitor is also considered. However, the correlation that the power transmission performance is better as the shift ratio S from the reference voltage of the voltage across the capacitor is closer to 1 is clear from FIG. By connecting one capacitor to each end of the power transmission coil 1, the shift ratio S is reduced, and from the above experimental results approaching 1, by connecting one capacitor to each end of the power transmission coil 1, It can be said that the power transmission performance can be improved. In the conventional example, these effects are not clarified at all.

  In addition, the case where two capacitors of 0.02 μF are used and the two capacitors are connected to both ends of the coil is compared with the case where one capacitor of 0.01 μF is used. When two capacitors were used, the value of Vp / Vn approached 1 and the power transmission performance was improved. Since both the coil and the capacitor have effective series resistance and there is a correlation between the symmetry of Vp and Vn and the power transmission performance, the connection method as shown in FIG. 91 improves the “symmetry” of Vp and Vn. There seems to be a possibility.

  The gist of the present embodiment is that a capacitor whose Vp / Vn value approaches 1 when one capacitor is connected to both ends of the reference coil as compared with the case where one capacitor is connected in series to the reference coil. By selecting, it is intended that a power transmission device with good power transmission performance can be realized. In addition, if the capacitors have the same configuration, there is an intention that the power transmission performance can be improved by connecting one capacitor to each end of the power transmission coil. There is no prior art that clarifies the effect of connecting one capacitor to each end of the power transmission coil in this way.

  Note that it is preferable to use two capacitors having the same capacitance because of the voltage applied to the capacitors. However, any capacitance with a nominal value within 20% is applicable to this embodiment. For example, when a capacitor having a nominal value of 0.01 μF is used on one side, a capacitor having a nominal value of 0.0082 μF to 0.12 μF can be used as the other capacitor. However, the nominal value uses a capacitor of about 5 to 10% or less.

  FIG. 92 is a circuit diagram for measuring the dielectric absorption of the capacitor.

  FIG. 92 shows an impedance conversion circuit using an operational amplifier. The inventor of the present application has created an impedance conversion circuit having an input impedance of 1010Ω or more as shown in FIG. 92, so that dielectric absorption can be measured with a digital tester generally used. In FIG. 92, an inverting input terminal and an output terminal of an operational amplifier 111 operating as an impedance conversion circuit are connected, and a DC voltmeter 112 is connected between the output terminal of the operational amplifier 111 and GND. The operational amplifier 111 and the DC voltmeter 112 operate as an electronic DC voltmeter having an input impedance of 1012Ω or more. Generally used digital testers and the like have an input impedance of several MΩ and cannot accurately measure dielectric absorption.

The switch 113 is provided to apply a DC voltage Vw = V1 (V) for accumulating electric charge to the measurement capacitor C. The switch 114 is connected to the capacitor C via a resistor R4 having an accuracy of 5Ω ± 10%. The switch 115 is provided to measure the dielectric absorption K by applying the charge accumulated in the capacitor C to the non-inverting input terminal of the operational amplifier 111. The operational amplifier 111 has a high impedance with a non-inverting input terminal bias current of about 1 pA and an input impedance of about 10 ¹² Ω.

  The procedure for measuring dielectric absorption with the measuring circuit shown in FIG. 92 is defined in JIS, but will be described below. First, the switch 113 is closed and a voltage of ± 10% of the rated voltage of the capacitor C, Vw = V1 (V), is applied to the capacitor C for 1 hour. Thereafter, the switch 113 is opened, the switch 114 is closed, and the resistor R4 is connected to both ends of the capacitor C for 10 seconds to discharge the charge stored in the capacitor C. After discharging for 10 seconds, the switch 114 is opened to open the capacitor C, the switch 115 is closed to apply the voltage Vb (V) across the capacitor C to the operational amplifier 111, and the voltmeter 112 measures it for 15 minutes. The maximum value of the voltage across the capacitor in the measurement time of 15 minutes is defined as Vb (V). The dielectric absorption K is obtained as K = Vb / Vw.

  The inventor of the present application measured a dielectric absorption K by applying a positive voltage to one terminal of a nonpolar capacitor. After that, when a negative voltage was applied to one terminal and the dielectric absorption K was measured, a result different from the case where the dielectric absorption K was measured by applying a positive voltage to one terminal was obtained. As will be described later, the asymmetry of dielectric absorption correlates with the power transmission performance. Hereinafter, dielectric absorption measured by applying a positive voltage to one terminal, for example, the terminal A of the capacitor C shown in FIG. 92 will be denoted as Kp. The dielectric absorption measured by applying a negative voltage to one terminal is expressed as Kn. For example, the dielectric absorption measured by applying a positive voltage to the B terminal of the capacitor C shown in FIG. 92 is expressed as Kn.

  As shown in Table 3, in JIS, dielectric absorption is specified only for nonpolar DC capacitors, and the specified value is 0.1%. However, in the embodiment of the present invention, an AC voltage is applied to a nonpolar capacitor to pass an AC current. Dielectric absorption is a direct current characteristic, and the impedance of a capacitor is theoretically infinite at direct current. A bias current flows through the input terminal of the operational amplifier 111 shown in FIG. According to JIS regulations, the insulation resistance of the capacitor is 1 GΩ to 30 GΩ. Even when an operational amplifier having a bias current value of about 1 pA is used, the capacitor is charged by the bias current, and the absolute value of the output terminal voltage of the operational amplifier 111 increases with time. Therefore, a high resistance Rh of 200 MΩ is attached between the GND at the non-inverting input terminal of the operational amplifier 111 in FIG.

Further, when the rated voltage of the capacitor is about 100V, actual measurement becomes dangerous. Therefore, a voltage of 20 V was applied to the capacitor C regardless of the rated voltage of the capacitor C. When the voltage applied to the capacitor C is high, the electric field strength in the dielectric increases and dielectric polarization is likely to occur. Therefore, the actual measurement value is within 0.25%, but it is necessary to allow a margin of about 5 times. Therefore, as the capacitor C of the power transmission device, the dielectric absorption K must be at least 1% or less. Actually, when the inventor of the present application forms a circuit with aerial wiring, the insulation resistance of the entire circuit is set to 10 ¹² Ω or more, the rated voltage of 100 V is applied to the capacitor C1m, and the dielectric absorption is measured. 93%. This corresponds to five times the actual measurement value shown in FIG.

  As can be seen from FIG. 93, a capacitor having a small dielectric absorption Kp has good power transmission, and a capacitor having a large dielectric absorption Kp has poor power transmission. Further, as described above, the inventor of the present application switches the terminal A of the capacitor C to which a positive voltage is applied and the terminal B of the capacitor C to which a negative voltage is applied in the circuit of FIG. saw.

  FIG. 94 is a graph in which the dielectric absorption Kn is measured by replacing the terminal A and the terminal B of the capacitor in FIG. 92, Kn is the X axis, and the power transmission performance is the Y axis. FIG. 95 is a graph with the ratio Kr of the dielectric absorption Kp and Kn of each capacitor shown in Table 2 as the X axis and the power transmission performance as the Y axis. The ratio Kr is defined such that when Kp> Kn, Kr = Kp / Kn, and when Kn> Kp, Kr = Kn / Kp, and Kr> 1.

  According to FIG. 95, the asymmetry of the dielectric absorption is clearly seen. A capacitor with good power transmission performance has low dielectric absorption asymmetry. In other words, the value of the ratio Kr is close to 1. A capacitor having poor power transmission performance has a large asymmetry of dielectric absorption. In other words, the value of the ratio Kr is greater than 1. In this way, a capacitor with good power transmission performance can be selected.

  That is, when one terminal A and the other terminal B of the capacitor C shown in FIG. 92 are interchanged, the dielectric absorption Kp and Kn have different values. This is remarkable in the ceramic capacitor C1r having the smallest step-up ratio. As can be seen from FIG. 95, a capacitor with a ratio Kr close to 1 has good power transmission, and a capacitor with a ratio Kr larger than 1 has poor power transmission. From the correlation between Kr and power transmission performance in FIG. 95, the ratio of Kp to Kn, Kr, Kr = Kp / Kn, is in the range of 1 <Kp / Kn <1.5. I understand that it is necessary.

(Explanation of the embodiment regarding the temperature rise of the capacitor)
FIG. 96 is a circuit configuration diagram for determining the performance of the capacitor C. The AC power supply 122 in FIG. 96 is configured such that the frequency can be varied, the output current Ia can be measured by the AC ammeter 123, and the output voltage Vt (V) can be measured by the AC voltmeter 124. When an AC voltage Vt (V) is applied to the capacitor C, a current Ia (A) due to reactive power flows through the capacitor. Ia (A) = Vt (V) / Xc (Ω). FIG. 97 is a circuit diagram for causing a current to flow through the capacitor Cx by configuring the LCR series resonant circuit shown in FIG. 73 described above.

  FIG. 98 is a graph in which, in the circuits of FIGS. 96 and 97, when an alternating current having a frequency of 200 kHz and an effective value of 0.5 A is passed through the capacitor, the rising temperature of the capacitor is taken as the X axis and the power transmission performance is taken as the Y axis. It is. In order to measure an accurate temperature, the temperature of the capacitor is measured with an infrared non-contact type thermometer.

  As is clear from FIG. 98, when the temperature of the capacitor C increases, the power transmission performance deteriorates. It can be seen that a predetermined power transmission performance can be obtained if the temperature rise of the capacitor is 10 ° C. or less. It is more preferable if the temperature rise of the capacitor is 5 ° C. or less.

(Description of performance evaluation method and performance evaluation device)
99A is a block diagram showing an example of measuring the performance of a capacitor, and FIG. 99B is an integration included in the double integration type A / D converter 125 shown in FIG. 99A. FIG.

  The A / D converter 125 operating as the pulse number measuring means and the AD converting means includes an integrating circuit shown in FIG. An integrating capacitor Cx is connected between the inverting input terminal b and the output terminal a of the operational amplifier 128. The integrating capacitor Cx is connected to one end of the resistor Rt and the inverting input terminal of the operational amplifier 128, and the other end of the resistor Rt is input. It is connected to the end VIN. The non-inverting input terminal of the operational amplifier 128 is connected to GND.

  The reference voltage Vref is connected to the reference voltage inputs REF_HI and REF_LO of the A / D converter 125. A display 126 is connected to the A / D converter 125. A switch 127 is connected to the measurement voltage input terminals IN_HI and IN_LO. The switch 127 switches whether the reference voltage Vref and the inverted reference voltage −Vref are applied to the measurement voltage input terminals IN_HI and IN_LO, or the short circuit state. When switched to the short-circuit state, the switch 127 operates as an initialization means for setting the voltage across the integrating capacitor to zero.

  In this embodiment, the A / D converter 125 causes the constant current Ip in the positive direction to flow through the integration capacitor Cx for a predetermined time T after the integration capacitor Cx is initialized, and integrates after the predetermined time T elapses. When a predetermined constant current In in the negative direction is passed through the capacitor Cx and the time until the voltage across the integrating capacitor Cx becomes zero is Tn, the number of pulses at the predetermined time T when Ip = In is counted. When the pulse count for the predetermined time T is N and the pulse count for the time Tn until the voltage across the integration capacitor Cx becomes zero is Nn, the absolute value of the difference between Nn and N is 0. Select a capacitor of .004 × N count or less.

  A double integration type A / D converter is used as the A / D converter 125, and the reference voltage and the input voltage of the A / D converter 125 are the same. When the output, for example, the display of the A / D converter 125 shows a deviation from the theoretical value 1, the performance of the power factor capacitor used for the power transmission measure can be determined. The A / D converter 125 has a resolution of 1000 counts or more. This resolution 1000 is the resolution for the theoretical value 1. That is, if the count number is 999, the deviation from the theoretical value 1 is 0.001. Therefore, if the count number is N and the deviation (digit) is 0.004 × N count or less, the power factor improvement performance of the capacitor is good. If the count number N is, for example, 10,000, a more accurate determination can be made. The theoretical value 1 may be 2 n bits, for example, 2047 if it is 11 bits.

  Further, a constant current In in the negative direction is passed through the capacitor for a predetermined time T, and after a predetermined time T has elapsed, a predetermined constant current Ip in the positive direction is passed through the capacitor until the voltage across the capacitor becomes zero. If the time is Tp, and | Ip | = | In |, where the pulse count number of T is N and the pulse count number of Tp is Np, the absolute value of the difference between Np and N is Select a capacitor of 0.004 x N counts or less.

  In this example, when the reference voltage of the A / D converter 125 is the same as the input voltage, the polarity of the input voltage is inverted. As described above in terms of dielectric absorption, even a nonpolar capacitor has a different display when the same positive voltage and the same negative voltage are applied to the A / D converter. By the above method, the power factor improvement performance of the capacitor can be accurately determined.

  Furthermore, if the absolute value of the difference between Np and Nn is 0.003 × N counts or less, the power factor improvement performance of the capacitor is good.

  In this example, the power factor improvement performance of the capacitor can be determined more accurately by comparing the absolute values of the differences between Np and Nn.

  FIG. 100 is a waveform showing the integration circuit output Vo of the A / D converter 125.

In the A / D converter 125 shown in FIG. 99 (B), when the voltage between the measurement voltage input terminals IN_HI and IN_LO, which are differential inputs, is Vm, the integrated current Ii (A) is
Ii (A) = Vm (V) / Rt (Ω). The output of the A / D converter 125 is displayed on the display 126. The input signal integration time is a fixed value of 1000 counts. Therefore, the peak voltage Vpeak (V) shown in FIG. 100 increases in proportion to Vm (V). The relationship between the integration capacitor Cx and Vpeak is Vpeak (V) = Ii (A) / Cx (F). This is because the input signal integration time is constant. After 1000 counts of input signal integration, switching to inverse integration is performed within the A / D converter 125. Inverse integration is performed from Vpeak with a constant negative slope according to currents Iref (A) and Iref (A) = Vref (V) / Rt (Ω) generated by reference voltage Vref operating as a current source. The counter integration time is counted, and the count value when the output of the integration circuit becomes zero is displayed as a 4-digit digital value.

  Assuming that an ideal capacitor Cx shown in FIGS. 99A and 99B is used, and assuming that the voltage input to the measurement voltage input terminals IN_NI and IN_LO of the A / D converter 125 is Vin, A / The display D displayed on the display 126 of the D converter 125 output is D = 1000 × (Vin / Vref). In this case, since Vin = Vref, the display D is always 1000. That is, in an ideal capacitor, the input voltage integration waveform shown in FIG. 100 and the reference voltage inverse integration waveform should be symmetric with respect to the peak voltage Vpeak. However, in a capacitor with poor power transmission performance, the input voltage integration waveform and the reference voltage inverse integration waveform are not symmetrical with respect to the peak voltage Vpeak, and a deviation occurs in the count number of integration time. If it is determined whether or not the deviation is within a predetermined value, it is possible to distinguish whether the power transmission performance of the capacitor is good or bad. This will be specifically described below.

  As shown in FIG. 99A, when the switch 127 is switched to the A side, the reference voltage Vref is input to the input Vin. At this time, the display Dp of the display 126 is 1000. When the switch 127 is switched to the B side, the input of the A / D converter 125 is short-circuited. At this time, the display Dz of the display 126 is ± 0. When the switch 127 is switched to the C side, the reference voltage −Vref is input to the input Vin. At this time, the display Dn of the display 126 becomes −1000. An example of the double integration type A / D converter 125 having such a function is ICL7106 manufactured by Intersil Corporation. In this embodiment, ICL7136, which is an improved version of ICL7106, is used. In order to perform more precise measurement, there are ICL7135 having a resolution of ± 19999, ICL7109 having a resolution of ± 12 bits in order to take data into a computer instead of a display and measure it. Both are double integration type A / D converters, and A / D converters other than the double integration type cannot be used in the present invention. It should be noted that, other than the double integration type A / D converter described above, those having the same operation principle can be used in the present invention.

  By configuring the circuit as shown in FIG. 99A, the above-described theoretical value is always displayed even if the reference power supply Vref varies. Therefore, the performance of the capacitor can be determined by seeing the deviation from this theoretical value. Although not shown in FIGS. 99A and 99B, the double integration type A / D converter 125 is provided with a circuit (auto zero circuit) for canceling the offset voltage of the operational amplifier 128 constituting the integration circuit. ing. This auto-zero circuit usually automatically adjusts the offset voltage of the integrating circuit to 50 μV or less. Therefore, if Vref is 100 mV or more, the display is always zero when the switch 77 is at the point B. In addition, Vref was set to 200 mV and measured. However, the double-integration A / D converter 125 is susceptible to noise, so the display will not be stable and accurate measurement will not be possible without careful attention to mounting, such as shielding by wiring and metal cases. Be careful as it becomes difficult.

  In FIG. 101, each capacitor shown in Table 2 is connected to the A / D converter 125, and when the switch 127 is a point B, the absolute value (unit digit) of the deviation from the zero point of each capacitor is taken as the X axis. It is a graph which made electric power transmission performance a Y-axis. In FIG. 101, when 0 digit and 1 digit display are repeated, 0 digit is set, and when 1 digit is displayed, 1 digit is set. When measuring Dp and Dn, which will be described later, the deviation from the zero point is corrected. When comparing the power transmission performance of each capacitor shown in Table 2 plotted in FIG. 101 and the digit value, there is no particular relationship between the displayed value and the power transmission performance when the input is short-circuited due to the difference in the capacitor. It is.

  First, the switch 127 shown in FIG. 99A is switched to the B side, and the input voltage of the A / D converter 125 is set to zero. The display at this time is Dz as a correction value. When 0 display and 1 display are repeated equally, Dz = + 0.5. When −0 display and −1 display are repeated equally, Dz = −0.5. This Dz is corrected by subtracting when measuring Dp and Dn. In the measurement of Dp and Dn, 999.5 is used as the measurement value when 0999 display and 1000 display are repeated equally.

  In FIG. 102, it is assumed that the switch 127 shown in FIG. 99 is switched to the A side, and the first deviation (digit) of the theoretical display 1000 and the actual display Dp when the input voltage = the reference voltage Vref is set. ) On the X axis, and the power transmission performance on the Y axis. FIG. 103 shows the absolute value of the second deviation (digit) of the theoretical display −1000 and the actual display Dn when the switch 127 is set to the position C in FIG. 99 and the input voltage = −reference voltage. Is a graph with X-axis and power transmission performance as Y-axis. FIG. 104 is a graph in which the absolute value of the difference (digit) between the absolute values of the actual displays Dp and Dn is the X axis, and the power transmission performance is the Y axis.

  The capacitors shown in Table 2 are connected to the A / D converter 125. If the absolute value of the display of the A / D converter 125 is between 0996 and 1002, both positive and negative, it can be determined that the capacitor has good performance. In FIG. 101 and FIG. 102, the reason why the maximum width of 4 digits is provided is that the A / D converter 125 normally has a basic error of ± 1 digit. It is also a value (digit) that satisfies the requirements of secondary power 3.25 W and transmission efficiency η 80% or more.

If both the actual displays Dp and Dn can be measured, as shown in FIG. 102, if the absolute value of the difference between the actual displays Dp and Dn is within 3 digits in the display of the A / D converter 125, the predetermined power transmission You can see the performance. For example, if the display of Dp is actually 1001, the display value of Dn only needs to satisfy 0998 <Dn <1004. Alternatively, a precision power supply different from the reference power supply Vref may be used, and the input voltage may be set so that the display of the A / D converter 125 becomes 1990. In this case, if the difference between the actual displays Dp and Dn is within 6 digits, a predetermined power transmission performance can be obtained. These 6 digits are
It is specified as 3 × (1990/1000) ≈6 digits.

  The above formula describes the case where the A / D converter 125 with the maximum count value and 19999 counts is used. For example, when the A / D converter with the maximum count value and 19999 counts is used, In the case of a circuit configuration, 3 × (10000/1000) ≈30 digits is the specified value. This is the same for Dp and Dn described above. When an A / D converter with 19999 counts is used, the specified value is 40 digits or less.

  Compared with the case where dielectric absorption is measured, by using the double integral type A / D converter 125, the performance of the power factor improving capacitor of the power transmission device can be judged in a short time. As described above, the performance of the power factor improving capacitor of the power transmission device can also be determined from the DC characteristics of the capacitor. In addition to the double integration type A / D converter, V / F converter (voltage frequency converter), F / V converter (frequency voltage converter), VCO (voltage controlled oscillator), sample hold amplifier (instantaneous voltage) Value holding circuit), RMS / DC converter (AC RMS-DC converter), analog circuits such as timers, analog ICs, etc., and circuits that use capacitors, judge the performance of power factor improvement capacitors for power transmission devices Is possible.

  In the V / F converter, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by looking at the linearity of the input voltage and the output frequency. An example of such an IC is AD654 from Analog Devices.

  In the F / V converter, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by looking at the linearity of the input frequency and the output voltage. An example of such an IC is LM2907 from National Semiconductor.

  Even in a VCO, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by looking at the linearity of the input voltage and the output frequency. An example of such an IC is C-MOS 4046 (manufactured by each company).

  In the sample and hold amplifier, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by inputting a constant DC voltage and looking at the difference between the output voltage and the input voltage. An example of such an IC is Analog Devices' AD585.

  In the RMS / DC converter, for example, the AC power is thermally converted, the accurate AC power is measured, and compared with the output of the RMS / DC converter, thereby determining the performance of the power factor improving capacitor of the power transmission device. Can do. An example of such an IC is AD637 from Analog Devices.

  In the timer, for example, it is possible to judge the performance of the power factor improving capacitor of the power transmission device by installing various capacitors and measuring and comparing the output with a frequency counter. An example of such an IC is a timer IC 555 (manufactured by each company).

  In the analog IC as described above, any capacitor affects the accuracy and function of each IC. Therefore, if the accuracy and function of a plurality of capacitors are compared in the IC as described above, the performance of the power factor improving capacitor of the power transmission device can be judged in the same manner as the double integration type A / D converter. The analog IC described above is an example, and there are many other analog ICs that can determine the performance of the capacitor.

  Thus, dielectric loss tangent at a predetermined frequency, symmetry with respect to zero potential, step-up ratio H, stability of capacitance when an alternating current flows through the capacitor, dielectric absorption, symmetry of dielectric absorption, double integral formula A By defining the actual display Dp of D / D converter 125, the deviation from the theoretical value of Dn, and the display difference between Dp and Dn, an optimum capacitor can be selected for improving the power factor of the power transmission device. Become. By using such a capacitor to improve the power factor of the power transmission device, it is possible to realize a power transmission device with good power transmission performance that was difficult to realize with the conventional technology.

  Next, the difference in power transmission performance due to the difference in capacitance will be examined for capacitors having exactly the same configuration using the same dielectric.

  FIG. 105 is a diagram showing the frequency characteristic of the absolute value | Z | of the complex impedance of each capacitor in a capacitor having the same configuration and different capacitance from the polypropylene capacitor C1c having the best power transmission characteristics. FIG. 106 is a diagram illustrating the frequency characteristics of the absolute value | Z | of the complex impedance of each capacitor in a capacitor having the same configuration and different capacitance from the ceramic capacitor C1r having the worst power transfer characteristics.

If the complex impedance Z of the capacitor is Z = Rc + jωC = Rc + Xc,
| Z | is represented by | Z | = √ (Rc2 + Xc2) (Ω).

  Referring to FIGS. 105 and 106, it can be seen that | Z | decreases as the frequency increases. In a capacitor having a large capacitance, | Z | is a minimum value Zb (Ω) in the vicinity of 1 MHz to 2 MHz. It can be seen that | Z | increases at higher frequencies. 105 and 106, the phase angle θ of the capacitor of 0.47 μF is plotted. At a frequency fb (Hz) or more at which | Z | is the minimum value Zb (Ω), the phase angle θ is shifted by 180 degrees, and it can be seen that the capacitor operates as an inductor. Therefore, the frequency fb (Hz) at which | Z | becomes the minimum value is the highest frequency at which the capacitor can be used. The frequency fb is a series resonance point determined by the capacitance C of the capacitor and the equivalent series inductance Le in the equivalent circuit of FIG. As will be described later, when the capacitors are connected in parallel, both the effective series resistance Rc and the equivalent series inductance Le decrease. Therefore, a large current can flow through the capacitor and heat generation can be suppressed.

  Referring to FIGS. 105 and 106, polypropylene capacitors C1c and ceramic capacitors C1r having different dielectric materials, structures, and characteristics have substantially the same frequency at which | Z | (Ω) becomes a minimum value if they have the same capacitance. You can see that Thus, by looking at the frequency characteristic of the impedance | Z | (Ω) of the capacitor, the maximum frequency at which the capacitor can be used is known. Alternatively, by looking at the frequency characteristic of the phase difference θ between the current flowing through the capacitor and the voltage across the capacitor, the maximum frequency fb (Hz) at which the capacitor can be used can be found.

  The frequency used for power transmission may be any frequency as long as it is equal to or lower than the maximum frequency fb (Hz). However, capacitor characteristics cannot be defined at any frequency. Therefore, the maximum frequency fb (Hz) is obtained with a capacitor having a predetermined capacitance Ca (F). The characteristic of the capacitor described above may be measured at a frequency equal to or lower than fb / 2 (Hz) with the lowest value of the maximum frequency fb (Hz) as a reference, and it may be confirmed whether or not the above-mentioned regulation is satisfied. Alternatively, the same frequency, such as 200 kHz or 500 kHz, that is equal to or lower than the maximum frequency fb may be used as a measurement frequency, and performance comparison of different capacitors may be performed. In this way, even when the components other than the capacitor and the power transmission frequency are changed, the power transmission performance can be ensured. From the above measurement results, the upper limit is about 0.47 μF that can be used as a single capacitor. As will be described later, when a capacitance of 0.47 μF or more is necessary, it is preferable to connect capacitors having a capacitance of 0.47 μF or less in parallel.

  In general, the minimum value of the effective series resistance Rc is regarded as a minimum value of impedance in FIGS. 105 and 106. However, as long as it is compared with FIGS. 86 and 88, the coincidence with FIGS. 105 and 106 is not seen. However, these are only differences between the definition and the measurement method. The present invention realizes a power transmission device with good power transmission performance by selecting a capacitor suitable for power factor improvement of a power transmission device that cannot be defined only by the effective series resistance of the capacitor or the dielectric loss tangent tan δ. As described above, the measurement of the capacitor characteristics using the shift ratio S from the zero point of the sine wave voltage across the capacitor, the specific value of the dielectric tangent, the dielectric absorption, the double integral type A / D converter, etc. Capacitors with good performance can be selected regardless of Thereafter, the frequency characteristics shown in FIGS. 86, 87, 88, 89, 105, and 106 are preferably measured and used as an indication of the frequency at which the capacitor can be used.

(Description of capacitor configuration)
107 and 108 are diagrams showing specific capacitor configurations according to the embodiment of the present invention. The capacitor C1 in FIG. 1 which is a circuit configuration diagram of the power transmission device includes a dielectric material 136 and a metal foil 135 as shown in FIGS. 107 (A), (B), 108 (A), and (B). Constructed by winding, or by alternately laminating dielectric material 136 and metal foil 135, and having a capacitance of 1000 pF or less per 1 cm ^{2 of} single layer made of dielectric material and metal foil Is preferably used. In FIG. 107, instead of the metal foil, an electrode may be formed by depositing metal on a dielectric film. A capacitor having such a configuration is called a metallized capacitor, and even if a pinhole is generated in the dielectric film, the metal vapor deposition layer around the pinhole is evaporated to return to a normal characteristic. This function is called self-healing action or self-healing function.

In the embodiment of the present invention, a capacitor in which a dielectric having a high insulation resistance and a high withstand voltage is installed in the conductive gap is used, and the capacitance per 1 cm ² composed of ^two conductive foils and a dielectric material is 1000 pF or less is selected. A capacitor using a dielectric has different characteristics depending on the structure of the capacitor depending on the capacitance value. The voltage at which dielectric breakdown occurs is as low as about 1000 V / mm. Therefore, a capacitor using a dielectric having a high voltage that causes dielectric breakdown must be used. For example, the dielectric breakdown voltage of a plastic film used for a film capacitor is about 1000 V / μm, which is about 1000 times that of air.

In addition, when the inventor of the present application conducted an experiment using various capacitors using the AC power source and the coil according to the embodiment of the present invention, the dielectric was a plastic film and the structure shown in FIGS. 107 and 108 was used. The capacitor had good power transmission performance. However, a capacitor C1z having an operable voltage of several hundred to 1,000 V or more, for example, using barium titanate as a dielectric, may have better power transmission performance than a capacitor using a plastic film as a dielectric. The C1z is a disk having a diameter of several centimeters or more, and the dielectric material is thick and the shape is large. However, when calculated backward from the capacitance value and the disk area, the capacitance per 1 cm ^{2 of} the single layer is about 200 pF or less. Thus, the above-mentioned condition of the electrostatic capacity of 1000 pF or less per 1 cm ^{2 of} the single layer was satisfied. Therefore, it is important to define the capacitance per unit area as a capacitor used in the power transmission device. As will be described later, the electrostatic capacity per unit area is defined as a condition for determining the other when one of the electrode plate interval and the relative dielectric constant of the dielectric is determined.

The inventor of the present application actually wound the foil-like conductor and film of FIG. 107 (B), and has the same dielectric capacity as that of the polypropylene capacitor C1c having the best power transmission performance in Table 2. When a 0.47 μF capacitor was disassembled and back-calculated from the surface area and nominal capacity, the capacitance per 1 cm ^{2 of} single layer area was about 1000 pF or less. The actual measurement results are shown below.

  When a polypropylene capacitor having the same configuration as the capacitor C1c of 0.47 μF was disassembled and the electrode plate dimensions were measured, it was about 2 cm × 250 cm.

The electrode plate area Sc is Sc = 2 cm × 250 cm = 500 cm ² .

Since the capacitance is 0.47 μF = 470000 pF, the capacitance per 1 cm ² is
470000 pF / 500 cm ² = 940 pF / cm ² .

The capacitance C is expressed as C = εo · εs · S / d. Here, εo is a dielectric constant in vacuum, and is a physical constant of εo = 8.85 × 10 ⁻¹² (F / m). εs is a relative dielectric constant (no unit), S is an area (m2) of the electrode plates, and d is a distance (m) between the electrode plates. Transform the above formula,
Substituting the above constants and measured values as d = εo · εs · S / C,
d = 8.85 × 10 ⁻¹² (F / m) · εs · 10 ⁻⁴ m ² / (940 × 10 ⁻¹² F)
d = (8.85 / 940) · εs · 10 ⁻⁴ m = εs · 9.41 × 10 ⁻⁷ m
Considering dimensional measurement error etc., if 9.41 × 10 ⁻⁷ m is 1 × 10 ⁻⁶ m,
According to the data, d = εs · 1 × 10 ⁻⁶ m = εs · μm, the relative dielectric constant εs of the same dielectric film as that of the capacitor C1c is 1.5 to 4, and between the electrode plates The distance d is required to be at least 1.5 μm. As far as the inventors of the present application have experimented, the polypropylene capacitor C1c having the best power transmission performance falls into a category with a large volume for the same capacitance. That is, the power transmission performance of the capacitor tends to improve in proportion to the volume. Therefore, it is preferable that the upper limit of the relative dielectric constant capable of reducing the electrode plate area S is 4.

  The thickness of the dielectric film measured with a micrometer was about 3 μm. Therefore, the dielectric constant εs of the dielectric film is estimated to be about 3 from the above-described induction formula, d = εs · μm, which is consistent with the numerical value of the material.

(Description of capacitor dielectric)
The capacitor dielectric is a polymer of polyimide, polyethylene terephthalate, polycarbonate, polysulfone, polyphenylene sulfide, polyethylene naphthalate, an olefin monomer represented by a general formula CR ₂ CQ ₂ , and CR ₂ CQ ₂ , R and Q are composed of a polymer which is an addition polymer of a monomer having a functional group containing H, or a mixture of at least two of the above-described polymers.

Here, R and Q are functional groups containing H (hydrogen), for example, a single atom such as Cl (chlorine), a functional group such as CH3 (methyl group), or C ₆ H ₅ (phenyl group). , And the like. For example, when all of R ₂ and Q in CR ₂ CQ ₂ are H, the monomer is ethylene and the polymer is polyethylene. When all of R and Q in CR ₂ CQ ₂ are F, the monomer is tetrafluoroethylene, and the polymer is polytetrafluoroethylene (Teflon (registered trademark)). When one of R in CR ₂ CQ ₂ is a phenyl group and the other R and Q are all H, the monomer is styrene and the polymer is polystyrene. When all of R in CR ₂ CQ ₂ are H and all of Q are F, the monomer is vinylidene fluoride and the polymer is polyvinylidene fluoride. In CR ₂ CQ ₂ , when one of R and Q in CH ₂ is CH ₃ and the remaining Q and R are all H, the monomer is propylene and the polymer is polypropylene. By selecting such a dielectric material and appropriately configuring the capacitor, a capacitor having characteristics suitable for improving the power factor of the power transmission device described above can be obtained. For example, a capacitor having good dielectric absorption characteristics and good symmetry of dielectric absorption characteristics can be obtained.

  The above notation is not an official compound nomenclature established by IUPAC (International Union of Pure and Applied Chemistry). The above notation is a commonly used compound name.

  As far as the inventors of the present application have experimented, if the capacitance is 0.01 μF and the transmission power is about 4 W under the above-mentioned power transmission circuit conditions, polypropylene (PP), polyphenylene sulfide (PPS), polystyrene (PS), polycarbonate ( The power transmission performance was good in the order of the capacitor using the PC) film as a dielectric. However, a capacitor having a different configuration using polypropylene (PP) as a dielectric has a lower power transmission performance than a capacitor using a polyethylene naphthalate (PEN) film as a dielectric. Next, a capacitor using polyethylene naphthalate (PEN) or polyethylene (PE) as a dielectric had good power transmission performance. A capacitor using polyethylene terephthalate (PET) as a dielectric is inferior in performance to a capacitor using polyethylene (PE) as a dielectric.

  However, as described above, the polystyrene capacitor may generate heat depending on the configuration and may not be used in the present invention. In such a polystyrene capacitor, the characteristics of the effective series resistance Rc and the dielectric loss tangent tan δ are close to the maximum performance. However, the shift values S and Vp (V) / Vn (V) from the zero point described above were larger than the polyethylene capacitor. Therefore, the power transmission performance by the dielectric of the capacitor is only an experimental result, and it is important to select the capacitor in accordance with the above-mentioned characteristic definition.

  Note that, as described above, a ferroelectric capacitor as a dielectric, for example, a ceramic capacitor, has a dielectric loss tangent, capacitance stability, capacitance temperature characteristics, dielectric absorption, etc. The performance as a capacitor that affects the power transmission performance is poor, and it is not suitable for a power factor improving capacitor of a power transmission device. However, as described above, when the capacitance increases, power transmission performance that is not significantly different from a capacitor using a dielectric having a small relative dielectric constant can be obtained. Even if the capacitors have exactly the same structure using the same dielectric, the power transmission performance differs depending on the capacitance. Further, as described above, there is an upper limit of the usable frequency due to the capacitance of the capacitor. Although it depends on the frequency, as a guideline, when the capacitance is about 0.1 μF or more, it seems that no significant difference is seen in the power transmission performance between the film capacitor and the ceramic capacitor.

  Referring to Table 2, the effective series resistance Rc at 200 kHz is 0.01Ω for the polystyrene capacitor C1a and 0.03Ω for the polypropylene capacitor C1c. That is, as long as the comparison is made with the value of the effective series resistance Rc, the performance of the polystyrene capacitor C1a is better. Similarly, as long as the values of the dielectric loss tangent tan δ, Q are compared, the performance of the polystyrene capacitor C1a is better. However, the value of Vp / Vn, the dielectric absorption K, the dielectric absorption Kp, the symmetry Kr of Kp, the actual power transmission performance, etc. are better for the polypropylene capacitor C1c. Therefore, it is not possible to judge the performance of the power factor improving capacitor of the power transmission device only by the dielectric loss tangent tan δ or Q. It is only a necessary condition that the dielectric loss tangent tan δ of the capacitor is equal to or less than a predetermined value at the predetermined frequency. It is necessary to select a capacitor having a large numerical value of the capacitor characteristic shown on the Y-axis (located on the upper side of the graph) in FIGS. 83 to 85, 93 to 95, 102 to 105, and the like.

  In the power transmission device having the circuit configuration shown in FIG. 1, even if a capacitor having good power transmission performance such as a polypropylene capacitor C1c is used, the power transmission control circuit 3a, the power transmission coil 1, the power reception coil 2, the relative position of both coils, the load Unless a component such as a resistance value is appropriately selected, power transmission performance as shown in FIG. 83 or the like cannot be obtained. Actually, in FIG. 83, the constituent elements other than the capacitor are exactly the same, but the power transmission performance differs depending on the capacitor. Therefore, in order to maintain the power transmission performance when the components other than the capacitor are changed, it is necessary to equip the power transmission device with the power factor improving capacitor of the present invention. Components other than the capacitor, particularly the coil and the load resistance value on the secondary side, have a great influence on the power transmission performance. However, in the embodiment of the present invention, if the capacitance is determined, a capacitor with good power transmission performance can be uniquely selected based on the above-mentioned rules. In particular, the capacitor selected by the shift value S from the zero point of the sine wave, the dielectric absorption K, and the symmetry Kr of dielectric absorption is always superior to other capacitors even if the components other than the capacitor change. Transmission performance can be maintained. By installing such a capacitor, a power transmission device with good power transmission performance can be realized.

  Note that the upper limit for using the capacitor can be defined by the measurement shown in FIGS. 104 and 105 described above. Then, capacitors having good characteristics shown on the Y axis (located on the upper side of the graph) such as those shown in FIGS. 83 to 85, 93 to 95, 98, and 101 to 104 are selected. As described above, the capacitors using the same dielectric have different power transmission performance. Naturally, the power transmission performance varies depending on the dielectric and configuration of the capacitor. However, it is not always possible to use the highest performance capacitor for mounting. Surface mount capacitors require heat resistance during mounting. When a dielectric is exposed to a temperature that exceeds the melting point of the dielectric (exactly the glass transition point, sometimes expressed as the softening point, it is expressed as the melting point), deformation occurs and electrostatic Capacity will fluctuate.

  For such applications, a polyphenylene sulfide capacitor having a high melting point is selected although the power transmission performance is slightly inferior. Alternatively, the mounting may be difficult due to the large physical dimensions of the capacitor. In such a case, a polyethylene naphthalate capacitor having a small physical dimension is selected instead of a polystyrene capacitor or a polycarbonate capacitor having a large physical dimension. As will be described later, the capacitor generates heat when an alternating current flows. In such a case, the capacitors are connected in parallel to ensure a current that can be passed through the capacitor. Since the capacitor connected in parallel has better characteristics than the single capacitor, the characteristics of the capacitor described above are measured by a composite capacitor connected in parallel. This is because at least the effective series resistance Rc can be reduced by connecting capacitors in parallel.

  Moreover, the ceramic capacitor has good heat resistance, and has a power transmission performance that is not so different from that of a film capacitor when the capacitance is 0.1 μF or more. Therefore, by connecting ceramic capacitors having an electrostatic capacity of 0.1 μF or more in series, parallel, and series-parallel, both heat resistance and size reduction can be satisfied. Furthermore, the operable voltage and the passable current can be increased. However, the ceramic capacitor needs to satisfy the above-mentioned regulations. However, the capacitance may be 0.1 μF or less as long as the above-described regulations are satisfied. In general, a chip-shaped (for surface mounting) ceramic capacitor has better characteristics as described above than a lead-shaped ceramic capacitor. The performance of the chip-shaped capacitor is better for the film capacitor.

  By adopting the method of using the capacitor as described above, it is possible to realize a power transmission device with good power transmission performance according to various specifications. A capacitor is selected according to the current flowing through the capacitor. The above-described capacitor having good power transmission performance is generally expensive. By selecting a capacitor by the above-described method, an inexpensive capacitor can be used according to the performance required by the power transmission measure. The embodiment of the present invention described above cannot be defined even in Patent Document 4 which simply describes a dielectric loss tangent or a dielectric.

(Example of circuit configuration)
FIG. 109 is a diagram showing another embodiment of the power transmission device of the present invention. A DC power source 12 and a switching element Q3 are connected in series to the power transmission coil 1, and the switching element Q3 is connected to the control circuit 201. When a unidirectional pulse current is passed through the power transmission coil 1 with the drive waveform VG as shown in FIG. 110 (A) by control, the voltage waveform VL (V) shown in FIG. appear.

  In FIG. 109, it is preferable to connect a capacitor C3 in parallel to the power transmission coil 1 to reduce the circuit current when there is no load and to bring the voltage and current waveform of the power transmission coil 1 closer to a sine wave. The switching element Q3 is driven with the drive waveform VG. The period (frequency) of the drive waveform is constant, and only the time for turning on the switching element Q3 can be changed. By setting it as such a structure, it is prevented that the electric current which flows into the power transmission coil 1 is saturated, and the power loss by the power transmission coil 1 is prevented.

  In addition, by installing the capacitor C3 in parallel with the power transmission coil 1, the voltage waveform and current waveform of the power transmission coil 1 can be brought close to a sine wave, and power loss due to the power transmission coil 1 can be prevented. The capacitor C3 provided in parallel with the power transmission coil 1 is not limited to the capacitor having the characteristics specified in the embodiment as described above, but it is more preferable to use a capacitor having the characteristics as described above. Alternatively, in order to prevent the negative spike voltage due to the counter electromotive force shown in the voltage waveform VL shown in FIG. 110 (B) when the switching element Q3 shown in FIG. It is preferable to connect a capacitor C3 in parallel.

  In the embodiment shown in FIG. 109, the power receiving coil 2 must be equipped with the capacitor C2. However, if the power receiving coil 2 is equipped with the capacitor C2, the power transmission control circuit 3a (drive circuit) of the power transmission coil 1 is not limited to that shown in FIG. Various things such as the embodiment shown in FIG. 72 and a sine wave output can be applied. When the AC power supply 3b of the embodiment shown in FIG. 72 is used, it is preferable to use a power supply whose duty is not fixed at 50% and whose duty is variable. Further, when the power transmission coil 1 is driven by a sine wave, the power transmission coil 1 does not consume any effective power when the power reception coil 2 is not inductively coupled to the power transmission coil 1. This is because all the power supplied to the power transmission coil 1 becomes reactive power. Therefore, there is an advantage that the power transmission coil 1 does not generate any heat even when AC power is supplied to the power transmission coil 1 alone even when power is not transmitted.

  FIG. 111 is a diagram showing another embodiment of the present invention. The circuit configuration is the same as that in FIG. 1, and a capacitor C2 is provided in series between the power receiving coil 2 and the load RL. With the circuit configuration as shown in FIG. 111, the impedance variation on the power transmission side due to the variation in the load resistance value RL can be reduced as compared with the circuit configuration of FIG. The capacitors in the embodiment of the present invention are used for C1 and C2 in FIG. However, C2 in FIG. 111 is mainly installed in a small mounting space such as a power receiving device. In the circuit configuration of FIG. 111, since the frequency range in which predetermined power can be transmitted is wide as described above, the capacitor described in the above embodiment is not necessarily used as the capacitor C2 if the passable current is satisfied. You don't have to.

(Explanation of two-terminal equivalent circuit depending on where the capacitor is equipped)
FIG. 112 is an equivalent circuit diagram in the case where the power transmission coil 1 is equipped with a capacitor in series and a simplified two-terminal equivalent circuit diagram.

(Explanation when a capacitor is installed on the power transmission side)
An equivalent circuit in which the capacitor C1 is connected in series to the power transmission coil 1 is shown in FIG. 112 (A), and a simplified two-terminal equivalent circuit is shown in FIG. 112 (B). The capacitor C1, the residual inductance Le, and , A series circuit of resistors Rx. The inductance of the power transmission coil 1 is indicated by L1, and the inductance of the power reception coil 2 is indicated by L2. This is the same as FIG.

  When a conducting coil is used, it is necessary to consider the two-terminal equivalent circuit of FIG. However, in the power transmission device of the present invention, the frequency fd (Hz) for driving the power transmission coil 1 is the series resonance point fr (Hz) determined by the inductance L1 (H) of the power transmission coil and the capacitance C1 (F) of the capacitor. ). However, as shown in FIG. 45, when the mutual inductance between the power transmission coil and the power reception coil cannot be ignored, the inductance of the power transmission coil decreases from the above-described equation (4). When operating under such conditions, the frequency fd (Hz) for driving the power transmission coil 1 is set higher than the previous period fr (Hz) in consideration of the decrease in inductance of the power transmission coil.

(Example of connecting multiple capacitors)
As described above, in the series circuit shown in FIG. 124, a boosting effect is generated in which a voltage equal to or higher than the power supply voltage Vt is generated across the capacitor. In the circuit configurations of FIGS. 1, 109, and 111, which are the same as the equivalent circuit of FIG. 124, when the operable voltage of the capacitor described in the embodiment of the present invention is low, the same type is used to ensure the operable voltage of the capacitor. Alternatively, a plurality of different types of capacitors may be connected in series. In addition, when the passable current of the capacitor described in the embodiment of the present invention is low, a plurality of same or different types of capacitors may be connected in parallel in order to secure the passable current of the capacitor. Alternatively, in order to secure both the operable voltage and the passable current of the capacitor, a plurality of capacitors of the same type or different types may be connected in series and parallel. As described above, the capacitors are connected in parallel in order to secure a current that can be passed through the capacitor and to suppress the heat generation of the capacitor to 5 ° C. to 10 ° C. or less.

  Preferably, when capacitors are connected in series and parallel, the number of capacitors connected in series is the same as the number of capacitors connected in parallel. FIG. 116 is a connection diagram when capacitors are connected in series and parallel. 116A, 116B, and 116C are connection examples when the number of capacitors connected in series is three and the number of capacitors connected in parallel is two.

  The reason why the value of the capacitor Cd connected in series and in parallel is within ± 20% is to make the voltage applied to each capacitor Cd and the current flowing through each capacitor Cd as close as possible. There are capacitors with an accuracy of 1% or less, but these are expensive. Since the capacitance described in a general capacitor has a deviation of about 5 to 10%, the nominal value of the capacitor Cd may be selected within about ± 20%. For example, on the basis of 0.01 μF, the upper limit is 0.012 μF, and the lower limit is 0.0082 μF. More preferably, capacitors having the same nominal value are used for all Cd. When capacitors of the same type and nominal value are connected in series and parallel, an excessive voltage is applied to the capacitor Cd1 when the connection is made as shown in FIG. 116D, so that such a connection method is not used. When capacitors are connected in series and parallel, there is provided a rule that the number of capacitors connected in series is the same as the number of capacitors connected in parallel.

  If more precise adjustment is required, the capacitor Cc is used as shown in FIG. In order to finely adjust the capacitance of the capacitor, the single capacitor Cd, the composite capacitor Cp, or the capacitor Cc connected in parallel to the Cd constituting the composite capacitor Cp must always satisfy the characteristic definition of the embodiment of the present invention. No, but preferably satisfied. However, the capacitor Cc connected in parallel is also connected to the capacitor Cc connected in parallel when the output frequency of the AC power supply is set to a point where the reactance is zero determined by Lp and C. Cc has an operable voltage performance higher than the voltage Vc (V), and a capacitor Cc connected in parallel with a passable current performance higher than the AC current Ia (A) flowing in the capacitor Cc connected in parallel. It is a condition that it is doing. Note that the composite capacitor Cp is composed of terminals on both the left and right ends shown in each drawing of FIG.

  At the point where the reactance of the series circuit of the coil and the capacitor becomes zero, the current Ia (A) flows through the capacitor due to the resonance action, and the voltage Vr (V) = Vs (V) × Qr is applied. It is important that the effective series resistance Rc of the capacitor is selected to be sufficiently low, and the frequency satisfying the thermal condition is selected for power transmission.

  The voltage Vc (V) across the capacitor may be measured when the maximum power is transmitted in the circuit of FIG. After the actual measurement, when the voltage Vc (V) at both ends exceeds the operable voltage of the capacitor, the capacitors are connected in series as described above. Similarly, the current flowing in the capacitor is measured when the maximum power is transmitted in the circuit of FIG. The alternating current that can be passed through the capacitor may be defined as a ripple current. However, if there is no regulation, the temperature rise of the capacitor is measured as described above. When the temperature rise exceeds 10 ° C., capacitors are connected in parallel. When both the operable voltage and the passable current of the capacitor are exceeded, the capacitors are connected in series and parallel. When the capacitors are connected in series, parallel, and series-parallel, the value of Cd constituting the composite capacitor is selected so that the capacitance of the composite capacitor becomes a predetermined value. And the above-mentioned characteristic definition is measured by the synthetic capacitor.

  Further, an LCR meter 4275A manufactured by Hewlett-Packard Company was used for measuring the effective resistance and inductance of each capacitor described above. In addition, since measurement can be performed only at each point of 1, 2, 4, and 10, intermediate points are interpolated by a graph. A Kenwood oscilloscope, CS-5370, was used for AC waveform measurement. In the AC wave diagram, the peak value is measured with the cursor and entered as a numerical value on the graph. CS-5370 is also used for measuring power transmission performance. For example, the voltage across the non-inductive load resistor connected to the power receiving coil 2 is measured, and the effective power transmitted to the non-inductive load resistor is obtained.

(Embodiment of AC power supply of power transmission device)
FIG. 114 is a circuit diagram showing a power transmission unit of the power transmission device according to the embodiment of the present invention. In FIG. 114, a power transmission unit 31 is connected in series between an AC power source 3b, a power transmission coil Lp that is a primary coil configured by winding a foil-like conductor into a flat spiral, and an AC power source 3b and the power transmission coil Lp. The power transmission capacitor Cp is included, and power is transmitted from the power transmission coil Lp to the power reception coil Lr (not shown). Hereinafter, a portion including the power transmission unit of the power transmission device is referred to as a power transmission device.

  The AC power supply 3b includes at least one DC power supply Vd, a control circuit 3c as control means, switching elements Qa and Qb as first and second semiconductor elements using, for example, a power MOS-FET, and an output terminal T1. , T2. A series circuit of a power transmission capacitor Cp and a power transmission coil Lp is connected between an output terminal T1, which is a first terminal as a common terminal, and an output terminal T2, which is a second terminal. The drains of the switching elements Qa and Qb and the control terminal FB of the control circuit 3 are connected to the output terminal T1. The control circuit 3c is supplied with a positive voltage and a ground voltage (0 V) from the DC power supply Vd. A + voltage is applied to the source of the switching element Qa from the control circuit 3c, the source of the switching element Qb is connected to the output terminal T2, and a ground potential (0 V) is applied from the control circuit 3c. .

  A control signal is supplied from the control terminal Ga of the control circuit 3c to the gate of the switching element Qa, and a control signal is supplied from the control terminal Gb of the control circuit 3c to the gate of the switching element Qb. The switching elements Qa and Qb are configured to be connected in a complementary manner. However, the present invention is not limited to this, and both the switching elements Qa and Qb use N-Ch FETs to connect the source of Qa and the drain of Qb. You may comprise. Moreover, a bipolar transistor and IGBT can also be used depending on the transmission power and the frequency used for power transmission. The same applies to other embodiments described later. In FIG. 114, if the frequency used for power transmission is fs (Hz), Zs is the output impedance of the AC power supply 3b at fs, Rc is the effective series resistance of the power transmission capacitor Cp alone at fs, and Rw is It is the effective series resistance of the power transmission coil Lp alone at fs.

  114 is the frequency at which the reactance of the series circuit of the power transmission coil Lp and the capacitor C determined by the power transmission coil Lp and the capacitor C in FIG. It is preferably set close to the point where the frequency is slightly higher than that, and the impedance of the series circuit is minimized.

  This is because, from the equation (4), when the power receiving coil connected to the load faces the power transmitting coil in a state where the coupling coefficient is high, the inductance on the power transmitting side decreases, so the frequency at which the reactance on the power transmitting side becomes zero is This is because the reactance of the Lp simple substance and the series circuit of C shown in FIG. 114 becomes higher than the frequency at which the reactance becomes zero. When the power receiving coil faces, the inductance of the power transmitting coil decreases.

  FIG. 115 is a diagram showing the control signal of switching elements Qa and Qb shown in FIG. 114 and the waveform of voltage Vout output between output terminals T1 and T2. FIG. 116 is a waveform diagram of a sine wave signal output between the output terminals T1 and T2.

  When the control signal output from the control terminal Ga of the control circuit 3c changes from OFF to ON as shown in FIG. 115 (A), the switching element Qa becomes conductive, and the potential between the output terminals T1 and T2 is changed to FIG. As shown in C), the voltage rises to the level of + Vd of the DC power supply Vd, and the current If flows out from the output terminal T1. When time t1 (S) shown in FIG. 115C elapses, the control signal output from the control terminal Ga is turned OFF, and the switching element Qa is turned off.

  Thereafter, when time t3 (S) elapses, the control signal output from the control terminal Gb changes from OFF to ON, as shown in FIG. As a result, the switching element Qb becomes conductive, the output terminal T1 becomes the ground potential, and the current Ir from the energy stored in the coil is drawn into the output terminal T1. When time t2 (S) elapses, the control signal output from the control terminal Gb changes from ON to OFF, and the switching element Qb becomes non-conductive.

  As described above, the gates of the switching elements Qa and Qb are connected to the control circuit 3c so that the time t1 (S) for flowing current from the output terminal T1 and the time t2 (S) for drawing current from the output terminal T1 are alternately present. And a square wave whose peak value changes between 0 V and + Vd is output between the output terminals T1 and T2.

  Normally, since the ON time and OFF time of the switching element are asymmetric, when a signal for turning on the switching element Qa and turning off the switching element Qb is applied to each gate at the same time, both the switching elements Qa and Qb Although it is a short time, it may become conductive at the same time. Therefore, the DC power supply Vd is short-circuited, an excessive current flows through the switching elements Qa and Qb, the ON resistance of the switching elements Qa and Qb consumes power, and the switching elements Qa and Qb generate heat.

  In order to avoid this state, a signal for turning off the switching element Qb is applied to the gate of the switching element Qb, and a signal for turning on the switching element Qa after the switching element Qb is completely turned off is supplied to the gate of the switching element Qa. Add to. Therefore, a time t3 ≧ 0 (S) in which both the switching elements Qa and Qb are turned off and the output of the output terminal T1 becomes high impedance is provided.

  By adjusting the time t3 (S) with respect to the time t1 (S) during which current is flowing out from the output terminal T1 and the time t2 (S) during which current is drawn into the output terminal T1, the action of the reactive element is achieved. As a result, the circuit voltage and current approach a sine wave, so that the power transmission coil Lp can be appropriately driven, and an alternating current that can generate sufficient magnetic flux through the power transmission coil Lp can flow through the power transmission coil Lp. Moreover, the electric power supplied to the power transmission coil Lp can be adjusted by increasing / decreasing the time of t3 (S).

  In the embodiment shown in FIGS. 114 and 115, t3 is provided for the time when the power supply output becomes high impedance. However, the switching speed of the switching elements Qa and Qb is higher than that of t1 and t2. When the speed is sufficiently high, approximately 50 times or more, it is not particularly necessary to provide t3. By not providing t3, the control circuit 3c can be simplified and the cost can be reduced. For example, when the output current is about 1 A, the switching element, Qa, Qb, having a high switching speed can be used, and the condition of Zs ≦ (Rw + Rc) described later may be satisfied. Therefore, it is not necessary to provide the t3.

  The output impedance Zs (Ω) at the frequency fs (Hz) for transmitting AC power of the AC power supply 3b shown in FIG. 114 is the effective series resistance of the power transmission coil Lp alone at fs, Rw (Ω), and the fs. When the effective series resistance of the power transmission capacitor Cp alone is Rc (Ω), at least the condition of Zs ≦ (Rw + Rc) is satisfied.

  If this condition is not satisfied, the loss due to the internal impedance Zs of the AC power supply 2 becomes larger than the power loss due to the effective series resistances Rw and Rc, and power cannot be efficiently transmitted from the AC power supply 3b to the power transmission coil Lp.

  When Zs = Rw + Rc, it can be used in a special case under the condition that the maximum power can be supplied from the power source having the internal impedance Zs according to the so-called maximum power supply law. However, when Zs> (Rw + Rc), the power consumed by the output impedance Zs exceeds the power consumed by Rw + Rc, and the power that can be supplied to the load decreases, so that Zs = (Rw + Rc) is satisfied. You must keep in mind.

  However, the operational effect of the present invention is long-distance power transmission, and is not intended to improve power transmission efficiency. Therefore, in general, the invention can be implemented as long as Zs is about 4 times or less of (Rw + Rc). In this case, only about 20% of the AC power can be supplied to the power transmission coil. However, in order to realize the effect of the present invention, there is no problem even if there is a power loss inside the AC power supply. However, in this case, the power source capacity of the DC power source supplied to the AC power source becomes excessive. Therefore, the value of Zs is preferably as small as possible.

  Preferably, in FIG. 114, Zs << (Rw + Rc) is satisfied.

For example, if the open circuit voltage of the AC power supply 3b is Vs, Zs = 0.1Ω, Rw + Rc = 0.9Ω, the load current Ir at the series resonance point is
Ir = Vs / (Zs + Rw + Rc) (A), and the load power Pr is
Pr = Ir ² × (Rw + Rc) (W).

The power Pd consumed by Zs of the AC power source 2 is the same as above.
Pd = Ir ² × Zs (W)
The load power, Pr, is Pr = 0.9 × Ir ² (W),
The power consumed in the AC power supply 3b, Pd, is Pd = 0.1 × Ir ² (W).

That is, 90% of the electric power output from the AC power supply 3b, (0.1 + 0.9) × Ir ² (W), is transmitted to the series circuit of the power transmission coil and the capacitor, and the AC power supply 2 inside The power that becomes the loss at 10% is 10%. That is, using the symbols Zs, Rw, Rc, the power transmission efficiency from the AC power source 2 to the power transmission coil, ηt,
ηt = (Rw + Rc) / (Zs + Rw + Rc)
= 1 / (Zs / ((Rw + Rc) +1)),
It is understood that the above-mentioned ηt increases as the value of Zs / (Rw + Rc) is satisfied and Zs << (Rw + Rc) is satisfied.

Therefore, preferably, Zs / (Rw + Rc) ≦ 4, (Zs loss 20%),
More preferably, Zs / (Rw + Rc) ≦ 9, (Zs loss 10%),
More preferably, it satisfies Zs / (Rw + Rc) ≦ 19 and (Zs loss 5%).

  Looking at the LC series circuit from the AC power source, it becomes a pure resistance, Rw + Rc at the resonance point. However, when the LC series circuit is driven by a square wave, the square wave has a harmonic component that is an odd multiple, Distortion occurs in the waveform.

  Therefore, the value of Zs / (Rw + Rc) is preferably as low as possible. The characteristics of the AC power supply are important characteristics of the AC power supply in the present invention and also in a coil capable of transmitting high power of other configurations with high efficiency. In the conventional technology not yet understood, the characteristics of the AC power supply are not mentioned at all.

  It is necessary to convert the output impedance of the transmitter into the impedance of the antenna inside the transmitter for a single unit such as an antenna having a specific resonance frequency and specific impedance. It has a fundamentally different operational effect from an antenna having a function of emitting electromagnetic waves of a specific wavelength (frequency) into the air or capturing electromagnetic waves of a specific wavelength (frequency). That is, in the case of an antenna, if the output impedance of the transmitter does not match the intrinsic impedance of the antenna, loss due to standing waves occurs. On the other hand, since an ideal coil alone has neither a specific resonance frequency nor a specific impedance, impedance matching is not required and no standing wave is generated.

  As long as the coil using the mutual induction action is used, as described above, matching with the intrinsic impedance of the antenna is performed so that half of the power supplied from the transmitter is lost (impedance conversion is performed). No need. At a frequency fs (Hz) used for power transmission at which reactance is zero, an effective impedance of the coil is Rw (Ω), an effective resistance of the capacitor is Rc (Ω), and the internal impedance Zs is sufficiently lower than (Rw + Rc). By using an alternating current power supply, the efficiency of transmitting power from the alternating current power supply to the power transmission coil can theoretically be close to 100%.

  The mutual induction action and the action of the antenna are fundamentally different in operation principle, but both are confused, and there is no prior art that explicitly mentions the difference. For example, in Japanese Patent Laid-Open No. 2000-348152, a loop-shaped coil that receives power by mutual induction is described as “antenna”, and the coil and the antenna are not clearly distinguished.

The control circuit 3c of FIG. 114 not only makes the switching elements Qa and Qb conductive / non-conductive, but also controls the gate voltages of the switching elements Qa and Qb, and generates a sine wave as shown in FIG. It can also be generated. In the sine wave shown in FIG. 116, since Qa and Qb are both conducting at a point where the output voltage is zero or other than the maximum value, the output impedance Zs of the AC power source is reduced, and the output terminal T1-T2 is connected. If an attempt is made to secure a flowing current, a current that short-circuits the DC power supply Vd flows through the ON resistances of the switching elements Qa and Qb, and a loss due to the ON resistances of the switching elements Qa and Qb occurs. In such a case, since the output impedance Zs cannot be obtained by calculation, for example, when a non-inductive resistor R (Ω) is connected between the output terminals T1 and T2 in FIG. 114 at the frequency fs at which power is transmitted. When the voltage between the output terminals T1 and T2 is Vr, and the voltage when the output terminals T1 and T2 are opened is Vn,
Since Vn = Vd and Vr = Vd (R / (R + Zs)),
Vr = Vn / (R / (R + Zs)), R + Zs = Vn / (R · Vr)
Zs = Vn / (R · Vr) −R
For example, if R = 1Ω, Zs = Vn / (Vr) −1 (Ω)
If R = 2Ω, Zs = Vn / (2 · Vr) −2 (Ω)
As a result, output impedance Zs is obtained. This is not limited to FIG. 116, and is the same for the square wave of FIG. 115 and the stepped wave shown in FIG. 122, which will be described later, and the output impedance Zs varies depending on the frequency. The same applies to other embodiments.

  An example of a method for reducing the output impedance of an AC power source that generates a sine wave is disclosed in Japanese Patent Laid-Open Nos. 5-22048 and 5-22049, but the advantage of using a sine wave is Circuit theory can be applied to elements and circuits, and the difference between measured values and theoretical values can be measured. A power supply with a sine wave output having a low output impedance Zs is also required to measure the effective resistance of the capacitor. However, it is necessary to measure the output impedance Zs of the sine wave output AC power supply by the method described above.

  In this case, since the first and second semiconductor elements operate in class A depending on the bias point, the DC-AC conversion efficiency is lowered, but since they are used in the non-saturation region, they do not have high-speed switching characteristics. Even in a semiconductor element, the output impedance can be lowered.

  Note that the class B operation and the class C operation make the output impedance somewhat higher than the class A operation and cause distortion in the output waveform, but the DC-AC conversion efficiency can be increased. The LC series circuit has a function of returning the waveform to a sine wave, and the action is particularly strong in the vicinity of the resonance point, and a class B operation or a class C operation can also be adopted. When the output frequency of the AC power supply is low, the DC-AC conversion efficiency can be further increased by adopting the class D operation.

  As described above, the present invention enables long-distance power transmission even if the power transmission efficiency is low. Therefore, the DC-AC conversion efficiency does not become a problem, and the output impedance is lowered and sufficient for the coil. In order to flow current, a technique of lowering the output impedance using an impedance converter or the like, or a technique of lowering the output impedance by performing feedback control from the output of the AC power source as shown in FIG. In FIG. 114, even when the ON resistances of the switching elements Qa and Qb are large and the output impedance Zs of T1 is large, Zs is defined as Zs = ΔE / ΔI. If the voltage is constant and ΔE is zero, Zs is theoretically zero. Actually, Zs does not become zero, but theoretically it is known as the operating principle of an operational amplifier or a constant voltage power supply. As described above, the actual Zs is obtained from the difference in output voltage between when the AC power supply terminal is open and when a non-inductive resistor is connected.

  FIG. 117 is a circuit diagram showing a power transmission section 32 in another embodiment of the present invention. In the embodiment shown in FIGS. 114 to 116, a square wave whose peak value changes between 0 V and + Vd is output from the AC power supply 3b, whereas the embodiment shown in FIG. The AC power supply 3b1 is configured to output a square wave whose peak value changes between −Vd and + Vd.

  That is, the control circuit 3c outputs control signals from the control terminals Ga and Gb and also outputs control signals from the control terminals Gc and Gd in the same manner as the control circuit 3c shown in FIG. Control signals output from the control terminals Gc and Gd are given to the gates of the switching elements Qc and Qd. For example, power MOS-FETs are used for the switching elements Qc and Qd.

  The drains of the switching elements Qc and Qd are connected to the control terminal FB2 of the control circuit 3c and to the output terminal T2. The source of the switching element Qc is connected to the source of the switching element Qa, and the source of the switching element Qd is connected to the source of the switching element Qb. Connections between the switching elements Qa and Qc and the control circuit 3a are the same as those in FIG. The switching elements Qa and Qc cause the current If to flow, and the switching elements Qb and Qd function to draw the current Ir.

  118 shows a waveform for controlling the gates of switching elements Qa to Qd in FIG. 117 and an output waveform output between output terminals T1 and T2.

  As shown in FIGS. 118A and 118D, when the control signals Ga and Gd change from OFF to ON, the switching elements Qa and Qd become conductive, and from the positive terminal of the DC power source Vd only for the time t1 (S). The voltage + Vd is output from the control circuit 3c to the output terminal T1 via the switching element Qa. A current flows in the direction of If from T1 to Cp, Lp, the output terminal T2, and from the switching element Qd to the negative terminal of the DC power supply Vd via the control circuit 3c. Since the control signals Ga and Gd are turned OFF after the elapse of time t1 (S), the switching elements Qa and Qd are turned off, and are in a high impedance state during the time t3 (S). Thereafter, as shown in FIGS. 118B and 118C, the control signals Gb and Gc change from OFF to ON. Then, the switching elements Qb and Qc become conductive, and the voltage + Vd is output from the + terminal of the DC power supply Vd to the output terminal T2 from the control circuit 3c via the switching element Qc only during time t2 (S) and from the output terminal T2. A current flows in the direction of Ir from Cp, Lp and the output terminal T1 to the negative terminal of the DC power supply Vd from the switching element Qb via the control circuit 3c. That is, at time t1, the output terminal T1 becomes + Vd and the output terminal T2 becomes GND, and at time t2, the output terminal T1 becomes GND and the output terminal T2 becomes + Vd. By repeating this operation, the output terminals T1 and T2 In the meantime, as shown in FIG. 118 (E), a peak value having two values of + Vd and -Vd is output as the output voltage Vout. As a result, a square wave having an amplitude twice that of the output voltage Vout shown in FIG. 115 (C) is obtained, so that the reactance of the series circuit of the power transmission coil Lp and the power transmission capacitor Cp becomes zero. A large current can be passed through the series circuit.

  In these embodiments, the output frequency of the AC power supply 3b is set close to the point where the reactance is zero or the impedance is minimized, which is determined by the power transmission coil Lp and the power transmission capacitor Cp.

  The point where the reactance is zero, which is determined by the power transmission coil Lp and the power transmission capacitor Cp, is a series resonance point. However, since an actual coil or capacitor has an effective series resistance or an effective parallel resistance, the point where the reactance is zero and the impedance Is slightly different, and the point where the impedance is minimized is slightly higher than the point where the reactance is zero. The frequency fs is a frequency equal to or less than f1, which is a frequency region in which the above-described coil can be used. The effect of setting fs to f1 or less is as described in the embodiment relating to the coil.

  The power transmission units 31 and 32 in the embodiment shown in FIGS. 114 and 117 can form a magnetic flux over a wide range around the power transmission coil Lp, and have an arbitrary configuration with an area smaller than the area of the power transmission coil Lp. The power receiving coil Lr has the effect of being a power transmission device capable of transmitting power over a long distance.

  In this case, since the coupling coefficient between the two coils is small, the power transmission side is not affected by the power reception side, so the frequency of the AC power supplies 3b and 3b1 is set so that the AC current flowing through the power transmission coil Lp is maximized. It may be set close to the point where the reactance determined by Lp and the power transmission capacitor Cp is zero or the impedance is minimized, and it is complicated as in a conventional power transmission device that uses a capacitor to improve the power factor. No control circuit is required.

In these embodiments, the thermal resistance of each element is θq (° C./W), the operable temperature of each element is Tq (° C.), the ambient temperature of the place where each element is installed is Ta (° C.), If the average effective current flowing through the element is Ia (A), and the ON resistance of each element Q is Rq (Ω), then all the elements are (Tq−Ta) ≧ θq (Rq × Ia ² ), When the voltage across the switching elements Qa, Qb, Qc, and Qd is Vq (V), all the elements satisfy (Tq−Ta) ≧ θq (Vq × Ia). .

The above conditions (Tq−Ta) ≧ θq (Rq × Ia ² ) and (Tq−Ta) ≧ θq (Vq × Ia) are thermal conditions that are also defined in the power transmission coil Lp. Regulates the usable thermal conditions.

The method for measuring the thermal resistance of the coil, θi, using a DC constant current source has already been described. First, the thermal resistance of the semiconductor element, θq, will be described. Usually, the thermal resistance needs to solve the thermal diffusion equation in consideration of the material and shape of the element. However, as described above, since it is often difficult to solve the thermal diffusion equation, when the initial element temperature is Ta, the element consumes a certain amount of power P (W), and a steady state of thermal equilibrium is reached. The difference in temperature between Ta and Ta is Tr,
Assuming that θq = Tr / P (° C./W), the thermal resistance is approximately obtained.

  In the case of an active element such as a transistor or FET, the thermal resistance θq is very large as a single unit and is 50 to 90 (° C./W). However, by providing a heat sink, the thermal resistance is several ° C./W, It can be reduced to: For active devices such as transistors and FETs, standards are defined, and the thermal resistance of a single unit is often specified without actually determining the thermal resistance by experiment, and what shape is used as a heat sink In many cases, it is also specified whether or not dimensions are required.

  For simplicity, since the TO-220 package has almost the same thermal resistance, it has a resistance of 5Ω on the output of the 7805 three-terminal regulator U1 (not shown) with an output voltage of 5V attached to the heat sink. Since a constant current of 1 A flows through the output of U1, if the input voltage is 15 V, the power P consumed by the 7805 is P = (15−5) V × 1A of about 10 W. If the temperature of 7805 rises by 30 ° C. in a steady state where thermal equilibrium is achieved, the thermal resistance θq becomes θq = 30 ° C./10 W = 3 (° C./W).

In the case of the FET, the product of the ON resistance Rq and the square of the current Ia flowing between the drain and source is the element loss Pq (W). Therefore, the thermal resistance θq thus obtained is added to the loss Pq. , The element temperature rise value Tr (° C.) is obtained. Tr = θq × Rq × Ia ² (° C.), where Tw (° C.) is the temperature at which the element can be operated, and Ta (° C.) is the ambient temperature where the element is installed, Tr = Tw−Ta. If (Tw−Ta) ≧ θq (Rw × Ia ² ) is not satisfied, the usable temperature of the element is exceeded, making it difficult to implement this embodiment.

Similarly, in the case of a bipolar transistor, the element loss Pq (W) is obtained by multiplying the collector-emitter saturation voltage, Vq (V), and the current Ia (A) flowing between the collector and emitter. When the thermal resistance θq is multiplied by Pq, the temperature rise value Tr (° C.) of the element is obtained. As in the case of FET, Tr = Tw−Ta, and the inequality,
(Tq−Ta) ≧ θq (Vq × Ia) must be satisfied.

  FIG. 119 is a diagram illustrating a circuit configuration of the AC power supply 3b2 of the power transmission unit 33 illustrating an example of an embodiment in which the problem of FIG. 9 of Patent Document 6 is solved. In FIG. 119, AC power supply 3b2 includes switching elements Q1 to Q4, reverse diodes D1 to D4, and diodes D11 to D14. Parasitic reverse diodes D1 to D4 exist between the drains and the sources of the switching elements Q1 to Q4, respectively, and the drains are connected to the output terminal T1 via the diodes D11 to D14. The output terminal T2 is grounded.

  When the switching elements Q1 to Q4 are turned on, the DC power sources V1 to V4 are prevented from being short-circuited by the reverse diode diodes D11 to D14. The DC power sources V1 and V2, V3, and V4 are connected in series, and the negative side of the DC power source V2 and the positive side of the DC power source V3 are grounded. Each source of switching elements Q1, Q2 is connected to the + side of DC power supplies V1, V2, and each source of switching elements Q3, Q4 is connected to the-side of DC power supplies V3, V4. Control signals G1 to G4 are given to the gates of the switching elements Q1 to Q4 from a control circuit (not shown).

  120 is a diagram showing a control signal of switching elements Q1 to Q4 shown in FIG. 119 and a waveform of voltage Vout output between output terminals T1 and T2.

  As shown in FIG. 120 (A), when the control signal G1 is turned from OFF to ON, the switching element Q1 becomes conductive, and a voltage + V × 2 obtained by adding the DC power sources V1 and V2 is output to the output terminal T1 through the switching element Q1. Is done. As shown in FIG. 120 (B), when the control signal G2 is turned ON, the switching element Q2 is turned on, and the DC voltage + V of the DC power supply V2 is output to the output terminal T1 via the switching element Q2.

  As shown in FIG. 120C, when the control signal G3 is turned ON, the switching element Q3 is turned on, and the DC voltage −V of the DC power source V3 is output to the output terminal T1 via the switching element Q3. As shown in FIG. 120D, when the control signal G4 is turned ON, the switching element Q4 is turned on, and the added voltage −V × 2 of the DC power sources V3 and V4 is output to the output terminal T1 via the switching element Q4. Is done.

  Therefore, as shown in FIG. 120 (E), a staircase waveform close to a sine wave is obtained between the output terminals T1 and T2, and the switching elements Q1 to Q4 operate only in conduction and non-conduction, so that the DC power supply The efficiency of converting DC power from V1 to V4 into AC power can be made extremely high.

  Note that the times ta, tb, and tc during which the control signals G1 to G4 are turned on and off are in high impedance, and when t3 = ta + tb + tc, the time t1 at which a potential of + appears between the output terminals T1 and T2. The time t2 when the potential of − appears is selected so that t3 <t1 and t3 <t2.

  FIG. 121 is a diagram illustrating a circuit configuration illustrating an example of the power transmission unit 34 obtained by modifying the AC power supply 3b2 of FIG. 119. This embodiment is an example in which a ground reference potential is provided, and is configured such that the output terminal T1 is connected to the ground potential and the output terminals T1 and T2 do not become high impedance.

  121, as in FIG. 119, AC power supply 3b2 includes switching elements Q1 to Q6, reverse diodes D1 to D6, and diodes D11 to D16. Reverse diodes D1 to D6 are connected between the drains and the sources of the switching elements Q1 to Q6, respectively, and the drains are connected to the output terminal T1 via the diodes D11 to D16. The output terminal T2 is grounded.

  The DC power sources V1 to V4 are respectively connected in series, and the negative side of the DC power source V2 and the positive side of the DC power source V3 are grounded. The sources of switching elements Q1 and Q2 are connected to the + side of DC power supplies V1 and V2, respectively. The sources of switching elements Q3 and Q4 are grounded. The sources of switching elements Q5 and Q6 are connected to DC. The power sources V3 and V4 are respectively connected to the negative side. Control signals G1 to G6 are given to the gates of the switching elements Q1 to Q6 from a control circuit (not shown).

  122 is a diagram showing a control signal of switching elements Q1 to Q6 shown in FIG. 121 and a waveform of voltage Vout output between output terminals T1 and T2. As shown in FIG. 122 (A), when the control signal G1 is turned from OFF to ON, the switching element Q1 becomes conductive, and a voltage + 2V obtained by adding the DC power sources V1 and V2 is output to the output terminal T1 through the switching element Q1. . When the control signal G2 is turned on as shown in FIG. 122 (B), the switching element Q2 is turned on, and the DC voltage + V of the DC power supply V2 is output to the output terminal T1 via the switching element Q2.

  As shown in FIG. 122C, when the control signal G3 is turned ON, the switching element Q3 becomes conductive, and the ground potential is output to the output terminal T1 via the switching element Q3. As shown in FIG. 122D, when the control signal G4 is turned ON, the switching element Q4 is turned on, and the ground potential is output to the output terminal T1 via the switching element Q4. As shown in FIG. 122 (E), when the control signal G5 is turned ON, the switching element Q5 becomes conductive, and the DC voltage −V of the DC power supply V3 is output to the output terminal T1 via the switching element Q5. When control signal G6 is turned on as shown in FIG. 122 (F), switching element Q6 is turned on, and voltage -2V obtained by adding DC power supplies V3 and V4 is output to output terminal T1 via switching element Q6. .

  Therefore, as shown in FIG. 122 (G), a staircase waveform close to a sine wave is obtained between the output terminals T1 and T2, and the switching elements Q1 to Q6 operate only in conduction and non-conduction, so that the DC power supply The efficiency of converting the DC voltage from V1 to V4 into the AC power source can be extremely increased, and the size of the heat sink provided in the semiconductor element can be reduced, or the semiconductor element can be used alone.

  Note that the times ta, tb, tc, td, te, and tf during which the control signals G1 to G6 are turned on and off are in a high impedance state. The time t1 (S) at which the potential appears and the time t2 (S) at which the − potential appears are selected so that t3 <t1 and t3 <t2.

  In the AC power supply of the embodiment of FIGS. 119 and 121, the output frequency fo (Hz) is fo = 1 / p, assuming that t = t1 + t2 + t3 (S) from t1, t2, and t3 in FIGS. And fo is set close to a point determined by the power transmission coil Lp and the power transmission capacitor Cp, where the reactance is zero or the impedance is minimized.

  In the above embodiment, since the power transmission coil of the present invention is designed based on the required performance, the inductance is a fixed value, but the frequency of the AC power supply and the capacitance of the capacitor can be varied, so the coil is driven. An approximate frequency may be determined and set to a point where reactance is zero or impedance is minimized with either an AC power supply or a capacitor.

  Further, if the output frequency of the AC power source can be varied and the frequency of the AC power source is varied in the vicinity of the resonance frequency fr, the current flowing through the coil changes, so that the power supplied to the coil can be adjusted. The power transmission unit is a simple series resonance circuit, is less affected by the load, and has a small coupling coefficient between the power reception coil and the power transmission coil. Therefore, the reactance of the power transmission coil is substantially constant.

  When a receiving coil whose area is about 1/5 of that of the power transmission coil is placed at the center of the power transmission coil, the resonance point is shifted and the current flowing through the power transmission coil is reduced. However, in such a state, since the coupling coefficient between the power transmission coil and the power reception coil is high, sufficient power can be sent to the power reception coil.

In the embodiment of FIGS. 119 and 121, it is necessary that D1 to D6 satisfy the condition of (Tq−Ta) ≧ θq (Vq × Ia), and (Tq−Ta) ≧ θq (Rq × Ia). ² ) Q1 to Q6 must be satisfied. As described above, Ia is an average effective current flowing through each element and flows separately in each element. However, it should be noted that Vq varies depending on element variations of D1 to D6, and Rq varies depending on element variations of Q1 to Q6.

  As described above, by using the AC power source of the embodiment as shown in FIGS. 114, 117, 119, and 121 for the power transmission unit, a sufficient AC current is caused to flow through the power transmission coil Lp in which the power transmission capacitors Cp are connected in series. The power transmission coil Lp can form a magnetic flux in a wide range.

  Note that the DC power sources Vd, V1 to V4 described in FIGS. 114, 117, 119, and 121 may use batteries or may be generated from a commercial AC power source.

  FIG. 123 is an example of a circuit diagram for generating DC power supplies V1-V4 in FIGS. 119 and 121. In FIG. 122, a commercial power supply voltage is applied to a power transmission coil (not shown) of the transformer PT, a predetermined voltage is stepped down from the secondary coil, and the AC voltage stepped down by the diodes D21 to D28 is rectified, whereby a direct current voltage + V × 2 , + V, 0V, −V, −V × 2 can be taken out.

  If a power source whose output is insulated from the commercial AC power source on the input side is used, a switching power source or the like may be used. In addition, since the power transmission coil and the power reception coil are essentially insulated in this method, the direct current power supply can be directly generated from the alternating current power supply by a PWM step-down method or the like.

The embodiments of FIGS. 114, 117, 119, and 121 are only examples of circuit configurations, and are clearly shown in FIGS. 114, 117, 119, and 121 in the transmission coil Lp in both directions of If and Ir. An AC power supply that allows a current to flow, wherein Zs, Rw, and Rc satisfy Zs ≦ (Rw + Rc) and satisfy the inequality, (Tw−Ta) ≧ θq (Rw × Ia ² ), When the inequality, (Tq−Ta) ≧ θq (Vq × Ia) is satisfied and the conversion efficiency from the DC power supply to the AC power supply does not matter, the output waveform of the AC power supply is a square wave or a staircase wave. The present invention is not limited, and the present invention is not limited to the above embodiment, and various embodiments can be realized.

For example, in FIG. 114, a self-excited oscillation can be performed by providing a third coil electromagnetically coupled to the power transmission coil Lp and applying positive feedback from the third coil to the control circuit 3. In the present invention, since a constant current is normally supplied to the power transmission coil Lp and no special control is required, a circuit configuration in which currents can be supplied to the transmission coil Lp in both directions of If and Ir, the Zs, The Rw and the Rc satisfy Zs ≦ (Rw + Rc) and satisfy the inequality, (Tw−Ta) ≧ θq (Rw × Ia ² ), or the inequality, (Tq−Ta) ≧ θq (Vq × Various self-excited oscillation circuits can be used as long as Ia) is satisfied.

  However, the provisions of t1, t2, and t3 are conditions that must be satisfied at the locations of the output terminals T1 and T2 of the AC power supply as shown in FIGS. 115, 118, 120, and 122. It is not a regulation of the time for controlling the gates of the switching elements Q1 to Q6. It is also possible to adjust the power supplied to the coil by changing the ratio of t1, t2, t3. In this case, it is not always necessary to satisfy the conditions of t3 <t1, t3 <t2. .

  Note that paragraph No. 0025 and FIG. 3 of Japanese Patent Laid-Open No. 2005-6459 disclose a control circuit that controls the gates of FETs that are switching elements and turns off the two FETs. 3 is that when the gate signal of one FET is turned OFF, the gate of the other FET is only turned ON. As shown in the embodiment of FIG. 114, the control circuit detects the signal from the drain output. Unless the gate signal is controlled in consideration of the ON time and OFF time of the FET, there is no guarantee that the time when both FETs are OFF exists reliably in FIG.

  That is, considering the delay time of a normal logic IC, in FIG. 3 of Japanese Patent Application Laid-Open No. 2005-6459, it is detected that the gate signal of one FET has been turned OFF, and then the gate of the other FET is turned ON. The time required for the power FET is 20 nS or less, and the power FET ON time and the OFF time are usually different by several tens of nS or more, although a capacitor delay circuit is provided.

  Japanese Patent Laying-Open No. 2005-6459 is an invention that does not prescribe a high impedance time, originally detects the state of the power receiving side, and changes the power transmission frequency to appropriately control the state of the power receiving side. Thus, the basic power transmission performance is not improved as in the present invention, and the power that can be supplied to the power transmission coil is 5 W at most. As will be described later, the present invention can be used even at a frequency of 1 MHz or higher. It is not a circuit suitable for supplying power of 10 W or more to the coil.

  According to the embodiment of the present invention, it is possible to generate magnetic flux spreading in all directions of the coil, and long-distance power transmission is possible. For example, a power transmission coil having a size of about B4 paper size can be used, and power can be transmitted to the power reception coil equipped with the LED load only for the LED to turn on at a distance of 50 cm or more. In the same plane as the coil winding surface, Power sufficient to light the LED can be transmitted to the power receiving coil equipped with the LED load in an area that is more than twice the area of the power transmitting coil, centering on the power transmitting coil. Alternatively, substantially uniform power can be supplied to the power receiving coil over almost the entire surface of the power transmitting coil. Moreover, even if the winding surface of a power transmission coil and the winding surface of a receiving coil are not parallel, and it is arbitrary angles and positional relationship, electric power can be transmitted.

  In order to realize the above performance, a square wave having a peak value of 10V is sufficient to drive the power transmission coil. For example, a general-purpose high-speed C-MOS logic buffer operating at 5V is connected in parallel. A sufficiently low power source impedance can be obtained, and an AC power source with low power loss can be realized at low cost. In order to achieve long-distance power transmission performance, it is necessary to drive the coil at a high frequency, but if an AC power supply with this configuration is used, an AC power supply that operates stably up to about 10 MHz can be realized at low cost. it can.

  In addition, according to the present invention, in order to improve the power factor, by defining the relationship between the effective series resistance, the effective parallel resistance, and the reactance of the capacitor that cancels the positive reactance component, the optimum capacitor at the frequency at which power is transmitted Or an optimum frequency for transmitting power can be selected.

  As described above, in the embodiment of the present invention, since power is transmitted between coils having a large area ratio, the coupling coefficient between the power transmitting coil and the power receiving coil is small. For this reason, since the power transmission coil is not affected by the power receiving side, the AC power source may be fixed at a frequency determined by the power transmission coil and the capacitor so that the reactance is zero, and feedback control, AC power source and power transmission coil Complicating means such as comparing the voltage phase, detecting the state of the power receiving side, and adjusting the AC voltage and frequency of the AC power supply is not required. Therefore, the power transmission device can be realized at low cost.

  Furthermore, the present invention provides a method for eliminating a phenomenon in which power transmission performance deteriorates when a metal body is close to an air-core coil. In addition, by installing a metal frame larger than the power transmission coil close to the power transmission coil, it is possible to expand the power transmission range to the size of the metal frame at least on the same plane as the winding surface of the coil.

  Specifically, for example, as described above, by supplying effective power of 10 W to a power transmission coil of about B4 size, a capacitor is connected to a power reception coil in which a formal wire is wound 50 turns in a disk shape of about 5 cm in diameter. Power can be transmitted over a distance of about 50 cm or more on the power transmission coil to a power receiving side device in which LEDs are connected in parallel. Further, the power receiving side device can transmit power at any angle with respect to the winding surface of the power transmission coil.

  Furthermore, a non-contact IC card for long-distance power transmission that is generally used can transmit several tens of watts of power with a transmission power of several tens of watts over a distance of several tens of centimeters. It is capable of transmitting near 0.5 W of power with a transmission power of about 10 W over a distance of several tens of centimeters.

  As mentioned above, although embodiment of this invention was described with reference to drawings, this invention is not limited to the thing of embodiment shown in figure. Various modifications and variations can be made to the illustrated embodiment within the same range or equivalent range as the present invention.

  The power transmission device, the power transmission device of the power transmission device, and the power reception device of the present invention can be used to transmit power from the power transmission unit to the power reception unit, and in particular, transmit power to a power reception unit that is separated from the power transmission unit. Available to: Moreover, even if the distance between the power transmission unit and the power reception unit is several tens of centimeters or more and the relative angle between the power transmission coil and the power reception coil varies, the present invention can be used for applications in which the transmittable power does not vary.

It is a block diagram of the electric power transmission apparatus in one Embodiment of this invention. It is a perspective view which shows the foil-like conductor coil used for the power transmission part of the electric power transmission apparatus in one Embodiment of this invention. It is a figure which expands and shows the cross section in alignment with line AA of the coil shown in FIG. It is a figure which shows an example of the receiving coil used for the power transmission apparatus of this invention. It is a figure which shows the other example of the receiving coil used for the power transmission apparatus of this invention. It is an equivalent circuit diagram for obtaining the input impedance of the transformer. It is an equivalent circuit diagram of the air-core coil simple substance of the electric power transmission apparatus in one Embodiment of this invention. FIG. 125 is an equivalent circuit diagram of a transformer portion configured as shown in FIG. 124 described in the conventional example. It is a figure showing the equivalent circuit of a transformer when a receiving coil is short-circuited. It is a figure showing the equivalent circuit of a transformer when load resistance RL is connected to a receiving coil. It is a figure which shows the relationship between Rw, Rs, Rn and frequency of the coil 1A which closely wound 25 turns of the 1 mm single conductor to the outer diameter of 70 mm. It is a figure which shows the relationship between Rw, Rs, Rn and the frequency of the coil 1B which closely wound 40 turns of the 0.6 mm single conductor wire to the outer diameter of 70 mm. It is a figure which shows the relationship between Rw, Rs, Rn and the frequency of the coil 1C which closely wound the 0.3-mm single conductor wire to the outer diameter of 70 mm for 70 turns. It is a figure showing the frequency characteristic of Rw, Rs, Rn when coils 10A are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when coils 10B are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when coils 10C are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when the coils 10D and 10C are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when the coils 10C and 10D are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn of the single-piece | unit of coil 10A, 10C, 10D. It is a figure showing the frequency characteristic of Rw, Rs, Rn when coils 10E are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when the coils 10F are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when the coils 10F and 10E are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, and Rn when coils 10A and 10F are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when the coils 10F and 10A are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, and Rn when coils 10A and 10J are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, Rn when coils 1A and 10F are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, and Rn when coils 10A and 1A are made to oppose. It is a figure showing the frequency characteristic of Rw, Rs, and Rn when coils 1A and 10A are made to oppose. It is a figure which shows the coil 10b of the electric power transmission apparatus in other embodiment of this invention. It is sectional drawing of the coil 10c formed in the umbrella shape. It is a figure which shows the other example of the coil 10d of the electric power transmission apparatus in other embodiment of this invention. It is a figure which expands and shows the cross section of the coil of the electric power transmission apparatus in other embodiment of this invention. It is a figure showing the magnetic flux direction and intensity | strength on a power transmission coil. It is a figure which shows the coil which used the power transmission device in further another embodiment of this invention, and angled only one side. It is a figure which shows a conductor cross section and magnetic flux shape. It is a figure which shows the coil 10e of the electric power transmission apparatus in further another embodiment of this invention. FIG. 37 is a cross-sectional view taken along line BB in FIG. 36. It is a figure which shows an example in which the material with a high dielectric constant exists in a copper gap | interval. It is a figure which shows the coil 10f of the electric power transmission apparatus in further another embodiment of this invention. It is a figure which shows the coil 10g of the electric power transmission apparatus in further another embodiment of this invention. It is a figure which shows the coil 10h of the electric power transmission apparatus in further another embodiment of this invention. It is a figure which shows the coil 10i of the electric power transmission apparatus in further another embodiment of this invention. It is a figure which shows the coil 10j of the electric power transmission apparatus in further another embodiment of this invention. It is a figure which shows the coil k of the electric power transmission apparatus in further another embodiment of this invention. FIG. 45 is a diagram illustrating a correspondence relationship between a power transmission coil and a power reception coil in the embodiment illustrated in FIG. 44. It is a figure which shows the example which installed the coil on the metal frame. It is a figure which shows the structure of the coil 10m which prevents unnecessary radiation. It is a figure which shows the coil 10n which provided the insulating board between the coil and the magnetic material board. It is a figure which shows the coil 10p which provided the two-layer magnetic material board of the magnetic material board and the magnetic material board in the one surface side of the coil. It is a figure which shows the coil 10q which provided the insulating board of thickness I between the two magnetic material boards which comprise the coil 10p shown in FIG. It is a figure which shows the structure of the coil 10r which prevents the characteristic fluctuation of a coil, when a metal body adjoins to a coil. It is a figure which shows the structure of coil 10s which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 10t which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 10u which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 10v which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 10w which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 10x which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 10y which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. FIG. 49 is a diagram showing the relationship between the effective series resistance Rw (Ω) in each coil when the coil 10F shown in FIG. 21 is used as the coil 10m shown in FIG. 47 and the coil 10n shown in FIG. 48, respectively. It is a figure which shows the relationship with the frequency of Q of each coil when using coil 10F as each of the coils 10m and 10n. It is a characteristic view which shows the effective series resistance Rw ((omega | ohm)) of the coil 10F single-piece | unit, and the inductance Lw ((micro | micron | mu) H) of the coil 10F single-piece | unit when various metal plates are made to oppose closely using the coil 10F. FIG. 62 is a characteristic diagram of the above-described coil 10Fa in which the various metal plates illustrated in FIG. 61 are opposed to the coil 10F via a 10 mm insulator. A characteristic diagram showing the effective series resistance Rw (Ω) of the coil 10Fb alone and the inductance Lw (μH) of the coil 10Fb alone when various metal plates are brought close to the coil 10F provided with one magnetic material plate 91. It is. When various metal plates are brought close to the coil 10Fc having the two magnetic material plates 911 and 912 provided on the coil 10F, the effective series resistance Rw (Ω) of the coil 10Fc alone and the inductance Lw (μH of the coil 10F alone) FIG. It is the characteristic view which measured effective series resistance Rw ((omega | ohm)) and inductance Lw ((micro | micron | mu) H) for the coil 10F in the structure equivalent to the coil 10r. It is the characteristic view which measured the effective series resistance Rw ((omega | ohm)) and the inductance Lw ((micro | micron | mu) H) for the coil 10E in the structure equivalent to the coil 10r. In coil 10y shown in FIG. 58 from coil 10r shown in FIG. 51, it is a figure which uses the metal plate with which the coil taken out from a coil inner peripheral part was equipped. It is a figure of a coil at the time of winding a thin foil-shaped conductor between the foil conductors of the coil of each structure, and taking out the winding line as a coupling wire 30d. It is a figure showing the circuit structure which is an example of embodiment of the electric power transmission apparatus in this invention. It is a figure showing the circuit structure which is an example of other embodiment in the electric power transmission apparatus in this invention. It is a figure which shows the circuit structure of the secondary side of FIG. 69 or FIG. It is a circuit diagram of the alternating current power supply contained in the power transmission control circuit of FIG. It is a circuit diagram which measures the characteristic of LC series resonance circuit. FIG. 74 is an equivalent circuit diagram including a resistance component existing in the circuit of FIG. 73. It is a principle circuit diagram of the measurement circuit which measures the electrostatic capacitance of a capacitor. It is a graph which shows the time change of the electrostatic capacitance of a capacitor. It is a graph which shows the relationship between the effective series resistance Rc of a capacitor, step-up ratio H, and drive circuit electric energy. FIG. 74 is a circuit diagram in which a resistor R3 is provided in series with the circuit of FIG. 73. It is a voltage waveform of the polypropylene capacitor C1c when the polypropylene capacitor C1c is equipped in the circuit of FIG. It is a voltage waveform of the ceramic capacitor C1r when the ceramic capacitor C1r is equipped in the circuit of FIG. 79 is a voltage waveform across a capacitor in the circuit of FIG. 78. It is a graph which shows the relationship between the effective series resistance Rc of a capacitor, and the shift ratio S from the zero point of the sine wave voltage waveform of both ends of a capacitor. It is a graph which shows the relationship between the shift ratio S from the zero point of the sine wave voltage waveform of a capacitor both ends, and electric power transmission performance. It is a graph which shows the relationship between the effective series resistance Rc of a capacitor, and electric power transmission performance. It is a graph which shows the relationship between the dielectric loss tangent tan-delta of a capacitor, and electric power transmission performance. It is a graph which shows the frequency characteristic of the effective series resistance Rc of the ceramic capacitor C1r. It is a graph which shows the frequency characteristic of the effective series resistance Rc of the ceramic capacitor C1r. It is a graph which shows the frequency characteristic of the dielectric loss tangent tan-delta of the polypropylene capacitor C1c. It is a graph which shows the frequency characteristic of the dielectric loss tangent tan-delta of the polypropylene capacitor C1c. It is a connection diagram in the case of measuring the voltage across the capacitor with an oscilloscope when the capacitor is connected to the VOUTH side. It is a connection diagram in the case of measuring the voltage across each capacitor with an oscilloscope when one capacitor is connected to each end of the power transmission coil. It is a circuit diagram which measures the dielectric absorption characteristic of a capacitor. It is a graph which shows the relationship between the dielectric absorption Kp of a capacitor, and electric power transmission performance. It is a graph which shows the relationship between the dielectric absorption Kn of a capacitor, and electric power transmission performance. It is a graph which shows the relationship between Kr which is the ratio of dielectric absorption Kp and Kn of a capacitor, and electric power transmission performance. It is an example of the circuit which measures the heat_generation | fever by the effective series resistance Rc of a capacitor with the electric current which flows into a capacitor. It is another example of the circuit which measures the heat_generation | fever of a capacitor by the effective series resistance Rc of a capacitor with the electric current which flows into a capacitor. It is a graph which shows the temperature rise of a capacitor, and the relationship between electric power transmission performance. It is a block diagram showing other embodiment which measures the performance of a capacitor. FIG. 100 is a diagram showing a voltage waveform of an output of the integrating circuit in FIG. 99. It is a graph which shows the relationship between the display Dz of A / D converter, and electric power transmission performance. It is a graph which shows the relationship between the display Dp of an A / D converter, and electric power transmission performance. It is a graph which shows the relationship between the display Dn of an A / D converter, and electric power transmission performance. It is a graph which shows the relationship between the difference Dd of the display Dp and Dn of an A / D converter, and electric power transmission performance. It is a graph which shows the frequency characteristic of the impedance of polypropylene capacitor C1c. It is a graph which shows the frequency characteristic of the impedance of ceramic capacitor C1r. It is an example which shows the structure of the capacitor in embodiment of this invention. It is another example which shows the structure of the capacitor in embodiment of this invention. It is a circuit diagram which shows other embodiment of the power transmission apparatus of this invention. It is a figure which shows the both ends waveform of the coil in FIG. It is a block diagram which shows other embodiment of the power transmission apparatus of this invention. It is the equivalent circuit at the time of equip | installing the capacitor | condenser with the power transmission coil 1 in series, and the simplified equivalent circuit diagram of 2 terminals. It is a circuit diagram which connects a some capacitor. It is a figure which shows the circuit structure of the electric power transmission apparatus in one Embodiment of this invention. FIG. 115 is a diagram showing a waveform for controlling the circuit of FIG. 114 and an output waveform. It is a figure which shows a waveform when the output of FIG. 114 is made into a sine wave. It is a figure which shows the circuit structure of the electric power transmission apparatus in other embodiment of this invention. FIG. 118 is a diagram showing a waveform for controlling the circuit of FIG. 117 and an output waveform. It is a figure which shows the circuit structure of the electric power transmission apparatus in further another embodiment of this invention. FIG. 119 is a diagram showing a waveform for controlling the circuit of FIG. 119 and an output waveform; It is a figure which shows the circuit structure of the electric power transmission apparatus in further another embodiment of this invention. It is a figure which shows the waveform which controls the circuit of FIG. 121, and an output waveform. It is a figure which shows an example of the circuit structure which produces | generates the direct current | flow electricity used with the power transmission apparatus shown to FIG. 119, FIG. It is a schematic block diagram of the conventional non-contact-type electric power transmission apparatus equipped with the capacitor on the primary side. It is an equivalent circuit schematic of the power transmission coil of the electric power transmission apparatus of this invention comprised as FIG. It is a figure showing the equivalent circuit of a capacitor.

Explanation of symbols

  1, Lp power transmission coil, 2, power reception coil, 3, 3d, 3e, power transmission unit, 3a power transmission control circuit, 3b, 3b1, 3b2 AC power supply, 3c, 201 control circuit, 4, 4d, 4e power reception unit, 4a power reception control Circuit, 12 DC power supply, 2, 2a, 2b, 2c AC power supply, 3c control circuit, 3, 3a power transmission unit, 5, 5a, 5b power receiving device 10a-10y, 10A-10J, 90 coil, 20a, 20b power receiving coil 20 20a, 20b, 21, 22, 50, 60, 64 Plate member, 26 Groove, 26, 29, 30, 34, 36, 30b, 30c, 30d, 30e, 51, 53, 55, 61, 63, 65 , 66, 67 Foil-shaped conductor, 30a insulator, 40, 50 covered electric wire, 22 flat surface, 23, 27, 33, 35, 40-44 inclined surface, 24, 30, 37, 45 side Surface, 51-54, 56, 62, 64 Conductor, 68 Solder, 71 Sparsely wound part, 72 Honey-rolled part, 75, 100 Power transmission coil, 73, 74, 76, 200 Power receiving coil, 80 Metal frame, 82 Aluminum Plate, 85, 85a, 85b Oscilloscope, 86 AC constant current power supply, 87 AC voltmeter, 91, 911, 912 Magnetic material plate, 92 Insulating plate, 95 Metal plate, 100, 100d, 100e Power transmission device, 125 A / D Converter, 126 display, 127 switch, 128 operational amplifier, 135 metal foil, 136 dielectric material, D1-D6 recovery diode, D11-D16 diode, Qa, Qb, Qc, Qd, Q1-Q6 switching element, T1, T2 output Terminal, Cp transmission capacitor, V1 to V4, Vd DC power supply.

Claims (26)

The power transmission unit and power reception unit are configured to be separable,
A power transmission coil used in a power transmission unit of a power transmission device that transmits power from the power transmission unit to the power reception unit by mutual induction,
The power transmission coil is an air-core coil configured in an air-core shape by winding a foil-like conductor in a spiral shape on an insulating material,
The effective series resistance of the power transmission coil alone is Rw (Ω),
In order to detect a change in the effective series resistance of the power transmission coil, the same coil as the power transmission coil is opposed to the power transmission coil to constitute a transformer, and both ends of the same coil opposed to the power transmission coil Rs (Ω), the effective series resistance of the power transmission coil when
Then,
The power transmission coil uses a predetermined foil-like conductor so that the maximum frequency f1 (Hz) satisfying the relationship of Rs ≧ Rw is at least 200 kHz or more. A power transmission coil with a diameter and an inner diameter. When the thickness of the foil conductor is t and the width is H,
At least the width H is not less than 10 times the thickness t;
The width H is 1 mm or more;
The power transmission coil is wound on the insulating material in a single-layer spiral shape so that the insulating material and the width H of the foil conductor are in contact with each other.
At least the maximum outer dimension of the power transmission coil is 10 times or more the width H,
The minimum outer shape of the power transmission coil is 10 cm or more;
And the power transmission coil of Claim 1 whose winding number of the said foil-like conductor is four or more. Furthermore, the thickness t of the foil conductor is 0.3 mm or less, 10 μm or more,
The power transmission coil according to claim 1, wherein a gap of 5 × t or more is provided between conductors of adjacent foil conductors. The width of the gap provided between adjacent foil conductors in the outermost peripheral portion of the air-core coil is w1,
The width of the gap provided between adjacent foil-like conductors in the innermost peripheral portion of the air-core coil is w2,
Then, w2>w1> 0 and
4. The power transmission coil according to claim 3, wherein the width of the gap increases from w <b> 1 as it goes from the outermost peripheral portion to the inner peripheral portion, and the width w <b> 2 of the gap is at least 5 × t. Winding at least one bond wire between the foil conductors;
The coupling wire is electrically insulated from the power transmission foil conductor of the air-core coil, and electromagnetically equivalent to the power transmission foil conductor and honeycomb coupling transformer of the air-core coil. The power transmission coil according to claim 3 or 4 configured to be in a state. Use a conductive wire for a part of the foil-shaped conductor forming the air-core coil,
The width H of a part of the foil-shaped conductor forming the air-core coil is different from other parts,
6. The power transmission coil according to claim 1, wherein at least a part of the foil conductor forming the air-core coil is composed of at least two electrically separated foil conductors. When the area of the power receiving coil is 1/4 or less of the area of the power transmitting coil, the power receiving coil has a rectangular shape having a short side and a long side, and the power transmitting coil and the power receiving coil are opposed to each other. Arranged,
The number of foil-like conductors intersecting the short side wound in a direction perpendicular to the short side and the number of foil-like conductors intersecting the long side wound in a direction perpendicular to the long side The power transmission coil according to any one of claims 1 to 6, wherein the foil-like conductor is wound so that values are close to each other.   The power transmission coil according to any one of claims 1 to 7, wherein a part or all of the foil-shaped conductor forming the air-core coil is wound with an angle with respect to a winding surface of the foil-shaped conductor.   The power transmission device according to any one of claims 1 to 8, further comprising at least one magnetic material plate on a side opposite to a winding surface of the foil-like conductor of the insulating material. Furthermore, the metal coil for shielding the influence when a metal approaches the power transmission coil according to any one of claims 1 to 9 and preventing characteristic fluctuations of the power transmission coil,
Alternatively, when the power transmission coil is installed on a metal material,
In the case where at least one of a magnetic material plate and an insulating material is equipped on the opposite side of the foil-shaped conductor winding surface of the insulating material forming the air-core coil of the power transmission coil,
・ Include only one magnetic material plate,
Whether the predetermined thickness M is (10 mm) or less,
When the outer diameter of the power transmission coil is D (mm) and the M is an insulating material including air, where M> D / 10,
The metal plate or metal material is a metal plate made of a metal or an alloy having a paramagnetic or diamagnetic magnetic property with a thickness M (mm) of 0.1 mm or more,
・ Contain at least two or more magnetic plates
-One magnetic material plate and an insulating material having a predetermined thickness (10 mm) or less,
-When it is a case where there is only an insulating material having a predetermined thickness (10 mm) or more,
The power transmission coil, wherein the metal plate or the metal material is a metal plate or a metal material made of a metal or an alloy having an arbitrary magnetic property with a thickness M (mm) of 0.01 mm or more. In order to prevent a change in inductance of the power transmission coil,
A lead wire to be taken out of the power transmission coil from the lead wire end of the inner periphery of the power transmission coil,
-Is it fixed to the winding surface of the foil-shaped conductor and used as one of the two terminals of the power transmission coil?-The foil-shaped conductor is used on the back surface of the insulating material on which the foil-shaped conductor is wound. A lead wire from the end of the inner peripheral wire is fixedly formed, and one of the two terminals of the power transmission coil is used,
-Alternatively, if a metal plate is provided on the back surface of the insulating material on which the foil-like conductor is formed, the conductor end of the inner peripheral portion of the power transmission coil is connected to the metal plate at the center of the coil. The power transmission coil according to any one of claims 1 to 10, wherein the metal plate is one of the two terminals of the power transmission coil. In order to expand the area where the air-core coil can transmit power,
Including a metallic frame larger than the outer dimension of the air-core coil,
The power transmission coil according to any one of claims 1 to 11, wherein a part of the metallic frame is configured to overlap a part of a winding surface of the power transmission coil. A power transmission coil according to any one of claims 1 to 12 ,
An AC power source comprising power conversion means for converting DC power into AC power;
A power transmission device comprising at least a capacitor (13) for canceling a positive reactance of the power transmission coil,
The capacitor is connected in series between the AC power source and a reference coil,
When a series resonance circuit is configured by the power transmission unit alone without facing another coil to the reference coil, the frequency (series resonance frequency) at which the reactance Xc of the capacitor and the reactance Xi of the power transmission coil become equal is fr. (Hz)
When the output frequency of the AC power supply is set to fr,
Q of the series resonant circuit is Qr,
When the resonance current flows through the series resonance circuit, the output voltage of the AC power supply is Vt,
The voltage across the capacitor is Vc,
If the boost ratio H for boosting the Vt to the Vc is H = Vc / Vt,
A power transmission device in which the capacitor (13) satisfies a condition of H ≧ 0.9 × Qr. A power transmission coil according to any one of claims 1 to 12 ,
An AC power source comprising power conversion means for converting DC power into AC power;
A power transmission device comprising at least a capacitor (13) for canceling a positive reactance of the power transmission coil,
When a reference coil is connected in series between the AC power supply and the capacitor (13), and another coil is not opposed to the reference coil, and a series resonance circuit is configured by a power transmission unit alone,
The output voltage of the AC power supply is a square wave with a duty of 50%, in which the voltage changes between a zero potential and a predetermined potential,
The frequency (series resonance frequency) at which the reactance Xc of the capacitor and the reactance Xi of the reference coil are equal is fr (Hz).
The effective series resistance of the coil is Ri (Ω),
If the output impedance Zs (Ω) of the AC power supply,
Zs <(Ri / 5) is satisfied,
Due to the DC current blocking action of the capacitor, the positive peak value and the negative peak value of the AC voltage waveform across the capacitor are symmetric with respect to zero potential. In an actual series resonance circuit, the AC voltage across the capacitor is The waveform shifts from the zero point,
When the output frequency of the AC power supply is set to fr,
The ratio of the positive peak value Vp and the negative peak value Vn to the zero potential of the voltage across the capacitor in which one capacitor is connected in series to the GND terminal of the AC power source and the reference coil,
S = Vp / Vn,
Positive peak value Vp with respect to zero potential of the voltage across the capacitor in which one capacitor of the same type (the same type, configuration and capacitance) is connected in series to the other terminal of the AC power source and the reference coil And the ratio of the negative peak value Vn,
S1 = Vp / Vn,
When the capacitor is connected in series between the GND terminal of the AC power supply and one terminal of the power transmission coil, and the same capacitor as the capacitor is connected in series between the other terminal of the AC power supply and the other terminal of the power transmission coil The ratio of the positive peak value Vp to the zero potential of the voltage across the capacitor connected to the other terminal and the negative peak value Vn is
When Sn = Vn / Vp, the capacitor (13)
1) The capacitance of the capacitor is C (μF),
The coefficient α is α = log (C / 0.001 μF),
(However, when C ≧ 0.01 μF, α ≧ 1; when C <0.01 μF, α = 1)
When the predetermined coefficient B is B = 1 + 0.06α (no unit),
Whether the capacitor (13) satisfies the relationship S ≦ B,
2) Alternatively, the capacitor (13) has all the following conditions:
Fa = (Sp-1) / (S-1) <0.75,
Fb = (Sp-1) / (S1-1) <0.75,
Fc = (Sn-1) / (S-1) <0.75,
Fd = (Sn-1) / (S1-1) <0.75,
Or at least one of the conditions is satisfied,
When the condition of 2) is satisfied, the capacitor (13) is connected in series between the GND terminal of the AC power supply and one terminal of the power transmission coil, and the other of the AC power supply is connected. A power transmission device in which the capacitor (13) is connected in series between a terminal and the other terminal of the power transmission coil. A power transmission coil according to any one of claims 1 to 12 ,
An AC power source comprising power conversion means for converting DC power into AC power;
A power transmission device comprising at least a capacitor (13) for canceling a positive reactance of the power transmission coil,
Of the two terminals of the capacitor (13), a positive voltage is applied to one terminal A, and a dielectric absorption when a negative voltage is applied to the other terminal B is Kp,
When a positive voltage is applied to the other terminal B and a positive voltage is applied to the one terminal A, the dielectric absorption is Kn,
Kr, which is the ratio of Kp and Kn,
When Kp ≧ Kn, Kr = Kp / Kn,
When Kp <Kn, and Kr = Kn / Kp,
The power transmission device in which the capacitor (13) satisfies a condition of 1 ≦ Kr <1.5.   Furthermore, the power transmission apparatus of Claim 15 whose value of at least any one of the said Kp or the said Kn is 1% or less. A power transmission coil according to any one of claims 1 to 12 ,
An AC power source comprising power conversion means for converting DC power into AC power;
A power transmission device comprising at least a capacitor (13) for canceling a positive reactance of the power transmission coil,
When an integrating circuit including the capacitor (13) is configured, the integrating circuit is
A reference pulse generating means;
A current source capable of flowing a predetermined constant current of the same value in both forward and reverse directions;
Initialization means for zeroing the output voltage of the integrating circuit;
Pulse number measuring means for measuring at least N = 1000 counts of the reference pulse for measuring the forward / reverse integration time,
1) When the output voltage of the integration circuit is set to zero by the initialization unit, after the pulse measurement value of the pulse number measurement unit is initialized to zero,
The constant current Ip in the positive direction is allowed to flow through the capacitor for a predetermined time T (S),
After the elapse of the predetermined time T, the time until the output voltage of the integrating circuit becomes zero when a predetermined negative constant current In of | Ip | = | In | ,
The pulse count number of the predetermined time T is N,
The pulse count number of time Tn until the voltage across the capacitor becomes zero is Nn,
2) When the output voltage of the integration circuit is set to zero by the initialization unit, after the pulse measurement value of the pulse number measurement unit is initialized to zero,
The constant current In in the reverse direction is passed through the capacitor for a predetermined time T,
After the elapse of the predetermined time T, when a predetermined constant current Ip in the positive direction of | Ip | = | In | is passed through the capacitor, the time until the output of the integration circuit becomes zero is Tp,
The pulse count number of the predetermined time T is N,
The pulse count number of time Tn until the voltage across the capacitor becomes zero is Np,
And when
A power transmission apparatus in which an absolute value of a difference between the Nn and the Np, | Nn−Np |, is a capacitor (13) having 0.003 × N counts or less. The absolute value of the difference between N and Nn, | N−Nn |
18. The power transmission device according to claim 17, wherein at least one of the absolute value of the difference between N and Np, | N−Np |, is a capacitor (13) of 0.004 × N counts or less. The effective series resistance at the time of the previous Symbol power transmission coil alone, Rw (Ω),
A transformer is configured by causing a power receiving coil configured to be inductively coupled to the power transmitting coil to face the power transmitting coil, and an effective series resistance of the power transmitting coil is Rs (Ω) when both ends of the power receiving coil are short-circuited. ,
F1 (Hz), the highest frequency satisfying the relationship that the power transmission coil is Rs ≧ Rw,
Then,
The capacitance of the capacitor (13) is selected so that the output frequency fa (Hz) of the AC power supply can be set to f1 or less, and the power transmission coil is driven by the AC power supply to receive power from the power transmission unit. The power transmission device according to claim 13, wherein power is transmitted to the unit. Furthermore, the effective series resistance of the power transmission coil when the power reception coil with both ends open is opposed to the power transmission coil, Rn (Ω),
When the power transmission coil has a maximum frequency satisfying the relationship of Rs ≧ Rn ≧ Rw, f2 (Hz),
The capacitance of the capacitor (13) is selected so that the power transmission coil can transmit power in a frequency region satisfying the relationship of Rs ≧ Rn ≧ Rw,
The output frequency fa (fa = fd), which is the frequency fd for driving the power transmission coil, is set to the frequency f2 or lower, and power is transmitted from the power transmitter to the power receiver at a frequency f2 or lower. The power transmission device according to any one of claims 13 to 19. Furthermore, the thermal resistance of the power transmission coil is θi (° C./W),
The allowable operating temperature of the power transmission coil is Tw (° C.),
The ambient temperature of the power transmission coil is Ta (° C.),
When transmitting the power from the power transmission unit to the power reception unit, when the effective value of the alternating current flowing through the power transmission coil is Ia (A),
In the fa (Hz),
Rw ≦ (Tw−Ta) / (Ia2 × θi),
The power transmission device according to claim 13, wherein power is transmitted from the power transmission unit to the power reception unit such that the power transmission coil satisfies the following relationship. The output frequency fa of the AC power supply is set to a frequency (series resonance frequency) fr (Hz) at which the reactance Xc of the capacitor becomes equal to the reactance Xi of the power transmission coil, and the power transmission coil is driven by the AC power supply,
The effective series resistance of the power transmission coil alone at the fa (Hz) is Rw (Ω),
The effective series resistance of the capacitor at fa (Hz) is Rc (Ω),
The output impedance of the AC power supply at the fa (Hz) is Zs (Ω),
The power transmission device according to claim 13, wherein the AC power supply satisfies at least Zs ≦ (Rw + Rc) × 4. A power receiving coil that receives power from the power transmitting device according to any one of claims 13 to 22 ,
The power receiving coil is formed on the air core to have a magnetic flux supplement area Sn,
The Sn is smaller than the area of the power transmission coil, and the Sn is approximately 2% or more of the area of the power transmission coil,
The effective series resistance of the single power transmission coil is Rw,
When the effective series resistance of the power transmission coil when the power reception coil is short-circuited and opposed to the power transmission coil is Rs,
The power receiving coil selected such that the frequency f1 satisfying the relationship of Rs ≧ Rw is 1 MHz or more.   A power receiving device that includes at least the power receiving coil according to claim 22 and a load and that receives power from the power transmitting device according to any of claims 13 to 21. The power receiving coil is selected such that its maximum outer dimension is smaller than the minimum outer dimension of the power transmitting coil,
The power receiving coil is installed at an arbitrary position on the winding surface of the power transmission coil at an arbitrary angle and an arbitrary direction with respect to the winding surface of the power transmission coil,
When a voltage higher than a certain level is required for the operation of the load connected to the power receiving coil and the electromotive force of the power receiving coil alone is insufficient, a capacitor is connected in parallel to the power receiving coil. The capacitance of the capacitor is selected so that the reactance of the two terminals in which the power receiving coil and the capacitor are connected in parallel, or the point where the impedance becomes maximum is close to the AC power supply frequency fs,
When a current of a certain level or more is required for the load connected to the power receiving coil to operate and the induced current in the power receiving coil alone is insufficient, The capacitance of the capacitor is selected so that the reactance of the two terminals in which the capacitor is connected in series and the power receiving coil and the capacitor are connected in series is zero, or the point where the impedance is a minimum value is close to the AC power supply frequency fa. The power receiving device according to claim 24.   A power transmission device comprising the power transmission device according to any one of claims 13 to 22 and the power reception device according to claim 24 or 25, wherein power is transmitted from the power transmission device to the power reception device.