JP4720468B2 - Nonlinear distortion compensation circuit and method, and wireless transmission system using the same - Google Patents

Nonlinear distortion compensation circuit and method, and wireless transmission system using the same Download PDF

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JP4720468B2
JP4720468B2 JP2005352794A JP2005352794A JP4720468B2 JP 4720468 B2 JP4720468 B2 JP 4720468B2 JP 2005352794 A JP2005352794 A JP 2005352794A JP 2005352794 A JP2005352794 A JP 2005352794A JP 4720468 B2 JP4720468 B2 JP 4720468B2
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雄三 鈴木
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本発明は非線形歪み補償回路及びその方法並びにそれを用いた無線送信システムに関し、特にディジタル無線通信システムにおける増幅器の非線形歪みを、復調信号から検出した歪みの度合いに基づいて適応的に補償制御するようにした非線形歪み補償方式に関するものである。   The present invention relates to a non-linear distortion compensation circuit and method and a radio transmission system using the same, and more particularly to adaptively control non-linear distortion of an amplifier in a digital radio communication system based on a degree of distortion detected from a demodulated signal. The present invention relates to a non-linear distortion compensation method.

ディジタル無線通信システムにおける変復調回路は、従来からＲＦ（Radio Frequency ）帯の信号を増幅する際に生じる非線形歪み補償が行われており、その概略ブロックを図１に示している。なお、この図１の構成は、本発明のそれと同等であ、この非線形歪みを補償するため回路は、非線形歪み補償部１１、直交変調器１２、高出力増幅器１３、直交復調器１４、非線形歪み補償係数演算部１５から構成される。   Conventional modulation / demodulation circuits in digital radio communication systems have been compensated for nonlinear distortion that occurs when amplifying RF (Radio Frequency) band signals, and a schematic block diagram thereof is shown in FIG. The configuration of FIG. 1 is equivalent to that of the present invention, and the circuit for compensating for this nonlinear distortion includes a nonlinear distortion compensator 11, a quadrature modulator 12, a high output amplifier 13, a quadrature demodulator 14, a nonlinear distortion. The compensation coefficient calculation unit 15 is configured.

なお、このシステムは、多値直交変調（ＱＡＭ：Quadrature Amplitude Modulation ）を想定しており、ディジタル復調方式としては、一般的なベースバンド準同期方式が適用されるものとし、同相と直交の各成分（チャネル）に対して、一般的な表記Ｉｃｈ、Ｑｃｈが用いられているものとする。   This system assumes multi-level quadrature modulation (QAM), and a general baseband quasi-synchronization method is applied as a digital demodulation method. In-phase and quadrature components are used. It is assumed that general notations Ich and Qch are used for (channel).

直交変調器１２は、非線形歪み補償が行われたベースバンド信号Ｉｃｈ、Ｑｃｈを直交変調して、直交変調信号を出力する。高出力増幅器１３は、直交変調信号を増幅し変調信号として外部に出力する。直交復調器１４は、高出力増幅器１３から出力された直交変調信号を直交復調することによって生成した直交復調信号Ｉ’ｃｈ、Ｑ’ｃｈを出力する。非線形歪み補償部１１は、ベースバンド信号Ｉｃｈ、Ｑｃｈに非線形歪み補償係数を乗算して、プリディストーション（非線形歪みの逆特性を付加）を行うことにより、非線形歪み補償を実行する。 The quadrature modulator 12 performs quadrature modulation on the baseband signals Ich and Qch on which nonlinear distortion compensation has been performed, and outputs a quadrature modulation signal. The high power amplifier 13 amplifies the quadrature modulation signal and outputs it as a modulation signal. The quadrature demodulator 14 outputs quadrature demodulated signals I′ch and Q′ch generated by performing quadrature demodulation on the quadrature modulated signal output from the high-power amplifier 13. The nonlinear distortion compensation unit 11 performs nonlinear distortion compensation by multiplying the baseband signals Ich and Qch by a nonlinear distortion compensation coefficient and performing predistortion (addition of inverse characteristics of nonlinear distortion).

ここで、ＲＦ帯の高出力増幅器１３で生じる非線形歪みについて説明する。なお、高出力増幅器１３に関する特性は、デシベル（ｄＢ）で表すものとする。高出力増幅器１３の入力電力をＰi 、出力電力をＰo 、増幅利得をＧ、飽和出力電力をＰsat と定義する。増幅器が理想的な特性を有しているとすると、出力電力Ｐo が、飽和出力電力Ｐsat 以上にならない限り、入力電力Ｐi に増幅利得Ｇを足した値が出力される。そのため、入出力特性は以下の式（１ａ），（１ｂ）で表される。   Here, the nonlinear distortion generated in the RF band high-power amplifier 13 will be described. Note that the characteristics relating to the high-power amplifier 13 are expressed in decibels (dB). The input power of the high output amplifier 13 is defined as Pi, the output power is defined as Po, the amplification gain is defined as G, and the saturated output power is defined as Psat. Assuming that the amplifier has ideal characteristics, a value obtained by adding the amplification gain G to the input power Pi is output unless the output power Po becomes equal to or higher than the saturated output power Psat. Therefore, the input / output characteristics are expressed by the following formulas (1a) and (1b).

ＰO ＝Ｐi ＋Ｇ （ＰO ＜Ｐsat ） ……（１ａ）
または、
ＰO ＝Ｐsat （ＰO ≧Ｐsat ） ……（１ｂ） PO = Pi + G (PO <Psat) (1a)
Or
PO = Psat (PO ≧ Psat) (1b)

ところが、実際の電気回路で送信増幅器を構成した場合には、出力電力Ｐ0 が飽和出力電力Ｐsat に近づくにしたがって徐々に圧縮され、実際の増幅器と理想の増幅器との特性差が大きくなる（非特許文献１参照）。   However, when the transmission amplifier is configured by an actual electric circuit, the output power P0 is gradually compressed as it approaches the saturation output power Psat, and the characteristic difference between the actual amplifier and the ideal amplifier becomes large (non-patented). Reference 1).

この非特許文献１によると、この圧縮効果をふまえた増幅器の入出力特性は、以下の式（２）で近似することができる。
ＰO ＝Ｐi ＋Ｇ−Ｋ・log 10｛１＋１０(Pi+G-Psat)/K ｝……（２） According to Non-Patent Document 1, the input / output characteristics of an amplifier based on this compression effect can be approximated by the following equation (2).
P0 = Pi + G- K.log 10 {1 + 10 (Pi + G-Psat) / K } (2)

Ｋは正の数であり、増幅器の特性を示す振幅圧縮係数である。Ｋが大きいほど増幅器の特性は悪化し、Ｋが０に近づくほど、先の理想増幅器の特性に近づく。さらに、式（２）に対して、飽和出力電力Ｐsat を基準点（０ｄＢ）とし、Ｐi ＋Ｇを増幅器の動作点Ｐopと定義すると、高出力増幅器の動作点と出力電力の関係は以下の式（３）で表される。
ＰO ＝Ｐop−Ｋ・log 10｛１＋１０Pop/K ｝ ……（３） K is a positive number and is an amplitude compression coefficient indicating the characteristics of the amplifier. The larger the K is, the worse the characteristics of the amplifier are. The closer K is to 0, the closer to the characteristics of the previous ideal amplifier. Further, when the saturated output power Psat is defined as the reference point (0 dB) and Pi + G is defined as the amplifier operating point Pop , the relationship between the operating point of the high output amplifier and the output power is expressed by the following equation (2): 3).
PO = Pop-K.log 10 {1 + 10 Pop / K } (3)

図６は、Ｋ→０，Ｋ＝３，５，７とした場合の増幅器の動作点対出力電力特性を示したものである。横軸は増幅器の動作点を示し、縦軸は出力電力を示す。同図より、理想的な増幅器（Ｋ→０）の場合には、動作点が飽和出力になるまで線形に動作し、飽和出力に達すると直ちに出力が飽和点にクリップされることが分かる。また、振幅圧縮係数Ｋが大きくなるにつれて、理想的な増幅器との特性差は大きくなり、動作点が飽和点を越える以前に、線形補償動作を行わなければならなくなる度合いが大きくなることがわかる。   FIG. 6 shows the operating point versus output power characteristics of the amplifier when K → 0, K = 3, 5, and 7. The horizontal axis indicates the operating point of the amplifier, and the vertical axis indicates the output power. From the figure, it can be seen that in the case of an ideal amplifier (K → 0), it operates linearly until the operating point reaches the saturation output, and the output is clipped to the saturation point as soon as the saturation output is reached. It can also be seen that as the amplitude compression coefficient K increases, the characteristic difference from the ideal amplifier increases, and the degree to which the linear compensation operation must be performed before the operating point exceeds the saturation point increases.

出力電力Ｐo を増幅器の動作点Ｐopの関数
Ｐo ＝ｆ（Ｐop）
とすると、式（３）に示すように、パラメータＫが固定されたとき、Ｐo とＰopの関係は１対１に対応しているため、Ｐo ＝ｆ（Ｐop）の逆関数
Ｐop＝ｆ-1（Ｐo ）
と表現することができる。 The output power Po is a function of the operating point Pop of the amplifier Po = f (Pop)
Then, as shown in the equation (3), when the parameter K is fixed, the relationship between Po and Pop corresponds to one-to-one, so that the inverse function of Po = f (Pop) Pop = f −1 (Po)
It can be expressed as

図７は、出力電力Ｐo ［ｄＢ］に対する増幅器の動作点Ｐop［ｄＢ］の特性を示したものである。同図より、横軸の出力電力は非線形歪み補償部１１への入力電力に相当し、縦軸である動作点電力はプリディストーション実行後の出力電力に対応する。   FIG. 7 shows the characteristic of the operating point Pop [dB] of the amplifier with respect to the output power Po [dB]. From the figure, the output power on the horizontal axis corresponds to the input power to the nonlinear distortion compensator 11, and the operating point power on the vertical axis corresponds to the output power after the predistortion.

さらに、出力電力と動作点電力との振幅比を振幅補償率Ｒe として定義し、出力電力の振幅を直交復調器の入力振幅にした場合、直交復調器１４の入力振幅に対する振幅補償率特性を図８のように表すことができる。図８の横軸は、増幅器の飽和電力の振幅に対する比率がデシベル表示されているので、高出力増幅器１３の出力信号の動作点を推定することができれば、非線形歪み補償部１１への入力電力をデシベル表現に変換することにより、振幅補償率が求まる。   Further, when the amplitude ratio between the output power and the operating point power is defined as the amplitude compensation rate Re, and the amplitude of the output power is set to the input amplitude of the quadrature demodulator, the amplitude compensation rate characteristic with respect to the input amplitude of the quadrature demodulator 14 is shown. It can be expressed as 8. The horizontal axis of FIG. 8 displays the ratio of the saturation power of the amplifier to the amplitude in decibels. Therefore, if the operating point of the output signal of the high-power amplifier 13 can be estimated, the input power to the nonlinear distortion compensator 11 can be calculated. The amplitude compensation rate can be obtained by converting to the decibel expression.

以上より、多値直交変調信号の平均信号電力の動作点を適応動作によって追従させ、検出した動作点と非線形歪み補償部１１への入力振幅より振幅補償率を導出し、入力信号に振幅補償率を乗じることによって振幅歪みの影響を補償することができる。   As described above, the operating point of the average signal power of the multilevel orthogonal modulation signal is made to follow by the adaptive operation, the amplitude compensation rate is derived from the detected operating point and the input amplitude to the nonlinear distortion compensator 11, and the amplitude compensation rate is input to the input signal. By multiplying by, the influence of the amplitude distortion can be compensated.

図４は、従来方式の非線形歪み補償係数演算部１５の構成を示すブロック図である。同図より、非線形歪み補償係数演算部１５は、２乗和根計算回路２１と振幅補償率演算表処理回路４１と、平均動作点推定回路４２と、補償極性検出回路４３と、判定回路２４とを備えている。２乗和根計算回路２１は、入力されたベースバンド信号Ｉｃｈ、Ｑｃｈのそれぞれの振幅における２乗和根を計算した結果を出力する。判定回路２４は、直交復調信号Ｉ’ｃｈ、Ｑ’ｃｈに基づいて送信シンボルを判定し、データ信号と誤差信号とを生成して出力する。補償極性検出回路４３は、入力された誤差信号のベクトルがデータ信号のベクトルと直角となる境界線で規定される補償極性領域に基づいて振幅歪み補償における補償量を調整するための制御信号を生成して出力する。   FIG. 4 is a block diagram showing a configuration of a conventional nonlinear distortion compensation coefficient calculation unit 15. From the figure, the nonlinear distortion compensation coefficient calculation unit 15 includes a square sum root calculation circuit 21, an amplitude compensation rate calculation table processing circuit 41, an average operating point estimation circuit 42, a compensation polarity detection circuit 43, and a determination circuit 24. It has. The square sum root calculation circuit 21 outputs the result of calculating the square sum root at each amplitude of the input baseband signals Ich and Qch. The determination circuit 24 determines a transmission symbol based on the orthogonal demodulated signals I′ch and Q′ch, generates a data signal and an error signal, and outputs them. The compensation polarity detection circuit 43 generates a control signal for adjusting a compensation amount in amplitude distortion compensation based on a compensation polarity region defined by a boundary line in which an input error signal vector is perpendicular to a data signal vector. And output.

図９は、信号点配置より歪みの影響を検出するための補償極性検出領域の一例を示したものである。同図は、１６ＱＡＭ変調信号の正規信号点配置において、その第一象限のみを取り出したものである。＋印が信号点の正規位置を示す。同図より、補償極性検出回路４３は、信号点配置の原点０と正規の信号点位置（＋印）とを直線で結び、その直線と直角に交わる直線を境界線とし、境界線の内側（斜線なし）を正の非線形歪みを受けた領域と判定し、境界線の外側（斜線あり）を負の非線形歪みの影響を受けた領域と判定し、両判定領域が等確率で発生するような適応動作が行われる。   FIG. 9 shows an example of a compensation polarity detection region for detecting the influence of distortion from the signal point arrangement. In the figure, only the first quadrant is extracted in the regular signal point arrangement of the 16QAM modulated signal. The + mark indicates the normal position of the signal point. From the figure, the compensation polarity detection circuit 43 connects the origin 0 of the signal point arrangement and the normal signal point position (+) with a straight line, and the straight line that intersects the straight line at a right angle is defined as a boundary line. It is determined that the area without the diagonal line is subjected to positive nonlinear distortion, and the area outside the boundary line (with the diagonal line) is determined to be an area affected by the negative nonlinear distortion. An adaptive action is performed.

平均動作点推定回路４２は、補償極性検出回路４３から出力された制御信号に従って平均信号電力の動作点推定値を適応変化した上で生成して出力する。振幅補償率演算表処理回路４１は、この動作点推定値と入力振幅とを代入することによって、振幅補償率導出可能な振幅補償率演算表テーブルを有しており、そのテーブルより導出された振幅補償率を出力する。   The average operating point estimation circuit 42 adaptively changes the operating point estimation value of the average signal power according to the control signal output from the compensation polarity detection circuit 43 and generates and outputs it. The amplitude compensation rate calculation table processing circuit 41 has an amplitude compensation rate calculation table table from which the amplitude compensation rate can be derived by substituting the operating point estimated value and the input amplitude, and the amplitude derived from the table. Output the compensation rate.

図５は、非線形歪み補償部１１を示したものである。同図より、非線形歪み補償部１１は、２つの乗算器５１から構成される。各乗算器５１は、ベースバンド信号Ｉｃｈ，Ｑｃｈと非線形歪み補償係数演算部１５から出力された振幅補償率とをそれぞれ乗算して出力する。   FIG. 5 shows the nonlinear distortion compensator 11. As shown in the figure, the nonlinear distortion compensator 11 is composed of two multipliers 51. Each multiplier 51 multiplies the baseband signals Ich and Qch by the amplitude compensation rate output from the nonlinear distortion compensation coefficient calculator 15 and outputs the result.

なお、歪み補償に関する従来例として下記の特許文献１〜３などがある。
Behavioral Modeling Of Nonlinear RF and Microwave Devices, Artech House 特開２００５−２７１１７３号公報 特開２００４−３６３７１３号公報 特開２００４−１１２１５１号公報 The following patent documents 1 to 3 are known as conventional examples relating to distortion compensation.
Behavioral Modeling Of Nonlinear RF and Microwave Devices, Artech House JP 2005-271173 A JP 2004-363713 A JP 2004-112151 A

ところで、以上に述べた非線形歪み補償動作は、高出力増幅器１３の期待特性に基づいて数値計算で求めた結果をテーブルとして利用するため、テーブルを保存するための膨大なメモリが必要になる。さらに、多値ＱＡＭ変調信号において各信号点の動作点（振幅補償率）は、原点からの信号点間距離によって異なるため、各々信号点に対して最適な振幅補償率をテーブルから選択するための回路構成が必要になり、回路規模が増大するという問題がある。   By the way, since the nonlinear distortion compensation operation described above uses a result obtained by numerical calculation based on the expected characteristics of the high-power amplifier 13 as a table, a huge memory for storing the table is required. Further, since the operating point (amplitude compensation rate) of each signal point in the multilevel QAM modulation signal varies depending on the distance between the signal points from the origin, the optimum amplitude compensation rate for each signal point is selected from the table. There is a problem that a circuit configuration is required and the circuit scale increases.

本発明の目的は、歪み補償を少ない回路規模で実現できる非線形歪み補償回路及びその方法並びにそれを用いた無線送信システム、プログラムを提供することである。   An object of the present invention is to provide a nonlinear distortion compensation circuit capable of realizing distortion compensation with a small circuit scale, a method thereof, and a radio transmission system and program using the same.

本発明による非線形歪み補償回路は、
無線送信装置における増幅器の非線形歪み補償回路であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成手段と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御手段と、
前記制御手段により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償手段とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御手段は、前記多値ＱＡＭ信号の復調信号の最外郭信号の誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。
本発明による他の非線形歪み補償回路は、
無線送信装置における増幅器の非線形歪み補償回路であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成手段と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御手段と、
前記制御手段により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償手段とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御手段は、前記多値ＱＡＭ信号の復調信号の全信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。
本発明による更に他の非線形歪み補償回路は、
無線送信装置における増幅器の非線形歪み補償回路であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成手段と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御手段と、
前記制御手段により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償手段とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記近似式生成手段は、前記近似式として３次と５次の多項式をも生成し、
前記制御手段は、前記多値ＱＡＭ信号の復調信号の最外郭信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。 The nonlinear distortion compensation circuit according to the present invention is
A non-linear distortion compensation circuit for an amplifier in a wireless transmission device, comprising:
An approximate expression generating means for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control means for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
Distortion compensation means for performing the nonlinear distortion compensation according to the approximate expression obtained by the control means ,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The control means determines the distortion amount based on an error signal of an outermost signal of the demodulated signal of the multilevel QAM signal, and controls the order of the approximate expression based on the distortion amount. .
Other nonlinear distortion compensation circuits according to the present invention are:
A non-linear distortion compensation circuit for an amplifier in a wireless transmission device, comprising:
An approximate expression generating means for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control means for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
Distortion compensation means for performing the nonlinear distortion compensation according to the approximate expression obtained by the control means,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The control means determines the distortion amount based on an average error signal of all the demodulated signals of the multilevel QAM signal, and controls the order of the approximate expression based on the distortion amount. .
Still another nonlinear distortion compensation circuit according to the present invention is:
A non-linear distortion compensation circuit for an amplifier in a wireless transmission device, comprising:
An approximate expression generating means for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control means for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
Distortion compensation means for performing the nonlinear distortion compensation according to the approximate expression obtained by the control means,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The approximate expression generating means also generates third and fifth order polynomials as the approximate expression,
The control means determines the distortion amount based on an average error signal of the outermost signal of the demodulated signal of the multilevel QAM signal, and controls the order of the approximate expression based on the distortion amount. To do.

本発明による非線形歪み補償方法は、
無線送信装置における増幅器の非線形歪み補償方法であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成ステップと、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御ステップと、
前記制御ステップにより得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償ステップとを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御ステップは、前記多値ＱＡＭ信号の復調信号の最外郭信号の誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。
本発明による他の非線形歪み補償方法は、
無線送信装置における増幅器の非線形歪み補償方法であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成ステップと、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御ステップと、
前記制御ステップにより得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償ステップとを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御ステップは、前記多値ＱＡＭ信号の復調信号の全信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。
本発明による更に他の非線形歪み補償方法は、
無線送信装置における増幅器の非線形歪み補償方法であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成ステップと、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御ステップと、
前記制御ステップにより得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償ステップとを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記近似式生成ステップは、前記近似式として３次と５次の多項式をも生成し、
前記制御ステップは、前記多値ＱＡＭ信号の復調信号の最外郭信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。 The nonlinear distortion compensation method according to the present invention includes:
A non-linear distortion compensation method for an amplifier in a wireless transmission device, comprising:
An approximate expression generating step for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
A control step for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal at the output of the amplifier;
A distortion compensation step for performing the nonlinear distortion compensation according to the approximate expression obtained by the control step ,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The control step performs the determination of the amount of distortion on the basis of the error signal of the outermost signal of the demodulated signal of the multilevel QAM signal, the feature that you control the order of the approximate expression based on this amount of strain To do.
Other nonlinear distortion compensation methods according to the present invention include:
A non-linear distortion compensation method for an amplifier in a wireless transmission device, comprising:
An approximate expression generating step for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
A control step for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal at the output of the amplifier;
A distortion compensation step for performing the nonlinear distortion compensation according to the approximate expression obtained by the control step,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
In the control step, the distortion amount is determined based on an average error signal of all the demodulated signals of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. .
Still another nonlinear distortion compensation method according to the present invention includes:
A non-linear distortion compensation method for an amplifier in a wireless transmission device, comprising:
An approximate expression generating step for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
A control step for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal at the output of the amplifier;
A distortion compensation step for performing the nonlinear distortion compensation according to the approximate expression obtained by the control step,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The approximate expression generation step also generates third-order and fifth-order polynomials as the approximate expression,
In the control step, the distortion amount is determined based on an average error signal of an outermost signal of the demodulated signal of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. To do.

本発明によるデジタル無線送信システムは、上記の非線形歪み補償回路を用いたことを特徴とする。   A digital wireless transmission system according to the present invention uses the above-described nonlinear distortion compensation circuit.

本発明によるプログラムは、
無線送信装置における増幅器の非線形歪み補償方法をコンピュータに実行させためのプログラムであって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成処理と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御処理と、
前記制御処理により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償処理とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御処理は、前記多値ＱＡＭ信号の復調信号の最外郭信号の誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。
本発明による他のプログラムは、
無線送信装置における増幅器の非線形歪み補償方法をコンピュータに実行させためのプログラムであって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成処理と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御処理と、
前記制御処理により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償処理とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御処理は、前記多値ＱＡＭ信号の復調信号の全信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。
本発明による更に他のプログラムは、
無線送信装置における増幅器の非線形歪み補償方法をコンピュータに実行させためのプログラムであって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成処理と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御処理と、
前記制御処理により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償処理とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記近似式生成処理は、前記近似式として３次と５次の多項式をも生成し、
前記制御処理は、前記多値ＱＡＭ信号の復調信号の最外郭信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする。 The program according to the present invention is:
A program for causing a computer to execute a nonlinear distortion compensation method for an amplifier in a wireless transmission device,
An approximate expression generating process for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control processing for adaptively updating and updating the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
A distortion compensation process that performs the nonlinear distortion compensation according to the approximate expression obtained by the control process ,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The control process is performed determination of the amount of distortion on the basis of the error signal of the outermost signal of the demodulated signal of the multilevel QAM signal, the feature that you control the order of the approximate expression based on this amount of strain To do.
Other programs according to the present invention are:
A program for causing a computer to execute a nonlinear distortion compensation method for an amplifier in a wireless transmission device,
An approximate expression generating process for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Based on the distortion amount of the demodulated signal at the output of the amplifier, a control process for adaptively updating and controlling the order of the approximate expression and its coefficient;
A distortion compensation process that performs the nonlinear distortion compensation according to the approximate expression obtained by the control process,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
In the control process, the distortion amount is determined based on an average error signal of all the demodulated signals of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. .
Still another program according to the present invention is:
A program for causing a computer to execute a nonlinear distortion compensation method for an amplifier in a wireless transmission device,
An approximate expression generation process for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Based on the distortion amount of the demodulated signal at the output of the amplifier, a control process for adaptively updating and controlling the order of the approximate expression and its coefficient;
A distortion compensation process that performs the nonlinear distortion compensation according to the approximate expression obtained by the control process,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The approximate expression generation process also generates cubic and quintic polynomials as the approximate expression,
In the control process, the distortion amount is determined based on an average error signal of an outermost signal of the demodulated signal of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. To do.

本発明の作用を述べる。振幅補償率を３次または５次の単項式で近似し、直交復調信号に基づいて推定される歪み補償の適正度に応じて、単項式次数の最適値を選択し、なおかつ単項式係数を適応的に制御する。高出力増幅器で加わる非線形歪みは、ＩＭ３（３次歪み）による影響が支配的であるが、歪みの大きさによっては３次だけではなく、５次の歪み成分も無視できなくなるため、歪みの逆特性を表現する振幅補償率を３次または５次の単項式で近似するようにして、上記目的を達成している。   The operation of the present invention will be described. The amplitude compensation rate is approximated by a third-order or fifth-order monomial, the optimum value of the monomial order is selected according to the appropriateness of distortion compensation estimated based on the orthogonal demodulation signal, and the monomial coefficient is adaptively controlled To do. The nonlinear distortion applied by the high-power amplifier is dominated by IM3 (third-order distortion). However, depending on the magnitude of distortion, not only the third-order distortion component but also the fifth-order distortion component cannot be ignored. The above-described object is achieved by approximating the amplitude compensation rate expressing the characteristic by a third-order or fifth-order monomial.

本発明によれば、ディジタル無線通信システムの送信装置側で使用され、一般に用いられるＲＦ帯の増幅器で生じる非線形歪みを補償するための歪み補償手段として、増幅器の歪み特性の逆特性を３次または５次の単項式からなる近似式で表現し、復調信号の歪み量に基づいて、近似式次数の最適値を選択し、近似式係数を適応動作により更新しながら歪み補償を行う。これにより、増幅器の入出力特性をテーブルに格納する必要がなくなるので、従来方式よりも、簡素な回路構成で送信側のみで充分な補償効果が得られる非線形歪み補償機能を構築することができるという効果が得られる。   According to the present invention, as a distortion compensation means for compensating for nonlinear distortion generated in a commonly used RF band amplifier used on the transmitter side of a digital radio communication system, the inverse characteristic of the distortion characteristic of the amplifier is obtained by a third order or Expressed by an approximate expression consisting of a fifth-order monomial, the optimum value of the approximate expression order is selected based on the distortion amount of the demodulated signal, and distortion compensation is performed while updating the approximate expression coefficient by an adaptive operation. This eliminates the need to store the input / output characteristics of the amplifier in a table, so that it is possible to construct a nonlinear distortion compensation function that can provide a sufficient compensation effect only on the transmission side with a simpler circuit configuration than the conventional method. An effect is obtained.

以下に、本発明の実施の形態について図面を参照しつつ詳細に説明する。以下の本発明の説明において、変調方式は多値直交変調（ＱＡＭ）を想定しており、ディジタル復調方式としては、一般的なベースバンド準同期方式が適用されるものとし、同相と直交の各成分（チャネル）に対して、一般的な表記Ｉｃｈ，Ｑｃｈが用いられているものとする。また、図１において、本発明の実施例の構成が従来方式の構成に対して異なる点は、非線形歪み補償係数演算部１５の内部構成のみであるため、その部分を重点的に説明する。   Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the following description of the present invention, it is assumed that the modulation method is multilevel quadrature modulation (QAM), and a general baseband quasi-synchronization method is applied as a digital demodulation method. It is assumed that general notations Ich and Qch are used for components (channels). In FIG. 1, the configuration of the embodiment of the present invention is different from the configuration of the conventional system only in the internal configuration of the nonlinear distortion compensation coefficient calculation unit 15.

図２は、図１における本発明の非線形歪み補償係数演算部１５の構成の一例を示したものである。図２において、非線形歪み補償係数演算部１５は、２乗和根計算回路２１と判定回路２４とに加えて、本発明の特徴である振幅補償率演算部２２と、振幅補償率近似式係数制回路２３と、振幅補償率近似式次数選択回路２５と、最外郭（最大振幅）信号点判定部２６とが設けられている。   FIG. 2 shows an example of the configuration of the nonlinear distortion compensation coefficient calculation unit 15 of the present invention in FIG. In FIG. 2, in addition to the square sum root calculation circuit 21 and the determination circuit 24, the nonlinear distortion compensation coefficient calculation unit 15 includes an amplitude compensation rate calculation unit 22 that is a feature of the present invention, and an amplitude compensation rate approximate equation coefficient control. A circuit 23, an amplitude compensation rate approximate expression order selection circuit 25, and an outermost (maximum amplitude) signal point determination unit 26 are provided.

２乗和根計算回路２１は、入力されたベースバンド信号Ｉｃｈ、Ｑｃｈのそれぞれの振幅における２乗和根を計算した結果を出力する。判定回路２４は、直交復調信号Ｉ’ｃｈ，Ｑ’ｃｈに基づいて送信シンボルを判定し、データ信号と誤差信号とを生成して出力する。すなわち、図１０に示す信号点配置（コンスタレーション）上の信号点において、入力される直交復調信号と最も距離が近い信号点が判定値であるデータ信号とされ、直交復調信号とこの判定値との差分が誤差信号となる。   The square sum root calculation circuit 21 outputs the result of calculating the square sum root at each amplitude of the input baseband signals Ich and Qch. The determination circuit 24 determines a transmission symbol based on the orthogonal demodulated signals I′ch and Q′ch, generates a data signal and an error signal, and outputs them. That is, at the signal point on the signal point arrangement (constellation) shown in FIG. 10, the signal point closest to the input quadrature demodulated signal is the data signal that is the judgment value, and the quadrature demodulated signal and the judgment value Is the error signal.

振幅補償率近似式係数制御回路２３は、従来方式と同様に、入力されるデータ信号と誤差信号とを用いて、図９に示される判定領域から歪み補償の極性を検出し、この極性に基づいて近似式係数を調整する。最外郭信号点判定部２６は、入力されるデータ信号に基づいて当該信号点の位置が最外郭か否かを判定し、判定結果を出力する。   Similar to the conventional method, the amplitude compensation rate approximate expression coefficient control circuit 23 detects the polarity of distortion compensation from the determination region shown in FIG. 9 using the input data signal and error signal, and based on this polarity. Adjust the approximate expression coefficient. The outermost contour signal point determination unit 26 determines whether the position of the signal point is the outermost contour based on the input data signal, and outputs a determination result.

振幅補償率近似式次数選択回路２５は、最外郭信号点判定部２６から入力される判定結果と、判定回路２４から入力される誤差信号とに基づいて、最外郭信号点における歪み補償の適正度を検出し、近似式次数選択信号を出力する。   The amplitude compensation rate approximate expression order selection circuit 25 determines the appropriateness of distortion compensation at the outermost signal point based on the determination result input from the outermost signal point determination unit 26 and the error signal input from the determination circuit 24. And an approximate expression order selection signal is output.

図３は、振幅補償率演算部２２の構成を示したものである。図３は、乗算器３４と３乗回路３１と、５乗回路３２と、セレクタ（ＳＥＬ）３３と、加算器３５から構成される。セレクタ（ＳＥＬ）３３は、近似式次数選択信号の極性に応じて３乗回路３１の演算結果と５乗回路３２の演算結果を選択して出力する。乗算器３４は、セレクタ（ＳＥＬ）３３の出力結果に近似式係数を乗じて出力する。加算器３５は、その出力結果に１を足し合わせる。   FIG. 3 shows the configuration of the amplitude compensation rate calculation unit 22. FIG. 3 includes a multiplier 34, a cube circuit 31, a fifth power circuit 32, a selector (SEL) 33, and an adder 35. The selector (SEL) 33 selects and outputs the calculation result of the third power circuit 31 and the calculation result of the fifth power circuit 32 according to the polarity of the approximate expression order selection signal. The multiplier 34 multiplies the output result of the selector (SEL) 33 by the approximate expression coefficient and outputs the result. The adder 35 adds 1 to the output result.

以下、本発明の歪み補償動作について説明する。図１０は、１６値直交変調信号の非線形歪みの影響に係わる信号点配置（コンスタレーション）の変遷を模式的に示したものである。同図において、信号点Ａ点、Ｂ点、Ｃ点、Ｄ点の正規位置を＋印で表し、非線形歪みを受けた各々の信号点の位置を黒丸印で表す。図１１は、振幅圧縮係数Ｋを７としたときの増幅器の振幅補償率特性と、非線形歪みの影響に係わる振幅補償率特性近似式の変遷を示したものである。   Hereinafter, the distortion compensation operation of the present invention will be described. FIG. 10 schematically shows the transition of the signal point arrangement (constellation) related to the influence of the nonlinear distortion of the 16-value quadrature modulation signal. In the figure, the normal positions of the signal points A, B, C, and D are represented by + marks, and the positions of the respective signal points subjected to nonlinear distortion are represented by black circle marks. FIG. 11 shows changes in the amplitude compensation rate characteristic of the amplifier when the amplitude compression coefficient K is 7, and changes in the approximate equation of the amplitude compensation rate characteristic related to the influence of nonlinear distortion.

同図において、増幅器の平均動作点はＸ点で運用されているものとし、信号点Ａ点、Ｂ点、Ｃ点、Ｄ点の振幅補償率は、それぞれＸＡ点、ＸＢ点、ＸＣ点、ＸＣ点であるものとする。ＸＢ点は、Ｘ点に対して、平均振幅に対するＢ点の瞬時振幅との比率（約２．５ｄＢ）の分だけ離れた場所に位置する。また、図１０（ａ）、図１１（ａ）は、非線形歪み影響大の場合に関するもの、図１０（ｂ）、図１１（ｂ）は、非線形歪み影響小の場合に関するもの、図１０（ｃ）、図１１（ｃ）は、非線形歪み影響なしの場合に関するもの、図１０（ｄ）、図１１（ｄ）は、非線形歪み過補償の場合に関するものである。   In the figure, it is assumed that the average operating point of the amplifier is operated at the point X, and the amplitude compensation rates at the signal points A point, B point, C point, and D point are XA point, XB point, XC point, XC, respectively. Suppose that it is a point. The point XB is located away from the point X by a ratio (about 2.5 dB) of the instantaneous amplitude of the point B with respect to the average amplitude (about 2.5 dB). 10 (a) and 11 (a) relate to the case where the nonlinear distortion influence is large, FIGS. 10 (b) and 11 (b) relate to the case where the nonlinear distortion influence is small, and FIG. 10 (c). 11 (c) relates to the case where there is no influence of nonlinear distortion, and FIGS. 10 (d) and 11 (d) relate to the case of nonlinear distortion overcompensation.

本システム初期時において、振幅補償率近似式係数制御回路２３で調整される近似式係数は０に、振幅補償率近似式次数選択回路２５で選択される近似式次数は３に設定されているので、振幅補償率近似式は、図１１（ａ）で示されるように入力振幅によらず、常に“１”となる（図３の加算器３５の“１”加算による）。これにより、振幅補償動作は行われず直交変調信号は非線形歪みの影響を強く受けるので、信号点配置は図１０（ａ）で示されるように非線形歪みの影響大の場合となる。   At the initial stage of the system, the approximate expression coefficient adjusted by the amplitude compensation rate approximate expression coefficient control circuit 23 is set to 0, and the approximate expression order selected by the amplitude compensation rate approximate expression order selection circuit 25 is set to 3. As shown in FIG. 11A, the amplitude compensation rate approximation expression is always “1” regardless of the input amplitude (by adding “1” of the adder 35 in FIG. 3). As a result, the amplitude compensation operation is not performed, and the quadrature modulation signal is strongly influenced by the nonlinear distortion. Therefore, the signal point arrangement is a case where the influence of the nonlinear distortion is large as shown in FIG.

このとき、振幅補償率近似式係数制御回路２３は、直交復調信号Ｉ’ｃｈ、Ｑ’ｃｈのデータ信号と誤差信号から、直交復調信号が図９に示される補償極性領域の斜線しなしの位置にある、すなわち、正の歪みを受けていることを検出し、近似式次数は３のままで、近似式係数を大きくする。その結果として、図１１（ｂ）に示されるように振幅補償率が大きくなり、信号点配置は、図１０（ｂ）に示されるように少しばかり非線形歪が補償された非線形歪み影響小の場合の信号点配置となる。このときの非線形歪みの逆特性を示す単項式は、ａ・ｘ3 となる。ａは近似式係数である。 In this case, the amplitude compensation rate approximate expression coefficient control circuit 23, the orthogonal demodulation signals I 'ch, from the data signal and the error signal Q'ch, the position of the hatched Shinashi compensation polar regions quadrature demodulation signal is shown in FIG. 9 In other words, it is detected that a positive distortion is applied, and the approximate expression coefficient is increased while the approximate expression order remains at 3. As a result, the amplitude compensation rate increases as shown in FIG. 11 (b), the signal point arrangement in the case of non-linear distortion effects small nonlinear distortion little as indicated is compensated in FIG. 10 (b) This is the signal point arrangement. A monomial expression indicating the inverse characteristic of the nonlinear distortion at this time is a · x 3 . a is an approximate expression coefficient.

引き続き、振幅補償率近似式係数制御回路２３が近似式係数を大きくする方向へ調整すると、図１１（ｃ）に示されるように近似曲線が増幅器の振幅補償率特性に近づくため、信号点配置は図１０（ｃ）で示されるように非線形歪みの影響無しの場合となる。   Subsequently, when the amplitude compensation ratio approximate expression coefficient control circuit 23 adjusts the approximation expression coefficient in the direction of increasing, the approximate curve approaches the amplitude compensation ratio characteristic of the amplifier as shown in FIG. As shown in FIG. 10C, there is no influence of nonlinear distortion.

更に、振幅補償率近似式係数制御回路２３が近似式係数を大きくして過制御し、図１１（ｄ）に示されるように近似曲線が増幅器の振幅補償率特性を超えてしまった場合、信号点配置は図１０（ｄ）で示されるように過補償の場合の信号点配置となる。このとき、振幅補償率近似式係数制御回路２３は、直交復調信号Ｉ’ｃｈ、Ｑ’ｃｈのデータ信号と誤差信号とから、直交復調信号が図９に示される補償極性領域の斜線部分にある、すなわち負の歪みを受けている（過補償の状態である）ことを検出し、近似式係数を小さくする。 Further, when the amplitude compensation rate approximate expression coefficient control circuit 23 over-controls by increasing the approximate expression coefficient, and the approximate curve exceeds the amplitude compensation ratio characteristics of the amplifier as shown in FIG. The point arrangement is a signal point arrangement in the case of overcompensation as shown in FIG. In this case, the amplitude compensation rate approximate expression coefficient control circuit 23, the orthogonal demodulation signals I 'ch, and a data signal and the error signal Q'ch, in the shaded portion of the compensation polarity region quadrature demodulation signal is shown in FIG. 9 That is, it is detected that a negative distortion is applied (overcompensation state), and the approximate expression coefficient is reduced.

その結果として、振幅補償率が小さくなり、再び図１０（ｃ）に示されるように非線形歪み影響無しの場合の信号点配置に戻される。このような帰還制御を繰り返すことにより、近似曲線を高出力増幅器１４の期待特性に近づけ、振幅補償率を適切な値に収束させることができる。   As a result, the amplitude compensation rate is reduced, and the signal point arrangement is returned to the case where there is no influence of nonlinear distortion as shown in FIG. By repeating such feedback control, the approximate curve can be brought close to the expected characteristics of the high-power amplifier 14 and the amplitude compensation rate can be converged to an appropriate value.

一方、増幅器の振幅補償率特性が３次で近似できない場合について説明する。図１３は、振幅圧縮係数Ｋを３としたときの増幅器の振幅補償率特性と、非線形歪みの影響に係わる振幅補償率特性の近似式の変遷を示したものである。同図において、増幅器の平均動作点（Ｘ点）は前記よりも高いレベルで運用されているものとし、信号点Ａ点、Ｂ点、Ｃ点、Ｄ点の振幅補償率は、それぞれＸＡ点、ＸＢ点、ＸＣ点、ＸＤ点であるものとする。図１２は、１６値直交変調信号の非線形歪みの影響に係わる信号点配置の変遷を模式的に示したものである。   On the other hand, the case where the amplitude compensation rate characteristic of the amplifier cannot be approximated by the third order will be described. FIG. 13 shows changes in the amplitude compensation rate characteristic of the amplifier when the amplitude compression coefficient K is 3 and the approximate expression of the amplitude compensation rate characteristic related to the influence of nonlinear distortion. In the figure, it is assumed that the average operating point (X point) of the amplifier is operated at a level higher than the above, and the amplitude compensation rates at the signal points A, B, C, and D are XA, It is assumed that the point is XB point, XC point, and XD point. FIG. 12 schematically shows the transition of the signal point arrangement related to the influence of nonlinear distortion of the 16-value quadrature modulation signal.

前記同様、本システム初期時において、振幅補償率近似式係数は０に、次数は３に設定される。歪み補償の過程において、振幅補償率が図１３（ａ）に示されるように近似された場合、最外郭信号点（Ｂ点）の推定振幅補償率が、正規の値であるＸＢ点に対して下方に位置し、Ａ点、Ｃ点における推定振幅補償率が、正規の値であるＸＡ点、ＸＣ点に対して上方に位置する。   Similar to the above, at the initial stage of this system, the amplitude compensation factor approximation equation coefficient is set to 0 and the order is set to 3. In the process of distortion compensation, when the amplitude compensation rate is approximated as shown in FIG. 13A, the estimated amplitude compensation rate of the outermost signal point (point B) is the normal value XB point. Located below, the estimated amplitude compensation rates at points A and C are above the normal values XA and XC.

信号点配置は、図１２（ａ）で示されるように、Ｂ点に対しては非線形歪み影響を受けた場合の信号点配置となり、Ａ点、Ｃ点に対しては歪み過補償の場合の信号点配置となる。振幅補償率近似式係数制御回路２３は、全信号点に対して歪による誤差が最小となる方向へ制御するため、最外郭の信号点Ｂ点に対しては、近似式係数を大きくし歪み補償を強めようとするが、Ａ点、Ｃ点に対しては近似式係数を小さくし歪み補償を弱める方向へ制御する。   As shown in FIG. 12A, the signal point arrangement is a signal point arrangement when the B point is affected by nonlinear distortion, and the A and C points are in the case of distortion overcompensation. Signal point arrangement. Since the amplitude compensation rate approximate expression coefficient control circuit 23 controls all signal points in such a direction that the error due to distortion is minimized, the approximate expression coefficient is increased for the outermost signal point B to compensate for distortion. However, for point A and point C, the approximate expression coefficient is reduced to control distortion compensation.

このように振幅補償率と近似曲線との差が不一致の状態で、各信号点における制御の方向が異なってしまうと、振幅補償率を適切な値に収束させることができず、誤差の抑圧が期待値を満たさなくなる。このときは、最も歪みの影響を受けやすい最外郭信号に着目する。振幅補償率近似式次数選択回路２５は、最外郭信号点判定部２６から入力される判定結果と、判定回路２４から入力される直交復調信号Ｉ’ｃｈ、Ｑ’ｃｈの誤差信号とに基づいて、最外郭信号点の平均誤差信号を計算する。この平均誤差信号がシステム初期時に固定値として与えられる閾値を長期にわたって連続して超えている場合、歪み補償が不充分であるとみなして近似式次数を３次から５次に変更する。 In this way, when the difference between the amplitude compensation rate and the approximate curve is inconsistent and the control direction at each signal point is different, the amplitude compensation rate cannot be converged to an appropriate value, and error suppression is suppressed. The expected value is not met. At this time, attention is focused on the outermost signal that is most susceptible to distortion. Amplitude compensation rate approximate equation order selection circuit 25, based the decision result received as input from the outermost signal points determining section 26, the orthogonal demodulation signals I 'ch inputted from the determination circuit 24, to the error signal Q'ch The average error signal of the outermost signal point is calculated. If this average error signal continuously exceeds a threshold value given as a fixed value at the initial stage of the system over a long period of time, it is considered that distortion compensation is insufficient and the approximate expression order is changed from the third order to the fifth order.

本例では、図１２（ａ）より、最外郭信号点が歪みを受けたままの状態であり、誤差信号が抑圧されないため、近似式次数が３次から５次へ変更される。なお、近似式次数を３次から５次に変更したとき、閾値を、近似式次数３次のときに計算した最外郭信号点の平均誤差信号に更新する。このときの非線形歪みの逆特性を示す単項式は、ｂ・ｘ5 となる。ｂは近似式係数である。 In this example, as shown in FIG. 12A, the outermost signal point remains in a distorted state and the error signal is not suppressed, so the approximate expression order is changed from the third order to the fifth order. When the approximate expression order is changed from the third order to the fifth order, the threshold is updated to the average error signal of the outermost signal point calculated when the approximate expression order is third order. A monomial expression indicating the inverse characteristic of the nonlinear distortion at this time is b · x 5 . b is an approximate expression coefficient.

近似式次数を５次に変更した後、振幅補償率近似式次数選択回路２５は、前と同様、最外郭信号点の平均誤差信号を計算する。この平均誤差信号が閾値（３次のときの平均誤差信号）を超えない場合には、近似式次数を５次のままとする。一方、閾値を長期にわたって連続して超える場合は、近似式次数３次が最適と判断して次数を５次から３次へ変更する。図１３（ｂ）は、近似式次数を３次から５次に変更したときの、近似曲線を示したものである。   After changing the approximate expression order to fifth, the amplitude compensation rate approximate expression order selection circuit 25 calculates the average error signal of the outermost signal points as before. When this average error signal does not exceed the threshold value (average error signal at the third order), the approximate expression order is kept at the fifth order. On the other hand, when the threshold value is continuously exceeded over a long period of time, it is determined that the third order of the approximate expression is optimum, and the order is changed from the fifth order to the third order. FIG. 13B shows an approximate curve when the order of the approximate expression is changed from the third order to the fifth order.

本例では、近似式次数を３次から５次へ変更することにより、最外郭信号点の誤差信号が抑圧されているため近似式次数は５のままとなる。図１２（ｂ）はこのときの信号点配置を示したものである。最外郭信号点（Ｂ点）において、歪みの影響が抑圧されている。   In this example, by changing the approximate expression order from the third order to the fifth order, the error signal of the outermost signal point is suppressed, so the approximate expression order remains at 5. FIG. 12B shows the signal point arrangement at this time. The influence of distortion is suppressed at the outermost signal point (point B).

この後は、前記と同様、振幅補償率近似式係数制御回路２３が近似式係数を調整することにより、近似曲線を図１３（ｃ）で示されるように増幅器の振幅補償率特性に収束させる。この結果、信号点配置は図１３（ｃ）で示されるように非線形歪みの影響無しの場合の配置に戻される。   Thereafter, as described above, the amplitude compensation rate approximate expression coefficient control circuit 23 adjusts the approximate expression coefficient so that the approximate curve converges to the amplitude compensation rate characteristic of the amplifier as shown in FIG. As a result, the signal point arrangement is returned to the arrangement when there is no influence of nonlinear distortion as shown in FIG.

図１４は、本発明の他の実施例を示したものである。平均誤差計算回路２７は、最外郭信号だけではなく、判定回路２４から入力される全ての信号から平均誤差信号を計算して出力する。振幅補償率近似式次数選択回路２５は、入力される平均誤差信号から最適な近似単項式の次数を選択する。   FIG. 14 shows another embodiment of the present invention. The average error calculation circuit 27 calculates and outputs an average error signal not only from the outermost signal but also from all signals input from the determination circuit 24. The amplitude compensation rate approximate expression order selection circuit 25 selects an optimal approximate monomial order from the input average error signal.

図１５、図１６は本発明の更に他の実施例を示したものである。図１５は非線形歪み補償係数演算部を、図１６は振幅補償率演算部を示したものである。図１５において、最外郭信号点判定部２６は、ベースバンド信号から最外郭信号を判定し、その判定結果を振幅補償率近似式制御回路３０へ出力する。振幅補償率近似式制御回路３０は、誤差信号と最外郭信号点判定部２６から入力される判定結果から最外郭信号点の平均誤差信号を監視する。   15 and 16 show still another embodiment of the present invention. FIG. 15 shows a nonlinear distortion compensation coefficient calculator, and FIG. 16 shows an amplitude compensation factor calculator. In FIG. 15, the outermost signal point determination unit 26 determines the outermost signal from the baseband signal, and outputs the determination result to the amplitude compensation rate approximate expression control circuit 30. The amplitude compensation rate approximate expression control circuit 30 monitors the average error signal of the outermost signal point from the error signal and the determination result input from the outermost signal point determination unit 26.

この回路の動作は振幅補償率近似式次数選択回路２５と同様で、最外郭信号点の平均誤差信号が閾値を長期にわたって連続して超えている場合、振幅補償率の近似式を３次の単項式から３次と５次の多項式に切り替えるように制御する。振幅補償率近似式係数制御回路２３は、最外郭信号点のデータ信号と誤差信号から５次の近似式係数を調整する。 The operation of this circuit is the same as that of the amplitude compensation rate approximate expression order selection circuit 25. When the average error signal of the outermost signal point continuously exceeds the threshold over a long period of time, the approximate expression of the amplitude compensation rate is changed to a cubic monomial. Is controlled so as to switch to a third-order and fifth-order polynomial. The amplitude compensation rate approximate expression coefficient control circuit 23 adjusts the fifth-order approximate expression coefficient from the data signal and error signal of the outermost signal point.

なお、この回路は、入力された近似式制御信号の極性が３次と５次の多項式を示すときのみ動作する。この３次と５次の多項式は、
ａ・ｘ3 ＋ｂ・ｘ5
となる。振幅補償率近似式係数制御回路２３は全ての信号点のデータ信号と誤差信号から３次の近似式係数を調整する。 This circuit operates only when the polarity of the input approximate expression control signal indicates third-order and fifth-order polynomials. The third and fifth order polynomials are
a · x 3 + b · x 5
It becomes. The amplitude compensation rate approximate expression coefficient control circuit 23 adjusts the third-order approximate expression coefficient from the data signals and error signals of all signal points.

図１６は、振幅補償率演算部２９の構成を示したものである。振幅補償率演算部２９は、ベースバンド信号の振幅値と、振幅補償率近似式係数制御回路２３から入力される３次の近似式係数と、振幅補償率近似式係数制御回路２３から入力される５次の近似式係数から、振幅補償率を計算する。 FIG. 16 shows the configuration of the amplitude compensation rate calculation unit 29. The amplitude compensation rate calculation unit 29 receives the amplitude value of the baseband signal, the third-order approximate equation coefficient input from the amplitude compensation rate approximate equation coefficient control circuit 23 , and the amplitude compensation rate approximate equation coefficient control circuit 23. The amplitude compensation rate is calculated from the fifth-order approximate expression coefficient.

なお、この回路は、最外郭信号点判定部２８から入力される判定結果に基づいて、ベースバンド信号が最外郭信号点のときは、３次と５次の多項式から振幅補償率を計算し、その他の信号点のときは３次の単項式から計算する。   This circuit calculates the amplitude compensation rate from the third and fifth order polynomials when the baseband signal is the outermost signal point based on the determination result input from the outermost signal point determination unit 28, For other signal points, calculation is performed from a cubic monomial.

本システム初期時において、３次と５次の近似単項式の係数はともに０である。また、近似式制御係数は３次に設定されているので、振幅補償率近似式係数制御回路２３は動作しない。これより、まずは全信号点に対して３次の近似曲線から求めた振幅補償率で歪み補償が行われる。ここで、振幅補償率近似式制御回路３０が最外郭信号点の誤差信号が抑圧されないと判断し、近似式を３次の単項式ではなく、３次と５次の多項式（加算器３５Ａによる）に変更するよう制御信号を出力したとき、振幅補償率近似式係数制御回路２３が５次の近似係数を調整し始める。 At the initial stage of the system, the coefficients of the third-order and fifth-order approximate mononomials are both zero. Further, since the approximate expression control coefficient is set to the third order, the amplitude compensation factor approximate expression coefficient control circuit 23 does not operate. Thus, first, distortion compensation is performed for all signal points with an amplitude compensation rate obtained from a third-order approximation curve. Here, the amplitude compensation rate approximate expression control circuit 30 determines that the error signal of the outermost signal point is not suppressed, and the approximate expression is not a cubic monomial but a cubic and quintic polynomial (by the adder 35A). When the control signal is output to be changed, the amplitude compensation rate approximate expression coefficient control circuit 23 starts to adjust the fifth-order approximate coefficient.

このとき、最外郭信号点に対して、今まで３次の近似単項式から求めた歪み補償量に加えて、５次の単項式から求めた歪み補償量を追加した振幅補償率で歪み補償が行われる。その他の信号点に対しては、今までどおり３次の近似単項式で歪み補償が行われる。   At this time, distortion compensation is performed on the outermost signal point with an amplitude compensation rate obtained by adding the distortion compensation amount obtained from the fifth-order monomial in addition to the distortion compensation amount obtained from the third-order approximate monomial so far. . For the other signal points, distortion compensation is performed by the third-order approximate monomial as before.

このように、本実施例は、回路規模が多少大きくなるものの、近似式次数を３次から５次へ変更する場合と比較して、近似曲線をスムーズに変化させながら所要の振幅補償率特性へ収束させる特徴をもつ。本発明は、ディジタル無線通信を行う送信装置全般に適用可能である。   As described above, in this embodiment, although the circuit scale is slightly increased, the required amplitude compensation rate characteristic is achieved while smoothly changing the approximate curve as compared with the case where the approximate expression order is changed from the third order to the fifth order. It has the characteristics to converge. The present invention is applicable to all transmission apparatuses that perform digital wireless communication.

上記各実施の形態における動作は、予めその動作手順をプログラムとしてＲＯＭなどの記録媒体に格納しておき、これをコンピュータ（ＣＰＵ）により読み取らせて実行させるようにすることも可能であることは勿論である。   The operation in each of the above embodiments can be stored in advance in a recording medium such as a ROM as a program and read by a computer (CPU) for execution. It is.

本発明の実施の形態（従来技術にも適用される）における全体システムブロック図である。It is a whole system block diagram in an embodiment (it is applied also to a prior art) of the present invention. 本発明の実施の形態における図１の非線形歪み補償係数演算部の例を示す図である。It is a figure which shows the example of the nonlinear distortion compensation coefficient calculating part of FIG. 1 in embodiment of this invention. 図２の振幅補償率演算部の例を示す図である。It is a figure which shows the example of the amplitude compensation factor calculating part of FIG. 従来の非線形歪み補償係数演算部の例を示す図である。It is a figure which shows the example of the conventional nonlinear distortion compensation coefficient calculating part. 従来の非線形歪み補償部の例を示す図である。It is a figure which shows the example of the conventional nonlinear distortion compensation part. 増幅器の動作点対出力レベル特性図の一例である。It is an example of an operating point versus output level characteristic diagram of an amplifier. 増幅器の出力レベル対動作点特性図である。It is an output level vs. operating point characteristic diagram of the amplifier. 増幅器の動作点対出力レベル特性図の他の例である。It is another example of the operating point versus output level characteristic diagram of the amplifier. 復調信号の信号点配置（コンスタレーション）より歪みの影響を検出するための補償極性検出領域の例を示す図である。It is a figure which shows the example of the compensation polarity detection area | region for detecting the influence of distortion from the signal point arrangement | positioning (constellation) of a demodulated signal. １６ＱＡＭ信号の非直線歪みの影響に係わる信号点配置の変遷を模式的に示した図である。It is the figure which showed typically the transition of the signal point arrangement | positioning in connection with the influence of the nonlinear distortion of 16QAM signal. 振幅圧縮係数Ｋを７としたときの増幅器の振幅補償率特性と、非線形歪みの影響に係わる振幅補償率特性近似式の変遷を示した図である。It is the figure which showed the transition of the amplitude compensation rate characteristic of an amplifier when the amplitude compression coefficient K is set to 7, and the amplitude compensation rate characteristic approximate expression regarding the influence of nonlinear distortion. １６ＱＡＭ信号の非直線歪みの影響に係わる信号点配置の変遷を模式的に示した図である。It is the figure which showed typically the transition of the signal point arrangement | positioning in connection with the influence of the nonlinear distortion of 16QAM signal. 振幅圧縮係数Ｋを３としたときの増幅器の振幅補償率特性と、非線形歪みの影響に係わる振幅補償率特性近似式の変遷を示した図である。It is the figure which showed the transition of the amplitude compensation rate characteristic of an amplifier when the amplitude compression coefficient K is set to 3, and the amplitude compensation rate characteristic approximate expression regarding the influence of nonlinear distortion. 本発明の実施の形態における図１の非線形歪み補償係数演算部の他の例を示す図である。It is a figure which shows the other example of the nonlinear distortion compensation coefficient calculating part of FIG. 1 in embodiment of this invention. 本発明の実施の形態における図１の非線形歪み補償係数演算部の更に他の例を示す図である。It is a figure which shows the further another example of the nonlinear distortion compensation coefficient calculating part of FIG. 1 in embodiment of this invention. 図１５の振幅補償率演算部の他の例を示す図である。It is a figure which shows the other example of the amplitude compensation factor calculating part of FIG.

符号の説明Explanation of symbols

１１ 非線形歪み補償部
１２ 直交変調器
１３ 高出力増幅器
１４ 直交復調器
１５ 非線形歪み補償係数演算部
２１ ２乗和根計算回路
２２，２９ 振幅補償率演算部
２３ 振幅補償率近似式係数制御回路
２４ 判定回路
２５ 振幅補償率近似式次数選択回路
２６，２８ 最外郭信号点判定部
２７ 平均誤差計算回路
３１ ３乗回路
３２ ５乗回路
３３ 選択回路
３４ 乗算器
３５ 加算器 DESCRIPTION OF SYMBOLS 11 Nonlinear distortion compensation part 12 Quadrature modulator 13 High power amplifier 14 Quadrature demodulator 15 Nonlinear distortion compensation coefficient calculating part 21 Square sum root calculation circuit 22, 29 Amplitude compensation rate calculating part 23 Amplitude compensation rate approximate expression coefficient control circuit 24 Determination Circuit 25 Amplitude compensation rate approximate expression order selection circuit 26, 28 Outermost signal point determination unit 27 Average error calculation circuit 31 3rd power circuit 32 5th power circuit 33 Selection circuit 34 Multiplier 35 Adder

Claims (12)

無線送信装置における増幅器の非線形歪み補償回路であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成手段と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御手段と、
前記制御手段により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償手段とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御手段は、前記多値ＱＡＭ信号の復調信号の最外郭信号の誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする非線形歪み補償回路。 A non-linear distortion compensation circuit for an amplifier in a wireless transmission device, comprising:
An approximate expression generating means for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control means for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
Distortion compensation means for performing the nonlinear distortion compensation according to the approximate expression obtained by the control means ,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The control means determines the distortion amount based on an error signal of an outermost signal of the demodulated signal of the multilevel QAM signal, and controls the order of the approximate expression based on the distortion amount. Nonlinear distortion compensation circuit. 無線送信装置における増幅器の非線形歪み補償回路であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成手段と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御手段と、
前記制御手段により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償手段とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御手段は、前記多値ＱＡＭ信号の復調信号の全信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする非線形歪み補償回路。 A non-linear distortion compensation circuit for an amplifier in a wireless transmission device, comprising:
An approximate expression generating means for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control means for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
Distortion compensation means for performing the nonlinear distortion compensation according to the approximate expression obtained by the control means,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The control means determines the distortion amount based on an average error signal of all the demodulated signals of the multilevel QAM signal, and controls the order of the approximate expression based on the distortion amount. Nonlinear distortion compensation circuit. 無線送信装置における増幅器の非線形歪み補償回路であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成手段と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御手段と、
前記制御手段により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償手段とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記近似式生成手段は、前記近似式として３次と５次の多項式をも生成し、
前記制御手段は、前記多値ＱＡＭ信号の復調信号の最外郭信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする非線形歪み補償回路。 A non-linear distortion compensation circuit for an amplifier in a wireless transmission device, comprising:
An approximate expression generating means for generating an inverse characteristic for compensation of nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control means for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
Distortion compensation means for performing the nonlinear distortion compensation according to the approximate expression obtained by the control means,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The approximate expression generating means also generates third and fifth order polynomials as the approximate expression,
The control means determines the distortion amount based on an average error signal of the outermost signal of the demodulated signal of the multilevel QAM signal, and controls the order of the approximate expression based on the distortion amount. non-linear distortion compensation circuit. 前記制御手段は、前記平均誤差信号が閾値を所定期間連続して超えた場合には、前記近似式として３次の単項式から前記多項式へ切り替えることを特徴とする請求項３記載の非線形歪み補償回路。 4. The nonlinear distortion compensation circuit according to claim 3, wherein when the average error signal exceeds a threshold value continuously for a predetermined period, the control means switches from a cubic monomial to the polynomial as the approximate expression. . 無線送信装置における増幅器の非線形歪み補償方法であって、A non-linear distortion compensation method for an amplifier in a wireless transmission device, comprising:
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成ステップと、An approximate expression generating step for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御ステップと、A control step for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal at the output of the amplifier;
前記制御ステップにより得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償ステップとを含み、A distortion compensation step for performing the nonlinear distortion compensation according to the approximate expression obtained by the control step,
無線送信装置の変調信号は多値ＱＡＭ信号であり、The modulation signal of the wireless transmission device is a multilevel QAM signal,
前記制御ステップは、前記多値ＱＡＭ信号の復調信号の最外郭信号の誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする非線形歪み補償方法。In the control step, the distortion amount is determined based on an error signal of an outermost signal of the demodulated signal of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. Nonlinear distortion compensation method. 無線送信装置における増幅器の非線形歪み補償方法であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成ステップと、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御ステップと、
前記制御ステップにより得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償ステップとを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記制御ステップは、前記多値ＱＡＭ信号の復調信号の全信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする非線形歪み補償方法。 A non-linear distortion compensation method for an amplifier in a wireless transmission device, comprising:
An approximate expression generating step for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
A control step for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal at the output of the amplifier;
A distortion compensation step for performing the nonlinear distortion compensation according to the approximate expression obtained by the control step ,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
In the control step, the distortion amount is determined based on an average error signal of all the demodulated signals of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. Nonlinear distortion compensation method. 無線送信装置における増幅器の非線形歪み補償方法であって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成ステップと、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御ステップと、
前記制御ステップにより得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償ステップとを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記近似式生成ステップは、前記近似式として３次と５次の多項式をも生成し、
前記制御ステップは、前記多値ＱＡＭ信号の復調信号の最外郭信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とする非線形歪み補償方法。 A non-linear distortion compensation method for an amplifier in a wireless transmission device, comprising:
An approximate expression generating step for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
A control step for adaptively updating and controlling the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal at the output of the amplifier;
A distortion compensation step for performing the nonlinear distortion compensation according to the approximate expression obtained by the control step,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The approximate expression generation step also generates third-order and fifth-order polynomials as the approximate expression,
In the control step, the distortion amount is determined based on an average error signal of an outermost signal of the demodulated signal of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. non-linear distortion compensation method to. 前記制御ステップは、前記平均誤差信号が閾値を所定期間連続して超えた場合には、前記近似式として３次の単項式から前記多項式へ切り替えることを特徴とする請求項７記載の非線形歪み補償方法。 8. The nonlinear distortion compensation method according to claim 7, wherein, when the average error signal exceeds a threshold continuously for a predetermined period, the control step switches from a cubic monomial to the polynomial as the approximate expression. . 請求項１〜４いずれか記載の非線形歪み補償回路を用いたことを特徴するデジタル無線送信システム。A digital radio transmission system using the nonlinear distortion compensation circuit according to claim 1. 無線送信装置における増幅器の非線形歪み補償方法をコンピュータに実行させためのプログラムであって、A program for causing a computer to execute a nonlinear distortion compensation method for an amplifier in a wireless transmission device,
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成処理と、An approximate expression generating process for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御処理と、Control processing for adaptively updating and updating the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
前記制御処理により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償処理とを含み、A distortion compensation process that performs the nonlinear distortion compensation according to the approximate expression obtained by the control process,
無線送信装置の変調信号は多値ＱＡＭ信号であり、The modulation signal of the wireless transmission device is a multilevel QAM signal,
前記制御処理は、前記多値ＱＡＭ信号の復調信号の最外郭信号の誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とするプログラム。In the control process, the distortion amount is determined based on an error signal of the outermost signal of the demodulated signal of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. program. 無線送信装置における増幅器の非線形歪み補償方法をコンピュータに実行させためのプログラムであって、A program for causing a computer to execute a nonlinear distortion compensation method for an amplifier in a wireless transmission device,
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成処理と、An approximate expression generating process for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御処理と、Control processing for adaptively updating and updating the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
前記制御処理により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償処理とを含み、A distortion compensation process that performs the nonlinear distortion compensation according to the approximate expression obtained by the control process,
無線送信装置の変調信号は多値ＱＡＭ信号であり、The modulation signal of the wireless transmission device is a multilevel QAM signal,
前記制御処理は、前記多値ＱＡＭ信号の復調信号の全信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とするプログラム。In the control process, the distortion amount is determined based on an average error signal of all the demodulated signals of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. program. 無線送信装置における増幅器の非線形歪み補償方法をコンピュータに実行させためのプログラムであって、
前記増幅器の非線形歪みの補償のための逆特性を３次または５次の単項式からなる近似式として生成する近似式生成処理と、
前記増幅器の出力の復調信号の歪み量に基づいて、前記近似式の次数及びその係数を適応的に更新制御する制御処理と、
前記制御処理により得られた前記近似式に応じて前記非線形歪み補償をなす歪み補償処理とを含み、
無線送信装置の変調信号は多値ＱＡＭ信号であり、
前記近似式生成処理は、前記近似式として３次と５次の多項式をも生成し、
前記制御処理は、前記多値ＱＡＭ信号の復調信号の最外郭信号の平均誤差信号に基づいて前記歪み量の判定を行って、この歪み量に基づき前記近似式の次数を制御することを特徴とするプログラム。 A program for causing a computer to execute a nonlinear distortion compensation method for an amplifier in a wireless transmission device,
An approximate expression generating process for generating an inverse characteristic for compensating for nonlinear distortion of the amplifier as an approximate expression composed of a third-order or fifth-order monomial;
Control processing for adaptively updating and updating the order of the approximate expression and its coefficient based on the distortion amount of the demodulated signal output from the amplifier;
A distortion compensation process that performs the nonlinear distortion compensation according to the approximate expression obtained by the control process ,
The modulation signal of the wireless transmission device is a multilevel QAM signal,
The approximate expression generation process also generates cubic and quintic polynomials as the approximate expression,
In the control process, the distortion amount is determined based on an average error signal of the outermost signal of the demodulated signal of the multilevel QAM signal, and the order of the approximate expression is controlled based on the distortion amount. Program to do.
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JP5499878B2 (en) * 2010-04-23 2014-05-21 富士通株式会社 Distortion compensation apparatus, radio communication apparatus, and distortion compensation method
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