JP4635584B2 - Switching power supply - Google Patents

Description

  The present invention relates to a full-bridge type switching power supply device using a phase shift method.

  Conventionally, various types of switching power supply devices have been proposed and put into practical use. One of them is a full-bridge type switching power supply device using a so-called phase shift method.

  The full-bridge type switching power supply device by this phase shift system includes four switching elements bridge-connected as a switching circuit, and a DC input voltage is converted into an AC voltage by these switching elements. The AC voltage is transformed by a transformer, and further rectified and smoothed by an output circuit to output a DC output voltage.

  These switching elements are all driven with the same ON width and frequency, but are divided into two switching element pairs in terms of the timing. The two switching elements constituting one switching element pair are both controlled to be turned on at a fixed timing on the time axis. The two switching elements constituting the other switching element pair are both controlled to turn on at variable timing on the time axis. By the switching operation by these two pairs of switching elements, the current flowing in the primary winding of the transformer is switched in both directions, and the transformer is excited. Here, the two pairs of combinations of these four switching elements are set so that the input terminal to which the DC input voltage is applied is not electrically short-circuited in any scene of the switching operation. In addition, the switching phase difference generated between the switching element pair controlled to be turned on at a fixed timing and the switching element pair controlled to be turned on at a variable timing is controlled based on the detected output voltage or the like. The output voltage is stabilized.

  Furthermore, capacitors are connected in parallel to these four switching elements, and a resonance circuit is configured by these capacitors and the leakage inductance of the transformer, and zero volt switching (ZVS) is achieved by utilizing the resonance characteristics. The operation is realized and the short-circuit loss of the switching element is reduced.

  For example, in Japanese Patent Application Laid-Open No. H10-228867, the present applicant provides a full bridge type switching by a phase shift method in which an independent inductor is provided in addition to the leakage inductance of the transformer as the inductance constituting the resonance circuit, and its resonance characteristics are adjusted. A power supply is proposed. According to this configuration, the ZVS operation can be realized more reliably, and the short-circuit loss of the switching element can be reduced.

WO 01/071896 pamphlet

  However, the switching power supply device disclosed in Patent Document 1 mainly targets operation in a rated load region (for example, a region where the output current is about 65 to 85 A), and a lighter load region (for example, output current). In the region of about 20 A), the energy stored in the inductor of the resonance circuit is small, so that the drain-source voltage cannot be lowered, and a large short-circuit loss occurs when the switching element is turned on. is there. Such a decrease in efficiency in the light load region makes it difficult to apply the switching power supply device in, for example, a hybrid vehicle that requires high efficiency in the light load region, and there is room for improvement.

  In the same document, a technique for changing the inductance of the resonance circuit in accordance with the size of the load is also proposed. However, as shown in FIG. 24 of the same document, the overall efficiency is improved by making the inductance of the resonance circuit variable. However, in the light load region, compared with the rated load region. Efficiency has been reduced.

  As described above, in the conventional technology, it is difficult to obtain a switching power supply device that can maintain high efficiency in the rated load region and can achieve high efficiency in a light load region where the load is smaller than that. There was room.

  The present invention has been made in view of such a problem, and an object of the present invention is to provide a switching power supply device that can maintain high efficiency in a rated load region and increase efficiency even in a light load region with a smaller load. It is to provide.

The switching power supply device of the present invention includes first to fourth switching elements, one end of each of the first and second switching elements is connected to each other to form a first connection point, and the third and fourth switching elements. One ends of the switching elements are connected to each other to form a second connection point, and are applied between the other ends of the first and third switching elements and the other ends of the second and fourth switching elements. A switching circuit for converting a DC input voltage to an AC voltage, a first capacitive element externally connected in parallel with at least one of the first and second switching elements, a primary input, and a secondary side The transformer has an output and the primary side input is connected between the first connection point and the second connection point, and transforms the alternating voltage, and the primary side input of the transformer from the first connection point. Is inserted in the path leading to the second connection point by reason, an inductor constituting a resonant circuit for ZVS operation with the first capacitive element is connected to the secondary output of the transformer is transformed by the transformer The first and second fixed-side switching elements for the output circuit that generates a DC output voltage based on the AC voltage and the first and fourth switching elements or the second and third switching elements, respectively. A switching drive circuit for driving these switching elements by phase shift control so as to form a switching phase difference between the fourth and third switching elements as the shift-side switching elements with reference to the switching elements of depending on the magnitude of the load of the circuit, so that at least one capacitance value of the first capacitive element is changed Is obtained by a capacitance element control circuit for controlling.

  Here, “external connection” means that the capacitive element is not configured only by the parasitic capacitances of the first to fourth switching elements, for example. Further, the “switching phase difference” means a timing difference between drive signals at which the switching elements are turned on.

In the switching power supply device of the present invention, the DC input voltage is converted into an AC voltage by the switching circuit including the first to fourth switching elements, the AC voltage is transformed by the transformer, and based on the transformed AC voltage, A DC output voltage is generated by the output circuit. The first to fourth switching elements are driven by the phase shift control so as to form a predetermined switching phase difference by the switching drive circuit, and the switching level is changed by the ZVS operation by the resonance circuit formed by the first capacitive element and the inductor. Short-circuit loss of switching elements that make a phase difference is suppressed. Also, the capacitance element control circuit changes the capacitance value of at least one of the first capacitance elements in accordance with the load size of the output circuit. That is, the characteristic of the resonance circuit is set to be appropriate according to the load of the output circuit.

In the switching power supply device of the present invention, a second capacitive element externally connected in parallel with at least one of the third and fourth switching elements is further provided, and the capacitive element control circuit includes the first and second capacitive elements. Of these capacitive elements, it is preferable to control so that only the capacitance value of the first capacitive element changes.

In the switching power supply device of the present invention, each of the first to fourth switching elements has a parasitic capacitance, and the capacitive element control circuit includes a first capacitive element and a switching element connected in parallel to the first capacitive element, It is possible to control so that the capacitance value of the first capacitor element changes by ON / OFF control of the connection between the two.

In the switching power supply device of the present invention, the capacitive element control circuit is configured to control so that the capacitance value of at least one of the first capacitive elements becomes smaller as the load on the output circuit becomes smaller. Is preferred.

In the switching power supply device of the present invention, at least one of the first capacitive elements includes a plurality of capacitors connected in parallel to each other, and the capacitive element control circuit uses the plurality of capacitors to form the first capacitive element. It is possible to configure so as to control the capacitance value to change. In this case, the capacitive element control circuit can be configured to control the capacitance value of the capacitive element to change by selectively turning on / off each of the plurality of capacitors. It is.

According to the switching power supply device of the present invention, the short-circuit loss of the switching element having a switching phase difference by the resonance circuit is suppressed, and at least one of the first capacitor elements constituting the resonance circuit is controlled by the capacitor element control circuit. Since one capacitance value is changed according to the load size of the output circuit, the characteristic of the resonance circuit can be made appropriate according to the load size of the output circuit, and in the rated load region While maintaining high efficiency, it is possible to achieve high efficiency in a wide area including a light load area.
Further, since the first capacitive element is externally connected in parallel with at least one of the first and second switching elements, a capacitor is connected to the third and fourth switching elements. Compared to the case, the efficiency can be further improved particularly in a light load region (for example, a region where the output current is less than 10 A).

  Hereinafter, the best mode for carrying out the present invention (hereinafter simply referred to as an embodiment) will be described in detail with reference to the drawings.

  FIG. 1 shows a configuration of a switching power supply apparatus according to an embodiment of the present invention. This switching power supply device functions as a DC-DC converter that converts a high-voltage DC input voltage Vin supplied from a high-voltage power supply (not shown) and outputs a lower DC output voltage Vout. This is a full-bridge type switching power supply by a shift method.

  This switching power supply device includes a switching circuit 1 provided between a primary high voltage line L1H and a primary low voltage line L1L, an inductor 2, a primary winding 31, and secondary windings 32 and 33. And one end of the primary winding 31 is provided with a transformer 3 connected to the switching circuit 1 via the inductor 2. A DC input voltage Vin is applied between a high-voltage power supply (not shown) between the input terminal T1 of the primary high-voltage line L1H and the input terminal T2 of the primary low-voltage line L1L. The switching power supply device also includes a rectifier circuit 4 provided on the secondary side of the transformer 3, a smoothing circuit 5 connected to the rectifier circuit 4, and a current disposed between the smoothing circuit 5 and the transformer 3. A detection circuit 61, a capacitive element control circuit 62 connected to the current detection circuit 61, and a switching drive circuit 7 connected to the smoothing circuit 5 are provided. The high-voltage power source may be a high-voltage battery, or a combination of an AC generator and a rectifier circuit, or a combination thereof.

  The switching circuit 1 includes switching elements S1A to S4A, capacitors C1 to C4 externally connected in parallel with the switching elements S1A to S4A, and capacitor disconnecting elements S1B to S1B to disconnect the capacitors C1 to C4, respectively. S4B. The switching elements S1A to S4A are bridge-connected to each other. Specifically, one end of the switching elements S1A and S2A is connected to each other to form a connection point P1, and one end of the switching elements S3A and S4A is mutually connected. Connected to form a connection point P2. The other ends of the switching elements S1A and S3A are connected to each other to form a connection point P3, and the other ends of the switching elements S2A and S4A are connected to each other to form a connection point P4. P4 is connected to input terminals T1 and T2, respectively. With such a configuration, the switching circuit 1 converts the DC input voltage Vin applied between the input terminals T1 and T2 into an AC voltage.

  The switching elements S1A to S4A are each composed of a three-terminal switch element such as a MOS-FET (Metal Oxide Semiconductor-Field Effect Transistor), and the capacitor disconnecting elements S1B to S4B are each a bipolar transistor or an IGBT (Insulated Gate), for example. Bipolar output switch elements such as bipolar transistors). The reason why the capacitor disconnecting elements S1B to S4B are constituted by bipolar output switch elements is that current can flow in both directions, and that the current can be completely cut off when turned off.

  One end of the inductor 2 is connected to one end of the switching elements S3A, S4A at the connection point P2, and the other end is connected to one end of the primary winding 31 of the transformer 3. The inductor 2 forms a resonance circuit together with the capacitors C1 to C4 in the switching circuit 1, and uses the resonance characteristics of the resonance circuit to suppress short-circuit loss in the switching elements S1A to S4A as described later. It has become. The arrangement of the inductor 2 is not limited to this position, and it may be inserted into the path from the connection point P1 to the connection point P2 via the primary winding 31 of the transformer. Further, instead of the inductor 2 or in addition to the inductor 2, a resonance circuit (not shown) of the primary side winding 31 in the transformer 3 may be used to form a resonance circuit.

  One end of the primary winding 31 of the transformer 3 is connected to the other end of the inductor 2, and the other end is connected to one ends of the switching elements S1A and S2A at a connection point P1. Further, one ends of the secondary windings 32 and 33 are connected to each other at a connection point P7 (center tap), and the other ends thereof are connected to the anodes of the diodes 41 and 42 at connection points P5 and P6, respectively. . Further, the connection point P7 (center tap) is led to the output terminal T4 through the ground line LG and the current detection circuit 61. That is, this switching power supply device is of a center tap type. The transformer 3 steps down the AC voltage converted by the switching circuit 1 and from the other ends (connection points P5 and P6) of the pair of secondary windings 32 and 33, AC voltages VO1 and VO1 that are 180 degrees out of phase with each other. VO2 is output. In this case, the degree of step-down is determined by the turn ratio between the primary side winding 31 and the secondary side windings 32 and 33.

  The rectifier circuit 4 is a double-wave rectifier type composed of a pair of diodes 41 and 42. The anode of the diode 41 is connected to the other end of the secondary winding 32 at the connection point P5, and the anode of the diode 42 is connected to the other end of the secondary winding 33 at the connection point P6. The cathodes of the diodes 41 and 42 are connected to each other at the connection point P8 and are connected to the output line LO. That is, this rectifier circuit 4 has a cathode common connection structure, and rectifies each half-wave period of the AC output voltages VO1 and VO2 of the transformer 3 by the diodes 41 and 42, respectively, to obtain a DC voltage. It has become.

  The smoothing circuit 5 includes a choke coil 51 and a smoothing capacitor 52. The choke coil 51 is inserted into the output line LO, one end of which is connected to the cathodes of the diodes 41 and 42 at the connection point P8, and the other end is connected to one end of the smoothing capacitor 52 and the output terminal T3 of the output line LO. They are connected at connection point P9. The smoothing capacitor 52 is connected between the output line LO (specifically, the connection point P9) and the ground line LG (specifically, the connection point P10). An output terminal T4 is provided at the end of the output line LO. With such a configuration, the smoothing circuit 5 smoothes the DC voltage rectified by the rectifying circuit 4 to generate a DC output voltage Vout, and supplies this to the load 8 from the output terminals T3 and T4.

  The current detection circuit 61 is inserted into the ground line LG (specifically, disposed between the connection points P7 and P10) and outputs a signal LIout corresponding to the output current Iout. Since the current flowing through the primary winding 31 is proportional to the output current Iout, the current detection circuit 61 is connected to the inductor 2 and the primary winding 31 from the connection point P1 unlike the configuration of FIG. It may be arranged so as to be inserted in a route reaching the connection point P2. Unlike the configuration of FIG. 1, the current detection circuit 61 may be arranged so as to detect a current flowing from the high-voltage power supply (not shown) to the switching circuit 1.

  The capacitive element control circuit 62 controls the capacitance values of the capacitors C1 to C4 in the switching circuit 1 to change based on the signal LIout output from the current detection circuit 61. Specifically, in the switching power supply device of the present embodiment, the capacitor cutting signals L1B to L4B are respectively supplied to the capacitor cutting elements S1B to S4B, and for example, the connection of the capacitors C1 to C4 is controlled on / off. The capacitance values parallel to the switching elements S1A to S4A are controlled to change. In this way, the capacitive element control circuit 62 appropriately changes the resonance characteristics of the resonance circuit according to the magnitude of the load 8 (the magnitude of the output current Iout), as will be described later. As described above, when the switching elements S1A to S4A are constituted by MOS-FETs, the parasitic capacitance of the MOS-FET can also be used as a capacitance parallel to the switching elements S1A to S4A.

  The switching drive circuit 7 drives the switching elements S1A to S4A in the switching circuit 1 based on a signal LVout (supplied from the connection point P9 of the output line LO) corresponding to the output voltage Vout. Specifically, in the switching power supply device of the present embodiment, the switching drive signals L1A to L4A are supplied to the switching elements S1A to S4A, respectively, and the switching elements S1A to S4A are controlled to be turned on / off. Yes. The switching drive circuit 7 stabilizes the DC output voltage Vout by performing switching phase control on the switching elements S1A to S4A as described later and appropriately setting the switching phase difference. Yes.

  Here, each of the switching elements S1A to S4A corresponds to a specific example of “first to fourth switching elements” in the present invention, and the capacitors C1 to C4 and the capacitor disconnecting elements S1B to S4B correspond to the present invention. Corresponds to a specific example of “capacitance element” in FIG. The primary side winding 31 and the secondary side windings 32 and 33 correspond to specific examples of “primary side input” and “secondary side output” in the present invention, respectively. The smoothing circuit 5 corresponds to a specific example of the “output circuit” in the present invention. Each of the connection points P1 and P2 corresponds to a specific example of the “first connection point” and the “second connection point” in the present invention, and each of the connection points P3 and P4 is in the present invention. This corresponds to a specific example of “the other ends of the first and third switching elements” and “the other ends of the second and fourth switching elements”.

  Next, referring to FIGS. 2 to 13, the operation of the switching power supply having the above-described configuration is performed according to the size of the load 8 (the magnitude of the output current Iout) (for example, the output load region). An area where the current Iout is 50 A or more), a light load area (for example, an area where the output current Iout is less than 30 A), and an intermediate load area (for example, an area where the output current Iout is 30 A or more and less than 50 A) will be described. . In the following circuit diagram, the switching elements S1A to S4A and the diodes D1 to D4 connected in parallel with the capacitors C1 to C4 are also considered. As the diodes D1 to D4, when the switching elements S1A to S4A are composed of MOS-FETs as described above, parasitic diodes of the MOS-FETs can be used, and other three diodes having no parasitic diodes can be used. When the terminal switch element is used, an externally connected one is used.

  First, the operation in the rated load region will be described.

  2A and 2B show voltage waveforms of respective parts of the switching power supply apparatus in the rated load region. FIG. 2A shows the voltage waveform of the switching drive signal L1A, and FIG. 2B shows switching elements S1A and diodes D1 connected in parallel. The waveform of the current I1 flowing through the circuit composed of the capacitor C1 is a solid line, and the voltage V1 across the switching element S1A (when the switching element S1A is composed of, for example, a MOS-FET, the voltage between the drain and the source. The same applies to the following.) (C) shows the voltage waveform of the switching drive signal L2A, and (D) shows the current I2 flowing through the circuit composed of the switching element S2A, the diode D2, and the capacitor C2 connected in parallel. (E) is indicated by a solid line, and the waveform of the voltage V2 across the switching element S2A is indicated by a dotted line. The voltage waveform of the switching drive signal L3A is shown, and (F) is a solid line showing the waveform of the current I3 flowing through the circuit composed of the switching element S3A, the diode D3, and the capacitor C3 connected in parallel, and the voltage V3 across the switching element S3A. (G) shows the voltage waveform of the switching drive signal L4A, and (H) shows the waveform of the current I4 flowing through the circuit composed of the switching element S4A, the diode D4, and the capacitor C4 connected in parallel. The waveform of the voltage V4 across the switching element S4A is indicated by a dotted line. In the example shown here, the maximum value of the voltages V1 to V4 = Vin.

  In the switching power supply device shown in FIG. 1, the bridge-connected switching elements S1A to S4A are all driven by the switching drive signals L1A to L4A supplied from the switching control circuit 7 so as to have the same ON width and frequency. The Then, by the switching operation of these switching elements S1A to S4A, the current flowing through the primary side winding 31 of the transformer 3 is switched bidirectionally, thereby exciting the transformer 3. The AC voltage generated in the secondary windings 32 and 33 of the transformer 3 is rectified by the rectifier circuit 4 and smoothed by the smoothing circuit 5, so that the DC output voltage Vout is output from the output terminals T 3 and T 4 to the load 8. To be supplied.

  Looking at the timing at which the switching elements S1A to S4A are driven, these switching elements are divided into two switching element pairs. Specifically, both the switching element S1A and the switching element S2A are controlled to be turned on at a fixed timing on the time axis, and are referred to as “fixed-side switching elements”. Further, both the switching element S3A and the switching element S4A are controlled to be turned on at a variable timing on the time axis, and are referred to as “shift-side switching elements”.

  Further, these switching elements S1A to S4A are driven with a combination and timing at which the input terminals T1 and T2 to which the DC input voltage Vin is applied are not electrically short-circuited in any state of the switching operation. Specifically, the switching element S1A and the switching element S2A (fixed side switching element) are not simultaneously turned on. Further, the switching element S3A and the switching element S4A (shift-side switching element) are not simultaneously turned on. The time interval taken to avoid turning them on at the same time is called the dead time Td.

  The switching element S1A and the switching element S4A have a period in which they are simultaneously turned on, and the primary winding 31 of the transformer 3 is excited during this period in which they are simultaneously turned on. The switching element S1A and the switching element S4A operate so as to have a switching phase difference φ with respect to the switching element S1A (fixed side switching element). The period during which these are simultaneously turned on is controlled by controlling the switching phase difference φ.

  Similarly, the switching element S2A and the switching element S3A have a period in which they are simultaneously turned on. During this period in which the switching elements S2A and S3A are simultaneously turned on, the primary winding 31 of the transformer 3 is excited in the opposite direction to the above case. The The switching element S2A and the switching element S3A operate so as to have a switching phase difference φ with respect to the switching element S2A (fixed-side switching element).

  Further, for controlling the output voltage Vout, for example, based on the signal LVout corresponding to the output voltage Vout, the switching phase difference φ between the switching element S1A and the switching element S4A, and the switching position between the switching element S2A and the switching element S3A. The phase difference φ is controlled, and the time during which the switching element S1A and the switching element S4A are simultaneously turned on and the time during which the switching element S2A and the switching element S3A are simultaneously turned on vary. As a result, the duty ratio of the voltage applied to the primary winding 31 of the transformer 3 changes, and the output voltage Vout is stabilized.

  The capacitor C1 connected in parallel to the switching element S1A, the capacitor C2 connected in parallel to the switching element S2A, and the inductor during the dead time Td from when the switching element S2A is turned off to when the switching element S1A is turned on A resonance circuit is configured by the excitation inductance of the transformer 2 and the transformer 3. By using the resonance action of this resonance circuit to turn on the switching element S1A after all the electric charge accumulated in the capacitor C1 is released, a ZVS operation can be achieved, and the short-circuit loss of the capacitor C1 and the noise associated therewith Is suppressed. The same applies to the switching element S2A, and the ZVS operation can be achieved by turning on the switching element S2A after all the charges accumulated in the capacitor C2 are released, and the short-circuit loss of the capacitor C2 and the accompanying noise are generated. Is suppressed.

  Further, during the dead time Td from when the switching element S3A is turned off to when the switching element S4A is turned on, the capacitor C3 connected in parallel to the switching element S3A and the capacitor C4 connected in parallel to the switching element S4A A resonant circuit is configured by an inductor obtained by converting the inductance of the inductor 2 and the choke coil 51 to the primary side of the transformer 3. By utilizing the resonance action of this resonance circuit, the switching element S4A is turned on after all the electric charge accumulated in the capacitor C4 is released, so that the ZVS operation can be achieved, and the short-circuit loss of the capacitor C4 and the noise associated therewith Is suppressed. The same applies to the switching element S3A, and the ZVS operation can be achieved by turning on the switching element S3A after all the electric charge accumulated in the capacitor C3 is released, and the short circuit loss of the capacitor C3 and the accompanying noise are generated. Is suppressed.

  The capacitors C1 to C4 connected in parallel to the switching elements S1A to S4A are respectively connected to the capacitor disconnection signal L1B generated by the capacitor element control circuit based on the signal LIout corresponding to the output current Iout output from the current detection circuit 61. The connection is on / off controlled by L4B. However, in the rated load region, these capacitors C1 to C4 are always connected to form a resonance circuit together with the inductor 2.

  Here, the operation of the switching power supply device will be described in more detail. In this switching power supply device, five stages (periods T1 to T5) can be roughly divided from the voltage / current waveform of each part and the switching drive signals L1A to L4A shown in FIG. Hereinafter, each stage will be described with reference to FIGS. 3 to 10 in addition to FIG. 2.

<First stage>
The first stage corresponds to the period T1 in FIG. FIG. 3 shows a circuit portion working in the first stage in the switching power supply device shown in FIG. 1 by a solid line. The first stage is a period in which the switching element S2A and the switching element S3A are turned on and power is transmitted from the input side to the output side of the switching power supply device. Switching element S1A and switching element S4A are in an off state. Here, the current Ia flowing through the switching element S2A and the switching element S3A is determined by the output current Iout and the turns ratio of the transformer 3. This first stage continues until switching element S3A is turned off.

<Second stage>
The second stage corresponds to the period T2 in FIG. FIG. 4 shows a circuit portion that works in the second stage in the switching power supply device shown in FIG. 1 by a solid line. In the second stage, the capacitor C3 connected in parallel to the switching element S3A is charged (+ q3, −q3), and the charge (+ q4, −q4) accumulated in the capacitor C4 connected in parallel to the switching element S4A. Is the period during which is discharged. In the second stage, the switching element S3A is turned off. Further, the switching element S1A and the switching element S4A continue to be in the off state. Since the switching element S3A is turned off in this way, the capacitor C3 is charged as described above, and the charge accumulated in the capacitor C4 is discharged. In the second stage, since the transformer 3 performs power transmission, the capacitor C3 is charged with a constant current Ib, and the accumulated charge in the capacitor C4 is discharged with a constant current Ic. Therefore, as shown in FIG. 2, the voltage V3 across the switching element S3A and the voltage V4 across the switching element S4A change linearly. This second stage continues until the diode D4 connected in parallel to the switching element S4A conducts and the voltage V3 across the switching element S3A is clamped by the DC input voltage Vin.

<Third stage>
The third stage corresponds to the period T3 in FIG. FIG. 5 and FIG. 6 show, in solid lines, circuit portions that work in the third stage in the switching power supply device shown in FIG. The third stage is a period in which the energy stored in the inductor 2 is released through the diode D4 and the switching element S2A connected in parallel to the switching element S4A (FIG. 5). Therefore, the current If flowing through the diode D4 and the switching element S2A hardly changes. As shown in FIG. 2, in the middle of the third stage, the switching element S4A is turned on, and the current If flowing in the diode D4 flows into the switching element S4A (FIG. 6). Does not change. The third stage continues until switching element S2A is turned off.

<Fourth stage>
The fourth stage corresponds to the period T4 in FIG. FIG. 7 shows a circuit portion working in the fourth stage in the switching power supply device shown in FIG. 1 by a solid line. In the fourth stage, unlike the second stage, since the transformer 3 is in a short circuit state, when the switching element S2A is turned off, the capacitor C1 connected in parallel to the switching element S1A and the switching element S2A are connected in parallel. The capacitor C2 and the inductor 2 resonate. Therefore, as shown in FIG. 2, the current Ik flowing through the capacitor C1, the voltage V1 across the switching element S1A, the current Ij flowing through the capacitor C2 and the voltage V2 across the switching element S2A change in a trigonometric manner. To do. The fourth stage continues until the diode D1 connected in parallel to the switching element S1A becomes conductive.

<5th stage>
The fifth stage corresponds to the period T5 in FIG. 8 to 10 show the circuit portion working in the fifth stage in the switching power supply device shown in FIG. 1 by solid lines. In the fifth stage, first, the diode D1 connected in parallel to the switching element S1A is turned on, so that the energy stored in the inductor 2 passes through the switching element S4A and the diode D1 to the high-voltage side power supply (not shown). It is regenerated (Fig. 8). In the middle of the fifth stage, the switching element S1A is turned on (FIG. 9), but the basic operation state does not change. When the current Is flowing through the inductor 2 becomes zero (FIG. 2), the charging current Iu flows through the inductor 2 (FIG. 10). The charging current Iu continues to flow until the current flowing through the diode 41 in the rectifier circuit 4 becomes equal to the output current Iout. Therefore, as shown in FIG. 2, the current flowing in the fifth stage changes linearly. Further, by turning on the switching element S1A or the switching element S2A before the current Is flowing through the inductor 2 becomes zero, re-resonance during the dead time Td is prevented.

  In this way, in the rated load region, the capacitors C1 to C4 are connected in parallel to the switching elements S1A to S4A as in the prior art, and these capacitors C1 to C4 and the inductor 2 constitute an appropriate resonance circuit, thereby resonating. ZVS operation is achieved using the characteristics, and short-circuit loss between switching elements whose switching phases are controlled is suppressed.

  Next, the operation in the light load region will be described.

  In this light load region, when the current detection circuit 61 detects that the output current Iout is a value in this region, the capacitor element control circuit 62 applies the capacitor C1 and the switching device S2A connected in parallel to the switching device S1A. The connection of the capacitor C2 connected in parallel, that is, the capacitor of the fixed-side switching element is disconnected. Therefore, for example, in the above-described fourth stage (FIG. 7), the circuit configuration is as shown in FIG. In the circuit configuration shown in FIG. 11, since the capacitors C1 and C2 are disconnected, the LC resonant circuit cannot be established as it is, unlike the circuit configuration shown in FIG. Thus, for example, when the switching elements S1A to S4A are configured by MOS-FETs as described above, the switching elements S1A to S4A are configured by other elements by utilizing the minute parasitic capacitance of the MOS-FETs. Use an external capacitor with a small capacity.

  In this way, by disconnecting the capacitors C1 and C2 of the fixed-side switching element, the capacitance value C of the resonance circuit can be reduced, and the impedance Z of the resonance circuit can be increased by the following equation (1). . Here, L in Formula (1) represents the inductance of the resonant circuit. Therefore, by increasing the impedance Z of the resonance circuit, the peak value V2 (peak) of the voltage V4 across the switching element S2A can be increased by the following equation (2), and both ends of the switching element S1A can be increased. The minimum value of the voltage V1 between them can also be reduced (arrows X1 and X2 in FIG. 12, respectively). Note that n in the expression (2) represents the turn ratio of the primary winding 31 and the secondary windings 32 and 33 in the transformer 3.

Z = {L / (2C)} ^1/2 (1)
V2 (peak) = n · Iout · Z (2)

  Here, as shown in FIG. 12, in order to facilitate the ZVS operation by the resonance circuit, the peak value V4 (peak) of the voltage V4 across the switching element S4A is obtained by the following equation (3). What is necessary is just to set it as the value more than DC input voltage Vin. Therefore, by disconnecting the capacitor C1 and the capacitor C2 and increasing the impedance Z of the resonance circuit, the ZVS operation by the resonance circuit is facilitated, and the switching element is short-circuited in the light load region as well as in the rated load region. Loss can be suppressed. Even if the ZVS operation cannot be achieved, since the capacitance value C of the resonant circuit is reduced, the short-circuit loss Ploss of the switching element can be reduced by the following equation (4). Note that V and f in Equation (4) represent the voltage across the capacitor and the resonance frequency of the resonance circuit, respectively.

V2 (peak) ≧ Vin (3)
Ploss = (1/2) · C · V ² · f (4)

  Finally, the operation in the intermediate load region will be described.

  In this intermediate load region, when the current detection circuit 61 detects that the output current Iout is a value in this region, the capacitor element control circuit 62 connects one of the capacitors C1 and C2 of the fixed-side switching element. Is disconnected. Therefore, for example, when the connection of the capacitor C2 is cut, for example, the circuit configuration as shown in FIG. 13 is obtained in the above-described fourth stage (FIG. 7).

  Unlike the light load region described above, the connection of only one of the capacitors C1 and C2 of the fixed side switching element is disconnected when both capacitors are disconnected in the intermediate load region. As a result, the resonance angular frequency ω of the resonance circuit becomes too high as compared with the light load region, and the energy at the time of resonance is originally larger than that in the light load region. This is because the voltage V1 across the switching element S1A tends to return to close to the DC input voltage Vin when turning off (in this case, as shown by the arrow X4 in FIG. 14, between the both ends of the switching element S2A). The voltage V2 is also reduced). As described above, when the voltage V1 across the switching element S1A is turned off and returned to near the DC input voltage Vin, the short-circuit loss Ploss of the switching element increases. Therefore, by disconnecting only one of the capacitors C1 and C2 of the fixed-side switching element, the impedance Z of the resonance circuit is increased and the resonance angular frequency ω of the resonance circuit is prevented from becoming too high, and the rated load Similarly to the region and the light load region, the short-circuit loss Ploss of the switching element can be suppressed also in the intermediate load region.

ω = {1 / (2LC)} ^1/2 (5)

  Next, as a specific example, in the switching power supply device having the configuration shown in FIG. 1, the switching elements S1A to S4A are configured by MOS-FETs, and the capacitor disconnecting elements S1B to S4B are configured by IGBTs. The case where C2 = 3300 pF and capacitors C3 and C4 = 1000 pF of the shift side switching element are externally connected will be described.

  FIGS. 15 to 18 show waveforms of the voltage V1 across the switching element S1A at the time of turn-on when the output current Iout = 20A, 10A, 5A, and 2A, respectively. In each figure, (A) shows a case where both capacitors C1 and C2 are connected, (B) shows a case where only capacitor C2 is disconnected, and (C) shows both capacitors C1 and C2. The case where it has cut | disconnected is shown. In each figure, the vertical axis represents voltage (V), and the horizontal axis represents time (nS).

  As described above, in any case of FIGS. 15 to 18, the impedance Z of the resonance circuit becomes smaller as the capacitance value C of the resonance circuit is decreased in the order of (A) → (B) → (C). The operation is performed so that the short circuit loss Ploss is increased. Therefore, as indicated by reference numerals G11 to G13, G21 to G23, G31 to G33, and G41 to G43 in the respective drawings, it can be seen that the waveform of the voltage V1 across the switching element S1A tends to decrease.

  FIG. 19 summarizes the efficiency of the switching power supply in each case of FIGS. 15 to 18 as an output current versus efficiency characteristic diagram. Here, the curves indicated by reference numerals G51 to G53 in the figure respectively cut only the capacitor C2 when both of the capacitors C1 and C2 are connected (corresponding to FIGS. 15A to 18A). (Corresponding to FIG. 15B to FIG. 18B) and when both capacitors C1 and C2 are disconnected (corresponding to FIG. 15C to FIG. 18C). ing. In each figure, the vertical axis represents efficiency (%) and the horizontal axis represents output current Iout (A). Also, in the figure, as an example, the rated load region is the region where the output current Iout is 50 A or more, the intermediate load region is the region where the output current Iout is 30 to 50 A, and the light load region is the output current Iout is 0 to 30 A. It is expressed as a region.

  Thus, in the light load region and the intermediate load region, the efficiency is improved in the case of the curves G52 and G53 compared to the curve G51, and in the light load region, the efficiency is particularly improved in the case of the curve G53. I understand that. Therefore, by controlling so that the capacitor is disconnected as the load decreases, that is, as the output current Iout decreases, the short-circuit loss Ploss of the switching element is suppressed and the light load region reaches the rated load region. High efficiency can be achieved in a wide area.

  As described above, in the present embodiment, by connecting the capacitors C1 to C4 in parallel to the switching elements S1A to S4A, an appropriate resonance circuit is configured by the capacitors C1 to C4 and the inductor 2, and the resonance circuit The short-circuit loss of the switching element having a switching phase difference due to the resonance characteristics is suppressed, and the capacitance value of the capacitors C1 and C2 of the fixed-side switching element is set by the capacitance element control circuit 62 to the magnitude of the load 8 (the magnitude of the output current Iout). Therefore, the characteristics of the resonant circuit can be made appropriate according to the size of the load 8, and while maintaining high efficiency in the rated load region, the light load region and the middle High efficiency can be achieved even in the load region.

  While the present invention has been described with reference to the embodiment, the present invention is not limited to this embodiment, and various modifications can be made.

  For example, in the above-described embodiment, the case where the capacitors C1 to C4 as the capacitive elements are connected in parallel to all the switching elements S1A to S4A has been described, but the capacitors as the capacitive elements are the switching elements S1A to S1A to S1A to S4A. What is necessary is just to be connected in parallel with at least one of S4A. Even when configured in this manner, the same effects as those of the above-described embodiment can be obtained.

  In the above-described embodiment, the case where control is performed so as to disconnect the capacitors C1 and C2 of the fixed-side switching element in the light load region and the intermediate load region has been described. As shown in FIG. 21 (intermediate load region), in addition to this, control may be performed so as to disconnect the capacitors C3 and C4 of the shift side switching element, and the capacitors C3 and C4 of the shift side switching element. It may be controlled to disconnect only the connection. That is, the capacitive element control circuit 62 may be controlled to change the capacitance value of at least one of the capacitors C1 to C4 connected in parallel. For example, FIG. 22 shows output current vs. efficiency characteristics when the switching power supply having the configuration of FIGS. 20 and 21 is made to correspond to the above-described specific example. Here, the curves indicated by reference numerals G53 to G55 in the figure are cases where both the capacitors C1 and C2 of the fixed-side switching element are disconnected and both the capacitors C3 and C4 of the shift-side switching element are connected. (Curve G53 in FIG. 19) shows a case where only the capacitor C3 of the shift side switching element is connected and a case where all of the capacitors C1 to C4 are disconnected. Thus, in the light load region, particularly in the region where the output current Iout is 5 A or less, the efficiency is improved in the case of the curves G54 and G55 compared to the curve G53, and the efficiency in the case of the curve G55 is particularly high. It can be seen that it has improved. Therefore, in addition to the capacitors C1 and C2 of the fixed side switching element, it is possible to improve the efficiency in the light load region by controlling the connection of the capacitors C3 and C4 of the shift side switching element. . Note that the degree of improvement in efficiency differs between the case where the capacitance values of the capacitors C1 and C2 of the fixed-side switching element are changed and the case where the capacitance values of the capacitors C3 and C4 of the shift-side switching element are changed. This is because the inductance value of the resonance circuit in each case is different as described above. That is, the inductance value is smaller in the case where the resonance circuit is constituted by the capacitors C1 and C2 of the fixed side switching element than in the case where the resonance circuit is constituted by the capacitors C3 and C4 of the shift side switching element. Changing the capacitance values of the capacitors C1 and C2 has a greater effect of improving efficiency.

  Further, in the above embodiment, the case where the capacitors C1 to C4 as the capacitive elements are connected in parallel to the switching elements S1A to S4A one by one has been described, but as shown in FIG. For example, a capacitor as a capacitive element connected in parallel to the switching element S5A may be composed of a plurality of capacitors C51 to C5n connected in parallel to each other. Here, the capacitor cutting elements S5B1 to S5Bn and the capacitor cutting signals L5B1 to L5Bn correspond to the plurality of capacitors C51 to C5n, respectively. When the capacitor as the capacitive element is configured as described above and the connection of the plurality of capacitors C51 to C5n is selectively controlled to be turned on / off, in addition to the effects in the embodiment, the capacitor as the capacitive element By changing the capacitance value in multiple stages, the efficiency can be improved more reliably.

  Furthermore, in the above embodiment, the circuit configuration of the switching power supply device has been specifically described. However, the circuit configuration is not limited to this, and for example, the transformer 3 may be a circuit configuration other than the center tap type. For example, the rectifier circuit 4 may have a circuit configuration other than the cathode common connection.

It is a circuit diagram showing the structure of the switching power supply device which concerns on one embodiment of this invention. FIG. 2 is a timing diagram for explaining the operation of the switching power supply device of FIG. 1. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram for demonstrating operation | movement of the switching power supply device of FIG. FIG. 2 is a circuit diagram for explaining an operation in a light load region of the switching power supply device of FIG. 1. It is a timing diagram for demonstrating operation | movement of a light load area | region. FIG. 2 is a circuit diagram for explaining an operation in an intermediate load region of the switching power supply device of FIG. 1. FIG. 6 is a timing diagram for explaining an operation in an intermediate load region. It is a characteristic view showing an example of the voltage waveform between the both ends of a switching element. It is a characteristic view showing an example of the voltage waveform between the both ends of a switching element. It is a characteristic view showing an example of the voltage waveform between the both ends of a switching element. It is a characteristic view showing an example of the voltage waveform between the both ends of a switching element. It is a characteristic view showing an example of the output current versus efficiency characteristic of the switching power supply. It is a circuit diagram showing the structure of the switching power supply device which concerns on the modification of this invention. It is a circuit diagram showing the structure of the switching power supply device which concerns on the modification of this invention. It is a characteristic view showing an example of the output current versus efficiency characteristic of the switching power supply according to a modification of the present invention. It is a circuit diagram showing a part of structure of the switching power supply device which concerns on the modification of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Switching circuit, 2 ... Inductor, 3 ... Transformer, 31 ... Primary side winding, 32, 33 ... Secondary side winding, 4 ... Rectification circuit, 5 ... Smoothing circuit, 61 ... Current detection circuit, 62 ... Capacity Element control circuit, 7 ... switching drive circuit, 8 ... load, S1A-S4A ... switching element, S1B-S4B ... capacitor disconnection element, C1-C4 ... capacitor, L1A-L4A ... switching drive signal, L1B-L4B ... capacitor disconnection signal , L1H: Primary side high-voltage line, L1L: Primary low-voltage line, LO ... Output line, LG ... Ground line, P1 to P10 ... Connection point, T1, T2 ... Input terminal, T3, T4 ... Output terminal, Iout ... Output Current, Vin ... DC input voltage, Vout ... DC output voltage.

Claims (6)

Including first to fourth switching elements, one ends of the first and second switching elements are connected to each other to form a first connection point, and one ends of the third and fourth switching elements are A DC input voltage applied between the other ends of the first and third switching elements and the other ends of the second and fourth switching elements is connected to each other to form a second connection point. A switching circuit that converts AC to AC voltage;
A first capacitive element externally connected in parallel with at least one of the first and second switching elements ;
A transformer having a primary side input and a secondary side output, the primary side input being connected between the first connection point and the second connection point, and transforming the AC voltage;
In order to perform ZVS (Zero Volt Switching) operation together with the first capacitive element, inserted into a path from the first connection point to the second connection point via the primary side input of the transformer. An inductor constituting the resonance circuit of
An output circuit connected to the secondary side output of the transformer and generating a DC output voltage based on the AC voltage transformed by the transformer;
With respect to the first and fourth switching elements, or the second and third switching elements, the first and second switching elements as fixed-side switching elements are used as references, and as shift-side switching elements, respectively. a switching drive circuit for driving the phase shift control so as to have a switching phase difference, these switching elements between said fourth and third switching elements,
A switching power supply device comprising: a capacitor element control circuit that controls the capacitance value of at least one of the first capacitor elements to change in accordance with the load of the output circuit. A second capacitive element externally connected in parallel with at least one of the third and fourth switching elements;
2. The switching power supply device according to claim 1, wherein the capacitive element control circuit performs control so that only a capacitance value of the first capacitive element among the first and second capacitive elements changes. . The capacitive element control circuit, according to the load of the output circuit is smaller, at least one capacitance value and controls so as to become smaller according to claim 1, wherein one of said first capacitive element Item 3. The switching power supply device according to Item 2 . Each of the first to fourth switching elements has a parasitic capacitance,
The capacitor element control circuit, by controlling on and off the connection between the first capacitor and the first capacitor in parallel-connected switching elements, the capacitance value of the first capacitor The switching power supply according to any one of claims 1 to 3, wherein the switching power supply is controlled to change. At least one of the first capacitive elements is composed of a plurality of capacitors connected in parallel to each other;
The said capacitive element control circuit controls so that the capacitance value of a said 1st capacitive element may change using these several capacitor | condensers. The Claim 1 thru | or 3 characterized by the above-mentioned. Switching power supply. The said capacitive element control circuit controls so that the capacitance value of a said 1st capacitive element may change by selectively turning on / off each connection of these several capacitor | condenser. 5. The switching power supply device according to 5.

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