JP4541120B2 - Radar equipment - Google Patents

Description

  The present invention relates to a radar apparatus having high resolution in a distance (range) direction and an azimuth (azimuth) direction.

The radar apparatus is mounted on a moving body such as an aircraft or an automobile, and generates a radar image in front of the moving body in the traveling direction.
Since the wavelength of the radar emitted from the radar device is longer than that of light, there is an advantage that an image can be generated even in a bad vision environment such as fog.
Therefore, for example, if a radar device is mounted on an aircraft, landing support is possible in an environment where visibility in the traveling direction is poor, which is beneficial in many situations.

As conventional image radar technology, synthetic aperture radar (SAR), Doppler beam sharpening (DBS), and the like are known (for example, see Non-Patent Document 1).
However, none of these techniques can image the front, even if the image can be imaged laterally or obliquely forward with respect to the traveling direction. The reason for this is that these technologies essentially use the movement of the platform to virtually increase the resolution in the direction parallel to the direction of travel by constructing a large array antenna that extends in the direction of travel of the platform. It is. In addition, since the Doppler frequency of the received signal is the same in a direction symmetric with respect to the traveling direction of the platform, when the vicinity of the front direction is imaged using these techniques, the obtained image is folded left and right with respect to the traveling direction of the platform. Become an image.

After all, in order to increase the resolution in the azimuth (azimuth) direction of the radar facing the front of the moving body, an array antenna having a large aperture is required.
However, in order to prevent the occurrence of grating lobes, it is necessary to arrange the element antennas at half-wavelength intervals. Therefore, when the aperture diameter of the array antenna increases, the number of element antennas constituting the array antenna increases. .
Therefore, the conventional radar apparatus employs a method of reducing the number of element antennas and suppressing generation of grating lobes at the same time by separately providing a transmitting array antenna and a receiving array antenna (for example, patents). Reference 1). Hereinafter, this method is referred to as a first method.

In this first method, the number of elements is reduced by widening the element spacing between the transmitting array antenna and the receiving array antenna beyond half a wavelength.
Then, by arranging the transmitting array antenna and the receiving array antenna so that the element spacing of the transmitting array antenna is different from the element spacing of the receiving array antenna, a transmitting antenna pattern (hereinafter referred to as “transmission pattern”). ) And the grating lobe generated in the reception antenna pattern (hereinafter referred to as “reception pattern”) do not coincide with each other so that the transmission / reception pattern (hereinafter referred to as “transmission / reception pattern”) is generated. Suppress the grating lobe.
Further, by scanning the transmission pattern and the reception pattern in synchronization, the region is scanned and observed.

  In addition, a second method is known in which the number of element antennas is reduced and the generation of grating lobes is suppressed by disposing element antennas at unequal intervals (see, for example, Non-Patent Document 2). .

US Pat. No. 3,825,928 “High Resolution Bistatic Rader System” D. R. Wehner, “High-Resolution Radar Second Edition,” Arttech House Inc, 1995. A. Manikas and C.M. Proukakis, “Modeling and Estimates of Ambientities in Linear Arrays,” IEEE Transactions on Signal Processing, Vol. 46, no. 8, August 1998

Since the conventional radar apparatus is configured as described above, in the first method, the arrangement of the element antennas is restricted in order to keep the grating lobe generated in the transmission / reception pattern sufficiently low. If the element spacing of the transmitting array antenna and the element spacing of the receiving array antenna do not satisfy a predetermined constraint, for example, the grating lobe generated in the reception pattern matches the direction of the side lobe of the transmission pattern, and the transmission / reception pattern is relatively There has been a problem that a high grating lobe may occur.
Further, in the first method, if it is intended to observe a wide region without gaps so that the observation ranges do not overlap, it is necessary to scan in angular increments corresponding to the angular resolution of the transmission / reception pattern. Therefore, there is a problem that the scanning time becomes long.
On the other hand, the second method has a problem that the side lobe level rises as a whole even if the grating lobe can be suppressed.

  The present invention has been made to solve the above-described problems, and can extend the observation area without increasing the scanning time, and can increase the reception antenna without causing a grating lobe. An object of the present invention is to obtain a radar apparatus capable of reducing the number of elements.

A radar apparatus according to the present invention is configured of a plurality of element antennas arranged at unequal intervals, and uses a receiving antenna having an aperture length longer than that of a transmitting antenna of a transmitting means, and outputs a pulse signal reflected to a target. Receiving means for repeatedly receiving, for forming a receiving beam in a direction within the main beam of the transmission pattern, and for forming a null in a direction symmetrical to the direction of forming the receiving beam with respect to the traveling direction of the moving body A load calculation means for calculating a load is provided, and a reception beam is formed using a pulse signal repeatedly received by the reception means and a load calculated by the load calculation means.
In addition, the load calculating means forms a reception beam in a direction within the main beam of the transmission pattern and forms a null in a direction symmetric to the direction in which the reception beam is formed with respect to the traveling direction of the moving body. The Doppler folding direction is obtained from the direction in which the received beam is formed, and the load for forming a null in the Doppler folding direction is calculated.

According to the present invention, a pulse signal reflected by a target is repeatedly received using a receiving antenna which is composed of a plurality of element antennas arranged at unequal intervals and has a longer aperture length than the transmitting antenna of the transmitting means. And receiving means for forming a reception beam in a direction within the main beam of the transmission pattern, and a load for forming a null in a direction symmetrical to the direction of formation of the reception beam with respect to the traveling direction of the moving body A load calculating means for calculating, and a reception beam is formed by using the pulse signal repeatedly received by the receiving means and the load calculated by the load calculating means. By combining the received beam null pattern with the symmetrical folding direction that occurs in the pulse Doppler processing for the obtained received signal. , While reducing the number of elements of the receiving antenna, can suppress this folding, also without causing prolonged scan time, it is possible to widen the observation area.
In addition, the load calculating means forms a reception beam in a direction within the main beam of the transmission pattern and forms a null in a direction symmetric to the direction in which the reception beam is formed with respect to the traveling direction of the moving body. In addition to calculating the load, the Doppler folding direction is obtained from the direction in which the received beam is formed, and the load for forming a null in the Doppler folding direction is calculated. There is an effect that the peak can be suppressed.

Embodiment 1 FIG.
FIG. 1 is a block diagram showing a radar apparatus according to Embodiment 1 of the present invention. In the figure, a transmitter 1 generates a reference pulse signal and repeatedly outputs the pulse signal to a transmission antenna 2. The transmission antenna 2 repeatedly transmits the pulse signal output from the transmitter 1 toward the target 24. The transmitter 1 and the transmission antenna 2 constitute transmission means.

The receiving antenna 3 is composed of a plurality of element antennas 4-1, 4-2,..., 4-N arranged at unequal intervals, and the opening length (array length) of the receiving antenna 3 is that of the transmitting antenna 2. It is comprised so that it may become longer than opening length.
The receivers 5-1, 5-2,..., 5-N use the element antennas 4-1, 4-2,. Receive and amplify the signal.
A / D converters 6-1, 6-2,..., 6-N convert the analog pulse signals amplified by the receivers 5-1, 5-2,. Convert to signal.
The storage unit 7 is a memory that temporarily stores the pulse signals A / D converted by the A / D converters 6-1, 6-2,.
The range compression unit 8 performs a pulse compression process on the pulse signal stored in the storage unit 7 to obtain a signal with a higher range resolution (hereinafter referred to as “range profile”).
In addition, the receiving antenna 3, the receivers 5-1, 5-2,..., 5-N, the A / D converters 6-1, 6-2,. The compression unit 8 constitutes reception means.

The symmetric direction suppression filter generation unit 9 forms a reception beam in a direction within the main beam of the transmission pattern, and forms a null in a direction symmetric to the direction in which the reception beam is formed with respect to the traveling direction of the moving body. The load calculating means for calculating the load for this is configured.
The beam combiner 10 combines the range profile acquired by the range compressor 8 and the load calculated by the symmetrical direction suppression filter generator 9 to constitute a receive beam forming means for forming a receive beam.

The own device position information collection unit 11 is equipped with, for example, an inertial navigation device or a GPS receiver, and collects position information indicating the current position of the moving body on which the radar device is mounted.
The synthetic aperture processing unit 12 constitutes synthetic aperture processing means for performing synthetic aperture processing on the received beam formed by the beam synthesizing unit 10 using the position information collected by the own position information collecting unit 11.
The motion compensation unit 13 of the synthetic aperture processing unit 12 uses the position information collected by the own position information collection unit 11 to compensate the phase of the received beam formed by the beam synthesis unit 10.
The coherent integrator 14 of the synthetic aperture processor 12 coherently integrates the received beam after phase compensation by the motion compensator 13.

Next, the operation will be described.
First, the transmitter 1 generates a reference pulse signal and repeatedly outputs the pulse signal to the transmission antenna 2 at intervals of the period T _pri to transmit P pulse signals toward the target 24.
Here, when the opening length (array length) of the transmission antenna 2 is L _t and the wavelength of the pulse signal is λ, the null to null width Δθ _t [rad] of the main beam of the transmission antenna 2 is expressed by the following equation (1). ).
Δθ _t = 2λ / L _t (1)
However, the transmitting antenna 2 is assumed to be scanning the center direction theta _c of main beam of an electronic or mechanical transmission pattern.

The P pulse signals transmitted from the transmitting antenna 2 are irradiated and reflected on the target 24 and arrive at the receiving antenna 3.
Here, as shown in FIG. 2, the receiving antenna 3 is assumed to have N element antennas 4-1, 4-2,..., 4-N arranged in a straight line. Element antennas 4-1 and 4-2, ..., the arrangement of the 4-N are expressed by the parameters _{d n.}
Thus, in the example of FIG. 2, element antennas 4-1, 4-2, ..., the interval _{4-N, d 1, d} 2 -d 1, ···, d N-1 -d N- ₂ unequal intervals.
When the opening length of the receiving antenna 3 is L _r , the following equation (2) is established in relation to parameters indicating the arrangement of the element antennas 4-1, 4-2,.
L _r = d _N−1 (2)
The main beam width Δθ _r [rad] of the receiving antenna 3 is given by the following equation (3).
Δθ _r = 2λ / L _r (3)

  In FIG. 2, N element antennas 4-1, 4-2,..., 4 -N are illustrated as being arranged in a straight line, but N element antennas 4-1, 4-2, and 4 -N are shown. .., 4-N need only be arranged at unequal intervals, for example, may be arranged in a two-dimensional manner.

The receivers 5-1, 5-2,..., 5-N are reflected by the target 24 using the element antennas 4-1, 4-2,. Receive and amplify the pulse signal.
When the A / D converters 6-1, 6-2,..., 6-N receive the analog pulse signals amplified by the receivers 5-1, 5-2,. The analog pulse signal is converted into a digital pulse signal.
The storage unit 7 temporarily stores the pulse signals A / D converted by the A / D converters 6-1, 6-2,.
The range compression unit 8 performs a pulse compression process on the P number of pulse signals stored in the storage unit 7 to obtain a signal with improved range resolution, that is, a range profile.
Here, assuming that the number of range bins is M, n is a natural number equal to or less than N, and m is a natural number equal to or less than M, the range profile output from the range compression unit 8 is X _{n, m} (p) (p = 1, 2,...・, P).

The symmetric direction suppression filter generation unit 9 forms a reception beam in a direction within the main beam of the transmission pattern, and forms a null in a direction symmetric to the direction in which the reception beam is formed with respect to the traveling direction of the moving object. Calculate the load to do.
Specifically, the load is calculated as follows.

First, a (θ _f ) is a steering vector corresponding to a desired direction (direction in the main beam of the transmission pattern) θ _{f in} which the beam is to be directed, and an azimuth direction in which a null is to be formed (for example, relative to the traveling direction of the moving body). The steering vector corresponding to θ _sg is a (θ _sg ) (g = 1, 2,..., G).
Here, for generalization, consider forming nulls in G different directions. These steering vectors are expressed by the following equations. Hereinafter, a desired direction in which the beam is aimed is referred to as a steering angle.
a (θ _f ) = [1 exp ((2πj · d ₁ · sin θ _f )) / λ)
... exp ((2πj · d _M−1 · sin θ _f ) / λ)] ^T (4)
a (θ _sg ) = [1 exp ((2πj · d ₁ · sin θ _sg ) / λ)
... exp ((2πj · d _M−1 · sin θ _sg ) / λ)] ^T (5)
g = 1, 2,..., G
However, the superscript T in the equations (4) and (5) represents transposition of the matrix.

Next, a load vector w in which loads w _n (n = 1, 2,..., N) for N element antennas 4-1, 4-2,. Define as follows.
w = [w ₁ w ₂ ... w _N ] ^T (6)
The symmetric direction suppression filter generation unit 9 determines the load w so as to minimize the square error σ defined by the following equation in order to generate a null in the azimuth direction to be suppressed with a gain in the steering angle direction set to 1. To do.
σ = ‖y−A · w‖ ² (7)

However, A is a (G + 1) × N matrix in which steering vectors corresponding to the steering angle and the azimuth direction forming the null are collected, and is represented by the following equation (8).
y is a (G + 1) -dimensional vector representing a desired gain pattern.
In order to generate a pattern having a peak in the direction of the steering angle θ _f and a null in the direction of θ _sg , the vector y is set as in the following equation (9).
A = [a (θ _f ) a (θ _s1 )... A (θ _sG )] ^{* T} (8)
y = [1 ε... ε] ^T (ε << 1) (9)
However, the superscript * represents a complex conjugate.

Thereby, the load w can be calculated | required by the least squares method, as shown in the following formula | equation (10).
w = A ⁺ y (10)
However, the superscript + in Expression (10) represents a pseudo inverse matrix.

The symmetric direction suppression filter generation unit 9 generates all the steering angles θ _{f, k} (k = 1, 2,..., K) and the direction θ _sg, By repeating the above calculation for _k combinations, the required number of loads w _k is calculated.
Here, K is the number of beams to be formed.
w _k = A _k ⁺ y (k = 1, 2,..., K) (11)
A _k = [a (θ _{f, k} ) a (θ _{s1, k} )... A (θ _{sG, k} )] ^{* T}
(12)

The symmetric direction suppression filter generation unit 9 sets the direction θ _{sg, k} to include −θ _{f, k} and forms a desired K beam pattern according to the equation (11). Calculate the load.
The received beam pattern formed by using the load w calculated by the symmetric direction suppression filter generation unit 9 has a peak in the desired direction θ _{f, k} , and the desired direction θ _{f, This} is a pattern in which a null is directed in a direction -θ _{f, k} symmetrical to _k .

When the symmetrical direction suppressing filter generation unit 9 calculates the load w _k as described above, the beam combining unit 10 calculates the range profile X acquired by the range compression unit 8 as shown in the following equation (13). _{n, m} (p) and the load w _k are combined to form a reception beam.
Y _m = W ^{* T} X _m (p) (13)
W = [w ₁ w ₂ ... w _K ] (14)
X _m (p) = [X _{1, m} (p) X _{2, m} (p)... X _{N, m} (p)] ^T
(15)
Y _m (p) = [Y _{1, m} (p) Y _{2, m} (p)... Y _{K, m} (p)] ^T
(16)
Y _{k, m} (p) in Expression (16) is a signal of the mth range cell of the range profile in the kth beam when the pth pulse signal is transmitted and received.
The beam synthesis unit 10 outputs the range profile Y _{k, m} (p) to the synthesis aperture processing unit 12.
Note that, for each range cell, a vector obtained by combining the signals of K beams is defined as Y _m (p).

The own position information collection unit 11 is equipped with, for example, an inertial navigation device or a GPS receiver, and collects position information indicating the current position of the moving body on which the radar device is mounted.
The synthetic aperture processing unit 12 performs synthetic aperture processing on the range profile Y _{k, m} (p) output from the beam synthesizing unit 10 using the position information collected by the own position information collecting unit 11.
Specifically, the synthetic aperture process is performed as follows.

The motion compensation unit 13 of the synthetic aperture processing unit 12 obtains the phase at the time t of the pulse signal reflected by the target 24 using the position information collected by the own position information collection unit 11.
FIG. 3 is an explanatory diagram for explaining a method of calculating the phase of the pulse signal.
In the figure, reference numerals 21 to 23 indicate points where the transmitting antenna 2 and the receiving antenna 3 mounted on the mobile object exist at each time. In the example of FIG. 3, it is assumed that the transmission antenna 2 and the reception antenna 3 are moving linearly at a speed v.
Further, it is assumed that time T elapses while moving from the point 21 to the point 23. In this case, the distance between the points 21 and 23 is v × T.
Further, it is assumed that the point 22 is an intermediate point between the point 21 and the point 23. Here, the point 22 is set to the origin O of the xy coordinate system, and the moving directions of the transmitting antenna 2 and the receiving antenna 3 are set to the positive y-axis. The direction.
Further, the target 24 exists at a position that forms an angle θ between the positive direction of the y-axis and the azimuth direction, and the distance between the origin O and the target 24 is R ₀ . In the following description, the angle θ is called a squint angle.

As is clear from FIG. 3, the distance R (θ, t) from the moving object to the target 24 at time t is calculated by the following equation (17) from the cosine theorem.
R (θ, t) = (R ₀ ² + (v · t) ² −2R ₀ · v · t · cos θ) ^1/2 (17)
In Expression (17), when the moving distance vt of the moving body is shorter than the distance R ₀ , the right side of Expression (17) can be approximated as Expression (18).
R (θ, t) ≈R ₀ −v · t · cos θ (18)

If the equation (18) is used, the pulse signal transmitted and received at the wavelength λ reciprocates the distance R (θ, t), and therefore the phase φ (θ, t) is given by the following equation (19).
Therefore, the motion compensation unit 13 obtains the phase φ (θ, t) of the pulse signal reflected by the target at time t by calculating the following equation (19).
φ (θ, t) = 4π · R (θ, t) / λ
≈4π (R ₀ −v · t · cos θ) / λ (19)

Next, the motion compensator 13 shifts the received P pulse signals, that is, the range profile Y _{k, m} (p) output from the beam combiner 10 as necessary, and the range bin including the target 24 is obtained. Adjust to the same range bin number m.
The shift amount here depends on the steering angle θ _{f, k} of the beam. Since the distance traveled by the moving body during transmission of p pulse signals with the repetition period T _pri is v × p × T _pri , the shift amount sk (p) of the range profile in the steering angle θ _{f, k} direction Can be calculated by the following equation (20).
sk (p) = v · p · T _pri · cos θ _{f, k} (20)

The range profile Y _{k, m} (p) is shifted by the shift amount sk (p) calculated by the equation (20) for each beam, and the range profile in which the range bins are aligned is _defined as Y ′ _{k, m} (p).
The motion compensation unit 13 compensates the phase φ (θ, t) calculated by the equation (19) according to the following equation (21).
Z _{q, m} (p) = Y ′ _{k, m} (p) exp {−jφ (θ _q , pT _pri )} (21)
p = 1, 2,..., P
q = 1, 2,..., Q
k = 1, 2,..., K
m = 1, 2,..., M
Here, Q is the number of samples in the azimuth direction after coherent integration. However, to ensure adequate reception gain in the theta _q direction can be taken, theta _q and theta _{f, k,} it is desirable to satisfy the equation (22) below.
| Θ _q −θ _{f, k} | ≦ Δθ _r / 2 (22)

When the motion compensation unit 13 compensates for the phase φ (θ, t) as described above, the coherent integration unit 14 of the synthetic aperture processing unit 12 starts from the motion compensation unit 13 as shown in the following equation (23). The output phase-compensated signal Z _{q, m} (p) is coherently integrated.
<Z _{q, m} > = ΣZ _{q, m} (p) (23)
q = 1, 2,..., Q
m = 1, 2,..., M
However, Σ in the equation (23) is a sum symbol, and the sum of Z _{q, m} (p) from p = 1 to P is calculated.

However, when the transmission pattern is scanning in the azimuth direction, the number of pulses that can be integrated in a certain azimuth direction θ _k is transmitted / received while the irradiation range of the main beam of the transmission pattern includes the azimuth direction θ _k . Determined by the number of pulses.
When the transmission pattern is scanned at the angular velocity ω, the number of pulses P ₀ that can be integrated in the azimuth direction θ _k can be calculated from the following equation (24).
P ₀ = [θ _max / (ω · T _pri )] (24)
P ₀ ≦ P

Here, θ _max is an angular width that can be regarded as having a sufficiently high gain with respect to the peak gain in the transmission pattern. Therefore, for example, the half width of the beam can be used as θ _max .
[] In Equation (24) is the maximum integer that does not exceed the value of the content equation, that is, θ _max / ω · T _pri .
The coherent integration unit 14 integrates the number of range profiles obtained based on the equation (13) based on the equation (23) in the same direction to obtain <Y _{k, m} >.

By the way, Equation (21) and Equation (23) represent SAR (Synthetic Aperture Radar) processing in the squint direction.
It is known that the azimuth resolution Δθ _d after the coherent integration by the equation (23) is calculated by the following equation (25).
Δθ _d = λ / (2v · T · sin θ) (25)

However, in equation (25), the resolution is defined as a 4 dB width.
As is generally known, the SAR processing does not improve the resolution in the azimuth direction near the front in the traveling direction. This can also be confirmed from the fact that the right side diverges when θ = 0 in Equation (25).
That is, the resolution of Expression (25) is not achieved over the entire observation region. Eventually, the overall azimuthal resolution is determined by the higher one of the resolution Δθ _r determined by the aperture length L _r of the receiving antenna 3 and the resolution Δθ _d of equation (25).
That is, if min {A, B} is an expression for selecting the smaller one of A and B, the resolution Δθ is given by the following expression (26).
Δθ = min {Δθ _r / 2, Δθ _d }
= Min {λ / L _r , λ / (2v · T · sin θ)} (26)

On the other hand, in the region where the squint angle θ is large, the angular resolution in the azimuth direction obtained as a result of the coherent integration of Expression (23) may be higher than the angular resolution determined by the thickness of the main beam of the received pattern.
Therefore, when determining the so theta _q satisfy equation (22), it can be seen that it is desirable to determine chopped finer than the θ _q θ _k.

Further, as shown in Expression (19), the phase φ (θ, t) of the pulse signal determined by the distance R (θ, t) from the moving body to the target 24 is a function with respect to the squint angle θ.
Thus, the results of the coherent integration of equation (23), at the same time that the value for the signal of squint angle theta _q direction increases, so does the symmetrical - [theta] _q direction of the signal for the traveling direction of the moving body.
Therefore, only the conventional squint mode SAR process can be used to obtain an image that is folded back and forth with the traveling direction as an axis.
However, in the first embodiment, since the received beam is formed using the load calculated by the symmetric direction suppressing filter generation unit 9, in the range profile Y _{k, m} (p) of each beam after the beam formation, The signal from the symmetrical direction has already been suppressed.

As is apparent from the above, according to the first embodiment, the antennas 4-1, 4-2,..., 4-N are arranged at unequal intervals, and the transmitting antenna 2 In addition to the receivers 5-1, 5-2,..., 5-N that repeatedly receive the pulse signal reflected by the target 24 using the receiving antenna 3 having a longer aperture length, A symmetric direction suppression filter generator 9 that calculates a load for forming a null in a direction symmetric to a direction in which the received beam is formed with respect to the traveling direction of the moving body. Since the received beam is formed using the pulse signal repeatedly received by the receiver and the load calculated by the symmetrical direction suppression filter generation unit 9, the scanning time can be increased. Observing area without inviting It is possible to widen the, without generating a grating lobe, an effect that it is possible to reduce the number of elements of the receiving antenna.

  That is, the element antennas 4-1, 4-2,..., 4-N are arranged at unequal intervals, and the range shift and phase change caused by the movement are compensated for coherent integration. With respect to, an angular resolution determined by the opening length of the receiving antenna 3 can be obtained, and a resolution equivalent to the squint mode SAR can be obtained with respect to the diagonally forward direction of the traveling direction. At the same time, a beam pattern having nulls in the direction symmetrical to the steering angle and the traveling direction is formed using the element antennas 4-1, 4-2,..., 4-N arranged at unequal intervals. Therefore, it is possible to suppress the folding of the signal in the left-right symmetric direction that has occurred in the conventional squint mode SAR.

In the first embodiment, the element antennas 4-1, 4-2,..., 4-N are shown as using the receiving antenna 3 in which the antennas are arranged in a straight line. 4-2,..., 4-N need only be arranged at unequal intervals. For example, the element antennas 4-1, 4-2,. A receiving antenna 3 may be used.
The antenna system of the transmitting antenna 2 is not particularly limited, and any antenna system such as a phased array antenna or an aperture antenna may be used.
In the first embodiment, the case in which the opening surface of the receiving antenna 3 faces the traveling direction of the moving body has been described, but it goes without saying that it can be easily expanded even when it faces the other direction.

Embodiment 2. FIG.
FIG. 4 is a block diagram showing a radar apparatus according to Embodiment 2 of the present invention. In the figure, the same reference numerals as those in FIG.
Doppler folding direction suppression filter generating unit 15 in the same manner as symmetrical direction suppression filter generating unit 9 of FIG. 1, a reception beam forming in the direction of the main beam of the transmission pattern, and to the traveling direction of the moving body In addition to calculating the load for forming a null in a direction symmetrical to the direction in which the reception beam is formed, the load for forming a null in the Doppler folding direction is calculated. The Doppler folding direction suppression filter generation unit 15 constitutes a load calculation unit.

Next, the operation will be described.
Since components other than the Doppler folding direction suppression filter generation unit 15 are the same as those in the first embodiment, only the operation of the Doppler folding direction suppression filter generation unit 15 will be described.

Here, the steering angle is represented by θ _{f, k} , and the direction in which the Doppler turns is represented by θ _{dh, k} .
When transmission / reception is performed with the pulse repetition period T _pri , the Doppler frequency aliasing repeatedly occurs with a width of 1 / T _pri . From the relationship between the angle in the azimuth direction and the Doppler frequency, the direction θ _{dh, k in which} the Doppler turns back when the steering angle is θ _{f, k} satisfies the relationship of the following equation (27). However, h is a natural number.
(2v · cos θ _{f, k} / λ) − (2v · cos θ _{dh, k} / λ) = h / T _pri
(27)

When Expression (27) is transformed, a direction θ _{dh, k} in which Doppler folding occurs is expressed as Expression (28) below.
θ _{dh, k} = cos ⁻¹ (cos θ _{f, k} −h · λ / (2v · T _pri )) (28)

First, the Doppler folding direction suppression filter generation unit 15 substitutes the value of the wavelength λ, the pulse repetition period T _pri, and the velocity v of the moving object output from the own position information collection unit 11 into Equation (28). Thus, the Doppler folding direction θ _{dh, k} is calculated.
Next, the Doppler folding direction suppression filter generation unit 15 calculates a load w _k that forms a null in the Doppler folding direction θ _{dh, k} .
The calculation method of the load w _k is the same as that of the symmetrical direction suppression filter generation unit 9 in the first embodiment.

That is, the Doppler folding direction suppression filter generation unit 15 sets the steering vector corresponding to the steering direction θ _{f, k} as a (θ _{f, k} ), and steering corresponding to the vicinity θ _{sg, k} in the symmetry direction forming the null. The vector is a (θ _{sg, k} ) (g = 1, 2,..., G), and the steering vector corresponding to the Doppler folding direction θ _{dh, k} forming the null is a (θ _{dh, k} ) (h = 1, 2, ..., H).
a (θ _f ) = [1 exp ((2πj · d ₁ · sin θ _f ) / λ)
... exp ((2πj · d _M−1 · sin θ _f ) / λ)] ^T (29)
a (θ _sg ) = [1 exp ((2πj · d ₁ · sin θ _sg ) / λ)
... exp ((2πj · d _M−1 · sin θ _sg ) / λ)] ^T (30)
g = 1, 2,..., G
a (θ _{dh, k} ) = [1 exp ((2πj · d ₁ · sin θ _{dh, k} ) / λ)
... exp ((2πj · d _M−1 · sin θ _{dh, k} ) / λ)] ^T
(31)
h = 1, 2,..., H

Here, H is the number of Doppler folding directions in which a null is formed and suppressed, and is determined by the width of the observation region, the pulse repetition period T _pri , the velocity v of the moving object, and the like.
The Doppler folding direction suppression filter generation unit 15 sets the gain in the steering angle direction to 1, and calculates a load w _k that generates null in the azimuth direction to be suppressed by the following equation (32).
w _k = A _k ⁺ y (32)
k = 1, 2,..., K
A _k = [a (θ _{f, k} ) a (θ _{s1, k} )... A (θ _{sG, k} )
a (θ _{d1, k} )... a (θ _{dH, k} )] ^{* T} (33)
y = [1 ε... ε] ^T (ε << 1) (34)

Here, A is a (G + H + 1) × N matrix in which the steering angle and the steering vector corresponding to the azimuth direction forming the null are collected, and y is a (G + H + 1) -dimensional vector representing a desired gain pattern.
Further, in order to generate a pattern having a peak in the direction of the steering angle θ _{f, k} and null in the directions of θ _{sg, k} , θ _{dh, k} , the vector y is set as shown in Expression (34).
The load w _k calculated by the Doppler folding direction suppression filter generation unit 15 is output to the beam synthesis unit 10. Other operations are the same as those in the first embodiment.

As is apparent from the above, according to the second embodiment, the load w _k for forming a null in the Doppler folding direction θ _{dh, k} is calculated, so the radar according to the first embodiment described above. In addition to the effects obtained by the apparatus, there is an effect of suppressing the peak generated in the Doppler folding direction.

Embodiment 3 FIG.
5 is a block diagram showing a radar apparatus according to Embodiment 3 of the present invention. In the figure, the same reference numerals as those in FIG.
Similar to the synthetic aperture processing unit 12 in FIGS. 1 and 4, the synthetic aperture processing unit 16 in FIG. 5 uses the position information collected by the own location information collecting unit 11 to receive the signal formed by the beam synthesizing unit 10. Synthetic aperture processing means for performing synthetic aperture processing on the beam is configured.
The range migration compensation unit 17 of the synthetic aperture processing unit 16 performs range migration compensation on the range profile output from the beam synthesis unit 10 (compensation for the shift of the range profile due to movement of the moving body).
The phase compensation unit 18 of the synthetic aperture processing unit 16 compensates for the phase of the focused scene center.
The fast Fourier transform unit 19 of the synthetic aperture processing unit 16 performs fast Fourier transform on the signal after phase compensation by the phase compensation unit 18 in the hit direction.

Next, the operation will be described.
Since the components other than the synthetic aperture processing unit 16 are the same as those in the first and second embodiments, only the operation of the synthetic aperture processing unit 16 will be described.

The processing contents of the range migration compensation unit 17 of the synthetic aperture processing unit 16 are the same as the range profile shift processing in the motion compensation unit 13 in the first embodiment, and the range profiles Y _k, P of received P pulse signals _{. m} (p) is shifted as necessary, and the range bin including the target 24 is adjusted to be the same range bin number m.
The shift amount here depends on the steering angle θ _{f, k} of the beam.
Since the moving distance of the moving body during transmission of the p number of pulse signals at the repetition period T _pri is v × p × T _pri , the shift amount sk (p of the range profile in the steering angle θ _{f, k} direction ) Can be calculated by equation (20) above.

The range migration compensator 17 shifts the range profile Y _{k, m} (p) by the shift amount sk (p) calculated according to the equation (20) for each beam, and the range profile Y ′ _{k, m} (p) is output to the phase compensation unit 18.

The phase compensation unit 18 of the synthetic aperture processing unit 16 compensates for second-order or more changes in the phase of the received beam. That is, the phase component in the center direction of the beam is compensated according to the following equation (35).
Z _{f, k, m} (p) = Y ′ _{k, m} (p) exp {−jφ (θ _{f, k} , pT _pri )}
(35)
p = 1, 2,..., P
k = 1, 2,..., K
m = 1, 2,..., M

As described above, when the phase compensator 18 compensates the phase of the beam in the center θ _{f, k} direction according to the equation (35), the phase change of the signal component from the angle adjacent to θ _{f, k} is approximately linear. Therefore, the resolution in the azimuth direction can be increased by performing Fourier transform in the pulse hit direction.
The fast Fourier transform unit 19 of the synthetic aperture processing unit 16 arranges the range bins at high speed for each range bin with respect to the range profile Z _{f, k, m} (p) obtained by performing the phase compensation processing of Expression (35). Perform a Fourier transform.

  As apparent from the above, according to the third embodiment, since the fast aperture transform is performed and the synthetic aperture processing is performed, the amount of calculation can be reduced and the processing time is shortened. There is an effect that can be done.

It is a block diagram which shows the radar apparatus by Embodiment 1 of this invention. It is explanatory drawing which shows arrangement | positioning of the aperture length of a receiving antenna, and the element antenna which comprises a receiving antenna. It is explanatory drawing explaining the method of calculating the phase of a pulse signal. It is a block diagram which shows the radar apparatus by Embodiment 2 of this invention. It is a block diagram which shows the radar apparatus by Embodiment 3 of this invention.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Transmitter (transmitting means), 2 Transmitting antenna (transmitting means), 3 Receiving antenna (receiving means), 4-1, 4-2, ..., 4-N element antenna, 5-1, 5-2 ..., 5-N receiver (reception means), 6-1, 6-2, ..., 6-N A / D converter (reception means), 7 storage unit (reception means), 8 range compression Unit (reception unit), 9 symmetry direction suppression filter generation unit (load calculation unit), 10 beam synthesis unit (reception beam forming unit), 11 own device position information collection unit, 12 synthetic aperture processing unit (synthetic aperture processing unit) , 13 Motion compensation unit, 14 Coherent integration unit, 15 Doppler folding direction suppression filter generation unit (load calculation unit), 16 Synthetic aperture processing unit (synthetic aperture processing unit), 17 Range migration compensation unit, 18 Phase compensation unit, 19 Fast Fourier Section, 21 point, 22 point, 23 point, 24 goals.

Claims (3)

A transmission means that repeatedly transmits a pulse signal toward a target and a plurality of element antennas arranged at unequal intervals, and using a reception antenna having an aperture length longer than the transmission antenna of the transmission means, A receiving means for repeatedly receiving a pulse signal reflected by a target, a receiving beam is formed in a direction within the main beam of the transmission pattern, and symmetrical to the direction in which the receiving beam is formed with respect to the traveling direction of the moving body A receiving beam for forming a receiving beam by using a load calculating means for calculating a load for forming a null in any direction, a pulse signal repeatedly received by the receiving means, and a load calculated by the load calculating means. and forming means, the radar device that includes a synthetic aperture processing means for performing the synthetic aperture processing on the received beams formed by the receiving beam forming unit The load calculation means forms a reception beam in a direction within the main beam of the transmission pattern, and forms a null in a direction symmetrical to the direction in which the reception beam is formed with respect to the traveling direction of the moving body. In addition to calculating a load for performing, a radar apparatus characterized in that a Doppler folding direction is obtained from a direction in which the reception beam is formed, and a load for forming a null in the Doppler folding direction is calculated. The radar apparatus according to claim 1 , wherein the synthetic aperture processing means compensates the phase of the reception beam formed by the reception beam forming means, and coherently integrates the reception beam after the phase compensation. The synthetic aperture processing unit performs range migration compensation on the reception beam formed by the reception beam forming unit, and then compensates for a second-order or more change in the phase of the reception beam, and Fourier transforms the phase-compensated reception beam. The radar apparatus according to claim 1, wherein:

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