JP3747405B2 - Antenna verification method and antenna verification processing apparatus - Google Patents

Description

【００１７】
この（１）式は、３ＧＰＰ（３ｒｄ Ｇｅｎｅｒａｔｉｏｎ Ｐａｒｔｎｅｒｓｈｉｐ Ｐｒｏｊｅｃｔ）によるアンテナベリフィケーションについてのＡｎｎｅｘ Ａの説明内容を基にしたものである。又（１）式のσ² は、
σ² ＝（Ｉ^D ）² ＋γ² （Ｉ^C ）²
であり、（Ｉ^D ）² ，（Ｉ^C ）² は、それぞれＤＰＣＨデータのパイロット部と、ＣＰＩＣＨパイロットとの分散を示す。
【００１８】
又フィードバック情報をアンテナベリフィケーションの値に反映させる程度を示す定数をχとする。この定数χは、χ＝０によりフィードバック情報を反映させないことを示し、又χ＝∞によりフィードバック情報のみを用いることを示す。そして、スロット番号が偶数の場合に、例えば、フィードバック情報がπであると、Ｔ＝χln（９６／４）＝３．１７８χ、又０であると、Ｔ＝χln（４／９６）＝−３．１７８χとすることができる。なお、（９６／４）は、フィードバック情報が正しく伝送される確率を９６％とし、又伝送誤りの確率を４％とした場合の比を示すもので、Ｗ−ＣＤＭＡシステムに於ける伝送誤り率は４％程度である。
【００１９】
そして、次の（２）式の条件が成立する場合は、ｘ₀ ＝０、成立しない場合は、ｘ₀ ＝πとする。
【数２】

ただし、（ｉ＝１）から（ｉ＝Ｎ）までのパス（ｐａｔｈ）についての前述の値Ｖ_i の累積値の実数部（Ｒｅ）を示す。
【００２０】
又スロット番号が奇数の場合、フィードバック情報が−π／２であると、Ｔ＝χln（９６／４）＝３．１７８χ、又π／２であると、Ｔ＝χln（４／９６）＝−３．１７８χとする。そして、（３）式の条件が成立する場合、ｘ₁ ＝π／２、成立しない場合、ｘ₁ ＝−π／２とする。
【数３】

ただし、（ｉ＝１）から（ｉ＝Ｎ）までのパス（ｐａｔｈ）についての前述の値Ｖ_i の累積値の虚数部（Ｉｍ）を示す。
【００２１】
前述のｘ₀ ，ｘ₁ により、共通パイロットチャネルＣＰＩＣＨの位相φは、次の（４）式に示すように推定することができる。
【数４】

【００２２】
前述の（１）式に従った従来のアンテナベリフィケーション処理部は、図１４に示す構成を有するものであり、１１１はＣＰＩＣＨ逆拡散部、１１２はＤＰＣＨ逆拡散部、１１３はアンテナ＃２平均化部、１１４はアンテナ＃２パイロット部平均化部、１１５，１１６は分散算出部、１１７，１１８は電力化処理部、１１９，１２４は逆数部（１／ｘ）、１２０，１２２，１２５は乗算器、１２１は平方根算出部（√）、１２３は加算器を示す。又分散算出部１１５，１１６は、下方に示すように、電力化部１３１，１３４と、平均化部１３２，１３３と、加算器１３５とから構成されている。
【００２３】
ＣＰＩＣＨ逆拡散部１１１によりＣＰＩＣＨコードで逆拡散し、アンテナ＃２平均化部１１３により第２のアンテナ＃２対応のパイロットパターンに従って平均化して、ＣＰＩＣＨデータの平均値ｈ^C を求め、その複素共役ｈ^C*を乗算器１２５に入力する。又ＤＰＣＨ逆拡散部１１２によりＤＰＣＨコードで逆拡散し、アンテナ＃２パイロット部平均化部１１４により第２のアンテナ＃２対応のＣＰＣＨパイロットパターンに従って平均化し、ＤＰＣＨパイロット部の平均値ｈ^D を求めて乗算器１２５に入力する。
【００２４】
又電力比を示すγは、γ＝（ＤＰＣＨ平均電力／ＣＰＩＣＨ平均電力）^1/2 により求めるもので、電力化部１１７，１１８によりｈ^C ，ｈ^D を電力値とし、ＣＰＩＣＨ平均電力を逆数部１１９により逆数とし、乗算器１２０により前述のγを求める式の括弧内の値を求め、平方根算出部１２１により平方根を求めて、乗算器１２５に入力する。従って、前述の（１）式の分母に相当する値を乗算器１２５に入力したことになる。なお、２^1/2 は定数であるから、図示を省略した手段により乗算器１２５に入力して乗算することができる。
【００２５】
又前述の（１）式の分母の雑音電力σ² は、
σ² ＝（Ｉ^D ）² ＋γ² （Ｉ^C ）²
として求めることができるもので、従って、分散算出部１１５，１１６と乗算器１２２と加算器１２３とにより求めることができる。分散算出部１１５，１１６は、電力化部１３１に電力値とし、平均化部１３３により平均化した値と、平均化部１３２により平均化し、電力化部１３４による電力値との差分を出力する。又分散出力部１１５の出力に、ＤＰＣＨ平均電力とＣＰＩＣＨ平均電力との電力比を示すγ² を乗算器１２２に於いて乗算し、加算器１２３に於いて加算することにより、雑音電力σ² を求め、逆数部１２４により１／σ² を求めて乗算器１２５に入力する。
【００２６】
図１５は従来例のシミュレーション結果の説明図であり、横軸は（２）式及び（３）式の条件式のＴを求める為の定数χ、縦軸はアンテナベリフィケーションの誤り率とし、各種の電波の伝搬環境や送信パワー等を想定したｃａｓｅ１〜ｃａｓｅ６のシミュレーション結果を示す。この場合、定数χの値を０から次第に増加するに従って誤り率が小さくなり、１．５〜２近傍に於いて最小となる。それより増加すると、図示を省略しているが、誤り率が大きくなる。従って、シミュレーション結果から、フィードバック情報を反映させる程度を示す定数χの最適値は、前述の従来例に於いては、１．５〜２近傍の値であることが判る。
【００２７】
【発明が解決しようとする課題】
送信ダイバーシチ方式を適用した従来例に於けるアンテナベリフィケーションは、前述の（１）式に従った構成及び処理を実行するものであり、従って、アンテナベリフィケーションの為の構成は、図１４に示すように、回路規模が大きくなる問題があった。又プロセッサの演算機能により処理する場合でも、ステップ数が多くなる問題があった。
本発明は、送信ダイバーシチ方式に於けるアンテナベリフィケーション処理を簡単化することを目的とする。
【００２８】
【課題を解決するための手段】
本発明のアンテナベリフィケーション方法は、図１を参照して説明すると、第１，第２のアンテナ＃１，＃２を有する基地局１を用いた送信ダイバーシチ方式に於けるアンテナベリフィケーション方法であって、基地局１を送信側、端末局を受信側として、送信側からの送信信号に多重化する共通パイロットチャネルと個別チャネルとにより送信する既知信号を、受信側に於いて受信して、前記共通パイロットチャネルと個別チャネルとに分離し、各チャネルの信号から前記既知信号を除去して求めた共通パイロットチャネルの推定値ｈ^Ｃの複素共役ｈ^Ｃ＊と、個別チャネルの推定値ｈ^Ｄとを基に、アンテナベリフィケーション処理部６によりパス対応の値Ｖ_ｉを
Ｖ_ｉ＝（ｈ^Ｄｈ^Ｃ＊）／｜（ｈ^Ｄｈ^Ｃ＊）｜
により求め、このパス対応の値Ｖ_ｉの加算値と、受信側から送信側に送信するフィードバック情報ｗ_ｆを基にした定数との条件を判定して、個別チャネルのデータに乗算する複素重み係数を求める過程を含むものである。
【００２９】
又受信側から送信側に、フィードバック情報ｗ_f を、Ｉ軸情報とＱ軸情報とし且つ相互に１スロット分シフトした２スロットを用いて送信し、受信側に於いて、フィードバック情報ｗ_f のＩ軸情報とＱ軸情報とを基に推定したＩ軸側の尤度情報とＱ軸側の尤度情報とを、それぞれ２スロット分について求めて加算し、フィードバック情報ｗ_f を基にした定数との条件を判定して、個別チャネルのデータに乗算する複素重み係数を求める過程を含むものである。又フィードバック情報ｗ_f のＩ軸情報とＱ軸情報との何れか一方は、位相情報の更新後の１スロット分についての尤度情報を用い、他方は、位相情報の更新前の２スロット分についての尤度情報を加算して、フィードバック情報ｗ_f を基にした定数との条件を判定して、個別チャネルのデータに乗算する複素重み係数を求める過程を含むことができる。
【００３０】
又本発明のアンテナベリフィケーション処理装置は、受信パス対応の共通パイロットチャネルデータを共通パイロットチャネルコードで逆拡散処理するＣＰＩＣＨ逆拡散部と、このＣＰＩＣＨ逆拡散部の出力信号を共通パイロットのシンボルパターンを用いて平均化する平均化部と、受信パス対応の個別チャネルデータを個別チャネルコードで逆拡散処理するＤＰＣＨ逆拡散部と、このＤＰＣＨ逆拡散部の出力信号を個別チャネルパイロット部のシンボルパターンを用いて平均化する平均化部と、この平均化部からの個別チャネルパイロット部の平均値と、共通パイロットデータの平均値の複素共役とを乗算する第１の乗算器と、この第１の乗算器の出力信号を電力値に変換して逆数を求める手段と、この手段からの逆数と第１の乗算器の出力信号とを乗算してパス対応のベリフィケーションの為のパス対応の値Ｖ_i を出力する第２の乗算器とを含む構成を有するものである。又スロット毎に求めたパス対応の値Ｖ_i のＩ軸成分とＱ軸成分とを２スロット分加算する手段を含むことができる。
【００３１】
【発明の実施の形態】
図１は本発明の実施の態様の要部説明図であり、１は送信側としての基地局、＃１，＃２は基地局の第１，第２のアンテナ、２は携帯電話機等の受信側としての端末局、３は端末局のアンテナ、４は送受信部、５は受信処理部、６はアンテナベリフィケーション処理部、７は送信処理部、８は送受信制御部を示す。
【００３２】
基地局１は、第１のアンテナ＃１から、個別チャネルデータｘ１^(D) と共通パイロットチャネルデータｘ１^(C) とを送信し、端末局２に対しては、第２のアンテナ＃２から、その端末局２からのフィードバック情報ｗ_f を基に決定した複素重み係数ｗを乗算した個別チャネルデータｘ２^(D) と、この複素重み係数ｗを乗算しない共通パイロットチャネルデータｘ２^(C) とを送信する。この場合、基地局１の第１のアンテナ＃１と端末局２のアンテナ３との間の伝搬路による複素振幅係数をα、基地局１の第２のアンテナ＃２と端末局２のアンテナ３との間の伝搬路による複素振幅係数をβとすると、端末局２のアンテナ３により受信される個別チャネルデータと、共通パイロットチャネルデータとは、αｘ１^(D) ＋βｘ２^(D) と、αｘ１^(C) ＋βｘ２^(C) として受信される。
【００３３】
この端末局２では、アンテナ３を介して送受信部４により受信した信号を、受信処理部５に於いて処理する。この受信処理部５は、複数のパス対応のフィンガー部，サーチャ部，ＲＡＫＥ合成部等を含み、前述のように、基地局１の第１，第２のアンテナ＃１，＃２対応のＣＰＩＣＨデータを分離して検出し、それぞれの複素振幅係数α，βを求めて、それらの位相差情報ｗ_f をフィードバック情報とし、送受信制御部８の制御に従って、送信処理部７から送受信部４を介してアンテナ３から基地局１へ送信する。
【００３４】
又端末局２に於いては、アンテナベリフィケーション処理部６により、基地局１の第２のアンテナ＃２から複素重み係数ｗを乗算されたＤＰＣＨデータに対する複素重み係数の推定値を求めるアンテナベリフィケーション処理を行うものであり、このアンテナベリフィケーション処理部６は、次の（５）式に示す構成を有するものである。
【数５】

【００３５】
即ち、σやγを含まない条件式に従った構成とすることにより、回路規模の縮小並びに処理ステップの簡単化を図ることができるもので、前述の（１）式から、前述の（５）式は次のようにして導出できる。先ず、（１）式を（６）式のように変形する。
【数６】

【００３６】
この（６）式の右辺の第１項を前述の（７）式とすると、γがｈ^D とｈ^C との電力比を示すから、Ａの平均値は前述の（８）式で表すことができる。このＡの平均値を示す（８）式は、雑音電力の平均値とＤＰＣＨパイロット部の平均値との比、即ち、Ｓ／Ｎを示すことになり、伝搬路を介した通信の動作点に於いてはＳ／Ｎはほぼ一定になるから、（７）式のＡの平均値を示す（８）式は、ほぼ定数と見做すことができる。従って、（６）式のように、アンテナベリフィケーションの為のｉ番目のパス対応の値Ｖ_i （チャネル推定値）を正規化した値として用いることができる。
【００３７】
図２は、本発明の実施の形態の要部説明図であり、図１のアンテナベリフィケーション処理部６の構成を示し、１１はＣＰＩＣＨ逆拡散部、１２はＤＰＣＨ逆拡散部、１３はアンテナ＃２平均化部、１４はアンテナ＃２パイロット部平均化部、１５は第１の乗算器、１６は電力化部、１７は平方根処理部（√）、１８は逆数部（１／ｘ）、１９は第２の乗算器を示す。
【００３８】
ＣＰＩＣＨ逆拡散部１１によりＣＰＩＣＨコードで逆拡散し、アンテナ＃２平均化部１３により、基地局の第２のアンテナ＃２対応のパイロットのシンボルパターンに従って平均化して、ＣＰＩＣＨデータの平均値ｈ^C を求め、その複素共役ｈ^C*を第１の乗算器１５に入力する。又ＤＰＣＨ逆拡散部１２によりＤＰＣＨコードで逆拡散し、アンテナ＃２パイロット部平均化部１４により、基地局の第２のアンテナ＃２対応のＤＰＣＨパイロットのシンボルパターンに従って平均化し、ＤＰＣＨパイロット部の平均値ｈ^D を求めて第１の乗算器１５に入力する。即ち、（５）式の分子の演算機能を示すものとなる。
【００３９】
又第１の乗算器１５の出力信号ｈ^D ｈ^C*を電力化部１６に入力して電力値とし、平方根算出部１７により平方根を求め、逆数部１８により逆数として第２の乗算器１９に入力する。即ち、第１の乗算器１５の出力を基にした（５）式の分母の算出手段を示すものとなる。従って、乗算器１９から（５）式のｉ番目のパス対応のアンテナベリフィケーションの為の値Ｖ_i を求めることができる。
【００４０】
前述のように、従来例の図１４に示す構成と、本発明の実施の形態の図２に示す構成とを比較すると明らかなように、本発明によれば、γやσを含まない処理でパス対応の値Ｖ_i を求めることができるから、回路規模を著しく縮小することが可能となる。
【００４１】
図３は本発明の実施の形態に於けるシミュレーション結果の説明図であり、横軸は定数χ、縦軸はアンテナベリフィケーションの誤り率として、各種の電波の伝搬環境や送信パワー等を想定したｃａｓｅ１〜ｃａｓｅ６のシミュレーション結果を示す。図１５に示す従来例のシミュレーション結果と比較して、定数χの値が小さいのは、前述の（５）式に示すように正規化を行ったことによるものであり、フィードバック情報を反映させる程度を示す定数χは、ほぼ０．１近傍で誤り率が最小となる。この定数χを用いて前述の（２）式と（３）式との条件式に従って、共通パイロットチャネルの位相を判定することができる。
【００４２】
又前述のように、クローズドループモード１に於いては、端末局から基地局へ送信するフィードバック情報に対して、１ビット分のみが割当てれらているから、端末局は、複素重み係数のＩ軸情報とＱ軸情報とを、１スロット分シフトした状態で、２スロットを用いて送信する。従って、複素重み係数は、Ｉ軸側とＱ軸側とが交互に更新されることになり、Ｉ軸側又はＱ軸側のみに着目すると、２スロットおきに変化することになる。そこで、各スロットの復調に用いる複素重み係数ｗ_v を求める時に、その前後のスロットも含めて、複素重み係数ｗ_v の推定精度を向上することができる。
【００４３】
例えば、図４の（Ａ）に示すように、Ｉ，Ｑ軸対応のフィードバック情報は、それぞれ２スロット分を用いて送信するから、それぞれ矢印で示すタイミングが位相情報の更新タイミングを示す。そして、スロット３２についてのアンテナベリフィケーション処理を行う為の値Ｖ_i ² を求める場合、その前後のスロット３１，３３のアンテナベリフィケーション処理を行う為の値Ｖ_i ¹ ，Ｖ_i ³ も用いるものである。即ち、Ｉ軸側の更新タイミング後のスロット３１，３２と、Ｑ軸側の更新タイミング後のスロット３２，３３との情報を用いる。
【００４４】
即ち、スロット３２に於けるＩ軸側の尤度情報は、前のスロット３１の尤度情報とスロット３２の尤度情報との和のＲｅ（Ｖ_i ¹ ＋Ｖ_i ² ）、Ｑ軸側の尤度情報は、スロット３２の尤度情報と次のスロット３３の尤度情報との和のＩｍ（Ｖ_i ² ＋Ｖ_i ³ ）とすることができる。なお、記号１，２，３は、指数ではなく、スロット３１，３２，３３対応のスロット番号を示す。又スロット３３に於いては、Ｉ軸側の尤度情報を、Ｒｅ（Ｖ_i ² ＋Ｖ_i ³ ）、Ｑ軸側の尤度情報を、Ｉｍ（Ｖ_i ³ ＋Ｖ_i ⁴ ）とすることができる。このような処理により、アンテナベリフィケーション処理の精度を向上することができる。なお、このような処理手段は、例えば、図２に於ける値Ｖ_i を３スロット分蓄積する手段と、加算手段とにより容易に実現することができる。
【００４５】
伝搬路の雑音がガウス雑音であると仮定すると、ＤＰＣＨデータはガウス分布に従った確率で変動する。従って、ＤＰＣＨデータの複素重み係数が、例えば、図１３に於ける複素重み係数（ａ，ａ），（ａ，−ａ），（−ａ，−ａ），（−ａ，ａ）の何れになるかを示す尤度は、その図１３の４点からのガウス分布に比例した値となる。このような点から、注目スロットの前後のスロットの推定値を用いることにより、推定精度を向上できるものである。その場合、例えば、前述の（２）式による条件式は、次の（９）式により表すことができる。
【数７】

同様に、（３）式のＩｍ（Ｖ_i ）は、Ｉｍ（Ｖ_i ² ＋Ｖ_i ³ ）とした条件式とすることができる。
【００４６】
又前述の注目スロット３２の次のスロット３３の情報を待つことになり、その場合の遅延を許容できない時は、図４の（Ｂ）に示すように、例えば、Ｑ軸側の矢印で示す更新タイミング後の注目スロット４２の前のスロット４１のＩ軸側の情報を用いることができる。この場合、Ｑ軸側の情報は、位相情報の更新タイミングによって更新されるから、Ｉ，Ｑ軸側の尤度情報は、それぞれＲｅ（Ｖ_i ¹ ＋Ｖ_i ² ）、Ｉｍ（Ｖ_i ² ）として表すことができる。なお、次のスロットに於いては、Ｒｅ（Ｖ_i ² ）、Ｉｍ（Ｖ_i ² ＋Ｖ_i ³ ）とする。
【００４７】
【発明の効果】
以上説明したように、本発明は、送信ダイバーシチ方式を適用した移動通信システムに於いて、基地局１等の第１，第２のアンテナ＃１，＃２からの送信信号を受信する端末局２等の受信側は、複数のパスを介して受信するから、パス対応のアンテナベリフィケーションの為の値Ｖ_i を、共通パイロットチャネルのパイロットデータの平均値と、個別チャネルのパイロット部の平均値とを用いて求めるもので、従来例のように、電力比γや雑音電力σ等を用いるものではないから、回路規模を縮小し、又処理ステップを削減することが可能となる。又複数スロットについての尤度情報を加算して前述の値Ｖ_i を求めることにより、アンテナベリフィケーションの精度を更に向上することができる。
【図面の簡単な説明】
【図１】本発明の実施の形態の原理説明図である。
【図２】本発明の実施の形態の要部説明図である。
【図３】本発明の実施の形態に於けるシミュレーション結果の説明図である。
【図４】アンテナベリフィケーション処理に用いるスロットの説明図である。
【図５】Ｗ−ＣＤＭＡシステムの基本構成の説明図である。
【図６】受信装置の要部説明図である。
【図７】送信ダイバーシチの説明図である。
【図８】受信信号ベクトルの説明図である。
【図９】フィードバック情報経路の説明図である。
【図１０】スロットフォーマットの説明図である。
【図１１】個別チャネル（ＤＰＣＨ）の復調手段の説明図である。
【図１２】送信ダイバーシチ適用時の個別チャネル（ＤＰＣＨ）の復調手段の説明図である。
【図１３】重み付けの説明図である。
【図１４】アンテナベリフィケーション処理部の説明図である。
【図１５】従来例のシミュレーション結果の説明図である。
【符号の説明】
１ 基地局
＃１，＃２ 第１，第２のアンテナ
２ 端末局
３ アンテナ
４ 送受信部
５ 受信処理部
６ アンテナベリフィケーション処理部
７ 送信処理部
８ 送受信制御部[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an antenna verification method and an antenna verification processing apparatus in a mobile communication system to which a transmission diversity method is applied.
[0002]
[Prior art]
FIG. 5 is an explanatory diagram of a basic configuration of a W-CDMA (Wideband Code Division Multiple Access) system, in which a transmission function of a base station and a terminal station (mobile station) when a transmission diversity method is not applied. An overview of the reception function is shown, 51 is an antenna of a base station, 52 is a combining unit, 53 and 54 are spreading units, 55 is a CPICH (common pilot channel) data unit, 56 is a DPCH (dedicated channel) data unit, and 61 is Terminal station antennas 62 and 63 indicate despreading units.
[0003]
The base station uses CPICH data x^(C)Is spread by the CPICH code in the spreading unit 53, and DPCH data x^(D)Is spread by the DPCH code in the spreading unit 54, synthesized (orthogonal modulation) by the synthesis unit 52, converted into a radio frequency, and transmitted from the antenna 51. If the complex amplitude coefficient of the propagation path between the base station and the terminal station is α, CPICH data x^(C)And DPCH data x^(D)The terminal station receives the same complex amplitude fluctuation, and the terminal station receives and orthogonally demodulates by the antenna 61, and despreads the data αx by the despreading unit 62 using the CPICH code.^(C)In the despreading unit 63, the data αx is despread with the DPCH code.^(D)Is obtained.
[0004]
FIG. 6 shows a main part of a receiving apparatus in the W-CDMA system, where 71 is an antenna, 72 is a down converter, 73 is an AGC amplifier, 74 is an AD converter (A / D), and 75 is channel estimation. , 76 denotes a despreading unit, and 77 denotes a synchronous detection unit. A radio frequency reception signal from the antenna 71 is converted into a baseband signal by the down converter 72, amplified to a certain level by the AGC amplifier 73, converted into a digital signal by the AD converter 74, and then despreading unit 76. A despreading process is performed using the despreading code, and channel estimation is performed in the channel estimation unit 75 based on the above-described CPICH data. Then, the synchronous detection unit 77 performs synchronous detection and outputs channel data.
[0005]
In the W-CDMA system, a transmission diversity method is generally applied. For example, as shown in FIG. 7, two first and second antennas # 1 and # 2 are provided in a base station. The same data is transmitted at predetermined intervals. Dedicated channel data x1 from the first antenna # 1^(D)And common pilot channel data x1^(C)And the dedicated channel data x2 from the second antenna # 2^(D)And common pilot channel data x2^(C)And send. In this case, the complex amplitude coefficient by the propagation path between the first antenna # 1 of the base station and the antenna of the terminal station is α, and the propagation path between the second antenna # 2 of the base station and the antenna of the terminal station When the complex amplitude coefficient by β is β, the dedicated channel data received by the antenna of the terminal station and the common pilot channel data are αx1^(D)+ Βx2^(D)And αx1^(C)+ Βx2^(C)As shown.
[0006]
In this case, when the complex amplitude coefficients α and β indicate the same vector direction, the reception gain is increased by applying the transmission diversity method. For example, the complex amplitude coefficients α and β indicated by the I and Q axes in FIG. When the direction of β is almost opposite, the reception gain is close to zero. Therefore, the phase difference information w of the complex amplitude coefficients α and β_fTo the base station as feedback information, the base station, for example, transmits dedicated channel data x2 transmitted from the second antenna # 2.^(D)Is multiplied by a complex weight coefficient w and transmitted.
[0007]
Thereby, individual channel data x2 received by the terminal station^(D)Is multiplied by w · β, and can be changed so as to be in substantially the same direction as the complex weighting factor α, as indicated by a one-dot chain line arrow. Therefore, the dedicated channel data αx1 by the first antenna # 1^(D)And dedicated channel data wβx2 by the second antenna # 2^(D)Means that they exist in the same quadrant, and the reception gain can be increased by combining them. In this case, only the dedicated channel data is multiplied by the complex weight coefficient w, and the common pilot channel data is transmitted from the second antenna # 2 of the base station without being multiplied by the complex weight coefficient w.
[0008]
As described above, in order to multiply the complex weighting factor w at the base station, the phase difference information w is received from the terminal station._fIs transmitted to the base station as feedback information. For example, as shown in FIG. 9, the despreading unit 83 of the terminal station 82 performs despreading with the CPICH code, and the averaging unit 84 performs despreading. To obtain the complex amplitude coefficient α by averaging with the pilot pattern by the antenna # 1 of the base station 81, and obtain the complex amplitude coefficient β by averaging by the pilot pattern by the antenna # 2 of the base station 81 in the averaging unit 85. The phase comparison unit 86 obtains the phase difference between the complex amplitude coefficients α and β, and the phase difference information w_fTo the base station 81 as feedback information. The base station 81 determines the above-described complex weighting coefficient w based on the feedback information, adjusts the phase and amplitude of the dedicated channel data, and transmits from the second antenna # 2. The common pilot channel data is transmitted from the first and second antennas # 1 and # 2 as the same phase and amplitude.
[0009]
The terminal station 82 needs to detect common pilot channel data corresponding to the first and second antennas # 1 and # 2 of the base station 81. However, since the same spreading code is used in the transmission diversity system, when the common pilot channel data having the same symbol pattern is transmitted from the first and second antennas # 1 and # 2, the despreading process is performed. It cannot be separated only by itself. Therefore, the common pilot channels corresponding to the first and second antennas # 1 and # 2 are assumed to be orthogonal symbol patterns.
[0010]
For example, as in the CPICH slot format shown in FIG. 10A, the symbol pattern is orthogonal with a period of 1 slot. Also, as in the DPCH slot format shown in (B), the data Data has the same symbol pattern, but the pilot pilot 1 corresponding to the first antenna # 1 and the pilot pilot 2 corresponding to the second antenna # 2 are orthogonal to each other. Symbol pattern.
[0011]
Therefore, despreading by the CPICH code is performed in the despreading unit 83 of the terminal station 82, and the first unit # 1 is multiplied and averaged by the symbol pattern corresponding to the first antenna # 1 in the averaging unit 84. CPICH data corresponding to the antenna # 1 can be obtained, and the averaging unit 85 multiplies the symbol pattern corresponding to the second antenna # 2 and averages the CPICH data corresponding to the second antenna # 2. Data can be obtained. Thus, since CPICH data corresponding to the first and second antennas # 1 and # 2 can be separated and detected, the respective complex amplitude coefficients α and β can be obtained. Then, in the phase comparator 86, the phase difference w_fAs feedback information.
[0012]
When the transmission diversity method is not applied, for example, as shown in FIG. 11, the DPCH despreading unit 91 performs despreading with the DPCH code, and inputs to the multiplier (synchronous detection unit) 94, or the CPICH despreading unit. In 92, the signal is despread by the CPICH code and input to the multiplier 94, whereby the despread output signal of the dedicated channel can be multiplied by the channel estimation value of the common pilot channel to output dedicated channel data. .
[0013]
On the other hand, when the transmission diversity method is applied, for example, as shown in FIG. 12, the DPCH despreading unit 101 despreads the DPCH code and inputs it to the multiplier (synchronous detection unit) 107. The CPICH despreading unit 102 performs despreading using the CPICH code, and the averaging units 103 and 104 perform averaging processing using pilot patterns corresponding to the first and second antennas # 1 and # 2. The common pilot channel data corresponding to the first antenna # 1 averaged by the averaging unit 104 corresponds to a value obtained by multiplying the complex amplitude coefficient α, and corresponding to the second antenna # 2 averaged by the averaging unit 105. This is equivalent to a value obtained by multiplying the common pilot channel data by the complex amplitude coefficient β. In this case, since α ≠ β, it corresponds to the complex weight coefficient w obtained by multiplying the reception common pilot channel data corresponding to the second antenna # 2 by the dedicated channel data corresponding to the second antenna # 2 in the base station. Complex weight coefficient w_vIs multiplied by α and β × w_vAre added by the adder 106 and input to the multiplier 107 as a complex conjugate number. Thereby, individual channel data can be output from the multiplier 107.
[0014]
In this case, in the base station, feedback information w from the terminal station_fThe complex weighting factor w is determined by using the base station. In a radio wave propagation environment where transmission errors cannot be ignored, the complex weighting factor w determined by the base station cannot be accurately estimated at the terminal station. Therefore, in the terminal station, feedback information w to be transmitted to the base station_fBased on the complex weight coefficient estimate w_vIs what you want. Antenna verification is based on the common pilot channel (CPICH) estimated value, the dedicated channel (DPCH) estimated value, and the feedback information._vIs a process for obtaining.
[0015]
In the transmission diversity system, STTD mode, closed loop mode 1, closed loop mode 2 and the like are known. For example, in closed loop mode 1, four types of complex weight coefficients shown in FIG. 13 are used. It has been. That is, it can be expressed by 2 bits as shown as (a, a), (a, -a), (-a, -a), (-a, a). However, since only 1 bit is allocated to the feedback information from the terminal station to the base station, the I-axis information and the Q-axis information are alternately transmitted for each slot. In that case, 2-bit I-axis information is transmitted using 2 slots, and similarly, 2-bit Q-axis information is shifted by 1 slot with respect to 2 slots of I-axis information. Send using.
[0016]
Also, the terminal station generally receives radio waves from the base station via multipaths, and therefore, a configuration is applied in which reception processing is performed for a predetermined number of paths in descending order of reception power. For the paths i = 1 to i = N in that case, the above-described complex weight coefficient w for each slot of the i-th path_vValue V for antenna verification equivalent to_i(Channel estimation value) can be obtained by the following equation (1).
[Expression 1]

[0017]
This equation (1) is based on the description of Annex A regarding antenna verification by 3GPP (3rd Generation Partnership Project). Also, σ in equation (1)²Is
σ²= (I^D)²+ Γ²(I^C)²
And (I^D)², (I^C)²Indicates the dispersion of the pilot part of the DPCH data and the CPICH pilot, respectively.
[0018]
A constant indicating the degree to which the feedback information is reflected in the antenna verification value is χ. This constant χ indicates that feedback information is not reflected when χ = 0, and indicates that only feedback information is used when χ = ∞. If the slot number is an even number, for example, if the feedback information is π, T = χln (96/4) = 3.178χ, and if 0, T = χln (4/96) = − 3 .178χ. Note that (96/4) indicates the ratio when the probability that the feedback information is correctly transmitted is 96% and the probability of transmission error is 4%, and the transmission error rate in the W-CDMA system. Is about 4%.
[0019]
If the condition of the following equation (2) is satisfied, x₀= 0, x if not true₀= Π.
[Expression 2]

However, the above-described value V for the path from (i = 1) to (i = N)_iThe real part (Re) of the cumulative value of
[0020]
When the slot number is an odd number, if the feedback information is -π / 2, T = χln (96/4) = 3.178χ, and if π / 2, T = χln (4/96) =- 3.178. When the condition of the expression (3) is satisfied, x₁= Π / 2, x if not true, x₁= −π / 2.
[Equation 3]

However, the above-described value V for the path from (i = 1) to (i = N)_iThe imaginary part (Im) of the cumulative value of
[0021]
X mentioned above₀, X₁Thus, the phase φ of the common pilot channel CPICH can be estimated as shown in the following equation (4).
[Expression 4]

[0022]
The conventional antenna verification processing unit according to the above equation (1) has the configuration shown in FIG. 14, where 111 is a CPICH despreading unit, 112 is a DPCH despreading unit, and 113 is an antenna # 2 average. , 114 is an antenna # 2 pilot unit averaging unit, 115 and 116 are dispersion calculation units, 117 and 118 are power processing units, 119 and 124 are reciprocal units (1 / x), and 120, 122, and 125 are multiplications. , 121 is a square root calculation unit (√), and 123 is an adder. As shown below, the variance calculation units 115 and 116 include power generation units 131 and 134, averaging units 132 and 133, and an adder 135.
[0023]
The CPICH despreading unit 111 despreads with the CPICH code, and the antenna # 2 averaging unit 113 averages it according to the pilot pattern corresponding to the second antenna # 2 to obtain the average value h of CPICH data.^CAnd its complex conjugate h^{C *}Is input to the multiplier 125. Further, the DPCH despreading unit 112 despreads the DPCH code, the antenna # 2 pilot unit averaging unit 114 averages it according to the CPCH pilot pattern corresponding to the second antenna # 2, and the DPCH pilot unit average value h^DIs input to the multiplier 125.
[0024]
Also, γ indicating the power ratio is γ = (DPCH average power / CPICH average power)^1/2Is obtained by the power generation units 117 and 118.^C, H^DIs the power value, the CPICH average power is the reciprocal by the reciprocal unit 119, the multiplier 120 determines the value in parentheses of the above equation for obtaining γ, the square root calculation unit 121 calculates the square root, and inputs the square root . Therefore, a value corresponding to the denominator of the above equation (1) is input to the multiplier 125. 2^1/2Is a constant, it can be input to the multiplier 125 and multiplied by means not shown.
[0025]
Also, the noise power σ of the denominator of the above equation (1)²Is
σ²= (I^D)²+ Γ²(I^C)²
Therefore, it can be obtained by the variance calculation units 115 and 116, the multiplier 122, and the adder 123. The variance calculation units 115 and 116 set the power value to the power generation unit 131, averages the value averaged by the averaging unit 133, averages the average value by the averaging unit 132, and outputs a difference between the power value by the powerization unit 134. In addition, the output of the distributed output unit 115 includes γ indicating the power ratio between the DPCH average power and the CPICH average power.²Is multiplied by the multiplier 122 and added by the adder 123, so that the noise power σ²And 1 / σ by the reciprocal part 124.²Is input to the multiplier 125.
[0026]
FIG. 15 is an explanatory diagram of a simulation result of a conventional example, where the horizontal axis is a constant χ for obtaining T in the conditional expressions (2) and (3), and the vertical axis is an error rate of antenna verification. The simulation result of case1-case6 which assumed the propagation environment of various electromagnetic waves, transmission power, etc. is shown. In this case, the error rate decreases as the value of the constant χ gradually increases from 0, and becomes the minimum in the vicinity of 1.5-2. If it is further increased, the error rate is increased although illustration is omitted. Therefore, it can be seen from the simulation results that the optimum value of the constant χ indicating the degree to which the feedback information is reflected is a value in the vicinity of 1.5 to 2 in the above-described conventional example.
[0027]
[Problems to be solved by the invention]
The antenna verification in the conventional example to which the transmission diversity method is applied executes the configuration and processing according to the above-described equation (1). Therefore, the configuration for antenna verification is as shown in FIG. As shown in FIG. 2, there is a problem that the circuit scale becomes large. In addition, there is a problem that the number of steps is increased even when processing is performed by the arithmetic function of the processor.
An object of the present invention is to simplify the antenna verification processing in the transmission diversity system.
[0028]
[Means for Solving the Problems]
  The antenna verification method of the present invention will be described with reference to FIG. 1. The antenna verification method in the transmission diversity system using the base station 1 having the first and second antennas # 1 and # 2 is described. BecauseWith the base station 1 as the transmission side and the terminal station as the reception side, the common pilot channel multiplexed on the transmission signal from the transmission side and a known signal transmitted by the dedicated channel are received at the reception side, and the common pilot is received. It was obtained by separating the channel into individual channels and removing the known signal from each channel signal.Estimated value h of common pilot channel^CComplex conjugate of^{C *}And the estimated value h of the dedicated channel^DBased on the above, the antenna verification processing unit 6 uses the path-corresponding value V_iThe
  V_i= (H^Dh^{C *}) / | (H^Dh^{C *}) ｜
The value V corresponding to this path_iAnd the feedback information w transmitted from the receiving side to the transmitting side_fAnd a process of determining a condition with a constant based on the above and obtaining a complex weight coefficient for multiplying the data of the dedicated channel.
[0029]
Also, feedback information w from the receiving side to the sending side_fAre transmitted as two I-axis information and Q-axis information and shifted by one slot each other, and at the receiving side, feedback information w_fThe I-axis side likelihood information and the Q-axis side likelihood information estimated based on the I-axis information and the Q-axis information are respectively obtained and added for two slots, and feedback information w_fAnd a process of determining a condition with a constant based on the above and obtaining a complex weight coefficient for multiplying the data of the dedicated channel. Also feedback information w_fOne of the I-axis information and the Q-axis information uses the likelihood information for one slot after the phase information is updated, and the other uses the likelihood information for two slots before the phase information is updated. Add the feedback information w_fA process of determining a condition with a constant based on the above and determining a complex weight coefficient for multiplying the data of the dedicated channel can be included.
[0030]
The antenna verification processing apparatus of the present invention also includes a CPICH despreading unit that despreads the common pilot channel data corresponding to the reception path with a common pilot channel code, and outputs the output signal of the CPICH despreading unit to the common pilot symbol pattern. An averaging unit that averages the received channel, a DPCH despreading unit that despreads the dedicated channel data corresponding to the reception path with a dedicated channel code, and a symbol pattern of the dedicated channel pilot unit for the output signal of the DPCH despreading unit A first multiplier that multiplies the averaging unit to be averaged by using the average value of the dedicated channel pilot unit from the averaging unit and the complex conjugate of the average value of the common pilot data, and the first multiplication. Means for converting the output signal of the multiplier into a power value to obtain an inverse, and the inverse of the means and the output of the first multiplier. Path correspondence of the value V for the path corresponding Verification by multiplying the signal_iAnd a second multiplier for outputting. The value V corresponding to the path obtained for each slot_iMeans for adding the I-axis component and the Q-axis component for two slots.
[0031]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 is an explanatory diagram of the main part of an embodiment of the present invention, where 1 is a base station as a transmission side, # 1 and # 2 are first and second antennas of the base station, and 2 is reception of a mobile phone or the like As a terminal station, 3 is an antenna of the terminal station, 4 is a transmission / reception unit, 5 is a reception processing unit, 6 is an antenna verification processing unit, 7 is a transmission processing unit, and 8 is a transmission / reception control unit.
[0032]
The base station 1 receives dedicated channel data x1 from the first antenna # 1.^(D)And common pilot channel data x1^(C)To the terminal station 2, feedback information w from the terminal station 2 is received from the second antenna # 2._fDedicated channel data x2 multiplied by a complex weighting factor w determined based on^(D)And common pilot channel data x2 not multiplied by the complex weight coefficient w^(C)And send. In this case, the complex amplitude coefficient due to the propagation path between the first antenna # 1 of the base station 1 and the antenna 3 of the terminal station 2 is α, the second antenna # 2 of the base station 1 and the antenna 3 of the terminal station 2 Is the complex amplitude coefficient by the propagation path between the dedicated channel data and the common pilot channel data received by the antenna 3 of the terminal station 2 is αx1^(D)+ Βx2^(D)And αx1^(C)+ Βx2^(C)As received.
[0033]
In the terminal station 2, a signal received by the transmission / reception unit 4 via the antenna 3 is processed in the reception processing unit 5. The reception processing unit 5 includes a plurality of path-corresponding finger units, searcher units, RAKE combining units, and the like. As described above, CPICH data corresponding to the first and second antennas # 1 and # 2 of the base station 1 Are detected separately, the respective complex amplitude coefficients α and β are obtained, and their phase difference information w_fIs transmitted from the antenna 3 to the base station 1 through the transmission / reception unit 4 according to the control of the transmission / reception control unit 8.
[0034]
In the terminal station 2, the antenna verification processing unit 6 obtains an estimated value of the complex weight coefficient for the DPCH data multiplied by the complex weight coefficient w from the second antenna # 2 of the base station 1. The antenna verification processing unit 6 has a configuration shown in the following equation (5).
[Equation 5]

[0035]
That is, by adopting a configuration according to the conditional expression that does not include σ and γ, the circuit scale can be reduced and the processing steps can be simplified. From the above equation (1), the above equation (5) can be achieved. The equation can be derived as follows. First, equation (1) is transformed into equation (6).
[Formula 6]

[0036]
If the first term on the right side of the equation (6) is the above equation (7), γ is h.^DAnd h^CThe average value of A can be expressed by the above-described equation (8). Equation (8) indicating the average value of A indicates the ratio between the average value of noise power and the average value of the DPCH pilot section, that is, S / N, and is used as an operating point for communication via the propagation path. In this case, since the S / N is substantially constant, the equation (8) indicating the average value of A in the equation (7) can be regarded as a substantially constant. Therefore, as shown in equation (6), the value V corresponding to the i-th path for antenna verification._i(Channel estimation value) can be used as a normalized value.
[0037]
FIG. 2 is an explanatory diagram of a main part of the embodiment of the present invention, showing the configuration of the antenna verification processing unit 6 of FIG. 1, wherein 11 is a CPICH despreading unit, 12 is a DPCH despreading unit, and 13 is an antenna. # 2 averaging unit, 14 is an antenna # 2 pilot unit averaging unit, 15 is a first multiplier, 16 is a power generation unit, 17 is a square root processing unit (√), 18 is a reciprocal unit (1 / x), Reference numeral 19 denotes a second multiplier.
[0038]
The CPICH despreading unit 11 performs despreading with the CPICH code, and the antenna # 2 averaging unit 13 performs averaging according to the pilot symbol pattern corresponding to the second antenna # 2 of the base station, thereby obtaining an average value h of CPICH data.^CAnd its complex conjugate h^{C *}Is input to the first multiplier 15. Further, the DPCH despreading unit 12 despreads the DPCH code, the antenna # 2 pilot unit averaging unit 14 averages according to the symbol pattern of the DPCH pilot corresponding to the second antenna # 2 of the base station, and the DPCH pilot unit average Value h^DIs input to the first multiplier 15. That is, the function of calculating the numerator of equation (5) is shown.
[0039]
The output signal h of the first multiplier 15^Dh^{C *}Is input to the power generation unit 16 to obtain the power value, the square root calculation unit 17 obtains the square root, and the reciprocal unit 18 inputs the reciprocal to the second multiplier 19. That is, the denominator calculating means of the equation (5) based on the output of the first multiplier 15 is shown. Accordingly, the value V for the antenna verification corresponding to the i-th path in the equation (5) from the multiplier 19 is obtained._iCan be requested.
[0040]
As described above, as is clear when comparing the configuration shown in FIG. 14 of the conventional example with the configuration shown in FIG. 2 of the embodiment of the present invention, according to the present invention, the processing without γ and σ is performed. Value V corresponding to the path_iTherefore, the circuit scale can be remarkably reduced.
[0041]
FIG. 3 is an explanatory diagram of the simulation results in the embodiment of the present invention, assuming that the horizontal axis is a constant χ, the vertical axis is an error rate of antenna verification, and various radio wave propagation environments and transmission powers are assumed. The simulation results of case 1 to case 6 are shown. Compared with the simulation result of the conventional example shown in FIG. 15, the value of the constant χ is small because the normalization is performed as shown in the above equation (5), and the extent to which feedback information is reflected. The constant χ indicating the minimum error rate is approximately in the vicinity of 0.1. Using this constant χ, the phase of the common pilot channel can be determined in accordance with the conditional expression of the above-described expressions (2) and (3).
[0042]
As described above, in closed loop mode 1, only one bit is allocated to the feedback information transmitted from the terminal station to the base station. Axis information and Q-axis information are transmitted using two slots in a state where the information is shifted by one slot. Accordingly, the complex weight coefficient is alternately updated on the I-axis side and the Q-axis side, and changes only every two slots when focusing only on the I-axis side or the Q-axis side. Therefore, the complex weight coefficient w used for demodulation of each slot_vComplex weight coefficient w including the slots before and after_vThe estimation accuracy of can be improved.
[0043]
For example, as shown in FIG. 4A, the feedback information corresponding to the I and Q axes is transmitted using two slots, and the timing indicated by the arrows indicates the update timing of the phase information. Then, the value V for performing antenna verification processing for the slot 32_i ², The value V for performing the antenna verification processing of the slots 31 and 33 before and after_i ¹, V_i ^ThreeIs also used. That is, the information of the slots 31, 32 after the update timing on the I axis side and the slots 32, 33 after the update timing on the Q axis side are used.
[0044]
That is, the likelihood information on the I-axis side in the slot 32 is the sum of the likelihood information of the previous slot 31 and the likelihood information of the slot 32, Re (V_i ¹+ V_i ²), The likelihood information on the Q-axis side is the sum Im (V of the likelihood information of the slot 32 and the likelihood information of the next slot 33_i ²+ V_i ^Three). Symbols 1, 2, and 3 indicate not slot numbers but slot numbers corresponding to slots 31, 32, and 33. In the slot 33, the likelihood information on the I-axis side is represented by Re (V_i ²+ V_i ^Three), Likelihood information on the Q-axis side is expressed as Im (V_i ^Three+ V_i ^Four). Such processing can improve the accuracy of the antenna verification processing. Such processing means is, for example, the value V in FIG._iCan be easily realized by means for accumulating three slots and adding means.
[0045]
Assuming that the propagation path noise is Gaussian noise, the DPCH data varies with a probability according to the Gaussian distribution. Therefore, the complex weighting coefficient of DPCH data is, for example, any of the complex weighting coefficients (a, a), (a, -a), (-a, -a), (-a, a) in FIG. The likelihood indicating whether or not is a value proportional to the Gaussian distribution from the four points in FIG. From this point, the estimation accuracy can be improved by using the estimated values of the slots before and after the slot of interest. In that case, for example, the conditional expression based on the above-described expression (2) can be expressed by the following expression (9).
[Expression 7]

Similarly, Im (V_i) Is Im (V_i ²+ V_i ^Three).
[0046]
If the information in the slot 33 next to the slot of interest 32 is awaited and the delay in that case cannot be tolerated, as shown in FIG. Information on the I-axis side of the slot 41 before the target slot 42 after the timing can be used. In this case, since the information on the Q-axis side is updated at the update timing of the phase information, the likelihood information on the I and Q-axis sides is Re (V_i ¹+ V_i ²), Im (V_i ²). In the next slot, Re (V_i ²), Im (V_i ²+ V_i ^Three).
[0047]
【The invention's effect】
As described above, according to the present invention, in the mobile communication system to which the transmission diversity method is applied, the terminal station 2 that receives transmission signals from the first and second antennas # 1 and # 2 of the base station 1 and the like. Since the receiving side receives the signal via a plurality of paths, the value V for antenna verification corresponding to the path is used._iIs obtained by using the average value of the pilot data of the common pilot channel and the average value of the pilot portion of the dedicated channel, and does not use the power ratio γ or the noise power σ as in the conventional example. It is possible to reduce the circuit scale and the processing steps. Also, the above-mentioned value V is added by adding likelihood information for a plurality of slots._iTherefore, the accuracy of antenna verification can be further improved.
[Brief description of the drawings]
FIG. 1 is a diagram illustrating the principle of an embodiment of the present invention.
FIG. 2 is an explanatory diagram of relevant parts of an embodiment of the present invention.
FIG. 3 is an explanatory diagram of a simulation result in the embodiment of the present invention.
FIG. 4 is an explanatory diagram of slots used for antenna verification processing;
FIG. 5 is an explanatory diagram of a basic configuration of a W-CDMA system.
FIG. 6 is an explanatory diagram of a main part of a receiving device.
FIG. 7 is an explanatory diagram of transmission diversity.
FIG. 8 is an explanatory diagram of a received signal vector.
FIG. 9 is an explanatory diagram of a feedback information path.
FIG. 10 is an explanatory diagram of a slot format.
FIG. 11 is an explanatory diagram of dedicated channel (DPCH) demodulation means.
FIG. 12 is an explanatory diagram of dedicated channel (DPCH) demodulation means when transmission diversity is applied;
FIG. 13 is an explanatory diagram of weighting.
FIG. 14 is an explanatory diagram of an antenna verification processing unit.
FIG. 15 is an explanatory diagram of a simulation result of a conventional example.
[Explanation of symbols]
1 base station
# 1, # 2 First and second antennas
2 terminal stations
3 Antenna
4 Transceiver
5 Reception processing section
6 Antenna verification processing section
7 Transmission processor
8 Transmission / reception controller

Claims (5)

In the antenna verification method in the transmission diversity system,
A known signal transmitted by the common pilot channel and the dedicated channel multiplexed on the transmission signal from the transmitting side is received at the receiving side, separated into the common pilot channel and the dedicated channel, and the signal of each channel Based on the complex conjugate h ^{C *} of the estimated value h ^C of the common pilot channel obtained by removing the known signal and the estimated value h ^D of the dedicated channel, the value V _i corresponding to the path is expressed as V _i = (h ^{D h C *) / | (h} D h C *) |
Sought by
A process of determining a condition of an addition value of the path-corresponding value V _i and a constant based on feedback information transmitted from the reception side to the transmission side, and obtaining a complex weight coefficient for multiplying the data of the dedicated channel An antenna verification method comprising:   The feedback information is transmitted from the reception side to the transmission side using two slots which are I-axis information and Q-axis information and shifted by one slot each other, and at the reception side, the feedback information I-axis side likelihood information and Q-axis side likelihood information estimated based on the I-axis information and the Q-axis information are respectively obtained for two slots, added, and a constant based on the feedback information 2. The antenna verification method according to claim 1, further comprising a step of determining a complex weight coefficient by which the data of the dedicated channel is multiplied by determining the condition of   The feedback information is transmitted from the reception side to the transmission side using two slots which are I-axis information and Q-axis information and shifted by one slot each other, and at the reception side, the feedback information Either one of the I-axis information and the Q-axis information uses the likelihood information for one slot after the phase information is updated, and the other uses the likelihood information for two slots before the phase information is updated. 2. The antenna verification according to claim 1, further comprising a step of determining a condition with a constant based on the feedback information and obtaining a complex weight coefficient by which the data of the dedicated channel is multiplied. Method. In an antenna verification processing apparatus in a mobile communication system to which a transmission diversity method is applied,
A CPICH despreading unit that despreads the common pilot channel data corresponding to the reception path with a common pilot channel code, an averaging unit that averages the output signal of the CPICH despreading unit using a symbol pattern of the common pilot, A DPCH despreading unit that despreads path-specific dedicated channel data with a dedicated channel code, an averaging unit that averages the output signal of the DPCH despreading unit using a symbol pattern of the dedicated channel pilot unit, and the average A first multiplier that multiplies the average value of the dedicated channel pilot section from the conversion section and the complex conjugate of the average value of the common pilot data, and converts the output signal of the first multiplier into a power value. A means for obtaining a reciprocal number, and a multiplication for the reciprocal number from the means and the output signal of the first multiplier for verification corresponding to the path. Antenna verification processing apparatus characterized by having a configuration including a second multiplier for outputting a path corresponding value V _i. Antenna verification apparatus according to claim 4, characterized in that it comprises means for adding two slots and I-axis component and a Q-axis component of the path corresponding value V _i obtained for each slot.

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