JP3304042B2 - Rotating body control filter circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a rotating body control filter circuit, a control circuit or a control system used for controlling a magnetic bearing used for a high-speed rotating machine, for example.

[0002]

2. Description of the Related Art In general, a rotating shaft of a rotating body supported by a magnetic bearing involves a whirling phenomenon of a motion of a center of gravity (translational motion) and a tilting motion (posture motion) around the center of gravity. In order to suppress such a whirling phenomenon, state quantities such as a position, a speed, and a tilt angle are detected, and a current applied to the electromagnet is controlled based on the detected state quantities. As a means of such control, for example, by passing state quantities related to two radial axes, x, and y axes orthogonal to the rotation axis detected by the sensor, through a control circuit having a transfer function suitable for the purpose of control. , A method of obtaining a predetermined output signal, or a method of using a state quantity estimator called an observer. At that time, the control circuit has conventionally been designed and constructed based on the transfer function of the real number coefficient.

[0003]

In the conventional vibration suppressing method as described above, it is difficult to separate vibration modes having opposite signs with equal or very close absolute values. In the flexible mode of the rotating body, there are often two modes of opposite signs whose absolute values are extremely close. Among them, one of the modes is given enough attenuation for some reason, so please do not touch that mode, but if only the other mode is unstable, only the unstable mode You have to choose. However, in the conventional method as described above, since only the mode cannot be selected, both modes are controlled simultaneously,
Modes that do not need to be touched could be adversely affected.

In a control of a high-speed rotating body, a filter circuit for passing or blocking only a narrow bandwidth near the rotating speed of the rotating body includes a coordinate conversion from stationary coordinates to rotating body coordinates to reduce the rotating speed to zero. It has been a conventional method to obtain a steep characteristic by substituting the above. To that end, Tokuhei 2-421
As disclosed in Japanese Patent Publication No. 25-205, two sets of coordinate conversion circuits configured to generate a positive cosine function corresponding to the rotation angle of the rotating body need to be configured in addition to the low-pass filter.

In the above-mentioned conventional method, it has been impossible to adjust the phase to an arbitrary value irrespective of the frequency similarly to the gain.

The present invention provides a control circuit for a rotating body that facilitates the separation of two modes of opposite sign that are equal or very close in absolute value, and adjusts gain and phase independently of frequency. It is an object of the present invention to provide a control circuit capable of performing such operations.

Incidentally, the conventional low-pass or high-pass filter has no distinction between the positive and negative angular frequencies Ω of the signal, and has a gain characteristic symmetrical with respect to the origin, that is, zero frequency if the negative frequency range is included. This means that the filter is used as a bandpass or band rejection filter having zero as a center frequency. Therefore, as shown in FIG. 16, the higher the breakpoint angular frequency (a, b) is, the wider the bandwidth is, and the abscissa is represented by a logarithmic scale lo of the angular frequency.
It was difficult to obtain a steep gain characteristic when a linear scale was used instead of gΩ.

In the gain characteristic of a normal Bode diagram, the slope (dB / dec) of the broken line approximation is apparently the same in any frequency region. It will be easily understood from the fact that the (increment) becomes much larger. For example, at break frequencies of 10 and 1000, the change per dec
Since 100−10 = 90 and the latter is 10000−1000 = 9000, the former has a steep slope of 100 times on the linear scale even with the same slope.

In the control of the rotating body, since the control is performed from two orthogonal radial axes, the x-axis related signal often uses a complex variable using an imaginary number multiplied by j for the y-axis related signal. . A complex coefficient transfer function including complex coefficients is a very effective means in that case. In such a complex system, the sign of the frequency should be distinguished sharply, and the characteristic root of the system is not necessarily a conjugate root.

For example, a first-order low-pass filter a / (a +
In s), it may be considered that a is a corner angular frequency and zero is a central angular frequency. Therefore, in a complex system, the central angular frequency is not limited to zero, and if, for example, ω, a /
{A + (s-jω)} changes to a complex coefficient transfer function with the symmetry point as ω. At the same time, a is not from zero, but a distance from ω or half the passband. Even if ω is a high value, it will be returned to the origin, so in the above example, ω is 1000,
If a is selected as 10, the slope of the polygonal line is about 100 on a logarithmic scale.
Double steepness is obtained, and of course (ω-a) on the opposite side of ω also becomes a break point, and a narrow steep bandpass characteristic is obtained.
(FIG. 17). Further, it has been conventionally impossible to sharply change the phase by ± 180 ° only during a specific angular frequency.
The present invention has been made based on the above findings.

[0011]

According to a first aspect of the present invention, there is provided a radial two-axis orthogonal to a rotation axis (z-axis) of a rotating body.
In a system for controlling (x, y axes) by an electromagnetic force or the like, the x, y
Axis-related input signals (the Laplace conversion amount is Ux,
Uy) is represented by one complex variable ( U = Ux + jUy),
The transfer function of a low-pass or high-pass filter with real coefficients is F (s), the central angular frequency irrespective of positive or negative to be passed or blocked is ω, the imaginary unit is j, and the Laplace operator s of the transfer function F (s) is (s the complex coefficient transfer function is replaced with -Jeiomega) and F (s-jω), the input U is complex-coefficient transfer function F
The x- and y-axis-related output signals (the Laplace transform amounts thereof are Vx and Vy, respectively) that have passed through (s-jω) are converted into one complex variable
In a transfer expression U · F (s−jω) = V (1) obtained by expressing ( V = Vx + jVy), the transfer after multiplying both sides of the expression by the denominator of F (s−jω). A rotating body control filter circuit, characterized in that the real number part and the imaginary part of the expression are connected by a transmission element of a real number coefficient so as to be equal to each other.

[0012] According to a second aspect of the invention, passage and blocking want positive or negative the center angular frequency and omega ₁ and omega _2, respectively, and omega ₁ as ≠ omega _2, before and after the bandwidth about the omega ₁ In order to pass the component including (2a) and block the component including the bandwidth (2b) before and after ω ₂ as a center, k is an arbitrary gain or coefficient, z is a dimensionless constant, The complex coefficient transfer function is k (s + b−jω ₂ ) / (s + a
−jω ₁ ) or k (s + b−jω ₂ ) / {(s−jω ₁ ) ² +2
^{2. The} filter circuit for controlling a rotating body according to claim 1, wherein the filter circuit is provided in a form of za (s−jω ₁ ) + a ² }.

According to a third aspect of the present invention, in the numerator and / or denominator of the transfer function F (s) according to the first aspect, k is an arbitrary gain or coefficient, and 2a is the center angular frequency. K (s ² + 2za)
s + a ² ), where s is replaced by (s−jω), and dimensionless number z is arbitrarily selected to obtain the vicinity of a break point at both ends of the bandwidth in the gain characteristic of the frequency characteristic of the filter. 2. The filter circuit for controlling a rotating body according to claim 1, wherein is set closer to an ideal filter.

According to a fourth aspect of the present invention, a plurality of the rotating body control filter circuits according to any one of the first to third aspects are combined to specify a filter in the entire frequency band including the negative frequency range. And a filter circuit for controlling the rotating body, wherein the filter circuit passes or blocks the band. Of course, not only the passing or blocking of the gain characteristic, but also the phase characteristic can be used.

According to a fifth aspect of the present invention, in the filter for controlling a rotating body according to any one of the first to fourth aspects, each instantaneous value is entered as a variable without using ω, ω ₁ and ω ₂ as constants. This is a tracking filter characterized by the following.

In a system for controlling two radial axes (x, y axes) orthogonal to a rotation axis (z axis) of a rotating body by an electromagnetic force or the like, a necessary angular frequency component or band component is obtained from an input signal regardless of positive or negative. And a part for giving a necessary phase angle to the extracted component, and an output passed through both parts is used for estimating other state quantities necessary for control or is fed back. And a control circuit for the rotating body for stabilizing the mode of the extracted component, wherein the portion that gives the phase angle is A where A and B are real constants.
+ a complex gain circuit represented by jB≡ Fc, Fc of the
between input (Ux, Uy) and output (Vx, Vy)
Connect so that Vx = AUx-BUy, Vy = BUx + AUy,
By arbitrarily selecting A and B, the phase of the output can be increased or decreased by an arbitrary phase angle irrespective of the input frequency, and any necessary only in a frequency region necessary for estimating a control or control-related signal can be obtained. A phase angle may be given.

Further, in a system for controlling two radial axes orthogonal to the rotation axis of the rotating body by electromagnetic force or the like, a related signal in the radial two-axis direction is input, and a positive or negative signal required for the configuration of the control signal in the two-axis direction is input. A rotating body control filter circuit for extracting a negative angular frequency component, wherein an integrator is provided in a signal path of each axis, and negative feedback with a gain of 1/2 of a pass band width is applied to each of the integrators. Feedback from each of the integrators so that the feedback from the first axis to the second axis is positive and the feedback from the second axis to the first axis is negative. The center angular frequency to be passed may be inserted as a gain to selectively pass the angular frequency component.

Further, in a system for controlling two radial axes orthogonal to the rotation axis of the rotating body by electromagnetic force or the like, a related signal in the radial two-axis direction is inputted, and a positive or negative signal required for forming a control signal in the two-axis direction is input. A rotary body control filter circuit for blocking or passing a negative angular frequency component, wherein a direct connection path having a gain of 1 is provided in a signal path of each axis, and a primary low-pass filter is provided in parallel with the direct connection path. The output is multiplied by a value obtained by subtracting half of the rejection bandwidth or the half of the tail band (difference in outer corner frequency) from the corner frequency of the low-pass filter, and the direct connection path. And a path that crosses the axes from the output of each low-pass filter and feeds back to the low-pass filter is provided. Insert as down, first at the connection point
The path from the axis to the second axis is added so that the path from the second axis to the first axis is subtracted, and the signal of the band component centered on the central angular frequency is blocked or passed. Is also good.

Further, in a system for controlling two radial axes orthogonal to the rotation axis of the rotating body by electromagnetic force or the like, a related signal in the radial two-axis direction is input, and a positive or negative signal required for forming a control signal in the two-axis direction is input. passed through a negative angular frequency components omega _1, a rotating system patronized filter circuit for blocking other positive or negative angular frequency components omega _2, a direct path to a gain of 1 to the signal path of each axis Two parallel circuits are provided by branching from each of the two parallel circuits. One of the two parallel circuits has an angular frequency of 阻止 of the rejection bandwidth and a breakpoint angular frequency of a primary low-pass filter connected to the rear thereof. 1 / of a certain pass bandwidth
2 is given as a gain to be an input to the low-pass filter, and ω ₁ −ω ₂ is given to the other parallel circuit as a gain, and the connection is cross-connected to the input of the low-pass filter; At the connection point, the path from the first axis to the second axis is added so that the path from the second axis to the first is subtracted, and one of the outputs of the low-pass filter is added and connected to the directly connected path. The other output is cross-feedback between the axes to the input of the low-pass filter to give ω ₁ as a gain. At the connection point at the input, the path from the first axis to the second axis is added, The path from the second axis to the first axis may be coupled in a subtractive manner to pass a bandwidth around ω ₁ and block a bandwidth around ω _2. Good.

Also, in a system for controlling two radial axes orthogonal to the rotation axis of the rotating body with an electromagnetic force or the like, a control circuit that inputs related signals in the two radial directions and outputs control signals in the two axial directions. A required angular frequency component or band is extracted from the input signal regardless of whether it is positive or negative, each path of the radial two axes is branched, and a first constant A is given as gain to one of the paths, and And the second path is given as a gain to the other path to cross the axes and connect to one path. At the connection point, the path from the first axis to the second axis is added, The path from the axis to the first axis may be combined in a subtractive manner.

Further, in claim 2, the complex coefficient transfer function is given by k (s + b−jω ₂ ) / (s + a−jω ₁ ), and when the angular frequencies ω ₁ and ω ₂ are arbitrarily selected, b May be set to zero or very small, so that a does not oscillate, so that the phase is sharply reversed over the bandwidth between the two angular frequencies.

[0022]

FIG. 1 is a diagram showing a filter circuit according to a first embodiment of the present invention. This is, for example,
A rotating body that is a supported body, an electromagnet that controls the rotating body in x and y axes orthogonal to the rotation axis, a displacement sensor that measures displacement in each axis direction, and a power amplifier that sends a drive signal to the electromagnet And the two-axis displacement signal U 2 = (U _x , U _y ) from the displacement sensor is input, and the control signal V 2 = (V _x , V _y ) in the two-axis direction is _supplied to the power amplifier. It is used as a filter circuit that constitutes a part of the output control circuit. This circuit is created based on the following transfer function F ₁ (s) of a low-pass filter of the first order real number coefficient. F ₁ (s) ≡1 / (s + σ) (2)

The Laplace operator s of this transfer function F ₁ (s)
Is replaced by (s−jω), and the complex coefficient transfer function is represented by F (s−
jω), the input U is the complex coefficient transfer function F (s−j)
ω), the output signals related to the x and y axes (the Laplace transform amounts thereof are Vx and Vy, respectively) are converted into one complex variable ( V = V
The transfer expression U · F (s−jω) = V (3) obtained by expressing the (x + jVy) is expressed as (U _x + jU _y ) / (s + σ−jω) = V _x + jV _y (4). Multiply both sides of the above equation by (s + σ−jω) and expand
When the real part and the imaginary part are associated with each other, U _x −σV _x −ωV _y = sV _x (5) U _y −σV _y + ωV _x = sV _y (6)

When this is materialized, a filter circuit as shown in FIG. 1 is obtained. That is, an integrator 1 / s is provided in a signal path of each of the x and y axes, and negative feedback is performed on each of the integrators 1 / s with a gain of σ which is の of the pass band width.
The following low-pass filter is configured. Then, feedback is performed from each integrator so as to cross the x and y axes so that the feedback from the x axis to the y axis is positive and the feedback from the y axis to the x axis is negative. In these feedback paths, an angular frequency ω (which may be positive or negative) to be passed is inserted as a gain, whereby the angular frequency component is selectively passed.

In this embodiment, ω = 3750 [rad /
s] and σ = 7.5 [rad / s] are shown in FIG. This shows an effect that a band including the center angular frequency regardless of positive or negative is passed with a sharp characteristic of selectivity.

Incidentally, in the conventional biquad filter corresponding to the filter having the configuration shown in FIG. 1, V = {2zωs / (s ² + 2zωs + ω ² )} U (7) z is a dimensionless number that gives the steepness of the filter, and corresponds to an attenuation ratio or a reciprocal of selectivity.
Equation (7) is a real number system and should be provided separately for the two axes x and y. FIG. 1 shows a specific block diagram thereof.
8 Integrators require a total of four for each two axes.
Further, when z is fixed, the slope of attenuation is constant in the Bode diagram, but when the horizontal axis is set to a linear scale, the steepness decreases in a high frequency region, and the bandwidth increases equivalently.

For this reason, in a use such as a tracking filter, an operation for reducing z in accordance with an increase in ω is required. In addition, since ω is a variable, ω in the figure needs to be a multiplier, and four of them are required, which increases cost or calculation time. Of course, equation (7) does not have a function of discriminating positive and negative ω. On the other hand, the complex filter of FIG. 1 has the advantage that the bandwidth remains unchanged at σ, is simple and clear, and requires about half the hardware, and the digital control requires a short calculation time.

FIG. 3 (a) shows a second embodiment of the present invention, which is a so-called notch filter for blocking only a specific frequency signal. The real coefficient transfer function that is the basis of this notch filter is: F ₂ (s) ≡k (s + b) / (s + a), a>b> 0 (8) The complex coefficient transfer function calculates s of this denominator and (s−j
ω) is replaced by F ₂ (s−jω). The specific connection method is omitted because it is a special case where ω ₁ = ω ₂ = ω in F ₃ described later.

When this is materialized, a filter circuit as shown in FIG. That is, in the signal path of each axis,
A direct connection path having a gain of k is provided, and a low-pass filter (LPF) is provided in parallel with a branch therefrom. On the output side, 1/1 of the rejection bandwidth is obtained from the corner frequency a of the low-pass filter. Value obtained by subtracting b that is 2
There are provided a path for multiplying (ab) as a gain and returning to the direct connection path for subtraction coupling, and a crossing path for crossing the axes from the output of each low-pass filter and feeding back to the low-pass filter. . The low-pass filter includes an integrator and a feedback of a gain a, as shown in FIG. In this crossing path, the angular frequency to be blocked is inserted as a gain, and at the connection point, the path from the x-axis to the y-axis is added, and the path from the y-axis to the x-axis is subtracted. I have. Thereby, as a result, the signal of the angular frequency component is canceled and the passage is prevented. This filter circuit can prevent the passage of only one of the positive and negative angular frequencies, and can compensate for maintaining a stable operation by giving an appropriate phase to the remaining signal. If b>a> 0, the filter becomes a pass filter. If k = a / b, the bottom of the bottom of the pass characteristic becomes 0.
dB. Since there is only a phase change with little change in gain near the stop or outside the pass band (see FIG. 4), this feature can be utilized for control.

FIG. 5 shows a third embodiment of the present invention, in which the first and second embodiments are combined. Functionally, the above two filter circuits are connected in series. Works the same as if you let it. That is, passed through a positive or negative angular frequency components omega ₁ required to configure two axial directions of the control signal,
A filter circuit for blocking other positive or negative angular frequency components omega _2.

From the transfer function F ₃ ≡k (s + b) / (s + a) (9) when the denominator and the denominator are both of the first order, the s of the denominator and the denominator are (s−jω ₁ ) and (s−jω ₁ ), respectively. jω
_The complex coefficient transfer function F ₃ ≡k (s + b−jω ₂ ) / (s + a−jω ₁ ) obtained by replacing with ( ₂ ) is expressed by (10). First, measure the dimension reduction of the molecule _{F 3 = k {s + a} -jω 1 -a + b + j (ω 1 -ω 2)} / (s + a-jω 1) = k + k [{b-a + j (ω 1 -ω 2) } / (S + a−jω ₁ )] (11). Therefore, multiplying the input by k and passing it through,
It can be decomposed into a first-order lag element including the complex constant of the second term or a low-pass filter. Here, the configuration method of only the second term will be described.

[{B−a + j (ω ₁ −
ω ₂ )} / (s + a−jω ₁ )] and input U ₂ = ( U ₂ x + j U ₂ y)
Is multiplied by the output V ₂ = V ₂ x + j V ₂ y, the real part and the imaginary part of the complex equation after multiplying both sides by the denominator are equal, so that U ₂ x (b−a ) −U ₂ y (ω ₁ −ω ₂ ) = V ₂ x (s + a) + V ₂ yω ₁ (12) U ₂ x (ω ₁ −ω ₂ ) + U ₂ y (ba) = V ₂ y (s + a) ) −V ₂ xω ₁ (13) is obtained, and from equation (12), V ₂ x = [U ₂ x (ba) −U ₂ y (ω ₁ −ω ₂ ) −V ₂ yω ₁ ] / (s + a) (14) Similarly _{(13) V 2 y = [} U 2 x (ω 1 -ω 2) + U 2 y (b-a) + V 2 xω 1] from equation / (s + a) (15 ) is obtained Can be Multiplying the denominator by the conjugate of the denominator in equation (10) to make the denominator a real number is not desirable because it increases the number of components, increases the order of the system, and increases the calculation time in digital control. The connections according to equations (14) and (15) are shown in FIG.
Is shown in the part other than the passage.

A signal path for each of the x and y axes is provided with a direct connection path having a gain of 1 and two parallel circuits branched from each of the paths. One of the parallel circuits includes a low-pass filter LP connected to a half of the stop band and a rear part thereof.
An amount (ba) obtained by subtracting the corner frequency a of F is given as a gain, and this is input to the low-pass filter LPF. On the other hand, (ω ₁ −ω ₂ ) is given as a gain and connected to the input of the low-pass filter LPF crosswise. The connection is such that the path from the first axis to the second axis is added and the path from the second axis to the first is subtracted.

A circuit including the same low-pass filter LPF and the cross feedback of ω ₁ as in FIG. 3B constitutes a band-pass filter, and is the same as that described in FIG. 1. In FIG. ω is replaced by ω ₁ . That is, an integrator is provided in the signal path of each axis, and each of them is provided with negative feedback accompanied by a which is の of the pass bandwidth. Then, from each integrator, x,
The feedback crossing the y-axis is performed so that the feedback from the x-axis to the y-axis is positive, and the feedback from the y-axis to the x-axis is negative. ₁ is inserted as gain. The output of the bandpass filter as a whole is summed and connected to a direct connection path.

This example is particularly effective when ω ₁ ≒ ω ₂ . In other words, only the mode of ω ₁ is selected and ω ₂
This is the case when you do not want to touch the mode. a = b = 10 [r
ad / s], ω ₁ = 5000 [rad / s] and ω ₂ = 5100 [rad / s] are Bode diagrams in FIG. In spite of such close frequencies and considerable bandwidth, the gain difference between passing and blocking is about 40 [dB], demonstrating the effectiveness of this filter circuit.

FIG. 7 shows an example in which in the circuit of FIG. 5, the phase angle of only a certain angular frequency band is delayed by 180 degrees and the phase of other angular frequency regions is not affected. ω ₁ = 10000 [rad / s], ω ₂ = 40000 [rad / s
s], b = 0, a = 1 [rad / s], and a is sharply reversed over the bandwidth between the two angular frequencies by setting a so small that a does not oscillate. However, the gain characteristics are not completely flat. ω
_{If 1} and ω ₂ are reversed, the phase between them will be reversed by 180 degrees, and the slope of the gain will also be reversed.

Next, an example will be described in which a transfer function is a real number coefficient and a second-order low-pass filter represented by the following equation is used as a band-pass filter having a steep gain characteristic. F ₄ ≡ka ² / (s ² + 2zas + a ² ) (16) z of this filter is usually called an attenuation ratio, and z <
Normally, it is set to 1 to make the bump near the break point thinner and higher to give resonance characteristics. However, in such a usage, the high-frequency side of the bump has a somewhat steep characteristic,
Since the low-frequency side has a disadvantage that the bottom is shallow, it is common to multiply the numerator of equation (16) by s, but the shape design of the bump was complicated (see FIG. 19).

In the present invention, steep characteristics can be obtained even with the secondary filter, and not only the bandwidth but also the shape of the bump can be freely selected, and the sign of the pass band can be freely selected. If s in equation (16) is replaced with (s-jω) in order to take the center angular frequency ω irrespective of positive or negative as a reference, the relationship with inputs U and V is obtained by multiplying both sides by a denominator including complex numbers. , Ka ² U = {(s−jω) ² + 2za (s−jω) + a ² } V = {(s ² + 2zas + a ² −ω ² ) −j ² (ωs + zaω)} (Vx + jVy) (17) The real number of this equation {Ka ² Ux−2 (ωs + zaω) Vy} / (s ² + 2zas + a ² −ω ² ) = Vx (18) {ka ² Uy + 2 (ωs + zaω) Vx} / (s ² + 2zas + a ²⁾ −ω ² ) = Vy (19) FIG. 8 shows a circuit in which the denominator and the denominator of the equations (18) and (19) are subdivided by ω ² and are materialized. However, k is omitted.

Both x and y axes are secondary circuits.
It has extremely advanced functions and performance by crossing between them. Numerical examples are a = 250 [rad / s], ω = 5000 [rad /
FIG. 9 shows three Bode diagrams in which z = 0.5, 0.707, and 1.0 in the case of [s]. However, the abscissa was a linear scale of angular frequency. The bandwidth can be given by 2a, z
The proper choice of helps to improve the corners of the bandwidth. The gain characteristic becomes steeper as the bandwidth is smaller.

FIG. 10 shows still another embodiment of the present invention, in which a required angular frequency component or band is extracted from an input signal of a sensor output irrespective of whether it is positive or negative. FIG. 3 is a block diagram of a control device that can be used to stabilize the component or band by giving a necessary phase angle and feeding it back, or to estimate other information required for control.

This comprises a rotating body 1, which is a supported body, electromagnets 2a, 2b for controlling the rotating body in x, y2 axes orthogonal to the rotation axis thereof, and displacement sensors for measuring displacements in respective axial directions. In a magnetic bearing device including 3a, 3b and power amplifiers 4a to 4d (see FIG. 12) for sending a drive signal to the electromagnet, a two-axis displacement signal U =
(U _x , U _y ) is input and the control signal V = (V _x ,
V _y ) to a power amplifier. Since the electromagnets 2a and 2b can only be attracted, a pair is provided to face each other across the shaft, and control signals with opposite signs are supplied to them.

In the figures, reference numerals 5a, 5b,... Denote the passing or blocking or combination filters described with reference to FIGS. 1 to 3, respectively, for passing different frequencies or removing nearby disturbing frequency components. Is what you do. The complex gain circuits 6a, 6b,...
. Are connected to each other, as shown in FIG.
Each of the two radial axes is branched, and a constant A is given as a gain to one of the two axes, and a constant B is given to the other path as a gain. At this connection point, the path from the x-axis to the y-axis is added, and the path from the y-axis to x
The paths to the axes are connected in a subtractive manner.

Such complex gain circuits 6a, 6b,...
In (2), the phase of the output can be increased or decreased by an arbitrary phase angle tan ^-1 [B / A] regardless of the input frequency by arbitrarily selecting A and B. The gain at this time is (A ² +
B ² ) ^1/2 . Taking a specific example of the control circuit having the above-described configuration for controlling a magnetic bearing as an example, FIGS. 12 and 13 show an example of a configuration of a closed loop system for stabilizing the control.

In FIG. 12, pass filters 5a and 5b
Respectively select and pass frequencies (+ ω and −ω in the example). The complex gain circuits 6a and 6b have A ₁ = 0 and B ₁ = +
k ₊ , A ₂ = 0, B ₂ = −k ₋ , so that the frequency ω has a 90 ° phase change and the k ₊ gain, and the frequency −ω has a −90 ° phase change. receive a gain of k _over. FIG. 13 shows the dynamic characteristic of the control target in the example of FIG.
FIG. 3 is a complex expression block diagram expressed as (s ² + ω ² ).

The operation of the complex gain circuits 6a and 6b can be arbitrarily determined whether the operation is advanced or delayed. Since the gain does not require dynamic characteristics, even if the phase is advanced, the gain does not increase unless both are increased while maintaining the ratio B / A. A plurality of such frequency (band) selection paths may be provided as necessary to provide a gain and a phase necessary for stabilization.

FIG. 14 shows a case where the object to be controlled has the same dynamic characteristic as that of FIG. 13 but has a first-order lag with a large time constant T, for example, where the electromagnet that generates a control force has a transfer function of 1 / l. (1 + Ts), where the + jω mode is sufficiently attenuated by some influence, but only the −jω mode is underdamped. Therefore, in this case, only the −jω mode component is selected, and a constant complex gain
What is necessary is just to give k and feed back.

The Laplace transform of the motion equation of the whole system including the feedback is as follows: k is an unknown complex number, and s ² + ω ² = k / [(1 + Ts) (s + σ + jω)] (20) The characteristic equation of this system is F (s) = (s ² + ω ² ) (1 + Ts) (s + σ + jω)] − k = 0 (21) When Taylor expansion is performed near s = −jω, approximately F (s) | _{s = −j} ω = −2jωσ Since (1−jωT) (s + jω) −k (22), the solution in the vicinity is s = −jω + k / {− 2jωσ (1-jωT)} (23)

Since the second term on the right side is an attenuation term, it is desirable to be a negative real number. Therefore, if k = j (1−jωT) k (24), then s = −jω−k / 2ωσ (25) where k is a positive real number and corresponds to a normal feedback gain. Eventually, the expression (20) is as follows: s ² + ω ² = jk (1−jωT) / [(1 + Ts) (s + σ + jω)] (26) Here, (1-jωT) is newly added to the numerator in order to remove the influence of the first-order lag. The specific connection of the numerator, excluding the selection filter, is given by (x + ji) jk (1-jωT) = k [-y + xωT + j (x + yωT)] (27) Become. With such a configuration, in this control circuit, only -ω out of ± ω can be passed by the selection filter to compensate for the delay of the electromagnet and to provide damping.

FIG. 15 shows an example in which the present invention is applied to a tracking filter. The tracking filter is used, for example, to extract a rotation synchronization component to remove a vibration component close to the rotation speed component, or to increase the bearing stiffness only near the rotation speed. In the tracking filter of the present invention, in the pass filter described with reference to FIG. 1, the cross feedback ω is not a constant but a variable. That is, the output of the integrator is multiplied by a signal from the sensors 3a and 3b or a voltage signal corresponding to the rotation speed generated from a sensor dedicated to rotation speed detection as a variable. Further, σ, which is fed back from the output of the integrator, is 帯 域 of the pass band width, and at the same time as the operation of the low-pass filter, plays a role of preventing oscillation of the selection path. If σ is suppressed to a level that also includes an error in voltage generation corresponding to the rotation speed, unnecessary signals such as noise other than the rotation synchronization signal are reduced.

[0050]

According to the filter circuit for controlling a rotating body of the present invention, it is possible to separate two vibration modes having opposite signs of equal or extremely close absolute values existing in the flexible mode of the rotating body. By selecting only one mode, the state quantity can be estimated or a predetermined control rule can be applied. Therefore, control suitable for the operating conditions of the rotating body can be performed with relatively simple devices and software, and stable operation of the rotating body supported by a magnetic bearing or the like is enabled. Further, in the rotating body control circuit of the present invention,
It is possible to simultaneously adjust the gain and the phase, which have been impossible with the conventional method, independently of the frequency, whereby control suitable for the operating conditions of the rotating body can be performed.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a first embodiment of the present invention.

FIG. 2 is a graph showing characteristics of the circuit according to the embodiment of FIG. 1;

FIG. 3 is a block diagram showing a second embodiment of the present invention.

FIG. 4 is a graph showing characteristics of the circuit according to the embodiment of FIG. 3;

FIG. 5 is a block diagram showing a third embodiment of the present invention.

FIG. 6 is a graph showing characteristics of the circuit according to the embodiment of FIG. 5;

FIG. 7 is a graph showing characteristics of another example in which constants are changed in the circuit of the embodiment of FIG. 5;

FIG. 8 is a block diagram showing a fourth embodiment of the present invention.

FIG. 9 is a graph showing characteristics of the circuit according to the embodiment of FIG. 8;

FIG. 10 is a block diagram of a control device using a control circuit for a rotating body according to an embodiment of the present invention.

FIG. 11 is a block diagram for realizing a complex gain according to another embodiment of the present invention.

FIG. 12 is a block diagram of a control device using a control circuit for a rotating body according to another embodiment of the present invention.

FIG. 13 is a block diagram showing a transfer function of the control circuit of FIG.

FIG. 14 is a block diagram of a control device using a control circuit for a rotating body according to another embodiment of the present invention.

FIG. 15 is a block diagram of a tracking filter of the present invention.

FIG. 16 is a graph showing characteristics of a conventional filter circuit.

FIG. 17 is a diagram illustrating characteristics of the filter circuit of the present invention.

FIG. 18 is a diagram showing a conventional filter circuit.

FIG. 19 is a graph showing characteristics of a conventional secondary filter circuit.

[Explanation of symbols]

 1 Rotating body 2a, 2b Electromagnet 3a, 3b Displacement sensor 4a, 4b, 4c, 4d Power amplifier 5a, 5b Filter 6a, 6b Complex gain circuit

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) G05B 13/02 F16C 32/04 G05B 11/36 501

Claims (5)

(57) [Claims] In a system for controlling two radial axes (x, y axes) orthogonal to a rotation axis (z axis) of a rotating body by an electromagnetic force or the like, an input signal related to the x, y axes (a Laplace conversion amount thereof) is provided. Are represented by one complex variable ( U = Ux + jUy), F (s) is a transfer function of a low-pass or high-pass filter of real coefficients, and ω is a center angular frequency regardless of positive or negative to be passed or blocked. The imaginary unit is j, the transfer function F (s)
The complex coefficient transfer function obtained by replacing the Laplace operator s with (s−jω) is F (s−jω), and the input U passes through the complex coefficient transfer function F (s−jω). In a transfer expression U · F (s−jω) = V obtained by expressing a related output signal (the Laplace transform amounts thereof are Vx and Vy, respectively) by one complex variable ( V = Vx + jVy), F (s− (jω) is multiplied by both sides of the expression, and the transmission expression is connected by a transmission element of a real coefficient such that a real part and an imaginary part of the transmission expression are equal to each other. 2. The positive or negative central angular frequencies to be passed and blocked are ω ₁ and ω ₂ , respectively, and ω ₁ ≠ ω
_As k = 2, k is arbitrarily set to pass components including the bandwidth (2a) before and after ω ₁ and to block components including the bandwidth (2b) before and after ω _2. And the complex coefficient transfer function is k (s + b−jω ₂ ) / (s + a−jω ₁ ) or k (s + b−jω ₂ ) / {(s−jω ^{_{1) 2 + 2za (s-}} jω 1) + a 2 rotation regime patronized filter circuit according to claim 1, characterized in that given in the form of a}. 3. The numerator and / or denominator of the transfer function F (s) according to claim 1, wherein k is an arbitrary gain or coefficient, and 2a is a passing or blocking bandwidth centered on the central angular frequency. ² represented by k (s ² + 2zas + a ² )
Substituting s into (s-jω) and arbitrarily selecting a dimensionless number z so that the vicinity of the break point at both ends of the bandwidth in the gain characteristic of the frequency characteristic of the filter becomes closer to the ideal filter. The rotating body control filter circuit according to claim 1, wherein: 4. A combination of a plurality of rotator control filter circuits according to any one of claims 1 to 3 to pass or block a specific band in the entire frequency band including a negative frequency region. A filter circuit for controlling a rotating body, characterized in that: 5. The rotary body control filter circuit according to claim 1, wherein ω, ω ₁ , ω ₂ are not constants but each instantaneous value is input as a variable. Tracking filter.