JP3243666B2 - Resonant DC-DC converter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type D of a phase difference control system which supplies an AC voltage from a DC power supply via an inverter to a transformer, rectifies the output and supplies the DC voltage to a load.
More particularly, the present invention relates to a C-DC converter, in which a loss in the switching element is reduced to increase the efficiency, and a rate of change of a voltage applied to each switching element of the inverter is reduced to reduce noise, The present invention relates to a resonance type DC-DC converter for preventing a switching element from being broken due to a short circuit of a lossless snubber circuit at the start of an operation.

[0002]

2. Description of the Related Art As one of DC power supply devices used in X-ray devices and the like, DC-DC power supplies have been developed for miniaturization and weight reduction and high efficiency.
A switching type power supply device called a DC converter is widely used.

[0003] However, switching-type power supplies that are widely used at present are basically controlled by PWM, so that the voltage and current waveforms to be handled are square waves, and have the following problems. .

(1) Switching noise is likely to occur.

(2) The loss at the switching element during switching is large.

(3) The upper limit of the operating frequency of the converter greatly depends on the switching time of the switching element.

As a solution to these problems, in recent years, a DC-DC converter has been proposed in which a resonant element is inserted into a part of a converter to make a voltage waveform or a current waveform sinusoidal, thereby reducing the load on the switching element during switching. Development is in progress.

As a method of controlling the output voltage of such a resonance type DC-DC converter, there is a phase difference control method. A conventional resonance type DC-DC converter using this type of control method is disclosed in There is one described in -190556.

This resonance type DC-DC converter is shown in FIG.
As shown in the figure, a DC power supply 1 and a positive electrode of the DC power supply 1
And a first series connected body composed of a first switching element 2a connected to the negative electrode and a second switching element 2b connected to the negative electrode, and a third connected to the positive electrode +.
, And a second series connected body connected in parallel to the first series connected body, and the first to the first series connected elements.
Inverter 4 having first to fourth diodes 3a to 3d connected in anti-parallel to fourth switching elements 2a to 2d, respectively, for converting DC from DC power supply 1 to AC.
And an inductance 5 and a capacitor 6 connected in series on the output side of the inverter 4, a transformer 7 connected in series with the inductance 5 and the capacitor 6 to insulate the output, and convert the output of the transformer 7 to DC. A rectifier 8 for supplying a DC output to the load, and the first to fourth switching elements 2 according to a setting signal of a voltage applied to a load 9 connected to an output side of the rectifier 8 and a current flowing through the load 9.
means (not shown) for controlling the on / off timings of a to 2d to supply a DC output to the load at a desired voltage and current.

In FIG. 6, the first to fourth switching elements 2a to 2d and the diodes 3a to 3d respectively include a first arm 10a and a second arm 10a.
b, a third arm 10c, and a fourth arm 10d. The rectifier 8 is configured to perform full-wave rectification of the input voltage by the four diodes 11a, 11b, 11c, and 11d. Further, reference numeral 12 denotes the capacitance of a high-voltage cable for applying the output voltage of the rectifier 8 to the load 9, and functions as a capacitor for smoothing the output voltage from the rectifier 8.

Next, the operation of the conventional resonant DC-DC converter configured as described above will be briefly described with reference to FIG. In FIG. 7, (a), (b),
(C) and (d) respectively show the on and off periods of the first switching element 2a, the fourth switching element 2d, the second switching element 2b, and the third switching element 2c of the inverter 4 shown in FIG. Is shown. And
As is clear from FIG. 7, the first switching element 2a and the fourth switching element 2d are turned on with a phase difference α, and the second switching element 2b and the third switching element 2b are turned on.
The switching element 2c is turned on with a shift of the phase difference α. Further, the first switching element 2a
And the second switching element 2b, and the third and fourth switching elements 2c and 2d are alternately turned on with a phase difference of 180 °.

In the above operation, the periods (Tb3 to Tb4) in which the first switching element 2a and the fourth switching element 2d shown in (a) and (b) are simultaneously turned on, and (c) and (d) The second switching element 2b shown in FIG.
Since power is supplied from the DC power supply 1 shown in FIG. 6 only during the period (Tb6 to Tb7) during which the third switching element 2c is simultaneously turned on, the output power waveform Vt of the inverter 4 is shown in FIG. As described above, a square wave having positive and negative peak values of the voltage only during the above period is obtained.

Therefore, the first switching element 2a
Phase difference α between the second switching element 2b and the third switching element 2d.
By changing the phase difference α with respect to c, it is possible to change the period during which the respective switching elements 2a to 2d are simultaneously turned on, and to control the power supplied to the load 9 shown in FIG.

FIG. 8 shows the relationship between the phase difference α between the corresponding switching elements and the output voltage Vt in this case. In FIG. 8, the horizontal axis represents the phase difference α, and the vertical axis represents the output voltage Vt to the load 9.
The relationship with t is determined by the resistance values R1, R2, R3 (R1>
R2> R3) is a graph represented by a predetermined curve using a parameter as a parameter.

Incidentally, in the X-ray high-voltage device using the X-ray tube as the load 9, the rise of the voltage (tube voltage) applied to the X-ray tube requires high speed. For this reason, a method is employed in which X-ray irradiation is started with the voltage of the DC power supply 1 (normally obtained by rectifying a commercial power supply) shown in FIG. 3 set to a certain value.

[0016]

Problems to be solved by the conventional resonance type DC-DC converter configured as described above will be described below. First, a first arm 10a composed of a first switching element 2a whose turn-on is not delayed and a first diode 3a connected in anti-parallel to the first switching element 2a, and a second switching element 2b connected in anti-parallel to the first switching element 2b The operation of the second arm 10b including two diodes 3b will be considered.

As shown in FIG. 7E, the first arm 1
The current I1 flowing through the first switching element 2a
Is negative at time Tb1 when the ON signal is input. Therefore, at this time, the first switching element 2a
Is only the ON voltage of the first diode 3a and is almost zero. When the current changes from negative to positive and the current starts to flow through the first switching element 2a, the loss of the switching element 2a is zero because it is the product of the voltage and the current at that time. However, the first
The current I1 flowing through the first arm 10a at the time Tb4 when the switching element 2a of FIG.
It is positive as shown in (e).

FIG. 9 shows the operation until the first switching element 2a starts to turn off and the current becomes zero. As shown in FIG. 9, before the current becomes zero, the voltage of the switching element 2a becomes zero. Begin to increase, the first switching element 2a generates a loss corresponding to the shaded region by the current and the voltage. Such an operation is the same for the second arm 10b.

[0019] In order to reduce the loss as described above, for example, 10 resistors and capacitors 14 as shown in (a) 1
5 are connected in series configuration, or con as shown in FIG. (B)
A circuit called a lossless snubber circuit having a configuration in which a capacitor 14, a resistor 15, and a diode 16 are combined is used in parallel with a switching element 13 such as a transistor.

When such a lossless snubber circuit is provided in parallel with the first and second switching elements 2a and 2b, the rise of the voltage when each of the switching elements 2a and 2b is turned off is suppressed, and the turn-off time is reduced. Was able to reduce the switching loss.

However, in the above-described lossless snubber circuit, when the switching element 13 shown in FIG. 10 is turned off, the electric charge accumulated in the capacitor 14 turns on when the switching element 13 is turned on. Is discharged via the resistor 15 and the resistor 15 causes a loss. Since the resistance 15 controls the maximum value of the current at this time, the resistance 1
If there is no 5, an excessive current flows and the switching element 13 is destroyed.

Since the loss caused by the resistor 15 occurs every time the switching element 13 repeats turn-on and turn-off, the loss of each of the switching elements 2a and 2b in the inverter 4 shown in FIG. Increase in proportion. In particular, in such a resonance type DC-DC converter, it is common to increase the operating frequency in order to reduce the size and weight of the device, and the switching loss becomes extremely large.

Next, in the configuration and operation shown in FIGS. 6 and 7, the third switching element 2 whose turn-on is delayed
c and a third arm 10c composed of a third diode 3c connected thereto in anti-parallel, and a fourth switching element 2d and a fourth diode 3d connected thereto in anti-parallel.
Consider the operation of the fourth arm 10d composed of the following.

In the example shown in FIG. 7, the fourth type shown in FIG.
The current I4 of the fourth arm 10d becomes zero at time Tb5 in the period Tb3 to Tb6 during which the ON signal of the switching element 2d is output (see FIG. 7F), and after that time Tb5, a negative current is generated. Flows. That is, the fourth arm 10d
, A current flows through the fourth diode 3d connected in anti-parallel. Thereafter, at time Tb6, there is no ON signal to the fourth switching element 2d as shown in FIG. 7B, and the third switching element 2c starts to turn on as shown in FIG. 7D. As a result, the current that has been flowing through the fourth diode 3d is commutated to the third switching element 2c, and the fourth diode 3d
Is reverse biased and turned off.

However, at this time, the fourth diode 3
d cannot be turned off instantaneously, and a current called a recovery current flows through the diode 3d until the junction capacitance of the PN junction is charged. Therefore, while the recovery current is flowing, the third arm 10 shown in FIG.
c and the fourth arm 10d are equivalent to a short-circuited state, so that not only an excessive current flows to increase the switching loss, but also the third and fourth switching elements 2c and 2d.
d was sometimes destroyed. Such an operation is the same when the third switching element 2c is turned off.

As described above, in the phase difference control in the conventional resonance type DC-DC converter, the operation of the first and second arms 10a and 10b of the inverter 4 shown in FIG. The operation is different from that of the arms 10c and 10d. In the first and second arms 10a and 10b, the switching loss due to the lossless snubber circuit (see FIG. 10) increases, and the third and fourth arms 10c and 1d increase.
In the case of 0d, there is a problem that the switching elements 2c and 2d are destroyed by arm short-circuit due to the recovery current of the diodes 3c and 3d connected in anti-parallel to the switching elements 2c and 2d.

In addition, each of the switching elements 2a to 2
d, the switching elements 2a to 2
Since the time change rate of the current flowing through d is large, the generated electromagnetic interference noise increases, which may adversely affect the control system. In particular, such a resonance type DC-DC
When the converter is used in a power supply device such as an X-ray CT device (X-ray computed tomography device) as an X-ray device, a diagnostic image is formed immediately near the inverter 4 that switches several hundred volts and several hundred amps. Therefore, a small signal (several microvolts) must be detected, and the influence of the electromagnetic interference noise becomes very large, and a strong improvement has been demanded.

The present invention addresses such a problem and has a small change rate of the voltage applied to each switching element of the inverter, which can reduce the electromagnetic interference noise, and can reduce the loss in the switching element to increase the efficiency. And a resonance type DC-D capable of preventing a switching element from being destroyed due to a recovery current of a diode or a short circuit of a lossless snubber circuit at the start of an inverter operation.
It is intended to provide a C converter.

[0029]

Means for solving the problems are as follows. (1) The current flowing in each switching element of the inverter becomes negative (the state in which current flows in a diode connected in anti-parallel to the switching element) at the time of turn-on. Maintaining an operation mode of a phase difference and a frequency that flows in a positive direction at the time of turn-on; (2) in parallel with each switching element in order to suppress a sharp rise in voltage applied to the switching element when the switching element is off; Connect a lossless (lossless) snubber circuit using only a capacitor . (3) Reduce the phase difference by 1 from the point when the inverter input voltage is zero.
80 °, the first half of the inverter cycle
And the third switching element are turned on at the same time, the second switching element and the fourth switching element are turned on at the same time during the latter half period, and this operation is repeated until the DC output supply to the load is started. Thus, the voltage charged in the capacitor at the start of the inverter operation is made substantially zero to suppress the overcurrent flowing through the switching element.

The object of the present invention is to provide the above (1), (2),
This is achieved by performing switching (soft switching) control that satisfies the condition (3). That is,
It has a DC power supply having a neutral point, a first series connection composed of a first switching element connected to the positive electrode of the DC power supply and a second switching element connected to the negative electrode, and is connected to the positive electrode. A third switching element, a fourth switching element connected to the negative electrode, and a second series connected body connected in parallel to the first series connected body; and the first to fourth switching elements. An inverter having first to fourth diodes connected in anti-parallel to each other to convert DC from the DC power supply to AC;
An inverter output circuit including at least a transformer connected to the output side of the inverter, and a rectifier connected to the transformer and converting the output to DC, and connected to an output side of the rectifier. In a resonance type DC-DC converter for supplying a DC output with a desired voltage and current to a load, first to fourth capacitors respectively connected in parallel to the first to fourth switching elements; between the connection point and the neutral point of the DC power source of the second switching element, and is connected to both between the connection point and the neutral point of the DC power source of said third and fourth switching elements And a phase difference at which the fourth switching element is turned on after the first switching element is turned on and a phase difference at which the second switching element is turned on. The switching element controls the power supplied to the load by varying each phase difference to turn on, and in time the ON signal to the first to fourth switching elements is input, the first to
The first to fourth switching elements are driven at a frequency and a phase difference at which the fourth diode is energized, and from when the voltage of the DC power supply rises until the voltage of the DC power supply reaches a predetermined voltage. In the operation start period, the above-mentioned phase difference is set to 1 / the operation cycle of the first and second switching elements.
In a state where 2 was achieved by providing a control hand stage for driving the respective switching elements.

Further, during the operation start period, this is achieved by providing control means for short-circuiting the primary side of the transformer or the inverter output terminals.

The value of the capacitor used for the lossless snubber must be set within a range where charging and discharging can be completed during the dead time.

[0033]

According to the above construction, the first and second inverters are provided.
Between the neutral point of the DC power source and the connection point of the switching elements, as well as connecting the inductance as an auxiliary circuit to both between the neutral point of the DC power source and the connection point of the third and fourth switching elements Thereby, the current flowing through each arm of the inverter becomes negative at the time of turn-on, and can always maintain the operation mode of the phase difference that becomes positive at the time of turn-off,
The destruction of the switching element due to the recovery current of the diode is prevented.

Also, capacitors are connected in parallel as lossless snubber circuits to the first to fourth switching elements of the inverter, respectively, so that the voltage applied to each switching element by charging and discharging of the capacitors during the dead time period is reduced. A soft switching element with a small rate of change in time can be realized. Therefore, the rate of change of the voltage applied to the switching element is small, the noise is reduced, and the loss in the switching element is reduced to increase the efficiency.

Further, from the time when the voltage of the DC power supply rises,
Until the voltage of the DC power supply reaches a predetermined voltage and during the subsequent period, when the switching element is turned on, current always flows through the diode connected in anti-parallel to the switching element, and unnecessary charge is not performed on the lossless snubber capacitor. Therefore, short-circuiting of the lossless snubber capacitor is prevented, overcurrent of each switching element of the inverter is eliminated, and destruction of the switching elements is prevented.

[0036]

Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 is a circuit diagram showing one embodiment of a resonance type DC-DC converter according to the present invention.

This resonance type DC-DC converter sends an AC voltage from a DC power supply via an inverter to a transformer,
The power converter rectifies the output and supplies a DC voltage to a load. As shown in FIG. 1, a DC power supply 1, an inverter 4, an inductance 5 and a capacitor 6,
A high voltage transformer 7, a high voltage rectifier 8, and X as a load
From the wire tube 17, a frequency phase control circuit 71 as an inverter control means, a frequency determination circuit 76, a phase determination circuit 70 for giving setting values such as a frequency and a phase to the frequency phase control circuit 71, and a frequency phase control circuit 71 And the drive circuits 72 to 74 for driving the inverter 4 in response to the above signal, and is called an inverter type X-ray high voltage device.

The DC power supply 1 is, for example, a secondary battery. In FIG. 1, two power supplies E / 2 are shown symmetrically for the sake of convenience. The inverter 4 receives direct current from the direct current power source 1 and converts it into alternating current. The inverter 4 is connected to a transistor 20 a as a first switching element connected to the positive electrode + of the direct current power source 1 and to the negative electrode − of the direct current power source 1. Transistor 20 as a second switching element
b, and a transistor 20c as a third switching element connected to the positive electrode +
And a second series-connected body composed of a transistor 20d as a fourth switching element connected to the negative electrode and connected in parallel to the first series-connected body, and anti-parallel connected to the transistors 20a to 20d, respectively. And first to fourth diodes 3a to 3d.

The above transistors 20a to 20d
Are turned on by flowing a base current. And the first transistor 2
0a and the first diode 3a, the first arm 10a
With the second transistor 20b and the second diode 3b
And the second arm 10b is connected to the third transistor 20c.
And the third diode 3c constitute a third arm 10c, and the fourth transistor 20d and the fourth diode 3d constitute a fourth arm 10d.

An inductor 5 is connected to the output side of the inverter 4 and a capacitor 6 is connected to the inductor 5 in series. The inductance 5 and the capacitor 6 constitute a resonance circuit. A primary winding of a high-voltage transformer 7 is connected in series to the inductance 5 and the capacitor 6, and the high-voltage transformer 7 boosts the output voltage from the inverter 4 and insulates the output from the output.

The high voltage rectifier 8 converts the output voltage from the high voltage transformer 7 into a direct current by full-wave rectification, and includes four diodes 11a to 11d as shown in FIG. Further, an X-ray tube 17 is connected as a load to the output side of the high voltage rectifier 8. Reference numeral 12 denotes a capacitor for smoothing the output voltage of the high-voltage rectifier 8.

Here, in the present invention, the first to fourth transistors 20a to 20d of the inverter 4 include:
1st to 4th used as lossless (lossless) snubber circuit
With the capacitor 22a~22d are connected in parallel, first and second transistors 20a, 20b
, And the neutral point (potential E / 2) of the DC power supply 1
Further, between the connection point of the third and fourth transistors 20c and 20d and the neutral point of the DC power supply 1, inductances 23a and 23b are connected as auxiliary circuits, respectively.

The phase determining circuit 70 determines the first to the first signals based on a voltage applied to the X-ray tube 17 (hereinafter referred to as a tube voltage) and a signal corresponding to a current flowing through the X-ray tube 17 (hereinafter referred to as a tube current).
This determines the operation phase of the fourth transistors 20a to 20d. The frequency determining circuit 76 determines the operating frequency of the first to fourth transistors 20a to 20d based on a signal corresponding to the tube voltage. Frequency phase control circuit 7
Reference numeral 1 denotes an X-ray emission signal that outputs a signal corresponding to the phase and frequency at which the first to fourth transistors 20a to 20d operate according to the signals of the phase determination circuit 70 and the frequency determination circuit 76. The drive circuits 72 to 75 respectively perform first to fourth signals in accordance with a signal from the frequency phase control circuit 71.
Drive the transistors 20a to 20d.

Next, the resonance type DC-
The operation of the DC converter will be described.

First, the main circuit components of the resonance type DC-DC converter shown in FIG. 1 (DC power supply 1, inverter 4, inductance 5, capacitor 6, high voltage transformer 7, high voltage rectifier 8, X-ray tube 17) Is an equivalent circuit as shown in FIG. That is, the transistors 20a to 20d of the inverter 4 are represented as a first switching element 2a, a second switching element 2b, a third switching element 2c, and a fourth switching element 2d, respectively, as shown in FIG. , X-ray tube 17 is designated as load 9.
Therefore, the operation principle of the main circuit component will be described with reference to FIGS. 3 and 4 using the equivalent circuit shown in FIG.

In the equivalent circuit of FIG. 2, attention is paid to the first arm 10a and the second arm 10b of the inverter 4. The current flowing through the inductance 23a to the DC power supply 1 side is defined as Ia, and the first and second arms 10a and 10b are connected to each other.
a, 10b are output to the high-voltage transformer 7 side by I
r. In this state, when the first switching element 2a is turned on, the current I1 flowing through the first switching element 2a is represented by I1 = Ia + Ir (1).

Since the first switching element 2a and the second switching element 2b are turned on and off with a duty cycle of about 50%, the current I in the steady state is
The waveform of “a” becomes a triangular wave as shown in FIG.
When the switching element 2a is turned off (FIG. 3 (a)
Reference) The current Ia becomes the maximum value Ia (max). That is,
From the above equation (1), the current I1 (0) at the time of turn-off is as follows: I1 (0) = Ia (max) + Ir (0) (2) Here, Ir (0) means the current Ir at the time of turn-off.

At this time, the slope of the current Ia depends on the value La of the inductance 23a and the potential E / 2 at the neutral point of the DC power supply 1.
The maximum value Ia (max) is also determined by L
a and E / 2. Therefore, it is possible to always set the magnitude of the current I1 (0) at the time of turn-off to a certain value or more. That is, Ia (max) can be set so that I1 (0) = Ia (max) + Ir (0)> (constant value) (3).

With this setting, the current flowing through each arm 10a, 10b at the time of turn-on in each switching element 2a, 2b is negative (as anti-parallel connected to each switching element 2a, 2b) as described below. The current flows through the respective diodes 3a and 3b). At this time, as shown in FIGS. 3A and 3B, between the time when the first switching element 2a is turned off and the time when the second switching element 2b is turned on,
A dead time period Td in which both the switching elements 2a and 2b are off is set.

As described above, the dead time period T
In d, the capacitor as the lossless snubber circuit shown in Fig. 2
Sa 22a, 22b effectively soft switching utilizing becomes feasible. In this regard, FIGS.
(D) turns off the first switching element 2a from the state (a) of the first switching element 2a (b),
The mode until the second switching element 2b is turned on (d) after a predetermined dead time (b) to (c) is shown.

First, in FIG. 4A, only the first switching element 2a is on, and as shown in FIG. 3E, the current Ia increases almost linearly regardless of the polarity of the current Ir. I do. Further, the slope depends on the value La of the inductance 23a and the power supply potential E / 2. In this case, the voltage Vc1 = 0 volt of the first capacitor 22a, the second co
The voltage Vc2 of the capacitor 22b is E volt.

Next, in FIG. 4B, both the switching elements 2a and 2b are turned off. In this case, FIG.
As shown in (e), the current Ia = Ia (max). According to the setting of the maximum value Ia (max) and the above equation (3), the current I1 (0) of the switching element 2a at the time of turn-off is sufficiently large. It can be a positive current. Therefore, Con as an inductance 23a and other inductances 5 and lossless snubber circuit as an auxiliary circuit shown in FIG. 2
Due to the resonance phenomenon of the capacitors 22a and 22b, the capacitor
The battery 22a performs charging, and the battery 22b performs discharging.

Next, in FIG. 4C, the capacitor 2
The charging and discharging of the first capacitor 22a and 22b are completed.
The voltage Vc1 of the first capacitor 22 is changed from 0 to E volt.
The voltage Vc2 at b changes from E to 0 volts, and the second diode 3b conducts. At this time, the inductance 23a
The current Ia flowing through starts to decrease by the voltage of -E / 2.

Thereafter, in FIG. 4D, an ON signal is supplied to the second switching element 2b, and the current (Ia + I
When the polarity of r) reverses from positive to negative, the second switching element 2b is replaced with the second switching element 2b as shown in FIGS.
Proceed in the same way as. The changes in the voltages Vc1 and Vc2 of the first and second capacitors 22a and 22b during the operations of FIGS. 4A to 4D are as shown in the graph of FIG.

The above operation is the same for the third arm 10c and the fourth arm 10d of the inverter 4 shown in FIG.

From the above, the turn-off current I1
The value La of the inductance 23a as an auxiliary circuit is determined so that (0) is equal to or greater than the value required for charging and discharging the first and second capacitors 22a and 22b, and charging during the dead time shown in FIG. Capacitor 22 that completes discharge
By appropriately selecting the values of a and 22b, soft switching can be realized for all the switching elements 2a to 2d shown in FIG. 2 by effectively using the capacitors 22a to 22d as lossless snubber circuits.

Here, in the X-ray high-voltage device, since a high-speed rise of the tube voltage is required, usually, the output voltage of the rectifier circuit (not shown; corresponding to the DC power supply 1), that is, the inverter 4 is used. The irradiation is started by operating the switching elements 2a to 2d of the inverter 4 in a state where the input voltage is set to a certain value.

However, in such an operation method, when the lossless snubber circuit is used, the capacitors 22a to 22d are always short-circuited at the time of start-up (at the start of operation of the inverter 4), and the switching elements 2a to 2d are destroyed. Can happen.

Therefore, in the present invention, the operation of the inverter 4 is started in a state where the output voltage of the rectifier circuit (DC power supply 1) is from zero to the maximum phase difference, and the output voltage of the rectifier circuit thereafter becomes the set voltage. Continue the operation until it reaches.
Thereafter, in synchronization with the irradiation start signal, the phase difference of the inverter 4 is reduced to supply power to the load 9 and output X-rays.

According to the above-described operation principle, each of the switching elements 2a to 2d is connected to the switching element 2a through the inductances 23a and 23b as auxiliary circuits.
A current flows through the diodes 3a to 3d connected in anti-parallel to the switching elements 2a to 2d, and a zero volt turn-on is realized in each of the switching elements 2a to 2d, so that short circuits of the capacitors 22a to 22d can be prevented.

According to this method, since the phase difference is maximum before the start of exposure, no current flows through the load 9 (no power is supplied), and the current flows only through the inductances 23a and 23b as auxiliary circuits. Flows and the switching element 2
It can be operated without risk of destroying a to 2d.

Next, returning to the embodiment shown in FIG. 1, the operation of this embodiment will be described with reference to FIGS. 5, 8, 11 and 12. The switching element 2a shown in FIG.
1 to 2d correspond to the transistors 20a to 20d in FIG.

First, as can be seen from the time chart of FIG. 11, when the voltage of the DC power supply (inverter input voltage) rises in response to the X-ray irradiation preparation signal, the first half of the inverter cycle is connected to the first transistor 20a and the first transistor 20a. The third transistor 20c is turned on at the same time, and in the latter half cycle, the second transistor 20b and the fourth transistor 20 are turned on at the same time, and even if the inverter input voltage reaches a steady state, the above operation is repeated until X-ray irradiation starts. First to fourth capacitors 2
The voltage is not charged to 0a to 20d.

When the tube voltage and tube current applied to the X-ray tube 17 as a load are determined, a tube voltage setting signal G1 corresponding to the tube voltage and a tube current setting signal G2 corresponding to the tube current.
Is input to the phase determination circuit 70 and the frequency determination circuit 76.

In the phase determination circuit 70, a load resistance value is obtained from the tube voltage setting signal G1 and the tube current setting signal G2.
Each of the transistors 20a of the inverter 4 is determined from the load resistance value and the tube voltage by using the relationship of the graph shown in FIG.
-20d is determined. The frequency determining circuit 76 determines the operating frequency of the inverter 4 from the tube voltage setting signal G1 using the relationship shown in the graph of FIG.

A signal corresponding to the above-mentioned operation phase difference and operation frequency is input to the frequency / phase control circuit 71.
20d creates a drive signal that turns on and off.

Next, an X-ray emission signal G3 is supplied from a controller (not shown) in FIG.
11, a signal for turning on the transistor 20 a of the inverter 4 is output to the drive circuit 72, and FIG.
The transistor 20a is turned on as shown in FIG.

Thus, an auxiliary current flows through the circuit of the DC power supply 1 on the + side (upper side in the figure) and the inductance 23a. After the time when the phase difference becomes α, the transistor 20d
1 is output to the drive circuit 75,
As shown in FIG. 1, the transistor 20d is turned on, an auxiliary current flows through the circuit of the DC power supply 1 on the negative side (lower side in the figure) and the inductance 23b, and at the same time, the DC power supply 1 on the positive side (upper side in the figure) flows through the resonance circuit. And the negative voltage (lower side in the figure) of the DC power supply 1 is applied, and the resonance current starts to flow.

When a signal that turns off the transistor 20a and turns on the transistor 20b is output from the frequency phase control circuit 71 after a half cycle has elapsed, the transistor 20a is turned off by the drive circuit 72, and the voltage across the capacitor 2
2a, the current of the inductance 23a is commutated to the diode 3b connected in anti-parallel with the transistor 20b, and the voltage across the transistor 20b is clamped to almost zero. The inversion causes the transistor 20b to be turned on with zero voltage.

When the transistor 20d is turned on by 180 °,
When a signal for turning this off is output from the frequency phase control circuit 71, the transistor 20d is turned off by the drive circuit 75, the voltage across the transistor 20d is suppressed by the capacitor 22d and rises, and the current of the inductance 23b is opposite to that of the transistor 20c. The current is commutated to the diode 3c connected in parallel, the voltage between both ends of the transistor 20c is clamped to almost zero, and the transistor 20c is turned on by zero voltage due to the inversion of the resonance current flowing through the load 17.

As a result, transistors 20c and 20c
b is turned on, a current flows in the resonance circuit in a direction opposite to that in the past, and thereafter, the transistors 20a to 20d are sequentially driven in accordance with a signal from the frequency phase control circuit 71. As a result, a resonance current It as shown in FIG. 5I flows through the high-voltage transformer 7 shown in FIG. 1, and a set tube voltage is applied to the X-ray tube 17 and a tube current flows. At this time, looking at the currents I1 to I4 flowing through the first arm 10a to the fourth arm 10d, as shown in FIGS. 5E to 5H, the operation described with reference to FIGS. According to the principle, all the transistors 20a to 20d take a negative value at the time of turning on and take a positive value at the time of turning off. Further, it can be seen that the time rate of change of the voltage applied to each of the transistors 20a to 20d is small as described with reference to FIG. In the time chart shown in FIG. 5, the dead time between the transistors 20a to 20d is omitted for convenience.

Since the phase determination circuit 70 and the frequency determination circuit 76 can be easily configured using a memory, a function generator, an operational amplifier, or the like in which the relationships shown in FIGS. 8 and 12 are tabulated, detailed description is omitted. I do.

FIG. 13 shows a second embodiment of the resonant DC-DC converter according to the present invention. In this example, a switch (switching element) is provided in the auxiliary circuit, and the configuration other than the auxiliary circuit is the same as that in FIG.

In the first embodiment, during the operation of the inverter 4, the alternating current always flows through the auxiliary circuit. The values of the inductances 23a and 22b of the auxiliary circuit must be designed on the assumption that the phase difference and the load condition are the worst. However, depending on the phase difference and the type of the X-ray tube 17 (load 9),
Unnecessary current flows, and the transistors 20a to 20a
There is a wasteful state in terms of conduction loss of 20d and loss of inductances 23a and 22b. Therefore, in the second embodiment, the switching elements 110 to 113 are added to the auxiliary circuit.
In accordance with the current state of the transistors 20a to 20d, and only during the switching period of the transistors 20a to 20d, a circuit is formed to allow current to flow as much as necessary for realizing soft switching. With this configuration, operation in a more efficient state than in the first embodiment becomes possible at any phase difference and load range. Note that FIG.
, 100 to 103 are diodes.

The effective driving method of this auxiliary circuit is as follows. That is, the current when the transistor 20a is turned off may be set to a value more than necessary for charging and discharging the capacitor 22a.
The switching element 110 when the transistor 20b is turned off, the switching element 113 when the transistor 20c is turned off, and the switching element 112 when the transistor 20d is turned off. What is necessary is just to make it conductive. According to such a method, an efficient operation can always be realized even for an X-ray high-voltage device having a very wide range of X-ray tubes 17 (load range).

FIG. 14 is a circuit diagram showing a third embodiment of the resonant DC-DC converter according to the present invention.
A switch 200 is added to the same configuration as in FIG.
In this case, the switch 200 performs an operation start period from the time when the voltage of the DC power supply 1 rises to the time when the output DC voltage of the rectifier 8 reaches a predetermined voltage (the output DC voltage of the rectifier 8 before the start of X-ray exposure). 1 until the voltage reaches a predetermined voltage from zero volts), the primary side of the transformer 7 is short-circuited (FIG. 1).
5).

According to this, regardless of whether the operation of the frequency phase control circuit 71 is correct or not, there is no room for unnecessary charging of the capacitors during the above-mentioned period, and the capacitors 22a to 22d
Of each switching element 20 of the inverter 4
The overcurrent of the switching elements 20a to 20d is eliminated, and the destruction of the switching elements 20a to 20d is prevented.

FIG. 16 is a circuit diagram showing a specific configuration example of the switch 200. Here, a thyristor 201, 204 connected in anti-parallel is used as a main switch for short-circuiting the primary side of the transformer 7, and before the start of X-ray exposure, the switches 202, 205 are turned on to turn on the thyristor 201. , 204, a current flows through the gates of the thyristors 201, 204, and the thyristors 201, 204 are turned off by turning off the switches 202, 205 with the X-ray irradiation start signal G3. Turn on the X-ray tube 17
Inverter output (output DC voltage of rectifier 8) to (load)
Is to give. Note that, in FIG.
Reference numeral 206 denotes a protection resistor.

FIG. 17 is a circuit diagram showing a fourth embodiment of the resonant DC-DC converter according to the present invention. In this embodiment, the switch 200 is connected between the output terminals of the inverter 4 and has such a structure. However, the same effect as in the third embodiment can be obtained.

In the third and fourth embodiments as well, the auxiliary circuit shown in FIG. 13, that is, the auxiliary circuit in which diodes 100 to 103 and switching elements 110 to 113 are added to the inductances 23a and 23b. Circuits can be applied.

In each of the above embodiments, a transistor is used as a switching element used in the inverter 4 and the auxiliary circuit. However, the present invention is not limited to this, and a GTO may be used. IGBT,
Switching elements such as SI transistors and SI thyristors may be used. The DC power supply 1 of the inverter 4
May be a battery or a rectified commercial power supply. Further, the load is not limited to the X-ray tube 17, but may be another general load.

[0083] Also further, the frequency and phase control circuit 71 is generally proportional - although integral control is generally, it is also possible to apply the software control to convert once the digital value.

[0084]

As described above, according to the present invention, between the connection point of the first and second switching elements of the inverter and the neutral point of the DC power supply, and between the connection point of the third and fourth switching elements. by connecting the inductance as an auxiliary circuit to both between the connection point between the neutral point of the DC power source,
The current flowing through each arm of the inverter becomes negative at the time of turn-on, and can always maintain the operation mode of the phase difference that becomes positive at the time of turn-off, thereby preventing the switching element from being destroyed due to the recovery current of the diode. .

Further, since capacitors are connected in parallel as lossless snubber circuits to the first to fourth switching elements of the inverter, the voltage applied to each switching element by charging and discharging of the capacitors during the dead time period is reduced. A soft switching element with a small rate of change in time can be realized. Therefore, there is an effect that the rate of change of the voltage applied to the switching element is small, the noise can be reduced, and the loss in the switching element can be reduced to increase the efficiency.

[0086] Further, configuration at the start inverter operation
Since the capacitor is not charged unnecessarily, short-circuiting of the capacitor is prevented, overcurrent of each switching element of the inverter is eliminated, and there is also an effect that destruction of the switching elements can be prevented.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing an embodiment of a resonance type DC-DC converter according to the present invention.

FIG. 2 is an equivalent circuit diagram of a main circuit component of the resonance type DC-DC converter.

FIG. 3 is a time chart for explaining the operation principle of the main circuit component.

FIG. 4 is a diagram for explaining operation modes of a first switching element and a second switching element in the main circuit configuration unit.

FIG. 5 is a time chart for explaining the operation of the circuit shown in FIG. 1;

FIG. 6 is a circuit diagram of a conventional resonance type DC-DC converter.

FIG. 7 is a time chart for explaining the operation of a conventional resonance type DC-DC converter.

FIG. 8 is a graph showing a relationship between a phase difference and an output voltage in a conventional phase difference control type resonant DC-DC converter using load resistance as a parameter.

FIG. 9 is a diagram showing a turn-off waveform when a lossless snubber circuit is not used.

FIG. 10 is a diagram illustrating an example of a conventional lossless snubber circuit.

11 is a time chart for explaining drive control of each switching element of the inverter from preparation for X-ray irradiation of the X-ray tube in FIG. 1 to normal operation through X-ray irradiation.

FIG. 12 is a diagram illustrating a relationship between an inverter frequency and an output voltage using a phase difference α as a parameter.

FIG. 13 is a circuit diagram showing a second embodiment of the resonance type DC-DC converter according to the present invention.

FIG. 14 is a circuit diagram showing a third embodiment of the resonance type DC-DC converter according to the present invention.

FIG. 15 is a time chart showing the operation of each unit from preparation for X-ray irradiation of the X-ray tube in FIG. 14 to normal operation through X-ray irradiation.

FIG. 16 is a circuit diagram showing a specific example of a switch in FIG.

FIG. 17 is a circuit diagram showing a fourth embodiment of the resonance type DC-DC converter according to the present invention.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... DC power supply 3a-3d ... 1st-4th diode 4 ... Inverter 5 ... Inductance 6, 12 ... Capacitor 7 ... High voltage transformer 8 ... High voltage rectifier 10a-10d ... First to fourth arms 17 X-ray tubes (loads) 20a to 20d First to fourth transistors (first to fourth transistors)
Fourth switching element 22a to 22d Capacitors 23a and 23b Inductance (auxiliary circuit) 70 Phase determining circuit 71 Frequency phase control circuit 72 to 75 Transistor driving circuit 76 Frequency determining circuit 200 switch

──────────────────────────────────────────────────続 き Continuation of front page (72) Inventor Takanobu Hatakeyama 1-1-1 Uchikanda, Chiyoda-ku, Tokyo Inside Hitachi Medical Corporation (56) References JP-A-2-131364 (JP, A) JP-A Sho 63-190556 (JP, A) JP-A-6-22551 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H02M 3/28 H02M 3/335 H05G 1/20

Claims (3)

(57) [Claims] 1. A DC power supply having a neutral point, a first series connection body including a first switching element connected to a positive electrode of the DC power supply, and a second switching element connected to a negative electrode of the DC power supply. third fourth and a second series connection body connected in parallel with the first series connection formed of switching elements and the first to which is connected to the switching element and a negative electrode connected to the positive electrode It includes an inverter for converting a direct current into an alternating current from the direct current power source having a first through fourth diodes antiparallel connected to the fourth switching element, at least a transformer connected to the output side of the inverter I and the inverter output circuit, the variable <br/> connected to voltage divider made and a rectifier for converting the output into a direct current, a desired voltage to a load connected to the output side of the rectifier, the current It supplies a DC output resonant DC-DC
In the converter, between the neutral point of the first to fourth of the first to the fourth capacitor, the DC power supply and a connection point of said first and second switching elements connected in parallel respectively to the switching elements , and the third and the connection inductance to both provided between the fourth connection point and a neutral point of the DC power supply of the switching device, said from the first switching element is turned < phase difference before the fourth switching element is turned on and before
Serial with the second switching element to control the power supplied to the load by turning on and the after the third switching element is varied each phase difference to turn on, turned to the first to fourth switching elements at the time a signal is input, the first to fourth diodes first the first to the frequency and the phase difference to be in conduction 4
Of drives the switching element, wherein in the operation start period from the rise to the voltage of the DC power source reaches a predetermined voltage of the DC power supply voltage, the phase difference the first
And the second half with the <br/> state resonant DC-DC converter, characterized in that said <br/> having a control hand stage for driving the switching elements of the operation cycle of the switching element. 2. A DC power supply having a neutral point, and a first series connected body including a first switching element connected to a positive electrode of the DC power supply and a second switching element connected to a negative electrode of the DC power supply. third fourth and a second series connection body connected in parallel with the first series connection formed of switching elements and the first to which is connected to the switching element and a negative electrode connected to the positive electrode It includes an inverter for converting a direct current into an alternating current from the direct current power source having a first through fourth diodes antiparallel connected to the fourth switching element, at least a transformer connected to the output side of the inverter I and the inverter output circuit, the variable <br/> connected to voltage divider made and a rectifier for converting the output into a direct current, a desired voltage to a load connected to the output side of the rectifier, the current It supplies a DC output resonant DC-DC
In the converter, between the neutral point of the first to fourth of the first to the fourth capacitor, the DC power supply and a connection point of said first and second switching elements connected in parallel respectively to the switching elements , and the third and the connection inductance to both provided between the fourth connection point and a neutral point of the DC power supply of the switching device, said from the first switching element is turned < phase difference before the fourth switching element is turned on and before
Serial with the second switching element to control the power supplied to the load by turning on and the after the third switching element is varied each phase difference to turn on, turned to the first to fourth switching elements at the time a signal is input, the first to fourth diodes first the first to the frequency and the phase difference to be in conduction 4
Of drives the switching element, in the operation start period from the rising of the voltage of the DC power supply to the DC power supply voltage reaches a predetermined voltage, the transformer primary or
1 duty cycle of the phase difference while short circuit between the inverter output terminals cross the first and second switching elements
/ 2 and in a state characterized by including a control hand stage for driving the respective switching elements resonant DC-DC
converter. 3. The resonant DC-DC converter according to claim 1, wherein said load is an X-ray tube.