JP3017400B2 - Array antenna control method and control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method and an apparatus for controlling an array antenna.

[0002]

2. Description of the Related Art Conventionally, a phased array antenna for satellite communication (hereinafter referred to as a first conventional example) mounted on a vehicle or the like and automatically tracking the direction of a geostationary satellite has been prototyped by the Communications Research Laboratory of the Ministry of Posts and Telecommunications. ing. The antenna of the first conventional example is composed of 19 microstrip antenna elements, each of which has a total of 18 microwave phase shifters, one for each element except one, and is electrically driven without mechanical driving. The direction of the beam is scanned. Here, as a sensor for controlling the directivity of the antenna and tracking the direction of the incoming beam, a magnetic sensor for detecting the direction of the geomagnetism and calculating the direction of the geostationary satellite seen from the vehicle that is known in advance, and An optical fiber gyro is provided for detecting the rotational angular velocity of the vehicle and accurately keeping the beam direction constant. By combining these two sensors, regardless of the presence or absence of an incoming beam, the antenna directivity is directed in a certain direction, and the directivity is always maintained in the same direction even when the vehicle moves. I have.

Further, a phase detection method for capturing and tracking an incoming beam (hereinafter, referred to as a digital beamforming antenna for satellite communication using digital phase modulation).
This is referred to as a second conventional example. ) Has been proposed by the present applicant. The method of the second conventional example includes a carrier recovery circuit using a Costas loop for each antenna element of an array antenna, and controls the phase of a voltage controlled oscillator (VCO) so that all elements have the same phase. This is a method of obtaining an array output by performing in-phase synthesis. This method also causes a phase uncertainty in the carrier recovery circuit for each element, and if combined as it is, causes a large power loss. Therefore, the phase of the pull-in is detected from the baseband output of each element, and the phase is determined based on this. The amount of correction is calculated, and the phase uncertainty is corrected by the phase shifter prior to the in-phase synthesis. In the method of the second conventional example, if the received signal is a phase-modulated wave, the directivity of the antenna automatically points in the direction of the incoming beam, and a special sensor for knowing the direction of the incoming beam is provided. I don't need it.

[0004]

In the case of the first prior art phased array antenna, a magnetic sensor capable of detecting an absolute azimuth is used in order to direct the directivity of the antenna toward a satellite. In many cases, the vehicle body is made of metal and is magnetized in many cases, which causes an error in the directivity of the antenna. In order to avoid this, it is necessary to perform calibration using magnetic data obtained by rotating 360 degrees in a wide place where there is no magnetic structure. Even if the satellite can be calibrated successfully and the satellite direction can be acquired and tracked, geomagnetism is often disturbed by surrounding buildings and other vehicles, and it is difficult to track the incoming beam direction using only a magnetic sensor. is there. For this reason, after tracking in the satellite direction, tracking is mainly performed based on the data of the optical fiber gyro, but the optical fiber gyro does not detect the absolute azimuth like a magnetic sensor but detects only the angular velocity, so the azimuth Angle errors accumulate. To prevent this,
Although a method of calibrating the optical fiber gyro based on the information of the magnetic sensor at a certain period is adopted, the control algorithm is complicated and a precise control algorithm has not been developed yet.

The phased array antenna of the first prior art can direct a beam in the direction of a signal source if the direction of the signal source is known regardless of the presence or absence of an incoming beam, but the direction of the signal source is unknown. In the case or when the signal source itself moves such as a low orbit satellite, tracking is impossible unless all the movements are predictable. As described above, the acquisition and tracking method using the azimuth sensor has problems that the configuration is complicated and the performance is limited.

In the case of the phase detection method of the second conventional example, since the directivity is formed by performing carrier recovery for each antenna element, the azimuth provided in the phased array antenna of the first conventional example is considered. No sensor needed,
It has the feature that complicated control algorithms are not required, but the problem is that the initial capture of the incoming beam takes a certain amount of time to converge because the carrier recovery circuit uses a Costas loop circuit that performs closed-loop phase tracking. There was a point. In particular, when satellite communication is performed by mounting an antenna on a mobile object such as a vehicle, instantaneous signal interruptions due to trees and buildings frequently occur, so it is necessary to perform initial acquisition at high speed with about several symbols of received data. is there.

The phase detection method of the second prior art example also
When the number of antenna elements in the array antenna is large, the received signal-to-noise power ratio per element is low, so that a phase cycle slip occurs for each element, carrier recovery becomes difficult, and the gain of the array antenna cannot be utilized. There was a point.

SUMMARY OF THE INVENTION An object of the present invention is to solve the above-mentioned problems, and to perform high-speed and stable mechanical driving without a mechanical drive even in a state where the received signal-to-noise power ratio of each antenna element is relatively low. An object of the present invention is to provide a control method and a control device of an array antenna which can capture and track an incoming beam without using a sensor such as a sensor.

[0009]

According to a first aspect of the present invention, there is provided an array antenna control method for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape. In the antenna control method, a plurality of reception signals received by each antenna element of the array antenna are respectively used by using a common local oscillation signal, and an in-phase component having the same phase as the local oscillation signal, and the local oscillation signal Converting the quadrature signal into quadrature signals composed of orthogonal quadrature components; and calculating a product of quadrature signals between quadrature signals in two first and second antenna elements of each of the plurality of different antenna elements, and First data calculated by the sum of the products of the components, the in-phase component of the quadrature signal of the first antenna element, and the second antenna element And the quadrature component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the first antenna element.
And the second data calculated by the difference between the products of the in-phase components of the quadrature signals of the antenna elements are passed through a noise suppression filter having a predetermined transfer function to be filtered. And the filtered second data and the filtered second data obtained by multiplying the filtered second data by (−1). 3 is multiplied by the orthogonal signal of the second antenna element by another 2 × 2 transformation matrix having two other elements that are diagonal to each other, thereby obtaining the orthogonal signal of the second antenna element. Outputting a plurality of in-phase quadrature signals by repeating the step of in-phase with the quadrature signal of one antenna element and the processing of the in-phase step for the mutually different pairs of the quadrature signals in the plurality of antenna elements. And steps, characterized in that it comprises a step of outputting one quadrature signals after a plurality of orthogonal signals phase synthesis phase combining a that is in-phase reduction.

According to a second aspect of the present invention, there is provided a method of controlling an array antenna for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape. In the plurality of received signals received by the antenna elements of the array antenna, respectively, using a common local oscillation signal, an in-phase component having the same phase as the local oscillation signal, and a quadrature component orthogonal to the local oscillation signal. Converting the quadrature signal into quadrature signals each consisting of: a product of in-phase components of quadrature signals and a product of quadrature components of each of the two first and second antenna elements which are mutually different pairs of the plurality of antenna elements; The first data calculated by the sum of
And the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. Are passed through a noise suppression filter having a predetermined transfer function and are filtered, and then the filtered second data is calculated.
Is divided by the filtered first data, and by calculating the arc tangent of the result of the division, the orthogonal signal of the first antenna element and the orthogonal signal of the second antenna element are calculated. A quadrature signal of the second antenna element is calculated by calculating a correction phase amount for in-phase and shifting the quadrature signal of the second antenna element by the calculated correction phase amount. In-phase to the quadrature signal of the element, and outputting the plurality of quadrature signals in-phase by repeating the process of the in-phase step for the mutually different pairs of the quadrature signals in the plurality of antenna elements, In-phase synthesizing the plurality of in-phase quadrature signals and outputting one quadrature signal after the in-phase synthesis. further,
The control method of the array antenna according to the third aspect is the second aspect.
In the array antenna control method described above, the in-phase step is performed based on an arrangement shape of the array antenna, wherein the calculated correction phase amount is returned to the plane of the arrangement shape. The quadrature signal of the second antenna element is changed to the quadrature signal of the first antenna element by regressively correcting the quadrature signal of the second antenna element by phase-shifting the quadrature signal of the second antenna element by the regression-corrected phase amount. It is characterized by being in phase.

According to a fourth aspect of the present invention, there is provided a method of controlling an array antenna for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape. In the plurality of received signals received by the antenna elements of the array antenna, respectively, using a common local oscillation signal, an in-phase component having the same phase as the local oscillation signal, and a quadrature component orthogonal to the local oscillation signal. And a sum of the product of the in-phase components of the quadrature signals and the sum of the products of the quadrature components of the pair of first and second antenna elements of the plurality of antenna elements. The first data, the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, and The second data calculated by the difference between the product of the quadrature component of the quadrature signal of the first antenna element and the in-phase component of the quadrature signal of the second antenna element is converted into a noise suppression filter having a predetermined transfer function. After passing through and filtering, the filtered first data has two elements each diagonally to each other, and the filtered second data and the filtered second data (-1)
By multiplying the orthogonal signal of the second antenna element by a 2 × 2 transformation matrix having another two elements diagonal to each other with third data obtained by multiplying the second data by the second antenna element.
In-phase the quadrature signal of the antenna element to the quadrature signal of the first antenna element, and performing the step of in-phase for the mutually different pairs of the quadrature signals in the plurality of antenna elements. The in-phase combining process is performed on each of two different quadrature signals among the plurality of in-phase quadrature signals, and the process is repeated until the number of quadrature signals after the in-phase combination becomes one. And outputting one orthogonal signal.

Further, in the array antenna control method according to a fifth aspect of the present invention, in the array antenna control method according to the first aspect, immediately before executing the in-phase step processing, each of the antenna elements of the array antenna may be used. A plurality of orthogonal signals, a direction of each main beam of a plurality of predetermined beams to be formed, which are predetermined so as to be able to receive a desired wave within a predetermined radiation angle range, and a reception frequency of the reception signal. Calculating the plurality of beam electric field values, outputting a plurality of orthogonal beam signals having the respective beam electric field values, and outputting the orthogonal beam signal having the largest beam electric field value among the plurality of outputted orthogonal beam signals. A predetermined number of orthogonal beam signals having a larger beam electric field value including a signal are selected, and the orthogonal beam signal having the maximum beam electric field value is used as a reference. The process of making the quadrature signal of the first antenna element and the selected quadrature beam signal other than the quadrature beam signal having the maximum beam electric field value as the quadrature signal of the second antenna element in-phase is performed. It is characterized by executing.

[0013]

[0014]

[0015]

[0016]

According to a sixth aspect of the present invention, there is provided an array antenna control device for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape. Using a common local oscillation signal for each of the plurality of reception signals received by each antenna element of the array antenna, an in-phase component that is in phase with the local oscillation signal and a quadrature component that is orthogonal to the local oscillation signal And a noise suppression filter having a predetermined transfer function. The orthogonal signals in each of the two first and second antenna elements, which are different pairs among the plurality of antenna elements, are provided. And the first data calculated by the sum of the product of the in-phase components and the product of the quadrature components, and the quadrature signal of the first antenna element. It is calculated by the difference between the product of the in-phase component and the quadrature component of the quadrature signal of the second antenna element and the product of the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. After passing through the noise suppression filter and filtering the data, the filtered first data is included in two elements that are diagonal to each other, and the filtered first data is The second data obtained by multiplying the filtered second data by (−1) and another third data obtained by multiplying the filtered second data by (−1)
In-phase for making the quadrature signal of the second antenna element in-phase with the quadrature signal of the first antenna element by multiplying the quadrature signal of the second antenna element by a 2 × 2 transformation matrix of the two elements A plurality of quadrature signals that are in-phase by repeating the processing of the in-phase unit for the mutually different pairs of the quadrature signals in the plurality of antenna elements. Synthesizing means for in-phase synthesizing the plurality of quadrature signals and outputting one quadrature signal after the in-phase synthesis.

An array antenna control device for controlling an array antenna comprising a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape according to a seventh aspect of the present invention. In the plurality of received signals received by the antenna elements of the array antenna, respectively, using a common local oscillation signal, an in-phase component having the same phase as the local oscillation signal, and a quadrature component orthogonal to the local oscillation signal. And a noise suppression filter having a predetermined transfer function, wherein each of the first and second antenna elements is a different pair of the plurality of antenna elements. First data calculated by the sum of the product of the in-phase components and the product of the quadrature components of the quadrature signal; The difference between the product of the in-phase component of the signal and the quadrature component of the quadrature signal of the second antenna element and the product of the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. After the calculated second data and the second data are passed through the noise suppression filter and filtered, the filtered second data is divided by the filtered first data, and the division result is obtained. By calculating an arctangent value of the above, a correction phase amount for making the quadrature signal of the first antenna element and the quadrature signal of the second antenna element in-phase is calculated, and the calculated correction phase amount is calculated. Phase shifting means for shifting the quadrature signal of the second antenna element into phase by shifting only the quadrature signal of the second antenna element to the quadrature signal of the first antenna element. The above multiple en Control means for outputting a plurality of quadrature signals in-phase by repeating for different pairs of quadrature signals in the element, and in-phase synthesis of the plurality of quadrature signals in-phase by the control means, after in-phase synthesis Combining means for outputting one orthogonal signal. Further, in the array antenna control device according to the eighth aspect, in the array antenna control device according to the seventh aspect, the in-phase means may calculate the corrected phase amount based on an arrangement shape of the array antenna. The calculated correction phase amount is regressively corrected so as to return to the surface of the arrangement shape, and the second signal is phase-shifted by the regression-corrected correction phase amount. The quadrature signal of the antenna element is in-phase with the quadrature signal of the first antenna element.

An array antenna control apparatus according to a ninth aspect of the present invention is an array antenna control apparatus for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape. In the plurality of received signals received by the antenna elements of the array antenna, respectively, using a common local oscillation signal, an in-phase component having the same phase as the local oscillation signal, and a quadrature component orthogonal to the local oscillation signal. And a noise suppression filter having a predetermined transfer function. The in-phase component of the quadrature signal of the pair of first and second antenna elements of the plurality of antenna elements is provided. First data calculated by the sum of the product of the components and the product of the quadrature components; the in-phase component of the quadrature signal of the first antenna element; Second data calculated by the difference between the product of the quadrature components of the quadrature signals of the second antenna element and the product of the quadrature component of the quadrature signal of the first antenna element and the in-phase component of the quadrature signal of the second antenna element. Are respectively passed through the noise suppression filter to be filtered, and then the filtered first data is contained in two diagonal elements, respectively, and the filtered second data is And a 2 × 2 transformation matrix having, in another two elements that are diagonal to each other, third data obtained by multiplying the filtered second data by (−1), A multiplying operation by the quadrature signal of the second antenna element to make the quadrature signal of the second antenna element in-phase with the quadrature signal of the first antenna element; Of the orthogonal signals in the antenna elements , And the above-described step of performing the above-mentioned in-phase step is performed for each of two mutually different quadrature signals of the plurality of in-phase quadrature signals to reduce the number of quadrature signals after the above-mentioned in-phase synthesis to one. And control means for outputting one quadrature signal after the in-phase synthesis by repeating the above steps.

The control device for an array antenna according to a tenth aspect of the present invention is the control device for an array antenna according to the sixth aspect, provided between the conversion means and the in-phase means, and output from the conversion means. A plurality of orthogonal signals in each antenna element of the array antenna, the direction of each main beam of a predetermined plurality of beams to be formed predetermined to be able to receive a desired wave within a predetermined radiation angle range, Multi-beam forming means for calculating the plurality of beam electric field values based on the reception frequency of the received signal and outputting a plurality of orthogonal beam signals having the respective beam electric field values, the conversion means and the in-phase Beam signal having the largest beam electric field value among a plurality of orthogonal beam signals output from the multi-beam forming means, provided between A predetermined number of orthogonal beam signals having a larger beam electric field value are selected, the orthogonal beam signal having the largest beam electric field value is used as a reference orthogonal signal of the first antenna element, and the maximum beam electric field value is selected. Beam selecting means for outputting a selected quadrature beam signal other than a quadrature beam signal having a value to the in-phase means as a quadrature signal of the second antenna element.

[0021]

[0022]

[0023]

[0024]

[0025]

In the method of controlling an array antenna according to the first aspect of the present invention, a plurality of received signals respectively received by the antenna elements of the array antenna are used in common with the local oscillation signal by using a common local oscillation signal. The signal is converted into a quadrature signal composed of a certain in-phase component and a quadrature component orthogonal to the local oscillation signal. Then, each of the two first and second antenna elements of the plurality of antenna elements,
First data calculated by the sum of the product of the in-phase components and the product of the quadrature components of the quadrature signals in the antenna elements of
The product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, the quadrature component of the quadrature signal of the first antenna element, and the quadrature signal of the second antenna element And the second data calculated by the difference between the products of the in-phase components are filtered through a noise suppression filter having a predetermined transfer function, and then the filtered first data is paired with each other. The filtered second data, which is included in the two contiguous elements, and the third data obtained by multiplying the filtered second data by (−1), The orthogonal signal of the second antenna element is multiplied by the 2 × 2 transformation matrix of the other two diagonal elements to obtain the orthogonal signal of the second antenna element. In-phase with the signal. Then, a plurality of quadrature signals in phase are output by repeating the processing of the in-phase step for the mutually different pairs of the quadrature signals in the plurality of antenna elements. Further, the in-phase plurality of quadrature signals are in-phase synthesized to output one quadrature signal after the in-phase synthesis.

In the method of controlling an array antenna according to a second aspect of the present invention, a plurality of reception signals received by each of the antenna elements of the array antenna are respectively used as a local oscillation signal and a common local oscillation signal. The signal is converted into a quadrature signal composed of an in-phase component having the same phase and a quadrature component orthogonal to the local oscillation signal. Next, the first data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of each of the two first and second antenna elements that are different pairs from each other among the plurality of antenna elements. And the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. After passing the second data calculated by the difference between the products of the in-phase components of the quadrature signal and a noise suppression filter having a predetermined transfer function, and filtering,
Dividing the filtered second data by the filtered first data and calculating the arctangent of the result of the division, the orthogonal signal of the first antenna element and the second signal are calculated.
A correction phase amount for making the quadrature signal of the antenna element of the second phase in-phase is calculated, and the second correction amount is calculated by the calculated correction phase amount.
The quadrature signal of the second antenna element is in-phase with the quadrature signal of the first antenna element by shifting the phase of the quadrature signal of the first antenna element. Then, a plurality of quadrature signals in phase are output by repeating the processing of the in-phase step for the mutually different pairs of the quadrature signals in the plurality of antenna elements. Further, the in-phase plurality of quadrature signals are in-phase synthesized, and the
Output two orthogonal signals. Further, in the array antenna control method according to the third aspect, in the array antenna control method according to the second aspect, in the step of performing the in-phase, the calculated correction phase is calculated based on an arrangement shape of the array antenna. The calculated correction phase amount is regressively corrected to return the amount to the surface of the arrangement shape, and the quadrature signal of the second antenna element is phase-shifted by the regression-corrected correction phase amount. The quadrature signal of the second antenna element is made in phase with the quadrature signal of the first antenna element.

According to a fourth aspect of the present invention, in the method of controlling an array antenna, a plurality of received signals respectively received by the antenna elements of the array antenna are combined with the local oscillation signal by using a common local oscillation signal. The signal is converted into a quadrature signal composed of an in-phase component having the same phase and a quadrature component orthogonal to the local oscillation signal. Next, first data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of the pair of first and second antenna elements of the plurality of antenna elements; And the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. Are passed through a noise suppression filter having a predetermined transfer function to be filtered, and then the filtered first data is diagonally opposite to each other. And the filtered second data and the third data obtained by multiplying the filtered second data by (−1) are diagonal to each other. 2x2 transformation row in another two elements And in phase of the quadrature signal of the second antenna to the first antenna element a quadrature signal of the element by multiplying the quadrature signal of the second antenna element. further,
After the process of the in-phase step is performed for different pairs of the quadrature signals in the plurality of antenna elements, the two in-phase signals of the plurality of in-phase quadrature signals are in-phase with each other. One quadrature signal after the in-phase synthesis is output by repeating the processing of the steps until the number of the quadrature signals after the in-phase synthesis becomes one.

Further, in the array antenna control method according to a fifth aspect of the present invention, in the array antenna control method according to the first aspect, each antenna element of the array antenna is provided immediately before performing the in-phase step. A plurality of orthogonal signals, a direction of each main beam of a plurality of predetermined beams to be formed predetermined to be able to receive a desired wave within a predetermined radiation angle range, and a reception frequency of the reception signal. Calculating a plurality of orthogonal beam signals based on the plurality of beam electric field values, and outputting a plurality of orthogonal beam signals having the respective beam electric field values. A predetermined number of orthogonal beam signals having a larger beam electric field value including the beam signal are selected, and the orthogonal beam signal having the maximum beam electric field value is selected based on the selected orthogonal beam signal. Said the first and the quadrature signal of the antenna element, and the maximum of the beam orthogonal beam signals other than the selected orthogonal beam signal the second having a field value
Is performed as the quadrature signal of the antenna element.

[0029]

[0030]

[0031]

[0032]

According to a sixth aspect of the present invention, in the array antenna control device, the converting means converts the plurality of reception signals received by each antenna element of the array antenna into a local oscillation signal using a common local oscillation signal. The signal is converted into a quadrature signal composed of an in-phase component having the same phase as the oscillation signal and a quadrature component orthogonal to the local oscillation signal. Next, the in-phase means calculates the sum of the product of the in-phase components of the quadrature signals and the sum of the products of the quadrature components in each of the two first and second antenna elements that are mutually different pairs of the plurality of antenna elements. The first data to be obtained, the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, the quadrature component of the quadrature signal of the first antenna element, and After passing through the noise suppression filter and filtering the second data calculated based on the difference between the products of the in-phase components of the quadrature signals of the second antenna element, the filtered first data is Each of the two elements that are diagonal to each other, and the filtered second data and the filtered second data have (-1)
By multiplying the orthogonal signal of the second antenna element by a 2 × 2 transformation matrix having another two elements diagonal to each other with third data obtained by multiplying the second data by the second antenna element.
Are made in phase with the quadrature signal of the first antenna element. Then, the control means outputs a plurality of in-phase quadrature signals by repeating the processing of the in-phase means for the mutually different pairs of the quadrature signals of the plurality of antenna elements. Further, the synthesizing means in-phase synthesizes the plurality of quadrature signals in-phase by the control means, and outputs one quadrature signal after the in-phase synthesis.

Further, in the array antenna control device according to the present invention, the converting means converts a plurality of received signals respectively received by the antenna elements of the array antenna using a common local oscillation signal. The signal is converted into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal. Next, the in-phase means calculates the sum of the product of the in-phase components of the quadrature signals and the sum of the products of the quadrature components of the two first and second antenna elements, which are mutually different pairs among the plurality of antenna elements. The first data to be obtained, the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, the quadrature component of the quadrature signal of the first antenna element, and The second calculated by the difference between the products of the in-phase components of the quadrature signal of the second antenna element
Are passed through the noise suppression filter to be filtered, the filtered second data is divided by the filtered first data, and the inverse tangent value of the division result is calculated as follows. By calculating, a correction phase amount for making the quadrature signal of the first antenna element and the quadrature signal of the second antenna element in-phase is calculated, and the second correction amount is calculated by the calculated correction phase amount. By shifting the phase of the quadrature signal of the antenna element, the quadrature signal of the second antenna element is made in phase with the quadrature signal of the first antenna element. And
The control means outputs a plurality of in-phase quadrature signals by repeating the processing of the in-phase means for the mutually different pairs of the quadrature signals of the plurality of antenna elements. Further, the synthesizing means in-phase synthesizes the plurality of quadrature signals in-phase by the control means, and outputs one quadrature signal after the in-phase synthesis. Further, in the array antenna control device according to the eighth aspect, in the array antenna control device according to the seventh aspect, the in-phase means may include the calculated correction phase amount based on an arrangement shape of the array antenna. Is regressively corrected to return the calculated phase to the plane of the arrangement shape, and the quadrature signal of the second antenna element is phase-shifted by the corrected phase corrected. Are made in phase with the quadrature signal of the first antenna element.

According to a ninth aspect of the present invention, in the array antenna control device, the converting means converts a plurality of received signals received by each antenna element of the array antenna using a common local oscillation signal, respectively. The signal is converted into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal. Next, the in-phase means calculates a first signal calculated by the sum of the products of the in-phase components of the quadrature signals and the products of the quadrature components of the pair of first and second antenna elements of the plurality of antenna elements. Data, the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, the quadrature component of the quadrature signal of the first antenna element, and the second antenna element And the second data calculated by the difference between the products of the in-phase components of the quadrature signals are passed through the noise suppression filter and filtered, and then the filtered first data is diagonal to each other. The second data having two elements and being filtered,
The 2 × 2 transformation matrix having the other data obtained by multiplying the filtered second data by (−1) in another two elements that are diagonal to each other, The quadrature signal of the second antenna element is in-phase with the quadrature signal of the first antenna element by multiplying the quadrature signal of the antenna element. Further, the control unit executes the processing of the in-phase unit for the mutually different pairs of the quadrature signals of the plurality of antenna elements, and then executes two different quadrature signals of the plurality of in-phase quadrature signals. The signal is subjected to the in-phase processing and repeated until the number of quadrature signals after the in-phase synthesis becomes one, thereby outputting one quadrature signal after the in-phase synthesis.

Furthermore, in the array antenna control device according to the tenth aspect, in the array antenna control device according to the sixth aspect, the multi-beam forming means may include a plurality of antennas of the array antenna output from the conversion means. A plurality of orthogonal signals in the element, the direction of each main beam of a plurality of predetermined beams to be formed predetermined to be able to receive a desired wave within a predetermined radiation angle range, and the reception frequency of the reception signal and , And calculates a plurality of beam electric field values to output a plurality of orthogonal beam signals having the respective beam electric field values. Next, the beam selecting means includes a predetermined number of orthogonal beam signals having a larger beam electric field value including the orthogonal beam signal having the largest beam electric field value among the plurality of orthogonal beam signals output from the multi-beam forming means. Is selected, and the first beam reference signal based on the orthogonal beam signal having the maximum beam electric field value is selected.
And outputs a selected orthogonal beam signal other than the orthogonal beam signal having the maximum beam electric field value to the in-phase unit as the orthogonal signal of the second antenna element.

[0037]

[0038]

[0039]

[0040]

[0041]

Embodiments of the present invention will be described below with reference to the drawings. First Embodiment FIG. 1 is a block diagram of a receiving unit of an automatic beam acquisition and tracking device for a communication array antenna according to a first embodiment of the present invention. The automatic beam capture and tracking device of the communication array antenna according to the present embodiment includes a plurality of N antenna elements A1 and A arranged side by side on an arbitrary plane or curved surface at predetermined intervals.
The directivity of the array antenna 1 composed of 2,..., Ai,. Here, in particular, the acquisition and tracking apparatus of the present embodiment is characterized by including quasi-synchronous detection circuits QD-1 to QD-N and amplitude / phase difference correction circuits PC-1 to PC-N.

As shown in FIG. 1, the array antenna 1
It includes N antenna elements A1 to AN and circulators CI-1 to CI-N as transmission / reception separators. The receiving modules RM-1 to RM-N are respectively
Using the low-noise amplifier 2 and the common first local oscillation signal output from the first local oscillator 11, down-converts a received radio signal having a radio frequency into an intermediate frequency signal having a predetermined intermediate frequency. And a converter (D / C) 3.

The receiving unit of the acquisition and tracking device further uses N A / D converters AD-1 to AD-N and a common second local oscillation signal output from the second local oscillator 12, The intermediate frequency signal after the A / D conversion is quasi-synchronously detected and two baseband signals orthogonal to each other (hereinafter, these 2
One baseband signal is called a quadrature baseband signal. N) quasi-synchronous detection circuits QD-1 through QD
The phase difference estimation value between the adjacent antenna elements and the received signal strength at each of the antenna elements A1 to AN are calculated using the DN and the converted orthogonal baseband signal, and the amplitude phase is calculated for each of the antenna elements A1 to AN. Execute the correction process and weight all baseband signals to make them in-phase N
The amplitude / phase difference correction circuits PC-1 to PC-N, the in-phase synthesizer 4 for synthesizing the output signals from the amplitude / phase difference correction circuits PC-1 to PC-N in-phase, and output from the in-phase synthesizer 4 A demodulator 5 that performs synchronous detection or delay detection from a baseband signal by predetermined baseband demodulation processing, extracts desired digital data, and outputs the digital data as received data.

In the receiving section, the antenna elements A1 to AN in the array antenna 1 to the amplitude / phase difference correction circuits PC-1 to PC-N are cascaded for each antenna element system. Since signal processing for each system of each antenna element in the receiving unit is executed in the same manner, processing for a radio signal wave received by the antenna element Ai will be described.

The radio signal wave received by the antenna element Ai is transmitted to the circulator CI-i and the receiving module RM-
The signal is input to the down converter 3 via the low noise amplifier 2 in i. The down converter 3 in the receiving module RM-i is connected to the common first
Using the local oscillation signal, the input radio signal is frequency-converted into an intermediate frequency signal having a predetermined intermediate frequency.
Output to the quasi-synchronous detection circuit QD-i via the D converter AD-i. The quasi-synchronous detection circuit QD-i is connected to the second local oscillator 1
2, using a common second local oscillation signal output from the second, the input intermediate frequency signal after A / D conversion is quasi-synchronously detected and converted into two orthogonal baseband signals I _i , Q _i and amplitude Output to the phase difference correction circuit PC-i and the adjacent amplitude phase difference correction circuit PC- (i + 1). Amplitude / phase difference correction circuit PC
−i is a value between adjacent antenna elements using the input orthogonal baseband signals I _i , Q _i and the orthogonal base band signals I _i−1 , Q _i−1 regarding the adjacent antenna element A- (i−1). Estimated phase difference δc _{i−1, i} and antenna elements A1 to A
N, calculate the received signal strength at N, correct the phase difference (phase shift) so that all baseband signals are in phase based on the calculated phase difference estimated value, and are proportional to the calculated received signal strength. By performing weighting with the amplification gain, the amplitude / phase correction processing for the antenna element Ai is executed. The baseband signal after the processing is applied to the in-phase combiner 4.
Is input to The circuit processing of the amplitude / phase difference correction circuit PC-i will be described later in detail.

The in-phase synthesizer 4 includes an amplitude / phase difference correction circuit PC
After in-phase combining the baseband signals input from −1 to PC-N for each channel, the signals are output to the demodulator 5. The demodulator 5 performs synchronous detection or delay detection by a predetermined baseband demodulation process from the input baseband signal, extracts desired digital data, and outputs it as received data.

FIG. 2 is a block diagram of a transmission section of the acquisition and tracking device. The transmission unit includes N transmission modules TM-1 to TM-N, N quadrature modulation circuits QM-1 to QM-N, and an in-phase distributor 9 as shown in FIG. Here, each of the quadrature modulation circuits QM1 to QM-N includes a quadrature modulator 6 and a transmission local oscillator 10, respectively. Up-converter (U) that converts the frequency of the
/ C) 7 and a transmission power amplifier 8. Here, each of the transmission local oscillators 10 in the quadrature modulation circuits QM-1 to QM-N is, for example, a DDS (Diode) driven by the same clock.
rect Digital Synthesizer)
Phase correction amount Δφ input from least square regression correction unit 42
A transmission local oscillation signal having a phase corresponding to each phase correction amount is generated based on _{c1 to ΔφcN} .

The baseband signal, which is the transmission data, is input to the in-phase distributor 9 and then distributed in-phase to each of the quadrature modulators 6 of the quadrature modulation circuits QM-1 to QM-N. For example, the quadrature modulator 6 in the quadrature modulation circuit QM-1
In accordance with the transmission baseband signal input from the in-phase distributor 9, the transmission local oscillation signal is quadrature-modulated, for example, by QPSK, and then the quadrature-modulated intermediate frequency signal is transmitted to the up-converter 7 in the transmission module TM-1 by the transmission power Amplifier 8
And is input to the circulator CI-1 in the array antenna 1 as a transmission radio signal. Then, the transmission radio signal is transmitted and radiated from the antenna element A1.
Further, similar signal processing is performed in each system of the transmission unit connected to the antenna elements A2 to AN.

FIG. 3 shows the i-th antenna element Ai (i
= 1, 2, 3,..., N). The amplitude phase difference correction circuit PC-i estimates and determines the phase difference δci _{-1, i} between adjacent antenna elements of a received radio signal composed of a digital phase modulation wave or an unmodulated wave, and determines the phase difference. 0, that is, the phase is corrected for each antenna element so that they are in phase.
In order to improve the received signal-to-noise power ratio when the baseband signals are combined in phase, this circuit performs amplification for each system with a gain proportional to the signal strength of the received wireless signal.
As shown in FIG. 3, the amplitude / phase difference correction circuit PC-i includes a phase difference estimator 40, an adder 41, a least squares regression corrector 42, a delay buffer 43, a phase difference corrector 44, And an amplitude correction unit 45. Note that the amplitude / phase difference correction circuit P
In C-1, the phase difference estimating unit 40 and the adder 41 are not provided, and Δφ ₁ = 0.

The quadrature baseband signals I _i , Q _i (hereinafter, I I ₎ , which are reception signals input from the quasi-synchronous detection circuit QD-i,
_i is referred to as a baseband signal of I channel, the Q _i of the baseband signal of the Q channel. ) Are input to the phase difference estimating unit 40 and the delay buffer 43. The phase difference estimating unit 40 outputs the quadrature baseband signals (sample values) I output from the quasi-synchronous detection circuits QD-i and QD- (i-1) for the two adjacent antenna elements Ai and Ai-1.
_{Based on i} , Q _i and I _i−1 , Q _i−1 , the phase difference δc _{i−1, i} between the two adjacent antenna elements Ai, Ai−1 at each sample timing is estimated, and the adder 41 is estimated. Output to
The adder 41 includes an estimated phase difference δc _{i-1, i} input from the phase difference estimating unit 40 and an amplitude / phase difference correction circuit PC
- (i-1) in the by adding the cumulative correction phase amount [Delta] [phi _i-1 output from the adder 41 outputs the accumulated correction phase amount [Delta] [phi _i of the addition result to the least-squares regression correcting unit 42 ,
Adder 4 in next amplitude / phase difference correction circuit PC- (i + 1)
Output to 1.

The least-squares regression correction unit 42 accumulates the estimated phase difference δc _{i-1, i} for each antenna element system by the adder 41 sequentially for each antenna element system. Based on the amounts Δφ _{1 to} Δφ _N , the phase correction amounts Δφ _{c1 to} Δφ _cN of the reception phase differences for each of the antenna elements A 1 to AN for suppressing noise using the spatial property of the array antenna are represented by The signal is output to each of the phase difference correction units 44 in the phase difference correction circuits PC-1 to PC-N, and is also output to the transmission local oscillator 10 in the quadrature modulation circuits QM-1 to QM-N. The least squares regression correction unit 42
Is provided only in the receiving unit, for example, DS
It is composed of P (Digital Signal Processor).

On the other hand, the delay buffer 43 converts the quadrature baseband signals I _i , Q _i into a phase difference estimating delay corresponding to the calculation time in the phase difference estimating unit 40, the adder 41, and the least square regression correcting unit 42. The signal is delayed by the time and output to the phase difference correction unit 44. Next, the phase difference correction unit 44
Correction amount Δ of reception phase difference output from power regression correction unit 42
Based on φ _ci , the quadrature baseband signal output from the delay buffer 43 is phase-corrected by rotating the phase by a phase shift amount corresponding to the correction amount Δφ _ci and output to the amplitude correction unit 45. Further, the amplitude correction unit 45 converts the orthogonal baseband signal output from the phase difference correction unit 44 into
Its quadrature baseband signal of the signal strength proportional to quadrature baseband signals are amplified by a gain Ic _i, outputs as Qc _i in phase combiner 4.

Now, two adjacent antenna elements Ai-
Assuming that sample values at a certain point in time of the orthogonal baseband signal after quasi-synchronous detection of 1 and Ai are I _i−1 , Q _i−1 , I _i and Q _i , respectively, the phase difference estimating unit 40 calculates the values. Instantaneous phase difference δ
_{i-1, i} are two vectors (I _i-1 ,
Q _i-1 ) and (I _i , Q _i ). These I _i−1 , Q _i−1 , I _i , and Q _i are represented by the following equations 1 to 4 in the case of digital phase modulation.

[0054]

## EQU1 ## I _i-1 = a _i-1 cos (θ)

## EQU2 ## Q _i-1 = a _i-1 sin (θ)

## EQU3 ## I _i = a _i cos (θ + δ _{i-1, i} )

## EQU4 ## Q _i = a _i sin (θ + δ _{i-1, i} )

Here, a _i-1 and a _i are the amplitudes of the respective base band signals, and θ means an arbitrary phase angle of each base band signal which fluctuates according to the modulated phase data. Therefore, if the baseband processing shown in the following Expressions 5 and 6 is performed, it is possible to obtain a value proportional to the sine and cosine of the phase difference δ _{i-1, i} and not depending at all on the modulated phase data. .

[0056]

## EQU5 ## I _i-1 · I _i + Q _i-1 · Q _i = a _i-1 a _i cosδ _{i-1, i}

I _i−1 · Q _i −I _i · Q _i−1 = a _i−1 a _i sin δ _{i−1, i}

From this, two adjacent antenna elements A
The instantaneous phase difference δ _{i−1, i} of i−1, Ai is calculated by the following equation (7).

[0058]

(Equation 7)

This does not depend not only on the modulated phase data of each signal but also on the amplitudes a _i−1 and a _i . Therefore, the phase difference δ _{i−1, i} can be calculated regardless of the modulation.
Here, the conversion from Equations 1 to 4 to Equation 7 is a conversion from the mutually orthogonal I axis and Q axis to two mutually orthogonal axes defining the phase difference δ _{i−1, i.} , Means the rotation of the coordinates about the axis center. In Equation 7, the data of the denominator of the fraction on the right side is the left side of Equation 5, and as shown in Equation 5, is proportional to the cosine of the phase difference δi _{-1, i} . on the other hand,
In Expression 7, the data of the numerator of the fraction on the right side is the left side of Expression 6, and as shown in Expression 6, is proportional to the sine of the phase difference δi _{-1, i} .

Then, in order to suppress the noise (mainly the thermal noise of the receiver) included in the received radio signal and obtain a more accurate phase difference, the two data obtained by Expressions 5 and 6 are used. Each is passed through a predetermined digital filter included in the phase difference estimating unit 40 and filtered. Here, the reason why the filtering is performed before the division or the operation of tan ^-1 is to prevent an increase in error due to the operation. Phase difference δc _{i-1, i} after filtering
Is estimated using the following equation (8).

[0061]

(Equation 8)

Here, F (•) represents a transfer function of the digital filter. Various filters can be applied as the digital filter, from a simple cyclic adder to a transversal filter with adaptive tap coefficients. The phase difference estimating unit 40 calculates the phase difference δ after the above-described filtering using Expression 8.
Ci _{-1 and i} are calculated and output to the adder 41.

FIG. 4 shows an example of the configuration of an FIR (Finite Impulse Response) filter with fixed tap coefficients included in the phase difference estimating section 40. In the example of FIG. 4, the buffer size Buff = 7. As shown in FIG. 4, the input signal x is input to an adder 70 via a tap coefficient multiplier 60, and is connected to six cascade-connected delay circuits 51.
To 56 input terminals. Each delay circuit 51 to 5
6 output from each tap coefficient multiplier 6
The signal is input to the adder 70 via 1 to 66. here,
The multipliers 60 to 66 have tap coefficients k0 to k6, which are multiplication coefficients, respectively, multiply the input signal by the tap coefficients, and output the result to the adder 70. The adder 70 adds all the input signals and outputs the result as an output signal F (x). Here, if all the tap coefficients k0 to k6 are set to 1, this filter becomes a simple cyclic adder. Hereinafter, the size of the buffer of this filter is simply referred to as a buffer size Buff.

The estimated phase difference δc calculated by the above equation (8)
_{Based on i-1, i} , the amount of phase to be corrected in the system of each antenna element (hereinafter referred to as corrected phase amount) △ φ _i (i
= 1, 2,..., I,..., N) are expressed by the following equation 9 and calculated by the adder 41.

[0065]

Δφ ₁ = 0 Δφ ₂ = Δφ ₁ + δc _1,2 Δφ ₃ = Δφ ₂ + δc _2,3 ------------------------- Δφ _i = Δφ _i-1 + δc _{i-1, i} − −−−−−−−−− Δφ _N = Δφ _N-1 + δc _{N-1, N}

In equation (9), it is assumed that the antenna element A1 is used as a phase reference (zero phase), and that the phases of all the antenna elements A1 to AN are equal to the antenna element A1. As a method of calculating the correction phase amount,
Several choices can be made as shown below.

When the antenna elements A1 to AN are in a one-dimensional array, for example, as shown in FIG. As shown in FIG. 5B, there are a first method and a second method in which an antenna element Ai satisfying 1 <i <N is used as a phase reference and parallel calculation is performed on both sides thereof.
The latter is faster in calculation speed because it branches into two branches and performs parallel processing, but an element serving as a phase reference requires two outputs.

When the antenna elements A1 to AN are in a two-dimensional square array, assuming that the number of input / output ports of one element is up to three, as shown in FIG.
A method may be considered in which the antenna element A1 at one end of the diagonal line is used as a phase reference and the phase difference is added while branching from the antenna element A1. In this method, the cumulative number of additions for each branch is all three. Furthermore, in the case of another arbitrary antenna arrangement, parallel calculation can be performed according to these examples to speed up the calculation.

The calculated correction phase amount Δφ _i suppresses noise components using a digital filter in the phase estimator 40 for each antenna element system, but the cutoff characteristics of the filter are made too steep. Therefore, the noise suppression by the filter is limited because the response delay increases. For this reason, the least squares regression correction unit 42 performs a linear or flat surface regression correction of the correction phase amount by the array spatial signal processing using the least squares method as described below, so that the noise characteristic on the receiver side is obtained. To improve

For the sake of simplicity, it is now assumed that four antenna elements A1 to A4 are arranged on a straight line at arbitrary intervals, and that one incoming beam of a radio signal wave is received from a certain direction. Each antenna element A1 to A4 in case
FIG. 7 shows the reception phase at. However, it is assumed that there is no noise originally included in the incoming beam. In this case, if there is no receiver noise, each reception phase can be accurately obtained. Therefore, as shown at 71 in FIG. 7, the reception relative phase amount △ φ _i (x) of the i-th antenna element at the position x.
Is a linear function with respect to the antenna position x. However, actually, since there is an independent receiver noise (mainly thermal noise) for each system of the antenna elements A1 to AN, the calculated phase amount (estimated value) Δφ _i (x) is 72 in FIG.
become that way. Here, a regression line △ φ _ci (x) that minimizes the sum of the square errors from the received relative phase amounts (estimated values) Δφ _i (x) as shown at 73 in FIG. 7 is corrected. Then, the receiver noise can be suppressed.

The phase amount regression correction process can be similarly applied to the case where the antenna array is two-dimensional, and is applicable not only when the antenna array surface is a plane but also when the antenna array is an arbitrary curved surface. . In that case, the regression surface is obtained from the shape of the antenna array surface. In the regression correction processing, the least squares method is used. However, the present invention is not limited to this, and a numerical calculation method of regressing on one straight line or a curved surface to obtain an approximate straight line or a curved surface may be used. .

A calculation example in the case where the antenna element arrangement surface is a primary plane will be described below. I-th arbitrary natural number (1 ≦ i ≦
N) is the position of the antenna element on the xy plane (x, y).
The regression plane △ φ _ci (x, y) of the correction phase amount that minimizes the evaluation function J given by the following equation is calculated using the following equation (10).

[0073]

(Equation 10)

Here, △ φ _i (x, y) is the estimated value of the correction phase amount before the least squares regression (corresponding to 72 in FIG. 7). Here, the antenna element array is x _max × y
_max is assumed to be a square array at _regular intervals, and natural numbers N (= x
_max × y _max ) antenna elements are x = 1, 2,..., x
It is assumed that they are arranged at the intersections of the axes of _max and y = 1, 2,..., y _max . Since the antenna surface is a plane, the phase plane that is the least square regression plane of the correction phase amount is also a plane, and the regression plane △ φ _ci (x, y) of the correction phase amount can be expressed by the following equation 11. .

[0075]

Equation 11] _{Δφ ci (x, y) =} ax + by + c, x = 1,2, ..., x max; y = 1,2, ..., y max

Here, a, b and c are parameters for determining the position of the plane. At this time, the normalization equation that satisfies the condition that minimizes the evaluation function J is expressed by the following equation 12.

[0077]

１２J / ∂a = 0 ∂J / ∂b = 0 ∂J / ∂c = 0

By transforming this, the following equation 13 is derived.

[0079]

(Equation 13)

From the above equation (13), the following equation (14) is obtained.

[0081]

[Equation 14]

Here, the matrix A and the matrix Φ are represented by the following equation (15).

[0083]

(Equation 15)

Here, since A is a coefficient matrix determined only from the position coordinates of the antenna elements A1 to AN, its inverse matrix A ^-1 can be calculated in advance and need not be calculated in real time. . For example, when x _max = y _max = 4, the inverse matrix A ^-1 is represented by the following equation 16.

[0085]

(Equation 16)

Accordingly, the parameters a, b, and c for determining the position of the plane are given by the following equation (17).

[0087]

[Equation 17]

Therefore, the estimated value Δφ of the correction phase amount
_The regression plane △ φ _ci (x, y) is determined using _i (x, y), and the corrected phase amount △ φ _c1 (= Δφ _c1 (1, 1)) ~ △
φ _cN (= △ φ _cN (x _max , y _max )) can be calculated by the least squares regression correction unit 42. The above is a calculation example assuming that the antenna surface is a primary plane,
Similarly, the present invention can be applied to a case of a quadratic surface or the like.

In the processing by the above least square method, when there is no margin in the calculation speed, the correction phase amount △ φ _ci (x, y) = △ φ _i
It can be skipped as (x, y). The correction phase amount △ φ _ci (= △ φ _ci (x, y)) thus obtained.
Is used to correct the phase of the orthogonal baseband signal in all the antenna element systems according to the following equation (18). Hereinafter, △ φ _ci = △ φ _ci (x, y).

[0090]

(Equation 18)

Here, the left side of Equation 18 is a matrix representing the vector of the received baseband signal of the i-th antenna element after the phase correction, and the first term on the right side of Equation 18 is to make all the received baseband signals in-phase. The second term on the right side of the matrix is a matrix representing a vector of the received baseband signal before the phase correction.

Next, when the received power at some antenna elements may be reduced due to multipath fading or blocking, etc., equal-gain in-phase combining in which the received signals of all antenna elements are combined with equal weights may result in poor quality. Since the good signal and the bad signal are added with the same weight, the signal-to-noise power ratio after the in-phase synthesis deteriorates. In order to suppress this, the amplitude correction unit 45 amplifies the reception baseband signal in the system of each of the antenna elements A1 to AN with a gain G proportional to the reception intensity as shown in the following Expression 19 to perform amplitude correction. . This means increasing the contribution of good quality signals and reducing the contribution of poor quality signals.

[0093]

[Equation 19]

Here, k is a proportional constant, and Ave (
) Indicates a time average value. If the signals after the amplitude correction are in-phase synthesized by all the antenna elements A1 to AN, the relative in-phase synthesized output of the quadrature baseband signal is expressed by the following equation (20).

[0095]

(Equation 20)

The amplitude correction process in the amplitude correction unit 45 is as follows.
If the difference in the received power among the antenna elements A1 to AN does not matter much, G = 1 can be set and this can be skipped. If this in-phase combined output is input to an arbitrary baseband processing type demodulator 5 that performs synchronous detection or delay detection, desired digital data can be obtained.

On the other hand, since the weight for controlling the directivity of the transmitting array antenna does not include the amplitude component but only the phase component, the phase correction amount △ φ _ci calculated by the least-squares regression correction unit 42 remains unchanged. It can be used as a weight for directivity control of a transmission array antenna, and can automatically direct a transmission beam in the direction of an incoming beam. However, in some cases, it is necessary to perform a simple conversion as necessary, as described below.

For example, when the wavelength of the radio wave is different between transmission and reception and the array antenna 1 is shared for transmission and reception, etc., the phase shift amount Δφ _Ti (x, y ) Is represented by the following equation 21.

[0099]

(Equation 21)

Here, λ _T and λ _R are free space wavelengths for transmission and reception, respectively. This conversion is unnecessary when the transmitting and receiving antenna elements are separate and the element intervals are equal in terms of wavelength, or when the antenna elements are shared for transmission and receiving and the frequencies are the same.

A description will be given of a calculation result of a simulation performed to confirm an effect when an incoming beam is received by using the automatic beam capturing and tracking apparatus of the array antenna according to the present embodiment configured as described above. Table 1 shows conditions for the simulation.

[0102]

[Table 1] ―――――――――――――――――――――――――――――――― Modulation method QPSK ――――――――― ――――――――――――――――――――――――― Bit rate 16kbps ―――――――――――――――――――――― ―――――――――――― Modulation frequency 32kHz ―――――――――――――――――――――――――――――――― Sampling Rate 128kHz ―――――――――――――――――――――――――――――――― Additional noise Gaussian noise ―――――――――― ―――――――――――――――――――――――― Array antenna Point radiation source, 4-element linear array ―――――――――――――――― ―――――――――――――――――― Antenna element spacing 1/2 Long ―――――――――――――――――――――――――――――――― Transmission low-pass filter 10-tap FIR filter, cut-off frequency = 8kHz ―― ―――――――――――――――――――――――――――――― Transmission band filter 51 tap FIR filter, cut-off frequency = 16kHz ―――――― ―――――――――――――――――――――――――――― Reception bandpass filter 51-tap FIR filter, cut-off frequency = 16kHz ―――――――――― ―――――――――――――――――――――――― Receive low-pass filter 10 tap FIR filter, cut-off frequency = 8kHz ――――――――――――― ――――――――――――――――――――― Remarks No interference wave or frequency fluctuation ―――――――――――――― -------------------

The digital filter for estimating the correction phase amount is a simple cyclic adder (FIR filter with all tap coefficients = 1), and the effect is examined by changing the addition buffer size Buff corresponding to the number of filter taps. Was. However, the power received by each antenna element is assumed to be equal, and no amplitude correction is performed. Also, least squares regression was not performed. Further, in the simulation, the phase difference correction is not performed for each sample, and the calculation frequency is reduced to once every nine samples. This not only reduces the computational load on a DSP (Digital Signal Processor), but also reduces the correlation of noise signals between computational samples, thus enabling more effective noise suppression by a digital filter.

FIG. 8 shows a time change of the antenna relative gain in the arrival direction of the signal beam when the phase difference estimation calculation is performed for each sampling (sampling frequency = 128 kHz).
It is shown together with an I-channel demodulated baseband signal (demodulated data). Here, FIG. 8A shows the case where the reception C / N per antenna element is 4 dB, and FIG. 8B shows the case where C / N = -2 dB. Where C /
N represents a ratio between carrier power and noise power (hereinafter, referred to as carrier power to noise power ratio). As shown in FIG. 8, the output generation of the transmission signal starts at the total number of samplings = 0, and then the input and the operation of the reception signal starts at the total number of samplings = 100, and then, at the total number of samplings = 700 to 1000. It is assumed that shadowing (blocking of the received signal) is received and the direction of the incoming signal beam changes at 90 degrees / second. Here, the operation from the start of the calculation until the antenna relative gain exceeds -3 dB is called "coarse capture", and the operation until the antenna relative gain exceeds -0.5 dB is called "fine capture". The total number of samplings required for acquisition is about 80 in the case of FIG. 8A and about 300 in the case of FIG. 8B. Therefore,
The total number of samplings required for fine acquisition depends on the carrier power to noise power ratio C / N. On the other hand, the total number of samplings required for the coarse acquisition is the carrier power-to-noise power ratio C
/ N, and the total number of samplings is about 30 to 50, and the incoming signal beam is captured. After the acquisition, when the carrier power to noise power ratio C / N is low as shown in FIG. 8B, the fluctuation of the antenna relative gain also increases. That is, it can be seen that in both cases (a) and (b) of FIG. 8, the incoming signal beam is stably tracked. Thus, the received carrier power to noise power ratio C /
Even when N is low, it is possible to capture at high speed and to track stably because the phase control of the system of each of the antenna elements A1 to AN is performed by the feedforward method.

FIGS. 9A and 9B show the time change of the antenna pattern at the time of capturing the signal beam under the same conditions as in FIG. In FIG. 9, the dotted line indicates the case where the total number of samplings is 8, the dashed line indicates the case where the total number of samplings is 26, and the solid line indicates that the total number of samplings is 35 ((a) in FIG. 9) or 1
25 (FIG. 9B). As is clear from FIG. 9, a state in which the signal beam of −45 degrees is captured from a state where the antenna pattern is random (when the total number of samplings = 8) (a total number of samplings = 35 ((a) in FIG. 9)) or 125 It can be seen from FIG. 9B that the antenna pattern is quickly converged.

FIGS. 10 (a) and 10 (b) also show FIG.
Under the same conditions as above, 90 ° per second is assumed as the maximum rotation speed assumed for a normal land moving body, and the change in the antenna pattern when the incoming signal beam direction changes at this speed is shown. In FIG. 10, the dashed-dotted antenna pattern is one-third second after the dotted line, and the solid-line antenna pattern is one-third second after the dashed-dotted line. As is clear from FIG. 10, even when the direction of the incoming signal beam changes, the main beam of the array antenna tracks the incoming signal beam almost accurately.

FIG. 11 shows tracking characteristics at the time of coarse capture and fine capture of an incoming signal beam with respect to the carrier power to noise power ratio C / N when the buffer size Buff is used as a parameter. Here, the operation cycle Topr is fixed at 1. As is evident from FIG. 11, the coarse acquisition has a carrier power to noise power ratio C / N and a buffer size B
It can be seen that there is almost no dependence on uff and stable capture characteristics can always be obtained. On the other hand, with respect to the fine acquisition, the total number of samplings before the acquisition increases with the deterioration of the carrier power to noise power ratio C / N, that is, the time until the acquisition becomes longer, and the acquisition becomes dull.
It largely depends on the carrier power to noise power ratio C / N.
In this case, the smaller the buffer size Buff is, the quicker the acquisition is. However, as will be described later in detail, the tracking becomes unstable. (Consideration of some conditions that cannot be satisfied at the same time) is required.

FIG. 12 shows the tracking characteristics with respect to the carrier power-to-noise power ratio C / N when the buffer size Buff is used as a parameter. −
The number of samplings is less than 0.5 dB, and represents the number of times the formed main beam deviates from the target direction. Here, the operation cycle Topr is fixed at 1.
As is apparent from FIG. 12, the stability of tracking at a relatively low carrier-to-noise power ratio C / N is significantly improved by increasing the buffer size Buff.

FIG. 13 shows the tracking characteristics at the time of coarse acquisition and fine acquisition with respect to the carrier power-to-noise power ratio C / N when the operation cycle Topr is used as a parameter.
Here, the buffer size Buff is fixed at 30. As is clear from FIG. 13, the tracking characteristics of the coarse capture hardly depend on the calculation cycle Topr, but on the other hand, regarding the tracking of the fine capture, the smaller the calculation cycle Topr is, the faster the capture is. However, in this case, as described later in detail, tracking becomes unstable,
In selecting the operation cycle Topr, a trade-off between acquisition and tracking in consideration of actual wireless communication line conditions is required.

FIG. 14 shows the tracking characteristics with respect to the carrier power to noise power ratio C / N when the operation period Topr is used as a parameter. The vertical axis indicates the relative gain of the array antenna until the total number of samplings = 8000. Is -0.
The number of samplings is less than 5 dB, and represents the number of times the formed main beam deviates from the target direction. Here, the buffer size Buff is fixed at 30. As is clear from FIG. 14, similarly to the case where the buffer size Buff is increased by increasing the operation cycle Topr (see FIG. 12), the tracking stability at a relatively low carrier-to-noise power ratio C / N is maintained. It can be seen that the properties are significantly improved. However, if the operation cycle Topr is set too long, the response to the change in the direction of the incoming signal beam becomes slow, which causes an increase in tracking error.

From the above simulation results, in the automatic beam acquisition and tracking apparatus of the present embodiment, the buffer size Buff and the operation cycle under the radio communication line conditions where the carrier power to noise power ratio C / N is relatively high. While Topr is set relatively small to speed up the acquisition, the buffer size Buff and the operation cycle Topr are changed under the radio communication line conditions where the carrier power to noise power ratio C / N is relatively low. It can be seen that stable tracking characteristics can be obtained by setting both of them relatively large.

As described above, the automatic beam tracking and capturing apparatus shown in this embodiment has the following specific effects. (1) The incoming beam is captured by correcting the phase difference between the received signals at each of the antenna elements A1 to AN by feedforward, and does not include a feedback loop unlike the second conventional example. Even at a power-to-noise power ratio C / N, an incoming beam of a radio signal composed of a digital phase-modulated wave or an unmodulated wave can be automatically and quickly captured, as shown in the second prior art method. The delay time for convergence is greatly reduced, and no training signal or reference signal for performing phase control is required. Therefore, the system configuration is simplified. (2) The incoming beam is tracked by correcting the phase difference between the received signals at each of the antenna elements A1 to AN by feedforward, and since a feedback loop is not included unlike the second conventional example, a relatively low carrier wave is used. Even when the power-to-noise power ratio is C / N and the direction of the incoming signal beam changes at high speed, the incoming beam of a radio signal composed of a digital phase modulated wave or an unmodulated wave can be tracked with high accuracy and stability. . Therefore, there is almost no influence of disturbance due to the phase slip and the surrounding electromagnetic environment and accumulation of tracking errors as seen in the conventional method. (3) Spatial information of the array antenna can be effectively used by further performing least-squares regression correction from the correction phase amount in each antenna element system, and this is a problem when the number of antenna elements is large. It is possible to suppress the influence of a decrease in the carrier power to noise power ratio C / N per antenna element. (4) Since the above acquisition and tracking are all performed on the received signal by, for example, signal processing such as digital signal processing, the microwave phase shifter or acquisition and tracking as seen in the phased array antenna of the first conventional example. No sensors or a motor for driving the machine are required.

The case where the regression correction by the least squares method is not performed in the first embodiment will be described below as a modification of the first embodiment. In this case, instead of calculating the phase difference between adjacent antenna elements in Equation 8, the numerator and denominator of Equation 8 are calculated with a predetermined reference antenna element, and the numerator of Equation 8 is substituted for sinΔφ _ci in Equation 18. And the denominator of Eq. 8 is also cosΔφ _ci
If the processing is performed by substituting into equation (8), the left side of equation (18) can be obtained without calculating tan ^-1 in equation (8) on the receiving side. Amplitude correction for ratio combining can be automatically performed. In this case, an equation for correcting the phase of the orthogonal baseband signal is expressed by the following equation (22).

[0114]

(Equation 22)

Here, the left side of Equation 22 is a matrix representing the vector of the received baseband signal of the i-th antenna element after the phase correction, and the first term on the right side is a phase rotation conversion matrix for phase correction, That is, it is a transformation matrix for in-phase, and the second term on the right side is a matrix representing the vector of the received baseband signal before phase correction. However, in this modified example, an operation is not performed between reception signals of two adjacent antenna elements, but an antenna element serving as a phase reference is set to, for example, A1, and this antenna element A1
And the other antenna elements A2 to AN
Is performed with each of the received signals, and in-phase processing is performed between them. In this modification, the reference antenna element is A1, but the present invention is not limited to this, and another single antenna element may be used. The advantage of executing this processing is that the operation of Equation 22 can not only convert the phase, but also convert the amplitude so that the amplitude also performs maximum ratio combining. That is, the following Expression 23 can be approximated from Expressions 5 and 6 using the approximate expression of Expression 24.

[0116]

(Equation 23)

As is apparent from Equation 23, the product F (a ₁ ) · F (a _i ) of the filtered amplitude coefficient with respect to the product of the two matrices of the third and fourth terms on the right side of Equation 23 Is hanging.
Here, the amplitude coefficient a ₁ , the amplitude coefficient a _i and the cosine c of the phase difference
If osδ1 _{, i} can be regarded as independent variables that fluctuate in time at random around an average value due to thermal noise in the short term _{, the following} equation 24 can be obtained. .

[0118]

F (a _{1 ai} cos δ _{1, i} ) ≒ F (a ₁ ) · F (a _i ) · F (cos δ _{1, i} ) F (a _{1 ai} sin δ _{1, i} ) ≒ F (a ₁ ) · F (a _i ) · F (sin δ _{1, i} )

Equation 24 is satisfied for the following reason.
Now, variables u and v are independent variables that fluctuate with time at random, and their average values are avr (u), av
If r (v) is set, it can be expressed as in the following Expression 25.

[0120]

[Mathematical formula-see original document] u = avr (u) + eu v = avr (v) + ev

Here, eu and ev are each a random component, and represent a component that fluctuates at random with an average value of 0. When the digital filter is, for example, a predetermined low-pass filter, F (·) is a transfer function of the low-pass filter, and therefore the following equation 26 is established from equation 25.

[0122]

F (u) ≒ avr (u) F (v) ≒ avr (v) F (eu) ≒ 0 F (ev) ≒ 0

If the following equation 27 holds between these variables u and v, then equation 24 holds.

[0124]

F (u · v) ≒ F (u) · F (v)

By substituting equation (25) into the left side of equation (27), equation (27)
Is transformed using Equation 26, the following Equation 28 can be obtained.

[0126]

F (u · v) = F ((avr (u) + eu) · (avr (V) + ev)) = F (avr (u) · avr (v) + ev · avr (u) + eu · avr (v) + eu · ev) = F (avr (u) · avr (v)) + F (ev · avr (u)) + F (eu · avr (v)) + F
(eu ・ ev) = avr (u) ・ avr (v) + avr (u) ・ F (ev) + avr (v) ・ F (eu) + F (eu ・
ev) ≒ F (u) ・ F (v) + F (eu ・ ev)

Here, it can be assumed that the random components eu and ev are independent of each other and have no correlation, and the cross-correlation function R (τ) between them is always 0. 29 holds.

[0128]

(Equation 29)

This means that the time average of (eu · ev) is almost 0
Is shown. Therefore, F (eu · ev) ≒
It becomes 0, and from this and Expression 28, Expression 27 is established, and Expression 24 is established. However, Equation 24 is satisfied with higher accuracy particularly in the case of the constant envelope modulation method in which the envelope is constant, and when the envelope fluctuates depending on the information symbol, the approximation accuracy deteriorates.

On the other hand, if the calculation of Expression 22 is to be performed between itself and the system of the reference antenna element A1, if the received signal-to-noise power ratio S / N is sufficiently high, the following Expression 30 is obtained. Holds.

[0131]

[Equation 30]

From Equations 23 and 30, the amplitude conversion coefficient of the received signal at each antenna element is proportional to the filter output F (a _i ) (i = 1, 2,..., N) of the amplitude of each received signal. You can see that there is. Combining the calculation results of Equations 22 and 30 in accordance with Equation 20 results in the same as performing maximum ratio combining, and greatly increases the received signal-to-noise power ratio after combining a plurality of received signals. Can be improved. In this case, the operation shown in Expression 19 is not required, and
The phase difference correction unit 44 and the amplitude correction unit 45 in FIG. 3 can be integrated. However, if the calculation of the filter output F (a ₁ ² ) is performed in the same manner as in Expression 28, with the random component of the amplitude coefficient a ₁ being ea _1, the following Expression 31 is obtained.

[0133]

Equation 31] ^{_{F (a 1 2) = F}} 2 (a 1) + F (ea 1 2)

That is, as is apparent from Equation 31, when the received signal power to noise power ratio S / N is low, Equation 31
There is a problem that the second term on the right side of can not be ignored, and therefore, the approximation error of Equation 30 increases. In addition, when there is no multipath and when the regression correction by the least squares method is not performed, the result is the same when using Equations 8 and 18 and when using Equations 22 and 30.

<Second Embodiment> FIG. 15 is a block diagram showing a part of a receiving section of an automatic beam acquisition and tracking device for a communication array antenna according to a second embodiment of the present invention. In the device of the second embodiment, a pair is formed by two adjacent antenna element systems,
After correcting the amplitude and phase difference so that the quadrature baseband signals between them have the same phase, in-phase synthesis between the two antenna element systems (that is, maximum ratio synthesis) is performed.
And a pair is formed again between the outputs to perform amplitude phase difference correction and in-phase synthesis (maximum ratio synthesis). By repeating this, finally, only one output of the array antenna is obtained in which the signals received by all the antenna elements are combined at the maximum ratio in-phase, and as a result, the array antenna performs acquisition tracking of the incoming signal beam. . The amount of calculation required for the amplitude / phase difference correction and the in-phase synthesis is substantially the same as that of the first embodiment. Here, maximum ratio combining or maximum ratio in-phase combining refers to performing in-phase combining such that the obtained received signal-to-noise power ratio is maximized.

The configuration shown in FIG. 15 is a configuration in which the device includes nine quasi-synchronous detection circuits QD-1 to QD-9. The figure shows a portion which is provided before the demodulator 5.

As shown in FIG. 15, the quasi-synchronous detection circuit QD
-1, the quadrature baseband signals I ₁ and Q ₁ relating to the antenna element A1 are input to the in-phase combiner 81 and the amplitude / phase difference correction circuit PCA-1, and the quasi-synchronous detection circuit QD-
The orthogonal baseband signals I ₂ and Q ₂ relating to the antenna element A2 output from the antenna element A2 are input to the amplitude / phase difference correction circuit PCA-1. Similarly, the quadrature baseband signals I ₃ and Q ₃ for the antenna element A3 output from the quasi-synchronous detection circuit QD-3 are input to the in-phase combiner 82 and the amplitude / phase difference correction circuit PCA-2, The quadrature baseband signals I ₄ and Q ₄ for the antenna element A4 output from the QD-4 are input to the amplitude / phase difference correction circuit PCA-2. Further, quadrature baseband signals I _5, Q ₅ relates to an antenna element A5 output from the quasi-synchronous detection circuit QD-5 are input to the phase combiner 83 and the amplitude phase difference correction circuit PCA-3, sub-synchronous detection circuit QD- The quadrature baseband signals I ₆ and Q ₆ related to the antenna element A6 output from ₆ are input to the amplitude / phase difference correction circuit PCA-3. Further, the antenna element A7 output from the quasi-synchronous detection circuit QD-7
The quadrature baseband signals I ₇ and Q ₇ for the in-phase combiner 8
4 and the quadrature baseband signals I ₈ and Q ₈ for the antenna element A8 output from the quasi-synchronous detection circuit QD-8 and input to the amplitude / phase difference correction circuit PCA-4. You. Further, the quadrature baseband signals I ₉ and Q ₉ for the antenna element A9 output from the quasi-synchronous detection circuit QD-9 are output from the amplitude / phase difference correction circuit P
Input to CA-5.

[0138] amplitude and phase difference correction circuit PCA-1 is a quasi synchronous detection circuit perpendicular to an antenna element A1 which is input from the QD-1 baseband signals I _1, Q ₁ and perpendicular to an antenna element A2 adjacent baseband signal I ₂ , Q _2, and a predetermined filter for noise removal, calculate a transformation matrix element (element of the transformation matrix of Expression 22) for in-phase of two received signals between adjacent antenna elements, and calculate the calculated transformation. Based on the transformation matrix (Equation 22) including the matrix elements, the phase difference correction (phase shift) is performed so that the baseband signals of the antenna elements A1 and A2 have the same phase. Similarly, by performing weighting with an amplification gain proportional to the calculated received signal strength, the amplitude / phase correction processing is performed, and the processed baseband signal is output to the in-phase combiner 81. You. The in-phase combiner 81 includes an antenna element A
And quadrature baseband signals I _1, Q ₁ for one-phase combiner 86 and a quadrature baseband signal outputted from the amplitude phase difference correction circuit PCA-1 phase combining for each channel
And the amplitude and phase difference correction circuit PCA-6. In addition,
All the in-phase synthesizers 81 to 88 in-phase synthesize the two pairs of input baseband signals for each channel.

The amplitude and phase difference correction circuit PCA-2 includes orthogonal baseband signals I ₃ and Q ₃ for the antenna element A3 input from the quasi-synchronous detection circuit QD- ₃ and an orthogonal baseband signal I ₄ for the adjacent antenna element A4. , Q ₄ , and performs an amplitude / phase correction process in the same manner as the amplitude / phase difference correction circuit PCA- 1, and outputs the processed baseband signal to the in-phase synthesizer 82. The in-phase combiner 82 performs in-phase synthesis on the quadrature baseband signals I ₃ and Q ₃ relating to the antenna element A3 and the quadrature baseband signal output from the amplitude / phase difference correction circuit PCA-2 to perform an amplitude / phase difference correction circuit PCA.
Output to -6.

[0140] amplitude and phase difference correction circuit PCA-3 are quasi-synchronous detection circuit QD-5 perpendicular to an antenna element A5 inputted from the baseband signal I _5, Q ₅ and quadrature baseband signals to an antenna element A6 adjacent I ₆ , with Q _6, performs similarly amplitude phase correction processing and the amplitude phase difference correction circuit PCA-1, and outputs the baseband signal after the processing to the in-phase combiner 83. The in-phase combiner 83 performs in-phase synthesis on the quadrature baseband signals I ₅ and Q ₅ related to the antenna element A5 and the quadrature baseband signal output from the amplitude / phase difference correction circuit PCA-3 to form the in-phase combiner 87 and the amplitude position. Output to the phase difference correction circuit PCA-7.

The amplitude / phase difference correction circuit PCA-4 includes the orthogonal baseband signals I ₇ and Q ₇ for the antenna element A 7 input from the quasi-synchronous detection circuit QD- ₇ and the orthogonal base band signals I ₈ for the adjacent antenna element A _8. , with Q _8, performs similarly amplitude phase correction processing and the amplitude phase difference correction circuit PCA-1, and outputs the baseband signal after the processing to the in-phase combiner 84. Phase combiner 84, a quadrature baseband signal I _7, Q ₇ an antenna element A7, phase combiner 85 and amplitude position and the quadrature baseband signal output from the amplitude phase difference correction circuit PCA-4 phase combining Output to the phase difference correction circuit PCA-5.

The amplitude / phase difference correction circuit PCA-5 includes a quadrature baseband signal output from the in-phase combiner 84 and quadrature baseband signals I ₉ and Q 9 for the antenna element A9 input from the quasi-synchronous detection circuit QD-9. ₉ to perform an amplitude / phase correction process in the same manner as in the amplitude / phase difference correction circuit PCA-1.
5 is output. The in-phase synthesizer 85 performs in-phase synthesis on the quadrature baseband signal output from the in-phase synthesizer 84 and the quadrature baseband signal output from the amplitude / phase difference correction circuit PCA-5 to perform an amplitude / phase difference correction circuit PCA-7. Output to

The amplitude / phase difference correction circuit PCA-6 uses the quadrature baseband signal output from the in-phase synthesizer 81 and the quadrature baseband signal output from the in-phase synthesizer 82 to generate an amplitude / phase difference correction circuit PCA. As in the case of -1, the amplitude and phase correction processing is executed, and the processed baseband signal is output to the in-phase synthesizer 86. The in-phase synthesizer 86 performs in-phase synthesis on the quadrature baseband signal output from the in-phase synthesizer 81 and the quadrature baseband signal output from the amplitude / phase difference correction circuit PCA-6 to form the in-phase synthesizer 88 and the amplitude / phase difference. Output to the correction circuit PCA-8.

The amplitude / phase difference correction circuit PCA-7 uses the quadrature baseband signal output from the in-phase synthesizer 83 and the quadrature baseband signal output from the in-phase synthesizer 85 to generate an amplitude / phase difference correction circuit PCA. As in the case of -1, the amplitude and phase correction processing is executed, and the baseband signal after the processing is output to the in-phase combiner 87. The in-phase synthesizer 87 performs in-phase synthesis on the quadrature baseband signal output from the in-phase synthesizer 83 and the quadrature baseband signal output from the amplitude / phase difference correction circuit PCA-7 to perform an amplitude / phase difference correction circuit PCA-8.
Output to

The amplitude / phase difference correction circuit PCA-8 uses the quadrature baseband signal output from the in-phase synthesizer 86 and the quadrature baseband signal output from the in-phase synthesizer 87 to generate an amplitude / phase difference correction circuit PCA. As in the case of −1, an amplitude / phase correction process is executed, and the processed baseband signal is output to the in-phase synthesizer 88. The in-phase synthesizer 88 performs in-phase synthesis on the quadrature baseband signal output from the in-phase synthesizer 86 and the quadrature baseband signal output from the amplitude / phase difference correction circuit PCA-8 and outputs the resultant to the demodulator 5. here,
The quadrature baseband signal output from the in-phase combiner 88 corresponds to the quadrature baseband signal output from the in-phase combiner 4 in the first embodiment of FIG. This is a quadrature baseband signal obtained by executing an amplitude / phase difference correction process based on a signal.

FIG. 16 shows the amplitude / phase difference correction circuit P of FIG.
It is a block diagram of CA-s (s = 1, 2,..., 8).
The amplitude and phase difference correction circuit PCA of FIG. 16 of the second embodiment.
−s is the amplitude / phase difference correction circuit PC of FIG. 3 of the first embodiment.
The following points are different from -i. (1) The phase difference estimator 40a calculates the orthogonal baseband signals I _i , Q _i, and I _{j for} the two antenna elements i and _j ,
Based on Q _j , a noise-free conversion matrix element (element of the conversion matrix of Equation 22) for in-phase of the received signals received by the two antenna elements i and j is calculated, and the calculated conversion matrix element is calculated. The conversion matrix including the conversion matrix is output to the phase difference correction unit 44a. (2) The phase difference correction unit 44a corrects the phase difference by shifting the phase of the quadrature baseband signal input from the delay buffer 43 based on the conversion matrix input from the phase difference correction unit 40a, and corrects the amplitude. 45. (3) The adder 41 and the least squares regression correction unit 42 are not provided. Note that the delay buffer 43 and the amplitude correction unit 45
The operation is the same as that of the embodiment.

Therefore, the amplitude / phase difference correction circuit PC shown in FIG.
A-s because of the quasi-synchronized orthogonal to an antenna element Ai inputted from the detection circuit QD-i baseband signals I _i, Q _i and quadrature baseband signals I _j on the neighboring antenna elements Aj, Q _j and denoising Calculates a transformation matrix element (element of the transformation matrix of Expression 22) for in-phase of the two received signals between adjacent antenna elements using the predetermined filter of, and based on the transformation matrix including the computed transformation matrix element The phase difference is corrected so that the two baseband signals of the antenna elements Ai and Aj are in phase, that is, the phase is shifted and the first
Similarly to Example amplitude correction unit 45 performs an amplitude phase correction processing by performing weighted by amplification gain proportional to the received signal strength calculated above, the baseband signal Ic _i after the treatment, the Qc _i In-phase synthesizer (1 of 81 to 88)
Output).

In the amplitude / phase difference correction circuit PCA-s of the second embodiment, the amplitude / phase difference correction circuit PC of FIG.
In A-1 to PCA-8, when the conversion operation using the conversion matrix for in-phase is configured to be performed using Equations 22 and 30, the phase difference correction unit 44a and the amplitude correction unit 45 in FIG. Can be integrated. In this integrated configuration, the phase difference correction and the amplitude correction for in-phase can be performed simultaneously, whereby the plurality of reception signals received by the array antenna 1 are combined at the maximum ratio and the amplitude is corrected. , And one combined received signal can be output. Further, as a modification of the second embodiment, similar to the processing of the first embodiment, the following configuration may be adopted. Phase difference estimation unit 40a includes quadrature baseband signals I _i for the two antenna elements i and j, Q _i and I _j, Q _j
, The instantaneous phase difference δ _{i, j} of the received signals received by the two antenna elements i, j is estimated using _Equation 7 and noise is removed, and the estimated phase difference δc _{i, j} after noise removal is obtained. (See Equation 8) is output to the phase difference correction unit 44a. Next, the phase difference corrector 44a receives the δ input from the phase difference corrector 40a.
c _i, and corrects the phase difference by the quadrature baseband signal input from the delay buffer 43 to only phase the estimated phase difference .delta.c _{i, j} based on the _j outputs to the amplitude correction unit 45.

The advantages of the second embodiment are as follows as compared with the first embodiment. In the first embodiment, the phases in each antenna element system with respect to a certain reference antenna are calculated by integrating the phase differences between all adjacent antenna element systems, and finally the maximum ratio in-phase synthesis is performed. When there is an antenna element having a low reception level or a defective antenna, not only the phase estimation for the antenna element cannot be performed, but also the phase estimation in another antenna element system may be affected. On the other hand, in the second embodiment, the maximum ratio in-phase synthesis is performed first between the two antenna systems without integrating the phase difference between the adjacent antenna elements. Even if there is a defective antenna element, this can be prevented from affecting the in-phase synthesis in other antenna element systems.
For this reason, it can be said that the second embodiment is more resistant to a failure of the antenna element and the circuit device connected thereto than the first embodiment. However, in the first embodiment, the phase difference correction can be performed in parallel in all the antenna element systems, whereas in the second embodiment, the serial processing needs to be performed approximately log ₂ (the number of antenna elements). As a result, the calculation time becomes longer.

<Embodiment 3> FIG. 17 is a block diagram showing a part of a receiving section of an automatic beam acquisition and tracking apparatus according to a third embodiment of the present invention. In the third embodiment, the received signal of each antenna element is set to 2
A plurality of M beam signals BE-1 to BE- input to a multi-beam forming circuit 90 based on a two-dimensional fast Fourier transform (FFT) or a discrete Fourier transform (DFT) are obtained.
From among M, a predetermined plurality of L beam signals B are arranged by the beam selection circuit 91 in order of the signal intensity, that is, the beam signal having the largest sum of squares of the beam electric field value, from the largest signal intensity.
After selecting ES-1 to BES-L, the amplitude / phase difference correction processing is performed between the plurality of beam signals BES-1 to BES-L by the amplitude / phase difference correction circuits PCA-1 to PCA- (L-1). Then, the in-phase synthesizer 92 performs in-phase synthesis (maximum ratio synthesis). As a result, the array antenna is characterized by capturing and tracking an incoming beam.

In FIG. 17, the multi-beam forming circuit 9
0 is the received quadrature baseband signals I _i , Q _i (i = 1, 2,...) From the quasi-synchronous detection circuits QD-1 to QD-N.
N), and a direction vector d _m representing the direction of each main beam of the predetermined plurality of M beam signals to be formed, which is predetermined so that the desired wave can be received within a predetermined radiation angle range,
Based on the reception frequency fr of the reception signal, each beam electric field value EI _m and EQ _m (m = 1, 2,..., M) of a multi-beam composed of a plurality of M beams is calculated, and the beam electric field value EI is calculated. A beam signal having _m and EQ _m is output to the beam selection circuit 91. That is, a plurality of M directions of each of the multi-beams to be formed are determined in advance corresponding to the arrival direction of the desired wave, and these directions are directional vectors d ₁ , d ₂ , and d ₂ when viewed from a predetermined origin. ..., d _M (hereinafter, representatively denoted by a reference sign d _m in.) represented by. Here, M is the number of direction vectors d _m is set so as to be able to receive the desired wave with an array antenna 1, the number of preferably a 4 or more and the antenna elements A1 to AN N or less. Also, position vectors r ₁ , r ₂ ,..., R _N of each of the antenna elements A1 to An of the array antenna 1
(Hereinafter, representatively denoted by a reference sign r _n in.) Is determined in advance as a direction vector as viewed from the predetermined origin. Then, the multi-beam forming circuit 90 calculates the following Expression 32 and Expression 3
3 with a plurality of 2N beam field strength EI _n and EQ _n corresponding to the represented above direction vector d _n in each composite electric field by computing, beam signal having a beam field strength EI _n and EQ _n Is output to the beam selection circuit 91.

[0152]

(Equation 32)

[Equation 33] here,

Equation 34] _{a mn = - (2π · fr} / c) · (d m · r n)

Here, c is the speed of light, and (d _m · r _n ) is the inner product of the direction vector d _m and the position vector r _n . Therefore, the phase a _mn is a scalar quantity.

Next, the beam selecting circuit 91 outputs a plurality of M beam electric field values EI _m and EQ _m of each of the beam signals BE-1 to BE-M output from the multi-beam forming circuit 90.
EI _m ² + EQ _m ² (m = 1, 2,..., M) and calculates the sum of the squares of the larger beam electric field values in descending order from the beam signal having the largest sum of the squares of the beam electric field values. Predetermined plurality of L beam signals BES-1 to BES-1
After selecting ES-L, the plurality of beam signals BES
-1 to BES-L are combined with the in-phase synthesizer 92 and the (L-1) amplitude / phase difference correction circuits PCA-1 to PCA- (L-
Output to 1). Here, L is a natural number equal to or less than the plurality M and is determined in advance. The beam selection circuit 91
Is provided to remove a received signal having a very low received signal level and a poor S / N. Further, in the above calculation, the sum of squares of the beam electric field value is calculated, but the present invention is not limited to this, and the square root of the sum of squares of the beam electric field value corresponding to the absolute value of the beam electric field value is calculated. It may be.

The quadrature baseband signal of the beam signal BES-1 having the square sum of the maximum beam electric field value serving as a reference beam signal is supplied to the in-phase combiner 92 and the amplitude / phase difference correction circuit P.
The orthogonal baseband signal of the beam signal BES-2 having the second largest sum of the squares of the beam electric field value input to the CA-1 is input to the amplitude / phase difference correction circuit PCA-1, and the signal having the third largest beam electric field value is obtained. Beam signal B having a sum of squares
The quadrature baseband signal of ES-3 is input to the amplitude / phase difference correction circuit PCA-2, and similarly, the beam signal BES-L having the L-th largest sum of squares of the beam electric field value is input.
Of the quadrature baseband signal is an amplitude phase difference correction circuit PCA-
(L-1). Here, the amplitude / phase difference correction circuit PCA-s (s = 1, 2,..., L−1) is configured similarly to the amplitude / phase difference correction circuit PCA-s of FIG. 16 in the second embodiment.

In the third embodiment, the amplitude / phase difference correction circuit PCA-1 uses the quadrature baseband signal of the reference maximum beam signal BES-1 and a predetermined filter for noise removal to generate these two beam signals. Is calculated, and based on the conversion matrix including the calculated conversion matrix element, the phase difference is corrected so that the baseband signals of these two beam signals are in phase, that is, the phase is shifted, Further, similarly to the amplitude correction unit 45 of the first embodiment, the amplitude and phase correction processing is executed by performing weighting with the amplification gain proportional to the calculated received signal strength, and the baseband signal after the processing is subjected to in-phase synthesis. Output to the output unit 92. Further, the amplitude / phase difference correction circuit PCA-2 uses the orthogonal baseband signal of the reference maximum beam signal BES-1 and the orthogonal baseband signal of the beam signal BES-3 to generate the amplitude / phase difference correction circuit PCA-1. Similarly, an amplitude / phase difference correction process is executed, and the processed baseband signal is output to the in-phase synthesizer 92. Hereinafter, similarly, the amplitude / phase difference correction circuit PCA- (L-1) outputs the reference maximum beam signal BE.
S-1 orthogonal baseband signal and beam signal BES-L
In the same manner as the amplitude / phase difference correction circuit PCA-1, the amplitude / phase difference correction processing is executed using the quadrature baseband signal, and the processed baseband signal is output to the in-phase synthesizer 92. The in-phase synthesizer 92 performs in-phase synthesis on the input plurality of L baseband signals for each channel, and outputs the resultant to the demodulator 5.

In the third embodiment, the phase of the beam signal having the highest signal intensity is adjusted to the phase of all other selected beam signals. That is, the beam signal having the largest signal strength is used as a reference reception signal, and the phase of another selected beam signal is corrected based on the reference signal. In the third embodiment, the calculation of the amplitude / phase difference correction and the in-phase synthesis only needs to be performed “(the number of selected beams L) −1” times. However, the multi-beam forming circuit 90 and the beam selecting circuit 91 are added. There is a need.

In the amplitude and phase difference correction circuit PCA-s of the third embodiment, the amplitude and phase difference correction circuit PC of FIG.
In A-1 to PCA- (L-1), when the conversion operation using the conversion matrix for in-phase is configured to be performed using Expressions 22 and 30, the phase difference correction unit 44 in FIG.
a and the amplitude correction unit 45 can be integrated. In this integrated configuration, the phase difference correction and the amplitude correction for in-phase can be performed simultaneously, whereby the plurality of reception signals received by the array antenna 1 are combined at the maximum ratio and the amplitude is corrected. , And one combined received signal can be output. Further, as a modified example of the third embodiment, the following configuration may be adopted similarly to the processing of the first embodiment. The phase difference estimator 40a calculates the instantaneous phase difference between the received signals received by the two antenna elements i and j based on the orthogonal baseband signals I _i and Q _i and I _j and Q _j for the two antenna elements i and j. δ _{i, j} is estimated using _Equation 7 and noise is removed, and the estimated phase difference δc after the noise is removed
_{i, j} (see Expression 8) is output to the phase difference correction unit 44a. Next, the phase difference correction unit 44a converts the orthogonal baseband signal input from the delay buffer 43 based on the δci _{, j} input from the phase difference correction unit 40a into the estimated phase difference δc
_The phase difference is corrected by shifting the phase by _{i and j,} and output to the amplitude correction unit 45.

The advantages of the third embodiment are as follows as compared with the first and second embodiments. In the first and second embodiments, as the number of antenna elements constituting the array antenna increases, the received signal-to-noise power ratio per element decreases, and the accuracy of phase difference correction deteriorates. On the other hand, in the third embodiment, since the multi-beam forming circuit 90 and the beam selecting circuit 91 form a beam having a high received signal-to-noise power ratio and then perform amplitude / phase difference correction, Even if the received signal-to-noise power ratio for each is low, there is no effect on the accuracy of the phase difference correction, and in principle there is no upper limit on the number of antenna elements. Also,
When a strong interference wave or the like arrives from another direction, the first and second embodiments all attempt to combine them, so that the combined reception signal is distorted or the directivity is disturbed. In this embodiment, these beams are spatially separated to some extent by beam selection, so that they are hardly affected by these interference waves. However, in the first and second embodiments, since the received signals input from all the antenna elements are effectively used and beam forming is performed so that the direction of the arriving beam always has the maximum gain, the arriving signal beam direction is In the third embodiment, if the number of beams to be selected is small, there is a loss of power at the time of performing beam selection, and the direction of the arriving signal beam is reduced. If it changes, there is a problem that the undulation occurs in the gain depending on the direction.

[0160]

As described above in detail, according to the present invention, there is provided an array antenna control method or apparatus for controlling an array antenna composed of a plurality of antenna elements closely arranged in a predetermined arrangement shape. Using a common local oscillation signal for each of the plurality of reception signals received by each antenna element of the array antenna, an in-phase component that is in phase with the local oscillation signal and a quadrature component that is orthogonal to the local oscillation signal , Respectively. Next, first data calculated by the sum of the products of the in-phase components of the quadrature signals and the products of the quadrature components of the two first and second antenna elements that are different pairs from each other among the plurality of antenna elements. And the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. The second data calculated by the difference between the products of the in-phase components of the quadrature signals is filtered through a noise suppression filter having a predetermined transfer function, and then the filtered first data is It is included in two elements that are diagonal to each other, and the filtered second data and the filtered second data have (−
By multiplying the orthogonal signal of the second antenna element by a 2 × 2 transformation matrix having another two elements diagonal to each other with the third data obtained by multiplying by 1), The quadrature signal of the second antenna element is made in phase with the quadrature signal of the first antenna element. The in-phase processing is repeated for mutually different pairs of the orthogonal signals in the plurality of antenna elements to output a plurality of in-phase quadrature signals. Further, the in-phase plurality of quadrature signals are in-phase synthesized to output one quadrature signal after the in-phase synthesis. According to another aspect of the present invention, there is provided an array antenna control method or apparatus for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape. Using a common local oscillation signal, each of the plurality of reception signals received in step (1) is converted into an in-phase component having the same phase as the local oscillation signal and a quadrature signal including an orthogonal component orthogonal to the local oscillation signal. . Next, first data calculated by the sum of the products of the in-phase components of the quadrature signals and the products of the quadrature components of the two first and second antenna elements that are different pairs from each other among the plurality of antenna elements. And the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. The second data calculated by the difference between the products of the in-phase components of the quadrature signals is passed through a noise suppression filter having a predetermined transfer function to be filtered, and then the filtered second data is subjected to the above-described processing. By dividing by the filtered first data and calculating the arctangent of the result of the division, the quadrature signal of the first antenna element and the quadrature signal of the second antenna element are made in phase. Correction phase Is calculated, and the quadrature signal of the second antenna element is shifted in phase by the calculated correction phase amount to make the quadrature signal of the second antenna element in-phase with the quadrature signal of the first antenna element. . The in-phase processing is repeated for different pairs of the orthogonal signals in the plurality of antenna elements to output a plurality of in-phase quadrature signals. Further, the in-phase plurality of quadrature signals are in-phase synthesized, and the
Output two orthogonal signals.

Therefore, the present invention has the following specific effects. (1) Since a feedback loop is not included as in the second conventional example, even when the received carrier power-to-noise power ratio C / N per antenna element is relatively low, a special direction sensor or a counterpart station can be used. An incoming signal beam of a radio signal can be automatically and quickly captured without using position information and the like, and data lost even when an instantaneous interruption of the signal beam due to an obstacle or the like can be minimized. Further, the preamble length can be reduced in a burst mode communication system such as packet communication. Further, for example, since a received signal modulated with communication data can be directly used, a special training signal or reference signal for performing phase control is not required, and the system configuration can be simplified. (2) Since a feedback loop is not included as in the second conventional example, a case where the carrier power to noise power ratio C / N per antenna element is relatively low and the direction of an incoming signal beam changes at high speed. Does not generate a phase slip, and does not have a direction sensor as in the first conventional example. Therefore, there is no influence of disturbance due to disturbance of the lines of magnetic force in the surroundings, accumulation of tracking errors, and the like. Beams can be tracked with high accuracy and stability, and for example, communication quality during movement can be improved. In addition, not only when the own station moves, but also when the other station moves, tracking can be performed without special information on the position of the other station. Further, in the case of a burst mode communication method such as packet communication, a tracking method using a training signal (preamble) cannot follow a change in the direction of an incoming beam in the middle of a burst. Since the modulated received signal can be used directly, it is possible to follow the signal in real time even during a burst.

According to another aspect, at the time of the in-phase, based on the arrangement shape of the array antenna, the calculated correction phase amount is returned so as to return to the plane of the arrangement shape. The orthogonal signal of the second antenna element is regressively corrected by the correction phase amount, and the orthogonal signal of the second antenna element is shifted by the phase amount of the regression-corrected correction phase. In-phase to quadrature signal. Therefore, since the spatial information of the array antenna can be effectively used, the effect of a decrease in the carrier-to-noise power ratio C / N per antenna element, which is a problem when the number of antenna elements is large, is suppressed. It is possible,
Deterioration of tracking characteristics and communication quality can be prevented.

Further, according to another aspect of the present invention, there is provided an array antenna control method or apparatus for controlling an array antenna including a plurality of closely arranged antenna elements in a predetermined arrangement shape. Using a common local oscillation signal, each of the plurality of reception signals received by each antenna element is converted into an in-phase component having the same phase as the local oscillation signal and a quadrature signal including an orthogonal component orthogonal to the local oscillation signal. Convert each one. Next, first data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of the pair of first and second antenna elements of the plurality of antenna elements;
And the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. And the second data calculated by the difference between the products are passed through a noise suppression filter having a predetermined transfer function, and are filtered.
Has two data, each of which is diagonal to each other,
Further, the filtered second data and the third data obtained by multiplying the filtered second data by (−1) are combined into another two diagonal elements. The quadrature signal of the second antenna element is in-phase with the quadrature signal of the first antenna element by multiplying the orthogonal signal of the second antenna element by the 2 × 2 transformation matrix. Further, after performing the in-phase processing on the mutually different pairs of the quadrature signals in the plurality of antenna elements, the in-phase processing is performed on each of two different quadrature signals among the plurality of in-phase quadrature signals. By executing the processing and repeating until the number of quadrature signals after the in-phase synthesis becomes one, one quadrature signal after the in-phase synthesis is output.
That is, in-phase synthesis is performed first between the two element systems without integrating the phase difference between adjacent antenna elements. Can be prevented from affecting the in-phase synthesis in other antenna element systems. Therefore, it can be said that the antenna element and the circuit device connected to the antenna element are resistant to a failure or the like.

According to another aspect, a plurality of orthogonal signals at each antenna element of the array antenna and a desired wave can be received within a predetermined radiation angle range immediately before executing the in-phase processing. Based on the direction of each main beam of the predetermined plurality of beams to be formed and the reception frequency of the reception signal, the plurality of beam electric field values are calculated to calculate each of the beam electric field values. Outputting a plurality of orthogonal beam signals respectively having a plurality of orthogonal beam signals, selecting a predetermined number of orthogonal beam signals having a larger beam electric field value including an orthogonal beam signal having a maximum beam electric field value among the plurality of outputted orthogonal beam signals; The orthogonal beam signal having the maximum beam electric field value as a reference orthogonal signal of the first antenna element, and the orthogonal beam signal having the maximum beam electric field value. Selected orthogonal beam signals other than performing the process of the phase as a quadrature signal of the second antenna element. That is, since a phase difference is corrected after a beam signal having a high received signal-to-noise power ratio is formed by multi-beam forming and beam selection, even if the received signal-to-noise power ratio of each antenna element is low, the position is low. There is no effect on the accuracy of phase difference correction, and there is in principle no upper limit on the number of antenna elements. In addition, when a strong interference wave or the like arrives from another direction, it is separated to some extent by beam selection, so that there is an advantage that it is hardly affected by these interference waves.

[0165]

[0166]

[0167]

Further, the present invention has the following specific effects. (1) Whether the arrangement intervals of the antenna elements are equal or unequal,
Alternatively, since the above operation can be performed regardless of whether the antenna surface is flat or curved, the arrangement of the antenna element has a large degree of freedom, and an array antenna that matches the shape of the moving object can be configured. (2) Since all of the above acquisition and tracking are performed by signal processing such as digital signal processing of the received signal, microwave phase shifters for the number of antenna elements, sensors for acquisition and tracking, or motors for mechanical drive are used. This is unnecessary, and the control device can be reduced in size and cost.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a receiving unit of an automatic beam acquisition and tracking device for a communication array antenna according to a first embodiment of the present invention.

FIG. 2 is a block diagram of a transmission unit of the automatic beam capturing and tracking apparatus of FIG.

FIG. 3 is a block diagram of the amplitude / phase difference correction circuit of FIG. 1;

FIG. 4 is a block diagram of a transversal filter included in the phase difference estimator of FIG. 3;

5A is a front view of each antenna element showing a calculation order of a correction phase amount in the first method for each antenna element of the array antenna, and FIG. 5B is a front view of each antenna element of the array antenna. FIG. 9 is a front view of each antenna element showing the order of calculation of the correction phase amount in the second method for calculating the amount of correction phase.

FIG. 6 is a front view of each antenna element showing the order of calculation of the correction phase amount in the third method for each antenna element of the array antenna.

FIG. 7 is a schematic diagram showing a relationship between an incoming beam and each antenna element and a graph showing a relationship between a position of each antenna element and a phase amount.

FIG. 8 shows a time change of an antenna relative gain in a signal arrival direction when an incoming signal beam direction is rotated at a beam rotation speed of 90 degrees / second in the automatic beam acquisition and tracking apparatus of FIG. It is a graph shown with.

9 is a graph showing a time change of an antenna pattern at the time of capturing a beam under the same conditions as in FIG.

FIG. 10 is a graph showing a change in an antenna pattern when an incoming signal beam direction is rotated at a beam rotation speed of 90 degrees / second under the same conditions as in FIG.

11 is a graph showing the total number of samplings until acquisition corresponding to the beam acquisition time with respect to the carrier power-to-noise power ratio C / N when the buffer size Buff is used as a parameter in the automatic beam acquisition and tracking apparatus of FIG. 1;

12 is a graph showing a tracking characteristic with respect to a carrier power-to-noise power ratio C / N when a buffer size Buff is used as a parameter in the automatic beam capture and tracking apparatus of FIG. 1;

13 is a graph showing tracking characteristics at the time of fine capture and coarse capture with respect to the carrier power-to-noise power ratio C / N when the operation cycle Topr is used as a parameter in the automatic beam capture and tracking apparatus of FIG. 1;

14 is a graph showing a tracking characteristic with respect to a carrier power-to-noise power ratio C / N when the operation cycle Topr is used as a parameter in the automatic beam acquisition and tracking apparatus of FIG. 1;

FIG. 15 is a block diagram showing a part of a receiving unit of the automatic beam acquisition and tracking device of the communication array antenna according to the second embodiment of the present invention.

16 is a block diagram of the amplitude / phase difference correction circuit of FIG.

FIG. 17 is a block diagram showing a part of a receiving unit of an automatic beam acquisition and tracking device for a communication array antenna according to a third embodiment of the present invention.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... Array antenna, 2 ... Low noise amplifier, 3 ... Down converter, 4 ... In-phase synthesizer, 5 ... Demodulator, 6 ... Quadrature modulator, 7 ... Up-converter, 8 ... Transmission power amplifier, 9 ... In-phase distributor, 10 ... Transmitting local oscillator, 11 ... First local oscillator, 12 ... Second local oscillator, 40, 40a ... Phase difference estimator, 41 ... Adder, 42 ... Least square regression corrector, 43 ... Delay buffer memory, 44, 44a: phase difference corrector, 45: amplitude corrector, 51 to 56: delay circuit, 60 to 66: tap coefficient multiplier, 70: adder, 81 to 88: in-phase combiner, 90: multi-beam forming circuit, 91 ... Beam selection circuit, 92 ... In-phase combiner, A1 to AN ... antenna elements, CI-1 to CI-N ... circulators, RM-1 to RM-N ... receiving modules, AD-1 to AD DN: A / D converter, QD-1 to QD-N: Semi-synchronous detection circuit, PC-1 to PC-N, PCA-1 to PCA-8: Amplitude / phase difference correction circuit, QM-1 to QM -N: quadrature modulation circuit, TM-1 to TM-N: transmission module.

────────────────────────────────────────────────── ─── Continuing on the front page (72) Inventor: Toyohisa Tanaka, 5: Sanraya, Daiya, Seika-cho, Soraku-cho, Kyoto Pref. No. 5, Hiratani, Seiya-cho, Gun-ri, Gunma, ATR Optical Co., Ltd. (56) References JP-A-61-137403 (JP, A) JP-A-2-141033 (JP, A) JP-A-5-63427 (JP, A) JP-A-4-113747 (JP, A) JP-A-3-85927 (JP, A) JP-A-3-234128 (JP, A) JP-A-6-90193 ( JP, A) JP-A-6-196921 (JP, A) 1992 IEICE Spring Conference Lecture Book [Part 2], Communication and Electronics, Lecture Number B-52, "DBF and MMIC Prototype of Active Conformal Array for Mobile Satellite Communication with GPS ", Wataru Nakajo et al., P2-52, Lecture Number SB-1-9," Prototype of DBF Antenna Signal Processor for Mobile Satellite Communication ", Otaki Yukio and 3 others, pp2-603 to 2-604, issued March 15, 1992 (58) Fields surveyed (Int. Cl. ⁷ , DB name) H01Q 3/00-3/46 H01Q 21/00- 25/04 G01S 7/00-7/46 G01S 13/00-13/95 H04B 7/00 H04B 7/02-7/12 H04L 1/02-1/04 JICST file (JOIS)

Claims (10)

(57) [Claims] An array antenna control method for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape, comprising: a plurality of antenna elements each of which is received by each antenna element of the array antenna. Using a common local oscillation signal to convert the received signal into a quadrature signal comprising an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal; First data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of the two orthogonal pairs of the first and second antenna elements that are different pairs of the elements; The product of the in-phase component of the quadrature signal of the antenna element and the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element And second data calculated from the difference between the products of the in-phase components of the quadrature signals of the second antenna element are passed through a noise suppression filter having a predetermined transfer function to be filtered, and then filtered. And the filtered first data is multiplied by (−1) by the filtered second data and the filtered second data, respectively. Another 2 data which is diagonal to each other with the obtained third data
Multiplying a quadrature signal of the second antenna element by a 2 × 2 transformation matrix of two elements to make the quadrature signal of the second antenna element in-phase with a quadrature signal of the first antenna element And outputting the plurality of in-phase quadrature signals by repeating the processing of the in-phase step for the mutually different pairs of the quadrature signals in the plurality of antenna elements; and the plurality of in-phase quadrature signals. And outputting one quadrature signal after the in-phase synthesis. 2. A method for controlling an array antenna for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape, the method comprising: receiving a plurality of antenna elements received by each of the antenna elements of the array antenna; Using a common local oscillation signal to convert the received signal into a quadrature signal comprising an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal; First data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of the two orthogonal pairs of the first and second antenna elements that are different pairs of the elements; The product of the in-phase component of the quadrature signal of the antenna element and the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element And second data calculated from the difference between the products of the in-phase components of the quadrature signals of the second antenna element are passed through a noise suppression filter having a predetermined transfer function to be filtered, and then filtered. Dividing the obtained second data by the filtered first data and calculating the arctangent of the result of the division, the orthogonal signal of the first antenna element and the orthogonal signal of the second antenna element are calculated. A quadrature signal of the second antenna element is calculated by calculating a correction phase amount for in-phase with the quadrature signal, and shifting the quadrature signal of the second antenna element by the calculated correction phase amount. The step of in-phase with the quadrature signal of the first antenna element and the processing of the step of in-phase are repeated for the mutually different pairs of the quadrature signals in the plurality of antenna elements. Steps and the control method for an array antenna, characterized in that it comprises a step of outputting one quadrature signal after phase synthesis phase combining a plurality of quadrature signals in-phase reduction which outputs a quadrature signal of a few. 3. The in-phase step includes regression-correcting the calculated correction phase amount based on the arrangement shape of the array antenna so as to return the calculated correction phase amount to a surface of the arrangement shape. And shifting the quadrature signal of the second antenna element to the quadrature signal of the first antenna element by shifting the phase of the quadrature signal of the second antenna element by the correction phase amount subjected to the regression correction. The method for controlling an array antenna according to claim 2, wherein: 4. A method of controlling an array antenna for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape, wherein the plurality of antenna elements received by each of the antenna elements of the array antenna are controlled. Using a common local oscillation signal to convert the received signal into a quadrature signal comprising an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal; First data calculated by the sum of the product of the in-phase components of the quadrature signals and the sum of the products of the quadrature components of the pair of first and second antenna elements, and the quadrature signal of the first antenna element; And the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element and the second antenna The second data calculated by the difference between the products of the in-phase components of the quadrature signals of the second signal and the second data are passed through a noise suppression filter having a predetermined transfer function to be filtered, and then the filtered first data is filtered. And the third data obtained by multiplying the filtered second data and the filtered second data by (−1). 2 × with data in another two elements diagonal to each other
Multiplying the quadrature signal of the second antenna element by the transform matrix of 2 to in-phase the quadrature signal of the second antenna element with the quadrature signal of the first antenna element; After performing the step processing for the mutually different pairs of the quadrature signals in the plurality of antenna elements, the in-phase processing for the two mutually different quadrature signals among the plurality of in-phase quadrature signals is performed. Outputting the one quadrature signal after the in-phase synthesis by repeating the process until the number of quadrature signals after the in-phase synthesis becomes one, thereby outputting one quadrature signal after the in-phase synthesis. 5. Immediately before executing the in-phase processing, a predetermined number of orthogonal signals and a desired wave at each antenna element of the array antenna are received in a predetermined radiation angle range. The plurality of beam electric field values are calculated based on the direction of each main beam of the predetermined plurality of beams to be formed and the reception frequency of the received signal, and the plurality of orthogonal electric fields having the respective beam electric field values are calculated. Outputting a beam signal, selecting a predetermined number of orthogonal beam signals having a larger beam electric field value including the orthogonal beam signal having the largest beam electric field value among the plurality of outputted orthogonal beam signals, and A quadrature beam signal having an electric field value is set as a reference quadrature signal of the first antenna element, and a signal other than the quadrature beam signal having the maximum beam electric field value is selected. 2. The array antenna control method according to claim 1, further comprising the step of: performing the in-phase step of converting the obtained quadrature beam signal into a quadrature signal of the second antenna element. 6. An array antenna control device for controlling an array antenna composed of a plurality of antenna elements closely arranged in a predetermined arrangement shape, wherein the plurality of antenna elements received by each antenna element of the array antenna are provided. A conversion means for converting the received signal into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal, using a common local oscillation signal; A noise suppression filter having a function, each of which is a different pair of the plurality of antenna elements.
First data calculated by the sum of the products of the in-phase components of the quadrature signals and the products of the quadrature components in the first and second antenna elements; the in-phase component of the quadrature signal of the first antenna element; A second calculated by the product of the quadrature component of the quadrature signal of the second antenna element and the product of the quadrature component of the quadrature signal of the first antenna element and the in-phase component of the quadrature signal of the second antenna element. And the data is passed through the noise suppression filter and filtered, and then the filtered first data is included in two elements that are diagonal to each other, and the filtered second data is 2 × having the data and the third data obtained by multiplying the filtered second data by (−1) in another two elements diagonally to each other
2 for multiplying the quadrature signal of the second antenna element by a transformation matrix of 2 to in-phase the quadrature signal of the second antenna element with the quadrature signal of the first antenna element. Control means for outputting a plurality of quadrature signals in-phase by repeating the processing of the quadrature signal for the mutually different pairs of the quadrature signals in the plurality of antenna elements, and a plurality of quadrature signals in-phase by the control means. A control unit for an array antenna, comprising: synthesizing means for performing in-phase synthesis and outputting one quadrature signal after in-phase synthesis. 7. An array antenna control device for controlling an array antenna composed of a plurality of antenna elements closely arranged in a predetermined arrangement shape, wherein the plurality of antenna elements are respectively received by the antenna elements of the array antenna. A conversion means for converting the received signal into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal, using a common local oscillation signal; A noise suppression filter having a function, each of which is a different pair of the plurality of antenna elements.
First data calculated by the sum of the products of the in-phase components of the quadrature signals and the products of the quadrature components in the first and second antenna elements; the in-phase component of the quadrature signal of the first antenna element; A second calculated by the product of the quadrature component of the quadrature signal of the second antenna element and the product of the quadrature component of the quadrature signal of the first antenna element and the in-phase component of the quadrature signal of the second antenna element. And the data is passed through the noise suppression filter and filtered, and then the filtered second data is divided by the filtered first data,
By calculating the arctangent value of the division result, a correction phase amount for making the quadrature signal of the first antenna element and the quadrature signal of the second antenna element in-phase is calculated,
In-phase means for shifting the quadrature signal of the second antenna element by the calculated correction phase amount to make the quadrature signal of the second antenna element in-phase with the quadrature signal of the first antenna element. A control unit that outputs a plurality of in-phase quadrature signals by repeating the processing of the in-phase unit with respect to mutually different pairs of the quadrature signals in the plurality of antenna elements; A controller for in-phase combining the quadrature signals and outputting one quadrature signal after the in-phase combination. 8. The in-phase unit performs a regression correction on the calculated correction phase amount based on the arrangement shape of the array antenna so as to return the calculated correction phase amount to a surface of the arrangement shape. Shifting the quadrature signal of the second antenna element to the quadrature signal of the first antenna element by shifting the phase of the quadrature signal of the second antenna element by the amount of the phase corrected by the regression correction. The control device for an array antenna according to claim 7, wherein: 9. An array antenna control device for controlling an array antenna composed of a plurality of antenna elements closely arranged in a predetermined arrangement shape, wherein the plurality of antenna elements are respectively received by the antenna elements of the array antenna. A conversion means for converting the received signal into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal, using a common local oscillation signal; A noise suppression filter having a function, wherein a sum of the product of the in-phase components of the quadrature signals and the sum of the products of the quadrature components of the pair of first and second antenna elements of the plurality of antenna elements is provided. 1 and the first
And the quadrature component of the quadrature signal of the second antenna element and the quadrature component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. And the second data calculated by the difference between the products is passed through the noise suppression filter and filtered, and then the filtered first data is stored in two diagonal elements. And the filtered second data and the filtered second data include (−
By multiplying the orthogonal signal of the second antenna element by a 2 × 2 transformation matrix having another two elements diagonal to each other with the third data obtained by multiplying by 1), The in-phase unit for in-phase of the quadrature signal of the second antenna element with the quadrature signal of the first antenna element, and the processing of the in-phase unit are executed for different pairs of the quadrature signals in the plurality of antenna elements. Thereafter, the in-phase step is performed for each of two different quadrature signals among the plurality of in-phase quadrature signals, and the process is repeated until the number of quadrature signals after the in-phase synthesis is reduced to one. A control device for outputting one orthogonal signal after the combination, the control device for an array antenna. 10. The control device for an array antenna is provided between the conversion means and the in-phase means, and outputs a plurality of orthogonal signals at each antenna element of the array antenna output from the conversion means. The plurality of beams are determined based on a direction of each main beam of a predetermined plurality of beams to be formed and a reception frequency of the reception signal, which are predetermined so that waves can be received within a predetermined radiation angle range. A multi-beam forming means for calculating an electric field value and outputting a plurality of orthogonal beam signals having the respective beam electric field values; provided between the converting means and the in-phase forming means; A predetermined number of orthogonal beam signals having a larger beam electric field value including the orthogonal beam signal having the maximum beam electric field value among the plurality of orthogonal beam signals The orthogonal beam signal having the maximum beam electric field value as a reference orthogonal signal of the first antenna element, and a selected orthogonal beam signal other than the orthogonal beam signal having the maximum beam electric field value 2
7. The control device for an array antenna according to claim 6, further comprising: a beam selecting unit that outputs a quadrature signal of the antenna element to the in-phase unit.