JP2568003B2 - High-sensitivity search signal detection radar system - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention has a high sensitivity for radiating a radio wave to a target object, receiving a reflected signal of the radiated radio wave, and detecting the position of the target object based on the round trip time and the directional characteristics of the antenna. The present invention relates to a search signal detection radar system.

[0002]

2. Description of the Related Art FIG. 7 shows a high-sensitivity search signal detection radar system 51 which is the basis of the invention of the present inventor. In the figure, IF is a received intermediate frequency input signal, and is a chirp FM modulated wave whose frequency changes linearly in the range of 5 MHZ, for example. This reception intermediate frequency input signal IF is determined by the pulse repetition frequency and pulse width of the transmission wave radiated from the local station. In this case,
The values are about 180 pps and 30 μsec, respectively.
Reference numeral 52 is a dynamic tracking filter (hereinafter referred to as DTF) whose center frequency can be controlled by an analog amount. This DTF is a filter whose center frequency is variable and which has a narrower band than the frequency band of an input signal. 53 is a frequency setting voltage Δf for capturing
Ac and 54 are level detectors DTF that detect the level appearing at the IF output of the DTF when the reception intermediate frequency input IF enters the DTF acquisition frequency. The level detector 54 outputs the signal Sset when detecting the level appearing at the IF output of the DTF, and outputs the signal Sreset when the level appearing at the IF output of the DTF is reduced. 55 is a sawtooth wave generator. The sawtooth wave generator 55 starts the oscillation of the sawtooth wave signal S1 when the signal Sset is input, and stops the oscillation of the sawtooth wave signal S1 when the signal Sreset is input. Reference numeral 56 is a frequency discriminator (hereinafter referred to as discriminator) that outputs a correction signal for correcting a control error, and 57 is an adder.
The adder 57 analog-adds the sawtooth wave signal S1, the correction signal output from the discriminator 56, and the capture frequency setting voltage Δfac, and outputs the calculation result to the DTF 52.

Next, the operation will be described. The reception intermediate frequency input signal IF is determined by the pulse repetition frequency and the pulse width of the transmission wave radiated from its own station. In this case, since the values are about 180 pps and 30 μsec, the sawtooth wave generator 55 outputs the reception intermediate frequency input signal IF. The sawtooth wave having the repeating period of the signal IF may be generated to control the DTF 52. However, the timing at which the received intermediate frequency input signal IF is generated is unknown because the distance between the own station and the target is unknown. Therefore, the center frequency of the DTF 52 is fixed to stand by, the level detector 54 detects the level appearing at the IF output when the frequency of the received intermediate frequency input signal IF matches the capture frequency of the DTF 52, and the signal Sset is saw-toothed. The signal is output to the generator 55 and the oscillation of the sawtooth wave signal S1 is started. Then, the sawtooth wave signal S1, the correction signal output from the discriminator 56, and the capture frequency setting voltage Δfac are analog-added by the adder 57, and the calculation result is output to the DTF 52 to control the DTF. When the level appearing at the IF output disappears, the signal Sreset is output to the sawtooth wave generator 55 to stop the oscillation of the sawtooth wave signal S1.

As described above, in this radar system, the control voltage of the DTF is fixed, and the frequency of the input signal is waited for to enter the set capture frequency.
The first sawtooth wave is generated when it comes in.
Therefore, the chirp frequency sweep rate of the input signal is K
(MHZ / Sec), the bandwidth of the filter is BW (MH
Z), the time t _det at which the input signal is detected is BW
/ K, the pulse width is 30 μsec and the sweep frequency width is 5
If the MHZ and the band of the filter are 300 KHZ, the time width t _det at which the input signal is detected is a short period of 1.8 μsec.

[0005]

Since the radar system which is the basis of the present invention is constructed as described above, the period during which the input signal is supplied to the level detector 54 is μsec.
Since the period is short, the acquisition probability at the time of initial acquisition of the received pulse is low. Furthermore, DTF
Since the center frequency of 52 is controlled in an analog manner and a wide frequency band is controlled, there is a problem that the DTF 52 is difficult to manufacture. Further, since the discriminator 56 is used for controlling the center frequency of the DTF 52, the DT is reduced from the frequency control error.
Phase rotation occurs at F52. Since this phase rotation is not included in the control loop, there is a problem that the pulse compression characteristic is deteriorated.

The present invention has been made in order to solve the above problems, has a high capture probability, is easy to manufacture a filter, and suppresses phase rotation in the filter by putting it in a control loop. It is an object of the present invention to provide a high-sensitivity search signal detection radar system capable of performing.

[0007]

A high-sensitivity search signal detection radar system according to the present invention includes a first mixer circuit.
Frequency and chirp frequency output from the voltage controlled oscillator
The modulated wave is mixed and the wavelength is different from the chirp frequency modulated wave.
Output mixed frequency, and fixed center frequency and bandwidth
Output from the first mixer circuit by the generated bandpass filter.
The second mixer circuit for limiting the mixed frequency within the bandwidth.
The mixed frequency and voltage control signal that passed through the bandpass filter.
Mix with the frequency output from the shaker and re-chirp
It is converted into a wave number modulated wave and output.

[0008]

The high-sensitivity search signal detection radar system according to the present invention
The stem has a wavelength different from that of the chirp frequency modulated wave.
Output a frequency with a control width ΔG larger than the control width ΔF.
Output voltage controlled oscillator and output from this voltage controlled oscillator
The mixed frequency and the chirp frequency modulated wave,
Outputs a mixed frequency whose wavelength is different from that of the frequency modulated wave
Output from the first mixer circuit and the first mixer circuit
A bandpass filter that limits the mixing frequency to within the bandwidth, and
Mixed frequency passed through bandpass filter and voltage controlled oscillator?
The frequency output from the
A second mixer circuit for converting into a harmonic wave and outputting the harmonic wave; and a first mixer circuit.
Pass the mixed frequency and bandpass filter output from the circuit
A phase detector that detects the phase difference from the mixed frequency
Suppresses the noise of the chirp frequency modulated wave.
Can reduce the control error and
Suppress phase rotation on filter to reduce range side lobe
Can be reduced .

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A first embodiment of the present invention will be described below with reference to FIG. FIG. 1 is a block diagram showing the configuration of a high-sensitivity search signal detection radar system 1 according to the first embodiment of the present invention. In FIG. 1, reference numeral 2 is an input terminal to which an input signal IF having a frequency of 30 MHZ + ΔF is applied. This intermediate frequency input signal IF is ΔF (5 MH in this embodiment) by the chirp frequency modulation method of the radar.
Z) range is pulse width T1 (30 μse in this embodiment).
The pulse signal is linearly frequency-modulated during the period c) and is repeated at the pulse repetition frequency PRF (180 pps in this embodiment) (see FIG. 2B). 3 is the first mixer circuit for frequency conversion, IF2 is 70M
70MHZ frequency-converted to HZ + (ΔG-ΔF) band
Band input signal, 5 is a bandpass filter circuit having a bandwidth BW (MHZ) (hereinafter referred to as BPF), 6 is a phase detector, and 7 is +
/ -Peak hold circuit, 8 is a low-pass filter circuit (hereinafter referred to as LPF), 9 is a voltage adder, 10 is a voltage controlled oscillator (hereinafter referred to as VCO), the frequency variable width is set to ΔG, and ΔG ≧ ΔF The condition of is satisfied. Δfac is a frequency setting voltage for capture, Sv is a VCO output signal, IF3 is a BPF output signal, 13 is an equalizer (hereinafter referred to as EQL), 14 is a second mixer for returning the frequency of the output signal of the EQL 13 to the 30 MHZ band. Circuit, 15
Is an output terminal, and the signal IF4 returned to the 30 MHZ band
Is output. 16 is a level detector for detecting a synchronization state, D1 is a frequency division ratio control signal, D2 is a counter initial value control signal for controlling a counter initial value, 17 is a clock signal generation circuit synchronized with a pulse repetition frequency, 18
Is a clock signal C output from the clock signal generation circuit 17.
A frequency divider that divides LK1 to output a clock signal CLK2, a latch circuit 19 that stores and outputs the counter initial value control signal D2, and a sawtooth wave generation circuit 20 that includes a counter or the like. The sawtooth wave 21 output from the sawtooth wave generation circuit 20 has a sweep rate determined by the frequency of the clock signal CLK2, and
The phase is controlled by the counter initial value control signal D2 output from the latch circuit 19.

Next, the operation of the frequency conversion function will be described. The input signal IF applied to the input terminal 2 and the VCO1
The output signal Sv of 0 is mixed in frequency by the first mixer circuit 3, converted into a signal IF2 of 70 MHZ + ΔF, and added to the BPF 5. The output signal of the BPF 5 is further equalized by the EQL 13 and added to the second mixer circuit 14. In the second mixer circuit 14, the EQ
The signal input via L13 and the output signal Sv of the VCO 10 are frequency mixed, and a signal IF4 of 30 MHZ + ΔF is obtained.
And output from the output terminal 15. That is, BP
F5 is the characteristic of frequency with fixed center frequency and bandwidth
Since it has, it follows the frequency change ΔF of the IF2 input.
The characteristics from the input of the first mixer circuit 3 to the second mixer circuit 3.
It is provided between the outputs of the output circuit 14. others
Therefore, before the BPF 5, the VCO 10 in the first mixer circuit 3 is
Output signal Sv (100MHZ + ΔG) of
By controlling ΔG, it is as if in BPF5
The circuit is as if the heart frequency had changed. This and
Frequency from the input terminal 2 to the output terminal 15 in FIG.
, First, in the first mixer circuit 3, the input signal to the input terminal 2 - (30MHZ + ΔF) output signal Sv from ·· VCO10 (100MHZ + ΔG) ·· where + Then, the input signal to BPF5 IF2 (70MHZ + ΔG- ΔF) the .... In the second mixer circuit 14, the output signal IF3 from the BPF 5 − (70 MHZ + ΔG−ΔF) , the output signal Sv from the VCO 10 (100 MHZ + ΔG) , and the output signal from the output terminal 15 (30 MHz + ΔF)・ ・ It becomes. Therefore, the formula of the input signal IF and the signal IF4
Although the frequency is exactly the same as the expression of , the band of the signal IF4 is limited by the BPF5 and the noise is suppressed. However, the frequency of the 70 MHZ band input signal IF2, 70 MHZ + (ΔG−ΔF), is within the band of BPF5 and must satisfy the conditions of | ΔG−ΔF | << BW and ΔG ≧ ΔF.

Next, the 70 MHZ band input signal IF2 input to the BPF 5 is applied to the input terminal 2 as the input signal IF.
The operation when following is described. In order to make the input frequency to the BPF 5 follow the frequency of the input signal IF, the VCO output signal Sv must be controlled. Capture is required to establish this tracking. Therefore, as shown in FIG. 2, the sweep rate K of the sawtooth wave 21 is
2 and the chirp frequency sweep rate K1 of the input signal IF are slightly shifted to be coincident with each other, and the relative frequency sweep rates of the two are reduced practically, and the time width T at which the input signal is detected.
_det = BW / a (K1-K2), it is sufficiently large compared to T _d et = BW / K in the radar system shown in FIG. Since the filter 5 that follows the input signal is a fixed frequency filter having a fixed center frequency, the filter can be manufactured easily and the EQL 13 can be inserted because it has a fixed frequency. The side rope can be improved. That is, by performing frequency conversion of the VCO output signal Sv whose frequency is variable before and after the filter as a local signal, the inside of the broken line dtf becomes equivalent to the DTF 52 shown in FIG. 7.

Next, the operation of the automatic frequency control loop will be described. In order to capture the input signal IF and enter the synchronization state, it suffices to control the frequency division ratio of the clock signal CLK2 and the counter initial value control signal D2 applied to the sawtooth wave generation circuit 20, but further improve the capture characteristic, In order to reduce the control error in the automatic frequency control loop, suppress the phase rotation of the BPF 5 and reduce the range side rope, etc., the phase detector 6 detects the phase difference between the I / O signals IF2 and IF3 of the BPF 5. , VCO10 as a control voltage, an automatic frequency control loop for adding a voltage to the sawtooth wave 21 is configured, and control is performed so that ΔG−ΔF≈0. FIG. 4 shows the characteristics obtained by combining the BPF 5 which is a detector and the phase detector 6 when performing automatic frequency control. The range of the sawtooth wave control error is set as the limit value of the +/- peak hold circuit 7, and FIG.
Set as shown in. The +/- peak hold circuit 7 excludes the frequency range in which only noise is output, which is distant from the band of the BPF 5, and focuses on the point where the detection voltage of the control pull-in point P is zero. Here, the input frequency of BPF5
When the wave number deviates from the center frequency, BPF5 is essentially
“Phase rotation when deviating from the center frequency” occurs
However, this phase rotation amount is detected by the phase detector 6 as two signals.
Easily detected as analog quantity by comparing signal
can do. This phase rotation is due to the center frequency of BPF5.
It occurs due to the difference between the number and the signal input frequency to the BPF 5. BP
Since F5 is centered around 70MHZ,
0 for ΔG-ΔF of heart frequency 70MHZ + (ΔG-ΔF)
It is necessary to That is, the output of the phase detector 6
To point P (zero voltage) in FIG.
It

Next, a second embodiment will be described with reference to FIG. FIG. 5 is a block diagram showing the configuration of a high-sensitivity search signal detection radar system according to the second embodiment of the present invention. In the first embodiment, the control voltage for performing the automatic frequency control is detected only from the phase rotation amount of the BPF 5, but in the threshold improvement type pulse compression radar device of this embodiment, the acquisition time is expanded in order to expand the acquisition range. Uses a first bandpass filter 30 having a wide bandwidth. Then, at the time of synchronization, the second band-pass filter 31 having a narrower band is used for BPF
It is added to 5 and used. Reference numeral 32 is a changeover switch for changing over between the first bandpass filter 30 and the second bandpass filter 31. At the same time when the changeover switch 32 is changed over, the set value of the +/- peak hold circuit 7 is also changed over. This embodiment further improves the capture range and accuracy.

Next, a third embodiment will be described with reference to FIG. FIG. 6 is a block diagram showing the configuration of a high-sensitivity search signal detection radar system according to the third embodiment of the present invention. Although the automatic frequency control for detecting the phase rotation of the BPF is used in the first and second embodiments described above, this embodiment has the configuration using the phase locked loop (PLL). 7
The frequency of the 0MHZ band intermediate frequency input signal IF2 is 70M.
The phase locked loop is configured so as to match the output frequency of the HZ oscillator 40. The loop characteristics are defined by the loop filter 41, a control voltage is generated based on the output of the phase detector 6, and this is added as the control voltage of the VCO 10. VC
The output signal of O10 is fed back to the mixer circuit 3 to form a loop. To determine the synchronization status of the phase locked loop,
The output of the 70 MHZ oscillator 40 is phase-shifted by the 90 ° phase shifter 42, the 70 MHZ band intermediate frequency input signal IF2 is detected by the first synchronous detector 43, and the second synchronous detector 44 compares and determines the voltage.

In any of the embodiments described above, the frequency of the sawtooth wave reset waveform changes rapidly, so it is necessary to use two circuits with the sawtooth wave timing shifted.

Further, as another embodiment, the sawtooth wave generating circuit may be omitted when a slight decrease in the trapping characteristics of the automatic frequency control loop and the phase locked loop is recognized.

[0017]

As described above, according to the present invention, the first
The frequency output from the voltage controlled oscillator by the mixer circuit of
The number and the chirp frequency modulated wave are mixed, and the chirp frequency
Outputs a mixed frequency whose wavelength is different from that of the modulated wave, and
And the bandwidth is fixed.
Limits the mixed frequency output from the mixer circuit within the bandwidth
The second mixer circuit passes the bandpass filter.
The mixed frequency and the frequency output from the voltage controlled oscillator.
When converted into chirp frequency modulated wave and output again
Both are output from the first mixer circuit by the phase detector.
And the mixed frequency passed through the bandpass filter
Since it is configured to detect the phase difference of, the radar system has a high acquisition probability, the filter can be easily manufactured, and the phase rotation in the filter can be suppressed.

[Brief description of drawings]

FIG. 1 is a block diagram showing the configuration of a high-sensitivity search signal detection radar system according to a first embodiment of the present invention.

FIG. 2 is an explanatory diagram of a sweep waveform according to the first embodiment of the present invention.

FIG. 3 is a diagram for explaining a capturing time and a synchronizing time according to the first embodiment of the present invention.

FIG. 4 is a diagram showing a phase detection characteristic in which a BPF which is a detector for automatic frequency control and a phase detector are combined according to the first embodiment of the present invention.

FIG. 5 is a block diagram showing a configuration of a high-sensitivity search signal detection radar system according to a second embodiment of the present invention.

FIG. 6 is a block diagram showing the configuration of a high sensitivity search signal detection radar system according to a third embodiment of the present invention.

FIG. 7 is a block diagram showing the configuration of a high-sensitivity search signal detection radar system which is the basis of the present invention.

[Explanation of symbols]

 5 BPF 6 phase detector 7 +/- peak hold circuit 8 LPF 10 VCO 13 EQL 16 level detector 18 frequency divider 19 latch circuit 20 sawtooth wave generation circuit

Front Page Continuation (72) Inventor Tomohiro Fujiwara 8-1-1 Tsukaguchi-cho, Amagasaki-shi Mitsubishi Electric Corporation

Claims (1)

(57) [Claims] 1. A control width ΔF having a pulse width T only.
Follows a chirp frequency modulated wave that is linearly frequency modulated
High-sensitivity search signal detection radar system
, The wavelength is different from the chirp frequency modulated wave,
The frequency having a control width ΔG larger than the control width ΔF of
Output voltage controlled oscillator and output from this voltage controlled oscillator
The mixed frequency and the chirp frequency modulated wave.
The mixed frequency with a wavelength different from that of the above chirp frequency modulated wave.
First mixer circuit for outputting wave number, center frequency and band
The bandwidth is fixed and the control width ΔG of the frequency is controlled.
The mixing output from the first mixer circuit.
A bandpass filter that limits the frequency within the above bandwidth, and
The mixed frequency passed through the bandpass filter and the voltage control
Mix with the above frequency output from the oscillator and repeat the above
Second mixer times to convert to chirp frequency modulated wave and output
And the mixed frequency output from the first mixer circuit
And the phase of the mixed frequency passed through the bandpass filter
A phase detector for detecting a difference is provided, and the above voltage controlled oscillator is provided.
Is to control the control width ΔG so that the phase difference becomes zero.
And a high-sensitivity search signal detection radar system.