JP2017504264A - Antenna system with excellent inter-sector interference mitigation - Google Patents

Abstract

An antenna system according to an example includes a radio base station that transmits an RF signal via a transmission port, an RF branching unit that receives an RF signal from the radio base station and branches the RF signal into two component signals, And at least two antennas transmitting corresponding component signals so as to produce an inferometric radiation gain pattern. When a radio base station communicates with at least one mobile terminal via a distributed multipath radio channel, an angle is spread by an angular spread of RF energy between the at least two antennas and the at least one mobile terminal. A null reduction of the inferometric radiation pattern is caused over the range.

Description

  The present disclosure relates generally to an antenna system, and more specifically to a base station antenna system that controls a roll-off rate of an azimuth (azimuth) radiation pattern in a state of being located at a sectorized base station site.

  A cellular base station site is typically configured with three sectors arranged to serve separate azimuth bearings, for example, each sector serving within a 120 ° angular range from the cell site location. Be placed. Each sector consists of an antenna whose coverage azimuth is defined by its azimuth radiation pattern. The optimum azimuth radiation pattern for the base station sector antenna is generally about 65 ° (with ± 3 dB beamwidth); this provides sufficient gain and there are multiple sites or cellular network areas in the network. This is because efficient tri-sector site tessellation is achieved for a group of sites serving the network.

  In most mobile data cellular network access technologies, including High Speed Packet Access (HSPA) and Long Term Evolution (LTE), a 1: 1 or full spectrum reuse scheme is employed to increase spectral efficiency and capacity. Yes. As given up from this aggressive spectrum reuse, it is necessary to suppress inter-sector and inter-cell interference to increase spectral efficiency. Inter-sector interference can be optimized, as opposed to antenna tilt, usually the tilt provided by the beam tilt of the electronic phase control array, which provides network optimization flexibility to handle inter-cell interference There is almost no means. Parameters indicating the amount of inter-sector interference include the antenna pattern front-to-back ratio (FTB ratio), front-to-side ratio (FTS ratio), and inter-sector power ratio (SPR). The higher the FTB and FTS, the smaller the SPR value. It is said that there is less inter-sector interference. A better measure for knowing inter-sector interference and potential throughput performance would be to calculate the resulting signal-to-interference ratio (C / I ratio) as a function of azimuth angle, thus widening the aperture as much as possible. Accordingly, it is desirable to increase the C / I ratio accordingly.

  Narrowing the 3 dB azimuth beam width to 60 ° or even 55 ° would typically improve SPR, but may affect cellular tessellation efficiency for basic service coverage, and In order to realize a narrow beam width, a wider antenna is inevitably sought, which in turn places an additional burden on the site in terms of zoning, wind load and rental. For example, the base station antenna can be of a variable azimuth beam width type, so that better inter-sector load balance can be achieved and sector-to-sector overlap can be adjusted. However, such a measure is not always appropriate in view of the conditions desired for a base station antenna to accommodate multiple arrays and thus support multiple spectral bands. Such variable beamwidth antennas can be large (such as the size is determined by the smallest beamwidth that can be achieved) for the kind of measures that require mechanical parts and active electronics, and therefore for placement and maintenance. Costs can potentially increase.

  An antenna system according to an example of the present disclosure includes at least one radio base station that transmits at least one RF signal via at least one transmission port, and at least one radio base station from the at least one radio base station. At least one RF branching means for receiving an RF signal and splitting the at least one RF signal into two component signals and separated by a distance exceeding one wavelength and connected to the at least one RF branching means And at least two antennas for transmitting corresponding component signals so as to produce an inferometric radiation gain pattern. When at least one radio base station listed above communicates with at least one mobile terminal via a distributed multipath radio channel, between the at least two antennas listed above and the at least one mobile terminal. The angular spread of the RF energy causes a reduction in the nulls of the inferometric radiation pattern described above over a range of angles.

  The teachings of the present disclosure can be understood quickly by referring to the detailed description given below in conjunction with the accompanying drawings.

In the examples of the present disclosure, the angular dispersion of the mobile radio channel is used, the intentionally generated inferometric radiation pattern is null-filled by the angular dispersion, and a strong null is retained at one side of the antenna arrangement. It is a figure which shows the azimuth radiation gain pattern at the time of a non-multipath among FIG. It is a figure which shows the azimuth radiation gain pattern example by this indication and a dispersive radio channel. FIG. 6 shows an effective azimuth radiation gain pattern produced in a dispersive radio channel. In a first example of the present disclosure, a 2T2R radio (a dual duplex radio unit having two transmit / receive (Tx / Rx) duplex ports) is used to generate an optimal azimuth in-plane inferometric radiation pattern. It is a figure which shows being connected to two polarized-wave base station antennas isolate | separated by RF through an RF splitter (branch). How the mobile radio channel angular dispersion, i.e., the angular dispersion that null-fills the intentionally generated inferometric radiation pattern and maintains a strong null on one side of the antenna arrangement, is used in the example of FIG. 3A to 3C showing azimuth radiation gain patterns during non-multipath. It is a figure which shows the azimuth radiation gain pattern in a dispersive radio channel. FIG. 3 is a diagram showing the C / I ratio resulting from placing the example of FIG. 2 on three sectors making up a tri-sector base station site as a function of azimuth angle. The 2T4R radio (a wireless unit having two Tx / Rx duplex ports and two Rx dedicated ports) provided in the second example of the present disclosure generates an optimum azimuth in-plane inferometric transmission (Tx) radiation pattern Therefore, it is a figure which shows being connected to two polarization sharing base station antennas spatially separated via RF hybrid coupler (coupler). The two 2T2R radios provided in the third example of the present disclosure are coupled to two spatially separated two polarization sharing base station antennas to generate an optimal azimuth in-plane inferometric Tx radiation pattern. It is a figure which shows that it is connected through and operates in a nearby spectrum band. The 2T2R radio included in the fourth example of the present disclosure is connected to three spatially separated polarization-sharing base station antennas via an RF splitter to generate an optimal azimuth in-plane inferometric radiation pattern. FIG. How the mobile radio channel angular dispersion, i.e., the angular dispersion that null-fills the intentionally generated inferometric radiation pattern and maintains a strong null on one side of the antenna arrangement, is used in the example of FIG. 7A and 7B showing the azimuth radiation gain pattern in the dispersive radio channel. FIG. 7 is a diagram showing the C / I ratio resulting from placing the example of FIG. 6 on three sectors constituting a tri-sector base station site as a function of azimuth angle.

  To facilitate understanding, identical reference numerals have been used, where possible, to designate identical elements that are common to the figures.

  Described in this disclosure is a base station antenna solution that controls the roll-off rate of the azimuth radiation pattern, particularly the front-to-back ratio (FSR) when deployed at the sectorized base station site. An example of the present disclosure is that two conventional directional polarized wave shared base station sector antennas having the same configuration are used, the antennas are directed in the same direction, that is, the same azimuth bearing, and are approximately odd multiples of a half wavelength in the lateral direction. This is an example in which the antennas are arranged apart from each other, and the same-polarity ports of both antennas are in phase for connection with the base station. The resulting radiation pattern is inferometric, i.e. it has a plurality of grating lobes and nulls, especially on one side of the antenna, a wide angular radiation pattern null, or ± about the boresight in the antenna horizontal plane. If the antenna arrangement is arranged on all sectors of, for example, a tri-sector base station cellular site, the FSR is lowered by the action of these nulls, and thus the inter-sector co-channel interference is generated. Decrease. Moreover, the multipath dispersion of the mobile radio channel is effectively utilized; that is, the ferromass azimuth radiation pattern generated over the target sector service area of ± 60 ° due to the angular spread of the radio channel is Null-filled by channel scattering and dispersion. In other words, the null of the inferometric radiation pattern is reduced over a range of angles and compared to the same communication scheme but without radio channel dispersion and scattering. This antenna solution is generally optimal for minimizing inter-sector interference because the angular spread in the azimuth plane of the radio channel is wider than the angular width of the inferometric null in the desired direction, but to a bearing of about ± 90 °. This is when the antenna separation distance is such that the angular dispersion is narrower than the width of the generated angle null.

  According to the present disclosure, an antenna arrangement and configuration solution suitable for a cellular base station, particularly a solution capable of reducing inter-sector interference or a solution capable of optimizing a cellular network configuration with adjustable inter-sector duplication. In the examples of the present disclosure, antennas selected by operators including multiband antennas can be used if desired, and in many cases, antenna replacement can be avoided altogether in light of existing antenna implementations. For example, examples of the present disclosure include base station sites having more than one horizontally positioned antenna position per sector, as is typical of most of the base station sites, particularly in North America. By using a plurality of antenna positions, for example, multiple spectrum bands, multiple input / multiple output (MIMO) antenna systems with spatial diversity, etc. can be supported.

  According to one embodiment of the present disclosure, a base station port that is a port that carries a transmission signal and is normally assumed to be connected to a single antenna, is not connected to it but is horizontal with more than one wavelength. Connected to an RF branching device for connection with at least two antennas that are spaced apart and have the same visibility bearing. In the examples of the present disclosure, an inferometric radiation pattern with multiple grating lobes and nulls across the sector is intentionally generated in the azimuth plane. Further, in some embodiments, the separation distance listed above is selected to be an odd multiple of a half wavelength. As a result, an inferometric null with a presence of a radiation pattern is generated at ± 90 ° with respect to the antenna azimuth visibility. As a result, the front-to-back (FTS) ratio is remarkably improved and the inter-sector interference is improved. It will be. In addition, in a system in which two spatial diversity antennas are spaced and in phase according to the present disclosure, the roll of edges can be made sharper and a wider azimuth angle (eg, a carrier to interference ratio greater than 15 dB). (C / I ratio)) can be ensured, and a high data rate (eg, a throughput gain of about 50%) can be provided.

  In the examples of the present disclosure, the fact that the mobile radio channel between the base station and the mobile terminal is a multipath channel and is therefore dispersive due to scattering. Multipath dispersion is the spread of RF energy between base stations and terminals over an angular range that is in the azimuth plane; angular dispersion or angular spread is typically several degrees in low dispersion channels to tens of degrees in high dispersion channels. It extends to values up to degrees. For example, as has been recognized, a normal macro cellular radio channel can exhibit an angular dispersion of 5-15 degrees. The effect of such dispersion is depicted in the examples of FIGS. 1A-1C and is described in more detail later.

  An example of the present disclosure is an example in which a base station site has an array of two or more antennas, and these antennas are arranged in the vertical direction (for example, at least two per sector). In such an example, an inferometric radiation pattern will occur in the elevation plane. The angular spread in the elevation plane due to multipath radio channels can vary as a function of the elevation angle (elevation angle), and the high elevation angle where the mobile device is close to the site and thus far from the horizontal plane. When there is a wide angular spread appears. At low elevation angles close to the horizon, the farther the mobile terminal is, the narrower the angular spread in the elevation plane will be. For example, the scattering volume appears to be a constant volume in the vicinity of a mobile terminal (for example, a house, a street, a building, etc.) that is likely to contribute to multipath extending over a certain angular range in the elevation plane, and the mobile terminal moves away from the base station site. As it gets smaller. An antenna arranged vertically in accordance with the present disclosure can be used to increase the upper part of the main beam roll-off rate that can be effectively used to reduce energy towards the horizon and thus inter-site interference.

  FIG. 1A is a diagram illustrating a mobile terminal (500) that is in approximately azimuth visibility as seen from a conventional base station sector, and a polar plot (700) showing antenna azimuth radiation or gain patterns. The communication link (uplink or downlink or both) is indicated by a dashed line (600) between the base station antenna and the mobile terminal.

FIG. 1B is a diagram illustrating a similar mobile terminal (500) communicating with a base station antenna arrangement (eg, the arrangement of FIG. 2) according to the present disclosure, and the azimuth radiation pattern (701) exhibited by the base station antenna arrangement is It has an inferometric pattern with multiple grating lobes and nulls. Scatterers (501 _{1 to} 501 _m ) in the wireless channel, usually close to and surrounding the mobile terminal (500) in a macrocellular wireless environment, are affected by a plurality of main propagation paths (601 ₁ to 601 _n ), that is, multipath propagation occurs. This multipath propagation will spread over a range of angles in the azimuth plane and thus exhibit angular dispersion or angular spread. Null fill will occur if the angular spread of the radio channel antagonizes or exceeds the grating lobe vs. grating lobe (or grating null vs. null) angular width (in other words, null in the inferometric radiation pattern). However, as far as the mobile terminal (500) is concerned, the appearance of the base station antenna pattern is, for example, a 65 ° beam over a certain angular range and under the same communication scheme without radio channel dispersion and scattering. The width is basically unchanged.

  FIG. 1C shows the azimuth radiation gain pattern of the base station for the mobile terminal (500) vs. base station link (602) in a distributed wireless channel. This base station antenna gain pattern (702) looks similar and similar to the antenna gain pattern (700) shown in FIG. 1A. For example, it exhibits some gain ripple across its opening. However, inferometric nulls occurring at ± 90 ° relative to visibility are much wider than inferometric nulls occurring within ± 60-65 ° sector aperture / beam width; Due to the fact that the lobe-to-lobe angular distance is a function of the cosine of the azimuth angle. The ideal is that the null at ± 90 ° is wider than the angular spread of the channel and can advantageously reduce inter-sector interference. Nonetheless, even with a wide angular spread, the presence of nulls at ± 90 ° helps to improve inter-sector interference compared to a single antenna case. In addition, the gain pattern (702) depicts a null fill in ± 60 ° sectors; this is in contrast to the gain pattern (701) of FIG. 1B.

  To further assist in understanding the present disclosure, a first system example (100) is shown in FIG. In FIG. 2, two dual cross-polarized antennas (170, 270) having an azimuth beam width of about 65 ° are arranged and configured in accordance with the present disclosure. For example, in the example of FIG. 2, an LTE frequency division duplex (FDD) service can be provided in the 700 MHz band, for example. Since the LTE base station radio (10) (eg “radio base station”) in this example is a conventional dual duplex radio unit with two transmit / receive (Tx / Rx) duplex ports (110, 210). 2T2R radio. The first Tx / Rx signal (110) from the 2T2R radio (10) passes through the connection with the common mode port (120) of the first 180 ° hybrid coupler (130) where the common mode branch appears at its ports (140, 141). Similarly, the second Tx / Rx signal (210) from the 2T2R radio (10) is connected to the in-phase port (220) of the second 180 ° hybrid coupler (230); An in-phase branch appears at ports (240, 241) of the same coupler. As shown in FIG. 2, the signals A and B on the ports of the 180 ° hybrid coupler (130, 230) are vector-synthesized in the same phase at the ports (140, 240) labeled A + B, and are labeled A-B. The vectors are synthesized in reverse phase at the ports (141, 241). It should be noted, however, that the signal is only connected to the port (120, 220) as the in-phase signal “A”. In this example, the anti-phase port (for signal “B”) is not used. It should also be noted that although the 180 ° hybrid coupler (130, 230) is shown in the example of FIG. 2, in other further and alternative embodiments, alternative or in addition RF splitters may be used. A 90 ° hybrid coupler or the like may be used. The branches (140, 141) of the first signal pair are connected to + 45 ° polarization ports (160, 260) of the two cross-polarized antennas (170, 270), and the second pair of branches (141). , 241) are connected to the -45 ° polarization ports (161, 261) of the two cross polarization antennas (170, 270).

  The antenna separation distance d in this disclosure is such that the resulting Tx signal radiation pattern grating lobe-to-lobe distance at a viewing beamwidth of 65 ° across is less than the angular spread of the radio channel and + 90 ° relative to the visibility. The nominal value should be an odd multiple of half-wavelength (eg, d≈ (n + 0.5) λ) so that strong nulls occur in the bearing and the −90 ° bearing. For example, if the angular spread of the radio channel is narrow, it may be desirable to increase the value of d so that the resulting lobe-to-lobe distance is small. Conversely, if the angular spread of the radio channel is wide, a larger lobe-to-lobe distance can be accepted, so a smaller value of d can be used. A distance d different from an odd multiple of the half-wavelength can be chosen to optimize C / I performance across the sector aperture; this will necessarily depend on the specific antenna azimuth pattern. Although not essential, the variable RF phase shifter (150, 151) is inserted in front of one connection port of the polarization sharing antenna (for example, in FIG. 2, the connection port (160, 161) of the polarization sharing antenna (170) is inserted). By adjusting the relative branch phase, it is possible to compensate for various phase variations brought in by differences in the wiring length in each branch, or 90 ° hybrid couplers can be used as RF branching means. If so, the phase can be compensated, and thus the Tx radiation pattern can be optimized to minimize inter-sector interference. Alternatively, if a wider sector overlap is desired, the sector-to-sector overlap may be changed using phase shifters (150, 151); by adding 180 ° phase delay using three phase shifters, ± 90 It is possible to generate side lobes instead of null at the azimuth angle of °.

  Since cellular data networks are generally limited to downlink interference, the example of FIG. 2 optimizes inter-sector interference for Tx signals. However, in an FDD system, the Rx signal will operate at different frequency ranges in the same spectral band, so the need for the Rx signal should also be considered. For example, when the Tx frequency and the Rx frequency are relatively close to each other, the antenna separation distance d may be set according to a frequency located in the middle of those frequencies. In another example, when the duplex distance between the Tx frequency and the Rx frequency can be large, a separation distance d that satisfies the condition that an odd multiple of a half wavelength is the same for both the Tx frequency and the Rx frequency is calculated. Selected and / or utilized. It should be noted that, in some embodiments, the example of FIG. 2 also allows polarization sharing so that more or less phase delay is introduced at the Tx frequency due to its delay / phase characteristics compared to that at the Rx frequency. An all-pass filter may be incorporated on the connection means with one of the antennas. In yet another embodiment, the hybrid coupler (130, 230) is removed and replaced with a group of RF components that separate the Tx / Rx line from the 2T2R radio (10) into two Tx and Rx line element pairs. The Tx and Rx signals are superimposed again after applying branching and phase shifting individually. In yet another embodiment, RF branching is performed at baseband, eg, prior to power amplification in a base station radio device.

  3A-3C show preferred results according to the example of FIG. FIG. 3A is a graph (320) showing the antenna radiation pattern and the inferometric pattern that results when the two antennas used are arranged at 4.5λ intervals and there is no channel dispersion. The first axis (323) represents the azimuth angle in degrees. The second axis (324) represents the visibility-based antenna gain in dB. In particular, the dashed line (321) in FIG. 3A represents the gain or radiation pattern or reference of a commercially available double cross-polarized 700 MHz band antenna that constitutes a + 45 ° polarization array and has an electrical tilt at 740 MHz of 2 °. Is expressed as a function of the azimuth angle. The solid line (322) shows the infero that results from the configuration described in the first example of FIG. 2 when the above antennas are arranged and separated at 4.5λ intervals and the radio channel has no dispersion. The metric radiation pattern is represented as a function of azimuth angle.

  FIG. 3B is a graph (330) showing the antenna radiation pattern, as well as the inferometric pattern that results when the two antennas used are spaced 4.5λ apart and the channel dispersion is 10 °. The first axis (333) represents the azimuth angle in degrees. The second axis (334) represents the visibility-based antenna gain in dB. In particular, in the graph (330), the reference radiation pattern (broken line (331)) is the same as that in FIG. 3A, but the solid line (332) is approximately 10 ° to the radio channel with the permission of the first example of FIG. Represents the radiation pattern produced when there is a variance (α). The angular dispersion (α) in the present application is an angular range in which 90% of the multipath energy is included. As can be clearly seen from FIG. 3B, although there are some ripples in the azimuth pattern, the roll-off rate of the azimuth pattern is greatly improved in comparison with a single antenna within a range exceeding ± 60 ° sector bearing. .

  FIG. 3C is a graph (340) showing the C / I response (where “I” is inter-sector interference (ISI)) when two antennas are used at 4.5λ intervals and the channel dispersion is 10 °. is there. The first axis (343) represents the azimuth angle in degrees. The second axis (344) represents C / I in dB. A solid line (342) in the graph (340) indicates C produced when the three sectors constituting the tri-sector site are arranged at intervals of 120 degrees and the antenna configuration is described and described with reference to FIG. / I response as a function of azimuth angle. A broken line (341) in FIG. 3C depicts C / I that is generated when one conventional antenna is used as a reference. FIG. 3C shows a significant C / I improvement over a fairly wide range of azimuth bearings, which can result in excellent spectral efficiency.

  FIG. 4 shows a second system example (200) according to the present disclosure. In FIG. 4, two double cross-polarized antennas (170, 270) having an azimuth beam width of about 65 ° are arranged and configured in accordance with the present disclosure. For example, in the example of FIG. 4, the LTE-FDD service can be provided in the 700 MHz band, for example. The LTE base station radio (20) (eg, “radio base station”) in this example is “2T4R” having two Tx / Rx duplex ports (110, 210) and two Rx dedicated ports (111, 211). Radio. The first Tx / Rx signal (110) from this 2T4R radio (20) is connected to the common-mode port (120) of the first 180 ° hybrid coupler (130) that provides the common-mode branch, thereby providing two common-mode branches or component signals. Similarly, the second Tx / Rx signal (210) from the 2T4R radio (20) is connected to the common mode port (220) of the second 180 ° hybrid coupler (230) and A branch or component signal “A” appears at the port (240, 241) of the coupler. It should be noted that although the 180 ° hybrid coupler (130, 230) is shown in the example of FIG. 2, in other further and alternative embodiments, an alternative or in addition RF splitter, 90 ° A hybrid coupler or the like may be employed. The branches (140, 141) of the first signal pair are connected to + 45 ° polarization ports (160, 260) of the two cross-polarized antennas (170, 270), and the second pair of branches (141). , 241) is connected to the -45 ° polarization port of the two cross-polarized antennas (170, 270). The first Rx dedicated signal (111) from the 2T4R radio is connected to the second (reverse phase) port (121) of the first 180 ° hybrid coupler (130), and that port is the same coupler for the Rx dedicated signal. Are the opposite phase vector sums of the component signals labeled “B” and “· B” at the ports (140, 141). Similarly, the second Rx dedicated signal (211) from the 2T4R radio is connected to the second (reverse phase) port (221) of the second 180 ° hybrid coupler (230), which port is connected to the Rx dedicated signal. , The phase vector sum of the component signals indicated by “B” and “· B” at the ports (240, 241) of the coupler. Therefore, an in-phase signal, that is, an A + B component signal appears at the ports (140, 240), while an anti-phase signal, that is, an A-B component signal appears at the ports (141, 241).

  The separation distance d of the antenna in this disclosure is + 90 ° so that the grating lobe-to-lobe distance of the resulting Tx signal radiation pattern at a visibility beamwidth of 65 ° across is less than the angular spread of the radio channel and to the visibility. In order to generate strong nulls in the bearings of −90 ° and −90 °, the spacing should be an odd multiple of a half wavelength (eg, d≈ (n + 0.5) λ). For example, if the angular spread of the radio channel is narrow, it may be desirable to increase the value of d so that the resulting lobe-to-lobe distance is small. Conversely, if the angular spread of the radio channel is wide, a larger lobe-to-lobe distance can be accepted, so a smaller value of d can be used. A distance d different from an odd multiple of the half-wavelength can be chosen to optimize C / I performance across the sector aperture; this will necessarily depend on the specific antenna azimuth pattern. Although not essential, the variable RF phase shifter (150, 151) is inserted in front of one connection port of the polarization sharing antenna (in FIG. 4, the connection ports (160, 161) of the polarization sharing antenna (170)). By adjusting the relative branch phase, it is possible to compensate for various phase variations brought in by differences in the wiring length in each branch, or 90 ° hybrid couplers can be used as RF branching means. If so, the phase can be compensated, and thus the Tx radiation pattern can be optimized to minimize inter-sector interference. Alternatively, if a wide sector overlap is desired, the phase shifters (150, 151) may be used to vary the sector overlap; by using these phase shifters and adding a 180 ° phase delay, ± 90 ° It is also possible to generate a side lobe instead of null at the azimuth angle.

  Since a 2T4R radio, such as that shown in FIG. 4 (20), would typically require connection with two shared antenna arrays (ie, four antenna ports), a second system example (200 (E.g., that shown in FIG. 4), inter-sector interference can be improved without adding any additional antennas or using any additional antenna positions. It should be noted that the 2T4R radio (20) employs 4-branch Rx synthesis in the baseband, such as maximum ratio synthesis (MRC) or interference cancellation synthesis (IRC). Therefore, since all Rx signals are vector-synthesized in the optimum manner in the baseband, it is necessary to work the separation distance so as to cover the Rx frequency as described based on the system example (100) of FIG. Not necessarily. It should be noted that in some embodiments, the example system (200) of FIG. 4 can be modified in the same or similar manner as described above in light of the example system (100) of FIG. 2; Replace hybrid couplers with RF components that use an all-pass filter on the connection to one of the polarization-sharing antennas so that more or less phase delay is introduced by delay / phase characteristics at the Tx frequency. Etc. are possible.

FIG. 5 shows a third system example (300) according to the present disclosure. In FIG. 5, two dual cross-polarized antennas (170, 270) having an azimuth beam width of about 65 ° are arranged and configured in accordance with the present disclosure. For example, in the example of FIG. 5, the LTE-FDD service can be provided in the 700 MHz band (f ₁ ), for example, and the HSPA-FDD service can be provided in the 850 MHz band (f ₂ ), for example. The LTE and HSPA base station radios (10, 30) (eg, “radio base stations”) are each 2T2R radios, each having two Tx / Rx duplex ports. The dual-polarized antenna (170, 270) has sufficient bandwidth to support the 700 MHz spectral band and the 850 MHz spectral band. The first Tx / Rx signal (110) from the LTE-2T2R radio (10) is a first 180 ° hybrid coupler (130) in which two in-phase branches or component signals “A” appear at its ports (140, 141). The second Tx / Rx signal (210) from the LTE-2T2R radio (10) is also connected to the second 180 ° hybrid coupler (230). ) And an in-phase branch or component signal “A” appears at ports (240, 241) of the same coupler. It should be noted that although the 180 ° hybrid coupler (130, 230) is shown in the example of FIG. 5, in other further and alternative embodiments, an alternative or additional RF splitter, 90 ° A hybrid coupler or the like may be employed. The branches (140, 141) of the first signal pair are connected to + 45 ° polarization ports (160, 260) of the two cross-polarized antennas (170, 270), and the second pair of branches (141). , 241) are connected to the -45 ° polarization ports (161, 261) of the two cross polarization antennas (170, 270). The first Tx / Rx signal (310) from the HSPA-2T2R radio (30) is connected to the second (reverse phase) port (121) of the first 180 ° hybrid coupler (130). The anti-phase component signals indicated by “B” and “−B” are generated at the ports (140, 141). Similarly, the second Tx / Rx signal from the HSPA-2T2R radio (30) is connected to the second (reverse phase) port (221) of the second 180 ° hybrid coupler (230), which also has Negative phase component signals indicated by “B” and “−B” are generated at the ports (240, 241). That is, an in-phase signal, that is, an A + B component signal appears at the ports (140, 240), while an anti-phase signal, that is, an AB component signal appears at the ports (141, 241).

The antenna separation distance d in this disclosure is such that the resulting Tx signal radiation pattern grating lobe-to-lobe distance at a viewing beamwidth of 65 ° across is less than the angular spread of the radio channel and + 90 ° relative to the visibility. The nominal value should be an odd multiple of half a wavelength for 700 MHz LTE service (eg, d≈ (n + 0.5) λ ₁ ) so that strong nulls are generated in the bearing and the −90 ° bearing. In addition, if minimization of inter-sector interference is desired also for the HSPA service, the distance d should be selected so that it is approximately an integral multiple of the wavelength (for example, d≈mλ ₂ ) for the Tx signal in the 850 MHz band. is there. In this case, an integer multiple of the wavelength (not an odd multiple of half wavelength) is desired because the 850 MHz band signal is branched through the second (reverse phase) port of the 180 ° hybrid coupler, and thus the resulting 850 MHz branch. This is because the signal is 180 ° out of phase. For example, if the angular spread of the radio channel is narrow, it may be desirable to increase the value of d so that the resulting lobe-to-lobe distance is small. Conversely, if the angular spread of the radio channel is wide, a larger lobe-to-lobe distance can be accepted, so a smaller value of d can be used. Although not required, by inserting a variable RF phase shifter (150, 151) into the antenna signal path (connected to the dual polarization antenna (170) in FIG. 5) and thereby adjusting the phase, The Tx radiation pattern can be optimized to minimize inter-sector interference, or alternatively, can cause changes in sector overlap if desired.

  Two 2T2R radios, such as that shown in FIG. 5 (10, 30), will typically require connection with two shared antenna arrays (ie, four antenna ports). Therefore, according to the third example of the present disclosure (for example, that shown in FIG. 5), improving inter-sector interference without adding any additional antennas or using any additional antenna positions. Can do. The example of FIG. 5 can be variously configured and modified so that two or more spectrum bands can be supported on the broadband antenna. This includes performing RF branching, RF synthesis, etc. using 90 ° hybrid couplers, duplexing synthesis filters, diplexing synthesis filters, and the like.

FIG. 6 shows a fourth system example (400) according to the present disclosure. In FIG. 6, three double cross-polarized antennas (170, 270, 370) having an azimuth beam width of about 65 ° are arranged and configured in accordance with the present disclosure. For example, in the example of FIG. 6, an LTE frequency division duplex (FDD) service can be provided in the 700 MHz band, for example. The LTE base station radio in this example is a conventional 2T2R duplex radio unit (10) (eg, “radio base station”) having two Tx / Rx duplex ports (110, 210). The first Tx / Rx signal (110) from the 2T2R radio (10) is branched into three component signals by the action of the first RF splitter (180) with the in-phase branch; similarly, from the 2T2R radio (10) The second Tx / Rx signal (210) is branched into three in-phase component signals by the action of the second RF splitter (380), thereby forming two groups of three in-phase component signal branches. . The split ratio (branch ratio) in the RF splitter (180, 380) can be set to different values as indicated by a ₁ , a ₂ , and a ₃ on each RF splitter. The first group of signals is connected to the + 45 ° polarization ports (160, 260, 360) of the three cross polarization antennas (170, 270, 370), and the second group of signals is the three. Are connected to the −45 ° polarization ports (161, 261, 361) of the cross polarization antennas (170, 270, 370). Although not essential, the variable RF phase shifters (150, 151) are arranged in front of the connection ports (160, 161) of the first dual-polarized antenna (170), and the variable RF phase shifters (350, 351) are third. Various phase variations introduced due to the difference in wiring length in each branch are compensated by inserting in the front stage from the connection port (360, 361) of the polarization sharing antenna (370) and thereby adjusting the relative signal branch phase. be able to. RF splitter (180, 380) branching ratio, first (170) -second (270) cross-polarized antenna separation distance (d ₁ ), second (270) -third (370) cross-polarized antenna The separation distance (d ₂ ), as well as the phase shifters (150, 151, 350, 351), if any, are both base station sector Tx and / or Rx signal radiation pattern grating lobes vs. lobes with a span of ± 60 °. Since the distance can be changed to be smaller than the angular spread of the radio channel, inter-sector interference can be minimized or adjusted accordingly. For example, if the angular spread of the radio channel is narrow, it may be desirable to increase the value of d so that the resulting lobe-to-lobe distance is small. Conversely, if the angular spread of the radio channel is wide, a larger lobe-to-lobe distance can be accepted, so a smaller value of d can be used. Three spatially separated antenna positions and dispersive radio channels are used, minimizing inter-sector power ratio (SPR), minimizing inter-sector interference (ISI) and design freedom The degree can be increased.

7A and 7B show favorable results according to the example of FIG. Shown in FIG. 7A is a graph (710) showing the antenna radiation pattern and the inferometric pattern resulting when the three antennas used are arranged at 4.66λ intervals and the channel dispersion is 10 °. The first axis (713) represents the azimuth angle in degrees. The second axis (714) represents the visibility-based antenna gain in dB. In particular, the dashed line (711) in FIG. 7A represents the gain or radiation pattern or reference of a commercially available double cross-polarized 700 MHz band antenna that constitutes a + 45 ° polarization array and has an electrical tilt at 740 MHz of 2 °. Is expressed as a function of the azimuth angle. In FIG. 7A, a solid line (712) indicates that the first and second antennas are arranged and separated by a 4.66λ interval (ie, d ₁ ), and the second and third antennas are arranged and separated by a 4.66λ interval (ie, d ₂ ). And when the wireless channel has 10 ° azimuth in-plane dispersion (α), the resulting radiation pattern resulting from the configuration described in the fourth example (eg, system (400) of FIG. 6) is It is expressed as a function of corners. The RF splitter (180, 380) has unequal branch loads of a ₁ = 0.2, a ₂ = 0.6, and a ₃ = 0.2, and the RF phase shifter (150, 151, 350). , 351) does not introduce a new phase delay. As can be clearly seen from the graph (720) in FIG. 7B, the azimuth pattern roll-off rate is higher than that of a single antenna in the range exceeding ± 60 ° sector bearing, although there are some residual ripples in the azimuth pattern. Greatly improved. Specifically, the graph (720) shows the C / I response when three antennas are used at intervals of 4.66λ and the channel dispersion is 10 ° (where “I” is inter-sector interference (ISI)). Is drawn. The first axis (723) represents the azimuth angle in degrees. The second axis (724) represents C / I in dB. A solid line (722) in FIG. 7B indicates C / I caused when the three sectors constituting the tri-sector site are arranged at intervals of 120 degrees and the antenna configuration is described and described with reference to FIG. The response (where I is inter-sector interference (ISI)) is a function of azimuth angle. A broken line (721) in FIG. 7B depicts C / I that is caused when one conventional antenna is used as a reference. FIG. 7B shows a significant C / I improvement over a fairly wide range of azimuth bearings, which can result in excellent spectral efficiency.

  Although various examples have been described according to one or more aspects of the present disclosure, other examples and further examples according to one or more aspects of the present disclosure are described in the claims below. It can be devised without departing from the technical scope of the present disclosure and the equivalent scope thereof.

Claims (24)

At least one radio base station transmitting at least one radio frequency signal via at least one transmission port;
At least one radio frequency branching means for receiving the at least one radio frequency signal from the at least one radio base station and branching the at least one radio frequency signal into two component signals;
At least two antennas separated by a distance of more than one wavelength and connected to the at least one radio frequency branching means for transmitting corresponding component signals to produce an inferometric radiation gain pattern;
When the at least one radio base station communicates with at least one mobile terminal via a distributed multipath radio channel, a radio frequency between the at least two antennas and the at least one mobile terminal. An antenna system in which angular spread of energy causes null reduction of the inferometric radiation pattern over a range of angles.   The antenna system according to claim 1, wherein the at least one radio base station further receives at least a second radio frequency signal.   2. The antenna system according to claim 1, wherein the at least two antennas are an array of a plurality of antenna elements, and the antenna elements are arranged so as to exhibit directivity and a specific radiation pattern. .   2. The antenna system according to claim 1, wherein the at least two antennas are arranged in a horizontal geometric plane so that an inferometric gain pattern is generated in the azimuth radiation plane.   The antenna system according to claim 1, wherein the at least one radio base station has at least two transmission ports, and the at least two antennas include at least two polarization sharing antennas.   2. The antenna system according to claim 1, wherein the at least two antennas include at least two polarization sharing antennas, and the at least one radio base station includes two transmission / reception duplex ports and two antennas. A receive-only port, wherein the at least one radio frequency branching means includes two hybrid combiners, and the two transmit / receive duplex ports are connected to corresponding in-phase ports of the two hybrid combiners An antenna system in which the two reception-only ports are connected to corresponding antiphase ports of the two hybrid combiners.   2. The antenna system according to claim 1, wherein a distance between the at least two antennas is an odd multiple of a half wavelength, and an azimuth radiation pattern null is generated in the azimuth plane of the at least two antennas. An antenna system that is selected such that the azimuth radiation pattern null includes at least two nulls in a ± 90 ° bearing in the azimuth plane of the at least two antennas.   2. The antenna system according to claim 1, wherein the distance between the antennas is an integral multiple of the wavelength, and the distance is selected so that an azimuth radiation pattern lobe occurs in the azimuth plane of the at least two antennas. The antenna system comprising at least two lobes in which the azimuth radiation pattern lobes are in a ± 90 ° bearing in the azimuth plane of the at least two antennas.   2. The antenna system according to claim 1, wherein the at least two antennas include at least two polarization sharing antennas, and the at least one radio base station has two transmission / reception duplex ports. A first radio base station operating in the first spectrum band; and a second radio base station operating in the second spectrum band having two transmission / reception duplex ports, the at least one radio frequency branch The means comprises two hybrid combiners, the two transmit / receive duplex ports of the first radio base station are connected to corresponding in-phase ports of the two hybrid combiners, and the second radio base station The two transmission / reception duplex ports are connected to the corresponding anti-phase ports of the two hybrid combiners.   10. The antenna system according to claim 9, wherein the distance between the at least two antennas is an odd number of a half wavelength according to the first spectrum band and an integer multiple of the wavelength according to the second spectrum band. Are further selected to produce azimuth radiation pattern nulls in the azimuth planes of the at least two antennas, the azimuth radiation pattern nulls for both the first spectral band and the second spectral band. An antenna system comprising at least two nulls in a ± 90 ° bearing in the azimuth plane of the single antenna.   2. The antenna system according to claim 1, wherein the at least one radio base station has two transmission ports, and the at least two antennas include three polarization sharing antennas. Is connected to the first three-way radio frequency splitter so that the first three-component signal group is generated, and the second port of the radio base station is the second three-component signal group so that the second three-component signal group is generated. Is connected to the way radio frequency splitter, and the first component signal in the first three component signal group and the first component signal in the second three component signal group are the first polarization signals of the three polarization sharing antennas. The second component signal in the first three-component signal group and the second component signal in the second three-component signal group are connected to the corresponding polarization ports of the wave-sharing antenna, and the three polarization-sharing antennas Out of the 2nd polarization antenna The third component signal in the first three-component signal group and the third component signal in the second three-component signal group are connected to the corresponding polarization port, and the third component signal in the three polarization-shared antennas is the third component signal. An antenna system connected to the corresponding polarization port of a dual-polarized antenna.   12. The antenna system according to claim 11, wherein a separation distance between the first polarization sharing antenna and the second polarization sharing antenna, a separation distance between the second polarization sharing antenna and the third polarization sharing antenna, and the first three-way radio. The split ratio of the frequency splitter, the split ratio of the second three-way radio frequency splitter, the first component signal in the first three-component signal group, the phase delay applied to the second and third component signals, and the second third The phase delay applied to the first component signal, the second component signal, and the third component signal in the component signal group is selected so that a null is generated in the azimuth plane of the three polarization sharing antennas, An antenna system in which a null in the azimuth plane of the three dual-polarized antennas includes at least two nulls in a ± 90 ° bearing in the azimuth plane of the three dual-polarized antennas. Transmitting at least one radio frequency signal via at least one transmission port of at least one radio base station;
Receiving the at least one radio frequency signal from the at least one radio base station via at least one radio frequency branching means;
Branching the at least one radio frequency signal into two component signals via the at least one radio frequency branching means;
Transmitting corresponding component signals via at least two antennas separated by a distance of more than one wavelength and connected to said at least one radio frequency branching means so as to produce an inferometric radiation gain pattern When,
When the at least one radio base station communicates with at least one mobile terminal via a distributed multipath radio channel, radio between the at least two antennas and the at least one mobile terminal A method in which angular spread of frequency energy causes null reduction of the inferometric radiation pattern over a range of angles. 14. The method of claim 13, further comprising:
Receiving at least a second radio frequency signal via the at least two antennas.   14. The method according to claim 13, wherein the at least two antennas are an array of a plurality of antenna elements, and the antenna elements are arranged to exhibit directivity and a specific radiation pattern.   14. The method of claim 13, wherein the at least two antennas are arranged in a horizontal geometric plane so that an inferometric gain pattern occurs in the azimuth radiation plane.   14. The method of claim 13, wherein the at least one radio base station has at least two transmission ports and the at least two antennas include at least two polarization sharing antennas.   14. The method of claim 13, wherein the at least two antennas include at least two dual polarization antennas, and the at least one radio base station has two transmit / receive duplex ports and two receive. The at least one radio frequency branching means includes two hybrid combiners, and the two transmit / receive duplex ports are connected to corresponding in-phase ports of the two hybrid combiners. And the two reception-only ports are connected to the corresponding antiphase ports of the two hybrid combiners.   14. The method of claim 13, wherein the distance between the at least two antennas is an odd multiple of a half wavelength, such that the azimuth radiation pattern null occurs in the azimuth plane of the at least two antennas. A method wherein the azimuth radiation pattern nulls are selected and include at least two nulls in a ± 90 ° bearing in the azimuth plane of the at least two antennas.   The method of claim 13, wherein the distance between the antennas is an integer multiple of the wavelength, and the distance is selected to produce azimuth radiation pattern lobes in the azimuth plane of the at least two antennas, The method wherein the azimuth radiation pattern lobes include at least two lobes in a ± 90 ° bearing in the azimuth plane of the at least two antennas.   14. The method according to claim 13, wherein the at least two antennas include at least two dual-polarization antennas, and the at least one radio base station has two transmission / reception duplex ports. A first radio base station operating in one spectrum band; and a second radio base station operating in a second spectrum band having two transmission / reception duplex ports, and the at least one radio frequency branching means Includes two hybrid synthesizers, two transmission / reception duplex ports of the first radio base station are connected to corresponding in-phase ports of the two hybrid synthesizers, and the second radio base station A method in which two transmission / reception duplex ports are connected to corresponding anti-phase ports of the two hybrid combiners.   14. The method of claim 13, wherein the distance between the at least two antennas is an odd number of half wavelengths associated with the first spectral band and an integer multiple of the wavelength associated with the second spectral band. Further, azimuth radiation pattern nulls are selected to occur in the azimuth plane of the at least two antennas, and the azimuth radiation pattern nulls are related to both the first spectral band and the second spectral band. Including at least two nulls in a ± 90 ° bearing in the azimuth plane of the antenna.   14. The method of claim 13, wherein the at least one radio base station has two transmission ports, the at least two antennas include three polarization sharing antennas, and The first port is connected to the first three-way radio frequency splitter so that the first three-component signal group is generated, and the second port of the radio base station is the second three-way signal so that the second three-component signal group is generated. The first component signal in the first three-component signal group and the first component signal in the second three-component signal group are connected to the radio frequency splitter, and the first polarization signal among the three polarization sharing antennas The second component signal in the first three-component signal group and the second component signal in the second three-component signal group are connected to the corresponding polarization ports of the common antenna. Of which the second polarized antenna The third component signal in the first three-component signal group and the third component signal in the second three-component signal group are connected to the wave port, and the third polarization signal is shared among the three polarization-sharing antennas. The method connected to the corresponding polarization port of the antenna.   14. The method according to claim 13, wherein a separation distance between the first polarization sharing antenna and the second polarization sharing antenna, a separation distance between the second polarization sharing antenna and the third polarization sharing antenna, and the first three-way radio frequency. Split ratio of splitter, split ratio of second three-way radio frequency splitter, first component signal in first first component signal group, phase delay applied to second component signal and third component signal, and second third component The phase delay applied to the first component signal, the second component signal, and the third component signal in the signal group is selected so that a null is generated in the azimuth plane of the three polarization sharing antennas. A method in which the nulls in the azimuth plane of each of the dual polarization antennas include at least two nulls in a ± 90 ° bearing in the azimuth plane of the three dual polarization antennas.