JP2017198474A - Pulse radar device and component of the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a short range pulse radar device that allows a simple configuration to measure a range-finding object at a shortest range up to 0 m as much as possible, as suppressing an influence of jamming waves due to transmission.SOLUTION: A local signal wave of a high frequency is configured to be distributed to a transmission side and a reception side by a rat-race type division wave modulation circuit 11. An internal reflection wave when a transmission signal wave is radiated from a transmission antenna 40 is configured to be attenuated by the branch wave modulation circuit 11. A phase-shift circuit 12 is configured to adjust a phase of the local signal wave distributed to the reception side to the same phase as a phase of a wraparound wave to a reception antenna 50 of the transmission antenna 40, switch the post-adjusted phase to 0 degree/90 degrees, and input the switched phase to a mixer 13. The local signal wave having the phase adjusted and a reception signal from the reception antenna 50 is configured to implement homodyne detection by the mixer 13, and thereby a jamming wave due to the internal reflection wave or wraparound wave is attenuated. Accordingly, a simple configuration makes it possible to highly accurately measure a shortest range up to 0 m as much as possible without adding a complex circuit although a radio wave radar is a pulse-type radio wave radar.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a pulse radar apparatus capable of measuring a close range of 0 m or more and parts thereof.

  As a radar device for measuring the distance to a distance measuring object, a pulse radar that is resistant to interference and can take high resolution is frequently used. In this type of pulse radar, an interference wave due to a transmission signal wave is generated at a close distance, and when this interference wave and a reflected signal wave from a distance measurement object overlap, it becomes difficult to separate signal components. The interference wave includes a sneak wave of the transmission signal wave to the reception antenna and an internal reflection wave generated in the pulse radar device. A sneak wave is one in which a transmission signal wave affects reception through a space from a transmission antenna to a reception antenna (leakage of a transmission signal wave). The internal reflected wave is generated by design mismatch or characteristic variation of the high-frequency part, which is transmitted between components of the high-frequency part or in the transmission line, and affects the receiving system circuit.

  As a technique for removing such an interference wave, the radar apparatus disclosed in Patent Document 1 blocks the interference wave by performing a mask process on a short-range received signal in the baseband portion. Further, in the radar device disclosed in Patent Document 2, the time axis is extended by sampling or integrating at a period slower than the pulse transmission period, thereby delaying the signal component of the received wave, etc. It corresponds to the measurement.

JP 2006-98167 A Japanese Patent No. 3294726

In the mask processing disclosed in Patent Document 1, since the masking distance is fixed, it is impossible to measure a close distance near 0 m from the reference position for distance measurement. Further, when the distance measuring object is a moving object, the reflected signal wave may be blocked depending on the position. In addition, the radar device disclosed in Patent Document 2 not only complicates the circuit configuration of the high-frequency unit, but it is not only difficult to detect a signal at a close distance of 0 m from the reference position in time axis expansion and integration. The problem remains that the distance measurement accuracy decreases as it approaches the reference position.
It is a main object of the present invention to provide a short-distance pulse radar device that enables measurement of a close range while suppressing the influence of an interference wave due to transmission with a simple configuration.

  According to the present invention, a high-frequency oscillator that outputs a local signal wave, a first rat race circuit that distributes the local signal wave in two, and a second rat race that performs pulse modulation on one of the distributed local signal waves A demultiplexing modulation circuit including a circuit, a transmission antenna that radiates a transmission signal wave pulse-modulated by the second rat race circuit, a reflected signal wave in which the transmission signal wave is reflected by an object to be measured, and the reflection A receiving antenna that receives a sneak wave other than a signal wave, a phase shift circuit that outputs the other distributed local signal wave with a predetermined phase difference from the sneak wave, and the reflected signal wave of the phase shift circuit A pulse radar device having a detection circuit that performs homodyne detection using an output signal can be provided.

  In addition, according to the present invention, it is possible to provide a high-frequency board that can be incorporated into a distance measuring device such as a pulse radar device. The high-frequency board is a high-frequency board that can be incorporated into a distance measuring device having a transmission antenna and a reception circuit used in, for example, a 24 [GHz] band or higher, and includes a high-frequency board and a first board mounted on the high-frequency board. The first rat race circuit includes an input port for inputting a local signal wave from the distance measuring device, and the input local signal wave to the second rat race circuit. A first port that leads to the rat race circuit; and a second port that guides the input local signal wave to the receiving circuit. The second rat race circuit receives the local signal input through the first port. A modulation circuit that modulates a signal wave and guides the signal wave to the transmission antenna, and an electrical length between the first port and the second port is determined from the second rat race circuit to the first port. Internal reflection waves transmitted to the second port through the bets is characterized in that it is set to a length to attenuate.

  According to the present invention, since a sneak wave other than a reflected signal wave received by a receiving antenna or an interference wave due to an internal reflected wave is attenuated, a simple circuit is not required even though it is a pulse radio wave radar. Thus, it is possible to perform a close range measurement as close as possible to 0 m with high accuracy.

The block diagram of the short-distance pulse radar apparatus to which this invention is applied. 1 is a configuration diagram of a demultiplexing modulation circuit according to the present embodiment. The block diagram of the phase-shift circuit of this embodiment. The timing chart of the control system by this embodiment. FIG. 4 is an exemplary diagram of a detection output waveform diagram when there is no input in the present embodiment. (A)-(c) is an external view of the case of a short-distance pulse radar apparatus.

  Embodiments of the present invention applied to a short-distance pulse radar device as an example of a distance measuring device will be described below. FIG. 1 is a configuration diagram of a short-range pulse radar device. The short-range pulse radar device 1 includes a high-frequency unit 10, a high-frequency oscillator 20, a control circuit 30, a transmission antenna 40, a reception antenna 50, a baseband circuit 60, and an A / D conversion circuit 70. These components are accommodated in a case described later.

  The high frequency unit 10 is a high frequency board that can be incorporated into a distance measuring device, and includes a demultiplexing modulation circuit 11, a phase shift circuit 12, and a mixer 13. That is, a transmission line pattern of a copper film is formed on a composite base material that is a high-frequency substrate, and circuit components are mounted on a predetermined portion. Although the material and size of the composite base material vary depending on the frequency to be used, as will be described later, in this embodiment, since the 24 GHz band is used, a composite base of thermosetting resin / ceramic filler / glass cloth having a thickness of 0.254 mm is used. An example of using a material will be described. A high-frequency board can be configured by mounting all circuit components on a single composite substrate.

  The high frequency oscillator 20 inputs a CW (unmodulated continuous wave) high frequency signal wave (referred to as a “local signal wave”) to the branching modulation circuit 11 of the high frequency unit 10. Here, an example in the case where the 24 GHz band is used will be described, but the present invention is used in a distance measuring device having a relatively low frequency band such as the 10 GHz band or a higher frequency band such as the 79 GHz band. It can also be applied to a pulse radar device.

The control circuit 30 generates a short pulse signal, a phase signal, and a control signal based on a reference signal (CLK) received from an external device (computer not shown) that uses distance measurement information (not shown). Further, signal processing for distance measurement is performed based on a short pulse signal and a digital detection signal described later. The control circuit 30 itself is a DSP (Digital Signal
Processor). The short pulse signal is a pulse signal having a pulse width of 100 nsec or less, and is used for pulse modulation. The short pulse signal is input to the demultiplexing modulation circuit 11 of the high frequency unit 10. The phase signal is a signal used for adjusting the amount of phase shift by the phase shift circuit 12 of the high frequency unit 10. The control signal is a signal that controls the operation of the demultiplexing modulation circuit 11 and the phase shift circuit 12.

  The transmission antenna 40 radiates the transmission signal wave output from the branching modulation circuit 11 toward the object to be measured. The receiving antenna 50 receives the reflected signal wave reflected from the object to be measured and the sneak wave other than the reflected signal wave, and inputs them to the mixer 13. In FIG. 1, the transmission antenna 40 and the reception antenna 50 are configured separately, but may be a shared antenna.

  The mixer 13 is an example of a detection circuit. The mixer 13 mixes the received reflected signal wave and sneak wave with the output signal of the phase shift circuit 12 and performs homodyne detection, and outputs the detection signal to the baseband circuit 60. The baseband circuit 60 amplifies the detection signal by pulse. The A / D conversion circuit 70 converts the detection signal amplified by the baseband circuit 60 into a digital signal and outputs it to the control circuit 30. The control circuit 30 generates distance measurement information to the object based on the digital signal of the detection signal and the short pulse signal, and outputs this to the external device.

  Next, each component of the high frequency unit 10 will be described. FIG. 2 is a configuration diagram of the branching modulation circuit 11. The demultiplexing modulation circuit 11 is one of the high-frequency boards described above, and a rat race circuit type distribution circuit (rat race circuit) is mounted on a composite substrate. A branch line type (branch line type) distribution circuit is known as a distribution circuit used at a high frequency, but this embodiment does not use it. This is because the rat race type distribution circuit has higher impedance than the branch line type, and the line width of the transmission line pattern can be narrowed. As a result, a small distribution circuit can be configured easily and easily even in a high frequency band such as 24 GHz, and since the impedance is stable, the pulse signal is not affected in the distribution circuit without affecting the performance. And supply lines (supply points) such as a bias can be easily connected.

The demultiplexing modulation circuit 11 includes a first rat race circuit 111 that distributes the local signal wave output from the high-frequency oscillator 20 in two, and a second rat race circuit 112 that performs pulse modulation on one of the distributed local signal waves. It is comprised including. The first rat race circuit 111 has a port (first port) for guiding one local signal wave to the second rat race circuit 112, and the other local signal wave on the receiving circuit side, in this example, phase shift. And a port for leading to the circuit 12 (second port). Between the first port and the second port, the electrical length (electrical length) at which the internally reflected wave transmitted from the second rat race circuit 112 to the second port through the first port attenuates is reduced. Is set. This will be described later.
The second rat race circuit 112 performs pulse modulation on the local signal wave input from the first rat race circuit 111 through the first port, and supplies the modulated local signal wave to the feeding point of the transmission antenna 40.

  Here, a port configuration example (including the electrical length between ports) of the first rat race circuit 111 and the second rat race circuit 112 will be described. The first rat race circuit 111 has ports P1, P2, P3, and P4. The second rat race circuit 112 has ports P5, P6, P7, P8, and P9. Port P4 and port P5 are configured as a common port. The port P9 serves as a bias point and receives a short pulse signal. Each port P1 to P9 is formed of a transmission line pattern having a line width of √2Zo (Zo is circuit impedance = 50Ω).

  Between the ports P1-P2, P2-P3, P3-P4, P5-P6, P6-P7, P7-P8, P8-P9, the electrical length is set to 1 / 4λ with respect to the wavelength λ of the operating frequency. Is done. On the other hand, an electrical length of 3 / 4λ is formed between each of the ports P1-P4 and P5-P8. When the port P1 of the first rat race circuit 111 is an input port, the port P4 is the first port and the port P2 is the second port. The port P3 is an isolation port.

In the first rat race circuit 111 configured as described above, the local signal wave input to the port P1 is in phase in the clockwise and counterclockwise directions at the ports P2 and P4. Therefore, a local signal wave can be output from the port P4. At the port P3, the local signal wave input to the port P1 is in the opposite phase between clockwise and counterclockwise. Therefore, the local signal wave input from the port P1 can be output from the port P2 to the phase shift circuit 12 side without considering the connection status of the port P3.
In the second rat race circuit 112, assuming that the port P5 is an input port (first port viewed from the first rat race circuit 111), in a normal rat race circuit, the ports P6 and P8 are output ports, and the ports P7 and P9. However, in the rat race circuit of this embodiment, the ports P6 and P8 are responsible for controlling the phase of the local signal wave, so that the ports P6, P8, and P9 are isolation ports and the port P7 is Output port. This port P7 is a port connected to the feeding point of the transmitting antenna 40.

  The port P1 of the first rat race circuit 111 is connected to the high-frequency oscillator 20 via a DC blocking capacitor C1. The port P2 is connected to the phase shift circuit 12 via the DC blocking capacitor C2. The port P3 is connected to one end of a termination resistor R2 (= 50Ω) via a DC blocking capacitor C3. The other end of the termination resistor R2 is connected to GND (ground end). A stub pattern ST is formed between the DC blocking capacitor C3 and the termination resistor R2. Port P4 is connected to port P5 of the second rat race circuit.

  Here, the stub pattern ST connected to the port P3 will be described. Internally reflected waves enter the port P4 of the first rat race circuit 111 through the port P5 of the second rat race circuit 112. As described above, such internally reflected waves affect the distance measurement accuracy. Since the port P2 is an isolation port when viewed from the port P4, the internal reflected wave is not originally large, but it is desirable to further reduce this in order to ensure higher ranging accuracy. A part of the internally reflected wave also enters the port P3. Internally reflected waves can be reduced by inserting a matching stub pattern between ports P4-P5, but on the other hand, since ports P4-P5 are a common line, they are greatly affected by high-frequency signal waves and loss is large. Become.

  Therefore, in this embodiment, the first rat race pays attention to the fact that the port P3 acts as a combiner of the port P2 and the port P4, and between the port P3 and the DC blocking capacitor C3 and the terminating resistor R2. A stub pattern ST is provided in the line so that the reflected signal wave entering the port P3 is reflected by the stub pattern ST, and the reflected signal wave cancels the internal reflected wave entering from P4. The stub pattern ST is set, for example, to a stub width of 0.68 mm and a length of 1.5 mm. As a result, the internal reflected wave from the second rat race circuit 112 reaching the port P2 can be reduced to 0 without affecting the local signal wave input from the port P1. Therefore, only the local signal wave input from the port P1 can be output to the phase shift circuit 12 side. As described above, the first rat race circuit 111 operates as a circuit that efficiently reduces the internal reflection wave to 0 and efficiently demultiplexes the local signal wave to the transmission antenna 40 side and the phase shift circuit 12 side.

  In the second rat race circuit 112, the local signal wave is input to the port P5 through the port P4 of the first rat race circuit 111 as described above. The port P6 is connected to GND (ground end) via the transmission line pattern M1, the diode D2, and the transmission line pattern M3. The port P8 is connected to the ground terminal via the transmission line pattern M2, the diode D1, and the transmission line pattern M4. The diodes D <b> 1 and D <b> 2 are switching elements for selectively performing conduction / non-conduction switching based on a control signal from the control circuit 30.

  The transmission line patterns M1 and M2 function as reactances for arbitrarily setting the phase of a signal wave propagating therethrough. In this example, the transmission line pattern M1 has a length that delays the phase by 270 degrees. The transmission line pattern M2 has a length that delays the phase by 90 degrees. The transmission line patterns M3 and M4 have lengths that cancel the capacitive reactances of the diodes D1 and D2, respectively. Thereby, the fluctuation | variation of the phase accompanying the switching operation | movement of conduction | electrical_connection / non-conduction of diode D1, D2 can be suppressed.

  In the second rat race circuit 112 configured as described above, the diodes D1 and D2 are turned on by the control signal from the control circuit 30, whereby the end portions of the transmission line patterns M1 and M2 are short-circuited (SHORT). Become. On the other hand, by turning off the diodes D1 and D2 with a control signal from the control circuit 30, the ends of the transmission line patterns M1 and M2 are opened (OPEN). As a result, the phase of the transmission signal wave radiated from the transmission antenna 40 changes according to the reactance of the transmission line patterns M1 and M2, so that there is no DC fluctuation and a constant amplitude phase modulation circuit with a simple configuration and control mode. Can be realized. Note that when viewed from the port P6, the port P8 is an isolation port, and when viewed from the port P8, the port P6 is an isolation port. Therefore, the second rat race circuit is not affected by the ports. The characteristic impedance of 112 becomes stable.

  An RF choke inductance element L1 and a diode current control resistor R1 are inserted into the port P9 so that the characteristic impedance is relatively high when viewed from the second rat race circuit 112. Then, the local signal wave input through the first rat race circuit 111 and the port P5 is subjected to pulse phase modulation by the short pulse signal input from the control circuit 30. The transmission signal wave subjected to pulse phase modulation in this way is output from the port P7 to the transmission antenna 40 via the DC blocking capacitor C4.

  The modulation operation by the second rat race circuit 112 will be described in more detail as follows. First, when the diodes D1 and D2 are non-conductive, the ports P6 and P8 are opened. At this time, in the clockwise direction between the ports P5 and P7, the phase of the input local signal wave is delayed by 180 degrees (2λ / 4) in a normal rat race circuit, but in the present embodiment, the transmission line pattern M1 Since the phase is delayed by 270 degrees (3λ / 4), the overall delay is 90 degrees (λ / 4). In the counterclockwise direction, there is no phase change in a normal rat race circuit, but in the present embodiment, the phase is delayed by 90 degrees (λ / 4) in the transmission line pattern M2, and therefore the entire delay is 90 degrees (λ / 4). . As a result, since both the clockwise and counterclockwise phases are in phase, a transmission signal wave whose phase is delayed by 90 degrees is output from the port P7 to the transmission antenna 40.

  On the other hand, when the diodes D1 and D2 are conductive, the ports P6 and P8 are short-circuited. At this time, the phase of the transmission signal wave is inverted (180 degrees). Between the ports P5 and P7, since the phase is delayed by 450 degrees in the transmission line pattern L1M1 in the clockwise direction, the phase is delayed by 270 degrees as a whole. On the other hand, in the counterclockwise direction, the phase is delayed by 270 degrees in the transmission line pattern L2M2. Therefore, as a whole, the phase is delayed by 270 degrees. As a result, both clockwise and counterclockwise are in phase, so that a transmission signal wave whose phase is delayed by 270 degrees is output from the port P7 to the transmission antenna 40. As described above, the conduction / non-conduction switching operation of the diodes D1 and D2 causes the phase difference of the transmission signal wave to be 180 degrees (= 270-90), and phase modulation with a difference of 0 degrees / 180 degrees is possible.

  As the diodes D1 and D2, Schottky barrier diodes having a quick response are used. Thereby, it can respond to a short pulse signal. Further, the inductance element L1 is configured by a pattern having a line width of 0.1 mm and a length of 1.55 mm. The diode current control resistor R1 is 120Ω. The transmission line pattern M1 is set to a width of 0.55 mm and a length of 3.0 mm, the transmission line pattern M2 is set to a width of 0.55 mm and a length of 1.2 mm, and M3 and M4 are set to a width of 0.55 mm and a length of 0.2 mm, respectively. ing. However, these numerical values may be slightly changed depending on the frequency used. For the short pulse signal, the Low level is set to 0 v and the High level is set to +3.3 v.

  Next, the phase shift circuit 12 will be described. The phase shift circuit 12 is a circuit that adjusts the phase of the local signal wave distributed at the port P2 of the branching modulation circuit 11 and the phase of the sneak wave to have a predetermined phase difference. The phase difference between the sneak wave and the local signal wave can be specified by calculation if the distance between the transmission antenna 40 and the reception antenna 50, the frequency band to be used, the output of the transmission signal wave, and the like are determined. In the present embodiment, the phase shift between the sneak wave and the local signal wave is eliminated, and the phase of the local signal wave is changed in the range of 0 degrees to 90 degrees. The reason why the phase shift between the local signal wave and the sneak wave is eliminated is that a null point is generated by the sneak wave when homodyne detection is performed by the mixer 13, thereby eliminating the influence of the interference wave due to the sneak wave. . The reason why the phase of the local signal wave is changed in the range of 0 to 90 degrees is to change the detection signal to the maximum amplitude. This is because when the distance measuring object moves, a NULL point is generated even in the reflected signal wave depending on the distance. That is, the phase shift between the reflected signal wave and the local signal wave is eliminated at the distance to the object to be measured (for each wavelength λ of the used frequency, approximately 12 mm in the 24G band), and a NULL point is generated, resulting in distance measurement accuracy. May be reduced. Therefore, the detected signal can be output as a significant value by changing the phase of the local signal wave input to the mixer 13 and changing the detected signal to the maximum amplitude.

  In order to operate as described above, in this embodiment, the phase shift circuit 12 is configured as shown in FIG. Referring to FIG. 3, the phase shift circuit 12 includes a DC cutoff capacitor C5, a variable phase shifter 121, a delay line (DL) 122, and a DC cutoff capacitor C6. The DC blocking capacitor C5 blocks the DC component of the local signal wave distributed from the branching modulation circuit 11. The variable phase shifter 121 sets the predetermined phase difference to 0, that is, eliminates the phase shift between the sneak wave and the local signal wave. The variable phase shifter 121 includes a hybrid coupled diode. The delay line (DL) 122 delays the phase of the local signal wave that passes therethrough. The DC blocking capacitor C6 blocks the DC component of the output signal to the mixer 13.

  A current adjustment resistor R3 is connected between the delay line (DL) 122 and the DC blocking capacitor C6 via an impedance element L2, and a phase signal is supplied to the current adjustment resistor R3. The phase signal is a signal supplied from the control circuit 30 and makes the diode of the variable phase shifter 121 conductive or non-conductive. The phase signal is a digital (square wave) signal or a sawtooth wave (triangular wave). When the phase signal is a digital signal, the variable phase shifter 122 selectively switches the phase of the local signal wave between 0 degrees / 90 degrees set in advance. When the phase signal is a sawtooth wave, the variable phase shifter 122 gradually switches the phase of the local signal wave in the range of 0 degrees and 90 degrees. This eliminates the NULL point at the time of detection of the reflected signal wave in the mixer 13.

  In the example of FIG. 3, an example in which the delay line (DL) 122 is used as the delay unit is shown, but reactance loading can also be handled. In the case of reactance loading, the reactance of the transmission path is varied using a switching element or the like to change the electrical length.

  Further, in the phase shift circuit 12 of this embodiment, the local signal wave is not included in the local signal wave by the branching modulation circuit 11 and only the phase of the sneak wave needs to be considered, so that the phase setting of the delay means is performed. And the adjustment of the phase shift becomes simple.

The mixer 13 may be mounted on a single high-frequency board together with the branching modulation circuit 11 and the phase shift circuit 12, or may be mounted as a single high-frequency board. At this time, the mixer 13 can be configured using a rat race circuit (third rat race circuit) as in the case of the demultiplexing modulation circuit 11.
When the mixer 13 is configured with a rat race circuit, the input port is connected to the feeding point of the receiving antenna 50 via a capacitor or the like. The other input port is formed at a site separated by an electrical length without mutual interference, and is connected to the phase shift circuit 12 to receive a phase-adjusted local signal wave. The output port is connected to the baseband circuit 60.

  The mixer 13 performs homodyne detection on the reflected signal wave and the sneak wave from the receiving antenna 50 with the local signal wave input from the phase shift circuit 12, but in the homodyne detection, the amplitude of the output detection signal is a reflected signal wave or the like. It changes according to the phase. The change in amplitude becomes a stationary wave, and “antinode” and “node” are repeated every 90 degrees in phase. Since the sneak wave has no phase shift from the local signal wave, the sneak wave disappears. Further, the transmission signal wave itself is subjected to 0/180 degree phase modulation, and the phase of the local signal wave changes in the range of 0 degree to 90 degrees, so that the NULL point of the reflected signal wave is avoided and 0 m from the reference position. It is possible to realize a state in which the distance can be accurately measured even at a close distance.

  FIG. 4 is a timing chart of the short pulse signal, the phase signal, and the detection signal according to the present embodiment. 4 is an enlarged view of a short pulse signal. In FIG. 4, T1 = 5 ns, T2 = 0.5 μs, and T3 = T4 = 500 μs. In the example shown in FIG. 4, the control circuit 30 controls so that the phase signal is high and the phase is 90 degrees when the amplitude of the detection signal is NULL point by homodyne detection. When the phase signal is set to Low, the phase becomes 0 degree and a detection signal appears. Therefore, it is possible to perform high-precision distance measurement without a NULL point by processing the detection signal temporally while switching High / Low of the phase signal. In this case, it is also possible to control the phase signal while monitoring the detection signal with software.

  FIG. 5 is a diagram showing an operation comparison example between the conventional pulse radar device and the short-range pulse radar device 1 of the present embodiment. Here, an output waveform of a detection signal when there is no input without an object is shown. As shown in the middle part of FIG. 5, the detection signal of the conventional pulse radar device outputs an interference wave caused by transmission at a close distance, which hinders close distance measurement. Therefore, for example, distance measurement at 0 m cannot be performed. On the other hand, in the case of the short-range pulse radar device 1 of the present embodiment, the interference wave in the detection signal is canceled. Therefore, even if the object to be measured exists at a close distance as close as possible to the reference position (0 m) for distance measurement, the distance can be measured with high accuracy. The reference position is an installation position of the transmission antenna 40, for example, and can be determined at the time of design.

  As described above, in the short-distance pulse radar device 1 according to the present embodiment, the null point (minimum signal point) peculiar to homodyne detection is used to eliminate the phase shift between the sneak wave and the local signal wave, thereby causing the sneak wave during transmission. Can be reduced. Further, by changing the phase of the local signal wave input to the mixer 13 in the range of 0 degrees to 90 degrees, it is possible to prevent the ranging accuracy from being deteriorated.

  The short-distance pulse radar device 1 of the present embodiment can be configured as a circuit board provided on a metal base, for example, and can be commercialized by being housed in a small case. FIG. 6A is a plan view of a metal base 100 mounted with a circuit board on which the short-range pulse radar device 1 is configured and a case for housing the metal base 100, FIG. 6B is a side view, and FIG. . The metal base 100 is made of die-casting and has a rectangular shape, and a rectangular RF substrate 110 that is slightly smaller than the metal base 100 is mounted on the upper surface thereof. Since the high-frequency unit 10 including the high-frequency oscillator 20 is mounted on the central portion of the RF substrate 110 as described later, a metal cover 120 for preventing electromagnetic radiation due to high frequency is covered. On the other hand, a signal processing board 130 on which various signal processing circuits including a control circuit 30, a baseband circuit 60, and an A / D conversion circuit 70 of the short-range pulse radar device 1 are mounted at the center of the lower surface of the metal base 100 is a metal. The shield cover 140 is attached in a state of being housed. The metal base 100, the signal processing board 130, and the shield cover 140 accommodate cables and connectors necessary for electrical connection between the RF board 110 and the signal processing board 130 and connection from the control circuit 30 to an external device. The communicating hole 150 is provided. In addition, the metal base 100 on which the RF substrate 110 and the signal processing substrate 130 are mounted is a case 160 having a radio wave transmission made of a hard insulator and slightly larger than the size of the metal base 100, as indicated by a dashed line. Is housed.

  As described above, the RF substrate 110 includes the high-frequency unit 10 including the high-frequency oscillator 20 at the center, and the transmission antenna 40 and the reception antenna 50 are formed on the left and right sides of the center, respectively. The In addition to the demultiplexing modulation circuit 11 shown in FIG. 2 and the phase shift circuit 12 shown in FIG. 3, the high-frequency unit 10 is also equipped with a mixer 13 shown in FIG. The transmission antenna 40 and the reception antenna 50 are planar antennas and have equivalent antenna performance with directivity upward. The RF board 110 and the signal processing board 130 are electrically connected by a wiring cable through the hole 150 or by a connector for connection provided in the hole 150. Further, the output of the short-distance pulse radar device 1, that is, the output signal from the control circuit 30 provided on the signal processing board 130, is connected to a wiring cable or a connector from an opening of the hole 150 provided below the shield cover 140. Is taken out and output to an external device.

  The case 160 having such a structure is a small one having a thickness of about 1 cm and a long side of about 15 cm in the 24 GHz band. Therefore, it can be installed in various places. For example, it can be applied to detection of a vehicle that is embedded in a predetermined area of a parking lot and exists in the area. Since the outer surface of the case 160 that is the distance measurement reference position and the position of the transmission antenna 40 are provided very close to each other, the distance measurement reference including the case where the distance measurement object is in contact with the case 160 is provided. It is possible to measure the distance with high accuracy even when it is located at a close distance as close as possible to the position.

  DESCRIPTION OF SYMBOLS 1 ... Short distance pulse radar apparatus, 10 ... High frequency part, 11 ... Demultiplexing modulation circuit, 111, 112 ... Rat race circuit, 12 ... Phase shift circuit, 13 ... Mixer, DESCRIPTION OF SYMBOLS 20 ... High frequency oscillator, 30 ... Control circuit, 40 ... Transmitting antenna, 50 ... Receiving antenna, 60 ... Baseband circuit, 70 ... A / D conversion circuit, 100 ... Metal base.

Claims (13)

A high-frequency oscillator that outputs local signal waves;
A demultiplexing modulation circuit including a first rat race circuit that distributes the local signal wave in two, and a second rat race circuit that performs pulse modulation on one of the distributed local signal waves;
A transmission antenna that radiates a transmission signal wave pulse-modulated by the second rat race circuit;
A receiving antenna that receives a reflected signal wave in which the transmission signal wave is reflected by a ranging object and a sneak wave other than the reflected signal wave;
A phase shift circuit for outputting the other distributed local signal wave with a predetermined phase difference from the wraparound wave; and
A detection circuit for homodyne detection of the reflected signal wave with an output signal of the phase shift circuit,
Pulse radar device. The first rat race circuit has a first port for guiding the one local signal wave to the second rat race circuit, and a second port for guiding the other local signal wave to the phase shift circuit. Including
The electrical length of the first port and the second port is set to a length that attenuates the internally reflected wave transmitted from the second rat race circuit to the second port through the first port. Yes,
The pulse radar device according to claim 1. In the first rat race circuit, an isolation port is formed between the first port and the second port;
The isolation port is provided with a stub pattern connected to the other end of a termination resistor whose one end is grounded, and attenuates the internal reflected wave that has reached the isolation port with the stub pattern.
The pulse radar device according to claim 2. The second rat race circuit includes modulation means for performing pulse phase modulation of the one local signal wave input through the first port with a phase difference of 0 degree or 180 degrees.
The pulse radar device according to claim 3. The modulation means includes a transmission line pattern through which the one local signal wave propagates, and a switching element that changes the electrical length of the transmission line pattern by conduction or non-conduction.
The pulse radar device according to claim 4. The switching element is a Schottky barrier diode;
The pulse radar device according to claim 5. The phase shift circuit includes delay means for delaying the phase of the other local signal wave input to the detection circuit, and phase adjustment means for changing the phase of the delayed local signal wave within a predetermined range. To be
The pulse radar device according to any one of claims 1 to 6. The phase adjusting means changes the phase of the local signal wave by selectively changing a plurality of preset lengths of the electrical length of the path through which the delayed local signal wave propagates.
The pulse radar device according to claim 7. The detection circuit includes a third rat race circuit that mixes the reflected signal wave and the sneak wave with the output signal of the phase shift circuit.
The pulse radar device according to claim 8. The transmitting antenna, the receiving antenna, the branching modulation circuit, the phase shift circuit, and the detection circuit are housed in a radio wave permeable case,
The distance measurement reference position is the surface of the case,
The distance measuring object can contact the case;
The pulse radar device according to any one of claims 1 to 9. The transmission signal wave and the reflected signal wave have a frequency of 24 [GHz] or higher,
The pulse radar device according to any one of claims 1 to 10. A high-frequency board that can be incorporated into a distance measuring device having a transmitting antenna and a receiving circuit used in a band of 24 [GHz] or higher,
A high frequency substrate;
A first rat race circuit and a second rat race circuit mounted on the high-frequency substrate;
The first rat race circuit includes an input port for inputting a local signal wave from the distance measuring device, a first port for guiding the input local signal wave to the second rat race circuit, and an input local signal. A second port for guiding a wave to the receiving circuit;
The second rat race circuit includes a modulation circuit that modulates a local signal wave input through the first port and guides the local signal wave to the transmission antenna;
The electrical length between the first port and the second port is set to such a length that the internally reflected wave transmitted from the second rat race circuit to the second port through the first port is attenuated. Being
High frequency board. The reception circuit includes a reception antenna that receives a reflected signal wave in which a transmission signal wave radiated from the transmission antenna is reflected by a distance measuring object and a sneak wave other than the reflected signal wave,
A phase shift circuit for outputting a local signal wave output from the second port to the high-frequency substrate with a predetermined phase difference from the wraparound wave, and homodyne detection of the reflected signal wave with an output signal of the phase shift circuit And a detector circuit to be implemented,
The high-frequency board according to claim 11.

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