JP2015019455A - Drive circuit for drive object switching element - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a drive circuit for a drive object switching element which can prevent the deterioration of reliability of a switching element S¥# caused by switching to an off state of the switching element S¥# not carrying out a soft interruption processing when the excess current flows into the switching element S¥#.SOLUTION: In a drive circuit DU, a gate and an emitter of a switching element S¥# are connected with each other via a path for clamp Lclamp. A resistor 32 is provided on the path for clamp Lclamp. In a state that the collector current Ic decreases, the path for clamp Lclamp and a main current flow path connected to the collector are magnetically coupled to each other so that gate charges move to the path for clamp Lclamp side.

Description

  The present invention relates to a drive circuit for a drive target switching element including constant current supply means for supplying a constant current to an open / close control terminal of the drive target switching element in order to switch the drive target switching element to an on state.

  2. Description of the Related Art Conventionally, as can be seen, for example, in Patent Document 1 below, a constant current driving circuit that supplies a constant current to the gate of a switching element (IGBT) is known.

Japanese Patent No. 4954290

  There is also known a drive circuit including an overcurrent protection circuit that protects a switching element when an overcurrent flows between a pair of main terminals (collector, emitter) of the switching element. The protection circuit will be described. First, in a situation where a constant current is supplied to the gate, the gate voltage is discharged by discharging the gate charge through the clamp path for a predetermined time before the gate voltage reaches the upper limit voltage. Is limited by a clamp voltage lower than the upper limit voltage. Here, the clamp path is a discharge path connected to the gate.

  Thereafter, under the situation where the gate voltage is limited by the clamp voltage, the gate charge is discharged through the soft cutoff path connected to the gate, thereby performing the soft cutoff for forcibly switching the switching element to the OFF state. Here, the soft cutoff path is a discharge path having a resistance value larger than that of the discharge path used for switching the switching element to the OFF state at the normal time. According to the soft shut-off, the discharge rate of the gate charge can be made lower than the discharge rate at the normal time, and the surge voltage generated when the switching element is switched to the off state can be reduced.

  Here, the present inventors have faced a problem that the switching element can be switched to the OFF state regardless of the soft cutoff despite the overcurrent flowing. In this case, there is a concern that the discharge rate of the gate charge is higher than the discharge rate at the time of soft shutoff, the surge voltage increases, and the reliability of the switching element is lowered.

  The present invention has been made to solve the above problems, and an object of the present invention is to provide a drive circuit for a drive target switching element capable of avoiding a decrease in reliability of the drive target switching element.

  In order to solve the above-mentioned problems, the present invention provides a constant current supply means (24; 24a) for supplying a constant current to the open / close control terminal of the drive target switching element in order to switch the drive target switching element (S ¥ #) to the ON state. 24b), a clamp path (Lclamp) connected to the open / close control terminal, and the clamp path when the charge of the open / close control terminal is discharged by the clamp path. In a situation where a constant current is supplied to the switching control terminal by the impedance part (32) having the highest impedance and the constant current supply means, the voltage of the switching control terminal reaches a predetermined time before reaching the upper limit voltage. Across the impedance part accompanying the flow of the output current of the constant current supply means to the impedance part Limiting means (34, 36, 38) for limiting the voltage of the switching control terminal with a clamp voltage lower than the upper limit voltage due to a voltage drop, and a current flowing between a pair of main terminals of the drive target switching element Is connected to at least one of the current flow path connected to the open / close control terminal and the pair of main terminals so that the charge of the open / close control terminal moves to the clamping path side. The main current flow path is magnetically coupled.

  In a situation where the current flowing between the pair of main terminals (hereinafter referred to as main current) decreases, the current flow path connected to the switching control terminal and the main switching path so that the charge of the switching control terminal moves to the clamping path side. There is a drive circuit in which a current flow path is magnetically coupled. In other words, the current flow path connected to the open / close control terminal and the main current flow path are magnetically coupled so that the charge moves from the clamping path side to the open / close control terminal in a situation where the main current increases. There is a drive circuit. In such a drive circuit, after a constant current starts to be supplied to the switching control terminal, the voltage at the switching control terminal once greatly exceeds the clamp voltage.

  Thereafter, when the voltage at the switching control terminal begins to decrease toward the clamp voltage, the main current also begins to decrease. Here, under the situation where the main current decreases, the electric charge of the switching control terminal is moved to the clamping path side by magnetic coupling. The transferred electric charge is discharged through the clamping path. As a result, the voltage at the open / close control terminal is significantly lower than the clamp voltage. Thereby, the problem that a drive object switching element switches to an OFF state arises.

  Therefore, in the above invention, the impedance portion is provided in the clamp path. In the above invention, the impedance unit is used to limit the voltage of the switching control terminal with the clamp voltage, and the impedance unit is also used to avoid the voltage of the switching control terminal being significantly lower than the clamp voltage. Here, the drop in the voltage of the switching control terminal can be avoided by the impedance unit even when the magnetic coupling occurs in a situation where the main current is reduced, the charge of the switching control terminal via the clamping path is reduced. This is because the discharge can be prevented by the impedance portion.

  Therefore, according to the above-described invention, it is possible to avoid switching the drive target switching element to the OFF state by the magnetic coupling in a situation where the voltage of the switching control terminal is limited by the clamp voltage. Thereby, the fall of the reliability of a drive object switching element can be avoided.

The lineblock diagram of the motor control system concerning a 1st embodiment. The block diagram of the drive circuit concerning the embodiment. The time chart which shows an example of the overcurrent protection process at the time of normal. The time chart which shows an example of the overcurrent protection process at the time of an upper and lower arm short circuit. The figure which shows the magnetic coupling aspect at the time of the overcurrent rise concerning 1st Embodiment. The figure which shows the magnetic coupling aspect at the time of the overcurrent reduction concerning the embodiment. The block diagram of the drive circuit concerning related technology. The time chart which shows an example of the overcurrent protection process concerning related technology. The time chart which shows an example of the overcurrent protection process concerning 1st Embodiment. The block diagram of the drive circuit concerning 2nd Embodiment. The time chart which shows an example of the overcurrent protection process concerning the embodiment. The block diagram of the drive circuit concerning 3rd Embodiment. The time chart which shows an example of the overcurrent protection process concerning the embodiment.

(First embodiment)
Hereinafter, a first embodiment in which a drive circuit of a drive target switching element according to the present invention is applied to a vehicle (for example, a hybrid vehicle or an electric vehicle) including a rotating machine as an in-vehicle main unit will be described with reference to the drawings.

  As shown in FIG. 1, the motor generator 10 is a multi-phase rotating machine (three-phase rotating machine) as an in-vehicle main machine, and is connected to driving wheels (not shown). The motor generator 10 is connected to a high voltage battery 12 as a “DC power supply” via an inverter 11. The output voltage of the high voltage battery 12 is, for example, 100 V or more. As the high voltage battery 12, for example, a lithium ion storage battery or a nickel hydride storage battery can be used. In the present embodiment, a synchronous machine (permanent magnet synchronous machine) is used as the motor generator 10.

  The inverter 11 includes a series connection body of a switching element S ¥ p (¥ = U, V, W) on the high potential side (upper arm side) and a switching element S ¥ n on the low potential side (lower arm side). . Specifically, the inverter 11 includes a series connection body of three sets of switching elements S ¥ p, S ¥ n, and the connection point of the switching elements S ¥ p, S ¥ n is connected to the ¥ phase of the motor generator 10. Yes. Incidentally, in the present embodiment, a voltage-controlled semiconductor switching element is used as the switching element S ¥ # (# = p, n), and more specifically, an IGBT is used. A free wheel diode D ¥ # is connected in reverse parallel to the switching element S ¥ #. In the present embodiment, the switching element S ¥ # corresponds to a “drive target switching element”.

  The control device 14 is composed mainly of a microcomputer using the low voltage battery 16 as a power source. The control device 14 operates the inverter 11 to control the control amount (for example, torque) of the motor generator 10 to the command value. Specifically, the control device 14 generates an operation signal g ¥ # and outputs it to the drive circuit DU corresponding to the switching element S ¥ # in order to operate the switching element S ¥ # constituting the inverter 11. Here, the high-potential side operation signal g ¥ p and the corresponding low-potential side operation signal g ¥ n are complementary to each other. That is, the switching element S ¥ p on the high potential side and the corresponding switching element S ¥ n on the low potential side are alternately turned on.

  The interface 18 has a function of transmitting signals between these high-voltage systems and low-voltage systems while electrically insulating them. Here, the high voltage system is a system including the high voltage battery 12, the inverter 11, and the motor generator 10. The low voltage system is a system including the low voltage battery 16 and the control device 14. In the present embodiment, the interface 18 includes an optical insulating element (photocoupler).

  Next, the configuration of the drive circuit DU will be described with reference to FIG.

  As shown in the figure, the drive circuit DU includes a drive IC 20 that is a one-chip semiconductor integrated circuit, a constant voltage power supply 22 having a predetermined output voltage Vom (for example, 15 V), and the constant voltage power supply 22 as a power supply source. The constant current power supply 24 is provided. Specifically, the constant current power supply 24 is connected to the drain of a P-channel MOSFET (hereinafter, charging switching element 26) via a first terminal T1 of the drive IC 20. The source of the charging switching element 26 is connected to the open / close control terminal (gate) of the switching element S ¥ # via the second terminal T2 of the drive IC 20. Here, in the present embodiment, an electric path from the constant current power supply 24 to the gate through the first terminal T1, the charging switching element 26, and the second terminal T2 is referred to as a “charging path Lcha”. And In the present embodiment, the constant current power supply 24 constitutes a “constant current supply unit”.

  The gate of the switching element S ¥ # is connected to the third terminal T3 of the drive IC 20 via the discharging resistor 28. The third terminal T3 is connected to the output terminal (emitter) of the switching element S ¥ # via an N-channel MOSFET (hereinafter, discharge switching element 30). Here, in the present embodiment, the path from the gate to the emitter via the discharge resistor 28, the third terminal T3, and the discharge switching element 30 is switched off in the normal state. It will be referred to as a “normal-time off route Ldis” used for switching to. The normal-time OFF path Ldis is closed by an ON operation (closing operation) of the discharging switching element 30 and is opened by an OFF operation (opening operation) of the discharging switching element 30. Here, the normal time is a time when a charging process or a discharging process, which will be described later, is performed based on an ON operation command or an OFF operation command.

  In the present embodiment, the emitter and input terminal (collector) of the switching element S ¥ # correspond to “a pair of main terminals”. The collector corresponds to the “first main terminal” and the emitter corresponds to the “second main terminal”.

  The gate of the switching element S ¥ # is also connected to the emitter via the second terminal T2, the resistor 32, and an N-channel MOSFET (hereinafter, clamping switching element 34). A connection point between the second terminal T2 and the resistor 32 is connected to a non-inverting input terminal of the clamping comparator 36, and an inverting input terminal of the clamping comparator 36 is connected to the first power supply 38. In the present embodiment, the clamping switching element 34, the clamping comparator 36, and the first power source 38 constitute a “limiter”. The clamp comparator 36 and the first power supply 38 constitute “clamp operating means”.

  Here, the output voltage of the first power supply 38 (hereinafter referred to as the clamp voltage Vclamp) is, for example, a voltage (for example, 12V) that does not flow a current that causes the reliability of the switching element S ¥ # to decrease excessively in a short time. ) Is set to a value that limits the applied voltage (gate voltage) of the switching control terminal of the switching element S ¥ #. In the present embodiment, the clamp voltage Vclamp is specifically a voltage equal to or higher than the threshold voltage Vth at which the switching element S ¥ # switches from the off state to the on state, and lower than the output voltage Vom of the constant voltage power source 22. Is set.

  Here, in the present embodiment, a path from the gate to the emitter via the second terminal T2, the resistor 32, and the clamp switching element 34 is referred to as a “clamp path Lclamp”. The clamp path Lclamp is closed when the clamp switching element 34 is turned on, and is opened when the clamp switching element 34 is turned off.

  According to such a configuration, when the voltage at the connection point between the second terminal T2 and the resistor 32 exceeds the clamp voltage Vclamp, the clamp switching element 34 is turned on. On the other hand, when the voltage at the connection point is lower than the clamp voltage Vclamp, the clamp switching element 34 is turned off. Thereby, the voltage at the connection point can be limited by the clamp voltage Vclamp.

  The gate of the switching element S ¥ # is further connected to the emitter via the soft cutoff resistor 40, the fourth terminal T4 of the drive IC 20, and an N-channel MOSFET (hereinafter, soft cutoff switching element 42). Here, in the present embodiment, a path from the gate to the emitter through the soft cutoff resistor 40, the fourth terminal T4, and the soft cutoff switching element 42 is referred to as a “soft cutoff path Lcut”. And The soft cutoff path Lcut is closed by turning on the soft cutoff switching element 42 and opened by turning off the soft cutoff switching element 42.

  The switching element S ¥ # includes a sense terminal St that outputs a minute current (for example, “1/10000” of the collector current Ic) having a correlation with a current flowing between the collector and the emitter (hereinafter, collector current Ic). . The sense terminal St is connected to the emitter via a resistor (sense resistor 44). As a result, a voltage drop occurs in the sense resistor 44 due to a minute current output from the sense terminal St, and therefore the potential on the sense terminal St side of the sense resistor 44 (hereinafter referred to as the sense voltage Vse) has a correlation with the collector current Ic. It can be an electrical state quantity. In this embodiment, the emitter potential is “0”, and the sense voltage Vse when the potential on the sense terminal St side of both ends of the sense resistor 44 is higher than the emitter potential is defined as positive.

  The sense terminal St side of both ends of the sense resistor 44 is connected to the non-inverting input terminal of the short-circuit detection comparator 46 via the fifth terminal T5 of the drive IC 20. The inverting input terminal of the short-circuit detection comparator 46 is connected to the second power supply 48. In the present embodiment, the output voltage of the second power supply 48 (hereinafter, short circuit threshold SC) is set to the sense voltage Vse corresponding to the upper limit value of the collector current Ic that can maintain the reliability of the switching element S ¥ #. Yes. Note that the output signal Sig of the short-circuit detection comparator 46 is input to the drive control unit 50 provided in the drive IC 20.

  Based on the operation signal g ¥ # input through the sixth terminal T6 of the drive IC 20, the drive control unit 50 alternately performs the charging process and the discharging process by operating the charging switching element 26 and the discharging switching element 30. To drive the switching element S ¥ #. Specifically, the charging process is a process of turning off the discharging switching element 30 and turning on the charging switching element 26 when it is determined that the operation signal g ¥ # is an on-operation command. That is, the charging process is a constant current control process for supplying a constant current to the gate. As a result, the gate voltage Vge becomes equal to or higher than the threshold voltage Vth due to the rise of the gate potential with respect to the emitter potential, whereby the switching element S ¥ # is switched to the on state.

  In addition, according to the constant current control process, it is possible to avoid an increase in the mirror period of the switching element S ¥ # due to a decrease in the gate charging current caused by the clamp process described later, thereby avoiding an increase in switching loss and the like. be able to.

  On the other hand, the discharging process is a process of switching the discharging switching element 30 to an on operation and switching the charging switching element 26 to an off operation when it is determined that the operation signal g ¥ # is an off operation command. Thereby, switching element S ¥ # is switched to the off state.

  The drive control unit 50 further performs overcurrent protection processing based on the gate voltage Vge, the output signal Sig of the short-circuit detection comparator 46, and the like. This process is a process including a clamp process and a soft shut-off process.

  First, the clamp process will be described. This process is performed for the clamp process over the clamp filter time Tclamp (corresponding to the “predetermined time”) from the timing when the gate voltage Vge reaches the predetermined voltage Vα when the charging process is performed. This is a process for turning on the switching element 34. That is, the clamping process is a process of limiting the gate voltage Vge with the clamp voltage Vclamp before the gate voltage Vge reaches the upper limit voltage (the output voltage Vom of the constant voltage power supply 22). Here, in the present embodiment, the predetermined voltage Vα is set to a voltage equal to or higher than the threshold voltage Vth and lower than the clamp voltage Vclamp. More specifically, the predetermined voltage Vα is set to a maximum value Vthmax within a range “Vthmin to Vthmax” that the threshold voltage Vth can take. This is a setting for starting the clamping process after the switching element S ¥ # is reliably switched to the on state. That is, the threshold voltage Vth varies depending on the collector current Ic, the individual difference of the switching element S ¥ #, and the like. Therefore, by setting the predetermined voltage Vα to the maximum value assumed as the threshold voltage Vth, the clamping process can be started after the switching element S ¥ # is reliably switched to the on state.

  According to the clamping process, for example, when the upper and lower arms are short-circuited and an overcurrent (short-circuit current) flows through the switching element S ¥ #, switching is performed until the switching element S ¥ # is switched to the OFF state by the soft cutoff process described later. The collector current Ic flowing through the element S ¥ # can be limited.

  Incidentally, the clamp filter time Tclamp may be set to a time slightly longer than the maximum value of the time from when the gate voltage Vge reaches the predetermined voltage Vα until the sense voltage Vse exceeds the short-circuit threshold value SC.

  Next, the soft cutoff process will be described. This process is performed when the logic of the output signal Sig of the short-circuit detection comparator 46 is determined to be “H” continuously for the short-circuit filter time Tsc. It is determined that an overcurrent flows in #. Then, the charging switching element 26 and the discharging switching element 30 are turned off, and the soft cutoff switching element 42 is turned on. By executing the soft shut-off process, the switching element S ¥ # is forcibly switched to the off state. In the present embodiment, the soft cutoff resistor 40, the soft cutoff switching element 42, the short-circuit detection comparator 46, and the second power supply 48 constitute “soft cutoff means”.

  Incidentally, the short circuit filter time Tsc is set in order to prevent the soft shut-off process from being erroneously executed due to noise mixed into the output signal Sig of the short circuit detection comparator 46. The soft blocking resistor 40 is provided to increase the resistance value of the discharge path of the gate charge. This is because, under a situation where the collector current Ic is excessive, if the switching element S ¥ # is switched from the on state to the off state (the discharge rate of the gate charge) is increased, the surge voltage may be excessive. This setting is based on the fact that In the present embodiment, the resistance value Ra of the soft cutoff resistor 40 is set higher than the resistance value Rb of the discharging resistor 28. As a result, the resistance value of the soft cutoff path Lcut is set to be larger than the resistance value of the normal OFF path Ldis. That is, the discharge rate of the gate charge by the soft cutoff path Lcut is lower than the discharge rate of the gate charge by the normal-time off path Ldis.

  When the soft shut-off process is performed, the drive control unit 50 performs a process of outputting the fail signal FL and a process of prohibiting driving of the charging switching element 26 and the discharging switching element 30. The fail signal FL is output to the low voltage system (control device 14) via the seventh terminal T7 of the drive IC 20. The inverter 11 is shut down by the fail signal FL.

  Next, FIG. 3 and FIG. 4 show an example of overcurrent protection processing. Specifically, FIG. 3 is an example of an overcurrent protection process in a normal state where no overcurrent flows, and FIG. 4 is an example of an overcurrent protection process when an upper and lower arm short circuit occurs.

  First, it demonstrates using FIG. 3A shows the transition of the gate voltage Vge, FIG. 3B shows the transition of the operation state of the charging switching element 26, and FIG. 3C shows the discharging switching element 30. The transition of the operation state is shown. 3D shows the transition of the operating state of the clamping switching element 34, FIG. 3E shows the transition of the operating state of the soft cutoff switching element 42, and FIG. The transition of the output signal Sig of the short-circuit detection comparator 46 is shown.

  In the illustrated example, the charging switching process 30 is started by switching the discharging switching element 30 to the off operation and switching the charging switching element 26 to the on operation at time t1. As a result, the gate voltage Vge starts to rise, and then the collector current Ic and the sense voltage Vse start to rise. Note that “Vmil” shown in FIG. 3A indicates the mirror voltage of the switching element S ¥ #.

  Then, after the lapse of the mirror period, it is determined that the gate voltage Vge has reached the predetermined voltage Vα at time t2. As a result, the clamping switching element 34 is switched to the on operation, and the clamping process is started. Here, in the present embodiment, the resistance value R of the resistor 32 is set so that the multiplication value of the constant current Ig output from the constant current power supply 24 and the resistance value R becomes the clamp voltage Vclamp.

  Then, at time t3 when the clamp filter time Tclamp elapses from time t2, the clamp switching element 34 is switched to the off operation. Thereby, after that, the gate voltage Vge reaches the output voltage Vom of the constant voltage power supply 22.

  Subsequently, an example of a case where the upper and lower arms are short-circuited will be described with reference to FIG. Here, FIGS. 4A to 4F correspond to FIGS. 3A to 3F described above.

  In the illustrated example, the gate voltage Vge starts to rise when the charging process is started at time t1. Thereafter, it is determined that the gate voltage Vge has reached the predetermined voltage Vα at time t2 without passing through the mirror period. As a result, the clamping switching element 34 is switched to the on operation, and the clamping process is started.

  Thereafter, at time t3, when the sense voltage Vse exceeds the short-circuit threshold SC, the logic of the output signal Sig of the short-circuit detection comparator 46 is inverted to “H”. In the present embodiment, when it is determined that the logic of the output signal Sig is inverted to “H” in the period from when the gate voltage Vge reaches the predetermined voltage Vα until the clamp filter time Tclamp elapses, at time t2 Even when the clamp filter time Tclamp elapses, the process of continuing the ON operation of the clamp switching element 34 is performed.

  Thereafter, at time t5 when it is determined that the logic of the output signal Sig of the short-circuit detection comparator 46 continues to be “H” for the short-circuit filter time Tsc, the soft-blocking switching element 42 is switched to the ON operation. Thereby, switching element S * # is forcibly turned off. Thereafter, the clamp switching element 34 is switched to the off operation.

  Here, the overcurrent protection process shown in FIG. 4 includes a current flow path connected to the gate (in this embodiment, a clamp path Lclamp) and a main current flow path connected to at least one of the collector and the emitter. This corresponds to the case of using a drive circuit DU that is not magnetically coupled to (the flow path of the collector current Ic). On the other hand, when the drive circuit DU in which the clamp path Lclamp and the main current flow path are magnetically coupled is used, the switching element S ¥ # can be switched to the off state without using the soft cutoff process. Can occur. Hereinafter, after describing the magnetic coupling, the problem of switching to the off state without depending on the soft shutoff process will be described.

  First, the mechanism of magnetic coupling will be described with reference to FIGS. Here, FIG. 5 and FIG. 6 are diagrams showing the configuration around the switching element S ¥ n on the low potential side in the configuration shown in FIG. 1 and FIG. 5 and 6, illustration of the charging switching element 26, the terminals of the drive IC 20, and the like are omitted.

  As shown in the figure, wiring inductance exists in the main current flow path connected to at least one of the collector and the emitter and the current flow path connected to the gate. 5 and 6 show that the wiring inductance “lm” exists in the path connected to the collector of the pair of main current flow paths, and that the wiring inductance “lg” exists in the clamp path Lclamp. Indicated. In this embodiment, these wiring inductances lm and lg are magnetically coupled. Specifically, as shown in FIG. 5, in a situation where the collector current Ic rises (the change rate of the collector current Ic becomes a positive value), the charge moves from the clamp path Lclamp side to the gate. The wiring inductances lm and lg are magnetically coupled. Here, the movement of the electric charge from the clamp path Lclamp side to the gate is caused by the current flowing from the gate side to the emitter side via the gate and emitter capacitance of the switching element S ¥ #. On the other hand, as shown in FIG. 6, under the situation where the collector current Ic decreases (the change rate of the collector current Ic becomes a negative value), the wiring inductance is increased so that the gate charge moves to the clamping path Lclamp side. lm and lg are magnetically coupled. Here, the gate charge moves to the clamp path Lclamp side because the current flows from the emitter side to the gate side via the gate-emitter capacitance.

  Such magnetic coupling occurs in the drive circuit DU according to the related art shown in FIG. 7, thereby causing a problem that the switching element S ¥ # can be switched to the off state without performing the soft cutoff process. Here, in FIG. 7, the same members as those shown in FIG. 2 are given the same reference numerals for the sake of convenience.

  As illustrated, the drive circuit DU according to the related art does not include the resistor 32. The drive circuit DU includes an operational amplifier 52 instead of the clamp comparator 36.

  A connection point between the clamp switching element 34 and the second terminal T 2 is connected to a non-inverting input terminal of the operational amplifier 52, and an inverting input terminal of the operational amplifier 52 is connected to the first power supply 38.

  In the related art, the clamp process is a process of outputting an enable signal to the operational amplifier 52 over the clamp filter time Tclamp after the gate voltage Vge reaches the predetermined voltage Vα. Thus, the on-resistance of the clamp switching element 34 is adjusted by the operation of the gate voltage of the clamp switching element 34, and the voltage of the second terminal T2 is limited by the clamp voltage Vclamp.

  With reference to FIG. 8, the above problem in the case where the related technique is used will be described. Here, FIG. 8 is an example of overcurrent protection processing according to the related art. 8A to 8F correspond to the previous FIG. 4A to FIG. 4F.

  In the illustrated example, the gate voltage Vge starts to rise when the charging process is started at time t1. Thereafter, it is determined that the gate voltage Vge has reached the predetermined voltage Vα at time t2. As a result, an enable signal is output to the operational amplifier 52 to start the clamping process.

  Thereafter, at time t3, when the sense voltage Vse exceeds the short-circuit threshold SC, the logic of the output signal Sig of the short-circuit detection comparator 46 is inverted to “H”. Thereafter, the gate voltage Vge greatly exceeds the clamp voltage Vclamp once due to magnetic coupling under the condition that the collector current Ic increases. As a result, in order to limit the gate voltage with the clamp voltage Vclamp, the clamp switching element 34 is brought into a full-on state, and the discharge of the gate charge via the clamp path Lclamp is started. Here, the full-on state of the clamping switching element 34 means that the gate voltage of the clamping switching element 34 when the clamping switching element 34 is turned on is a voltage that drives the clamping switching element 34 in the non-saturated region. This is the state to set. The non-saturated region is a region where the drain current Id increases as the drain-source voltage Vds increases in the output characteristics in which the drain-source voltage Vds and the drain current Id of the clamping switching element 34 are related. That is. As a result, the on-resistance of the clamping switching element 34 is substantially “0”.

  Thereafter, at time t4, the gate voltage Vge starts to decrease toward the clamp voltage Vclamp, and thus the collector current Ic also starts to decrease. Here, under the situation where the collector current Ic decreases, the gate charge is moved to the clamping path Lclamp side by magnetic coupling. The moved gate charge is discharged through the clamp path Lclamp. In particular, in the present embodiment, since the clamp switching element 34 is in the full-on state, the impedance of the clamp path Lclamp is lowered, and the discharge amount of the gate charge is increased. As a result, the gate voltage Vge drops sharply and greatly falls below the clamp voltage Vclamp. As a result, there arises a problem that the switching element S ¥ # is switched to the OFF state due to a decrease in the gate voltage Vge before the time Tsc elapses after the short-circuit filter time Tsc is started. Therefore, the surge voltage generated when switching element S ¥ # is switched to the off state increases, and the reliability of switching element S ¥ # decreases.

  In order to cope with such a problem, in this embodiment, as shown in FIG. 2, the resistor 32 is provided in the clamp path Lclamp. The resistor 32 corresponds to an “impedance part” having the highest impedance in the clamping path Lclamp when the gate charge is discharged through the clamping path Lclamp. Here, the resistance value R of the resistor 32 is set to a value obtained by dividing the clamp voltage Vclamp by the constant current Ig output from the constant current power supply 24.

  In the present embodiment, the resistor 32 is used to limit the gate voltage Vge with the clamp voltage Vclamp, and the resistor 32 is also used to avoid the gate voltage Vge being significantly lower than the clamp voltage Vclamp. Here, it is possible to prevent the gate voltage Vge from being lowered by the resistor 32 even if the magnetic coupling occurs in a situation where the collector current Ic is reduced, the resistor 32 prevents the gate charge from being discharged. This is because it can.

  FIG. 9 shows an example of overcurrent protection processing according to the present embodiment when magnetic coupling occurs. FIG. 9 shows the transition of the gate voltage Vge.

  As illustrated, when the charging process is started at time t1, the gate voltage Vge starts to increase. Thereafter, when it is determined that the gate voltage Vge has reached the predetermined voltage Vα at time t2, the clamping process is started.

  After the sense voltage Vse exceeds the short-circuit threshold SC, the gate voltage Vge once greatly exceeds the clamp voltage Vclamp in the vicinity of time t3. Thereafter, at time t4, the gate voltage Vge starts to decrease toward the clamp voltage Vclamp, and thus the collector current Ic also starts to decrease. Here, in the present embodiment, since the resistor 32 is provided in the clamp path Lclamp, the gate charge discharge is prevented even when the magnetic coupling occurs in a situation where the collector current Ic decreases. Can do. For this reason, the gate voltage Vge converges to the clamp voltage Vclamp without sharply dropping. After that, at time t6, the soft cutoff switching element 42 is turned on, and the switching element S ¥ # can be switched to the OFF state by the soft cutoff processing.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) In the drive circuit DU in which the constant current control process is performed, the resistor 32 is provided in the clamp path Lclamp. For this reason, even when magnetic coupling occurs in a situation where the collector current Ic decreases, the discharge of the gate charge via the clamping path Lclamp can be prevented by the resistor 32. As a result, the switching element S ¥ # can be switched to the off state by the soft shut-off process in a situation where an overcurrent flows through the switching element S ¥ #. Therefore, it is possible to avoid a decrease in the reliability of the switching element S ¥ #.

  (2) The predetermined voltage Vα is set to the maximum value Vthmax of the threshold voltage. If the gate voltage Vge is limited by the clamp voltage Vclamp before the switching element S ¥ # is turned on, the charge to be supplied from the constant current power supply 24 to the gate is discharged through the clamp path Lclamp, and the gate voltage Vge is It becomes difficult to rise. In this case, there is a problem that the switching element S ¥ # cannot be switched to the ON state or the reliability of the switching element S ¥ # is reduced due to heat generated by the switching element S ¥ # being in the half-ON state. obtain.

  Here, according to the setting of the predetermined voltage Vα, the clamping process can be started after the switching element S ¥ # is switched to the ON state. For this reason, generation | occurrence | production of the problem mentioned above can be avoided. The half-on state of the switching element S ¥ # is a state in which the gate voltage Vge when the switching element S ¥ # is turned on is set to a voltage that drives the switching element S ¥ # in the saturation region. Here, the saturation region is the output characteristic in which the collector-emitter voltage Vce of the switching element S ¥ # is related to the collector current Ic, and the collector current Ic is the same regardless of the magnitude of the collector-emitter voltage Vce. It is an area that is substantially constant.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 10 shows the drive circuit DU according to the present embodiment. In FIG. 10, the same members as those shown in FIG. 2 are given the same reference numerals for the sake of convenience.

  As shown in the drawing, in this embodiment, a capacitor 54 as a “charge storage element” is connected in parallel to the resistor 32.

  FIG. 11 shows an example of overcurrent protection processing according to the present embodiment. FIG. 11 corresponds to FIG. 9 above.

  As shown in the drawing, after the charging process is started at time t1, it is determined that the gate voltage Vge has reached the predetermined voltage Vα at time t2, and the clamping process is started.

  After the sense voltage Vse exceeds the short-circuit threshold SC, the gate voltage Vge exceeds the clamp voltage Vclamp in the vicinity of time t3. Here, in the present embodiment, even when the charge is about to move to the gate due to magnetic coupling under the situation where the collector current Ic increases, a part of the charge can be absorbed by the capacitor 54.

  For this reason, in this embodiment, in addition to the effects obtained in (1) and (2) of the first embodiment, an effect that the overshoot of the gate voltage Vge with respect to the clamp voltage Vclamp can be suppressed can be obtained. . Thereby, the gate voltage Vge can be brought close to the clamp voltage Vclamp, and the collector current Ic during the period in which the gate voltage Vge is limited by the clamping process can be accurately suppressed.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, the resistance value of the resistor 32 is set larger than the resistance value shown in the first embodiment. This is because the magnetic coupling in the present embodiment is larger than the magnetic coupling in the first embodiment, and the amount of movement of the gate charge due to the magnetic coupling is larger. Even if the amount of movement of the gate charge due to magnetic coupling is large, the discharge of the gate charge can be further prevented by setting the resistance value of the resistor 32 large.

  FIG. 12 shows a drive circuit DU according to the present embodiment. In FIG. 12, the same members as those shown in FIG. 2 are given the same reference numerals for the sake of convenience. In the present embodiment, the constant current power supply 24 is referred to as a first constant current power supply 24a.

  As shown in the drawing, the first constant current power supply 24a is connected to the first terminal T1. The second constant current power supply 24b using the constant voltage power supply 22 as a power supply source is connected to the eighth terminal T8 of the drive IC 20. In the present embodiment, the constant current output from the second constant current power supply 24b is set to “½” of the constant current output from the first constant current power supply 24a. In the present embodiment, the first constant current power supply 24a and the second constant current power supply 24b constitute “constant current supply means”.

  The first terminal T1 or the eighth terminal T8 is selectively connected to the drain of the charging switching element 26 by the switch 56. The switch 56 is energized by the drive control unit 50.

  In the present embodiment, the resistance value of the resistor 32 is set to “2R”, which is twice the resistance value shown in the first embodiment. This is because the gate charging current “Ig / 2” and the resistor are supplied during the period in which the gate charging current supply source is the second constant current power supply 24b (the period in which the gate voltage Vge is limited by the clamp voltage Vclamp by the limiting means). This is a setting for setting the multiplication value of the resistance value “2R” of 32 as the clamp voltage Vclamp. That is, the resistance value “2R” of the resistor 32 is set to a value obtained by dividing the clamp voltage Vclamp by the constant current “Ig / 2” output from the second constant current power supply 24b.

  Next, an example of overcurrent protection processing according to the present embodiment will be described with reference to FIG. Here, FIGS. 13 (a) to 13 (d), FIG. 13 (f), and FIG. 13 (g) correspond to the previous FIGS. 4 (a) to 4 (f). Moreover, FIG.13 (e) shows transition of the constant current which flows into charging path Lcha.

  In the illustrated example, the first constant current power supply 24a and the charging switching element 26 are connected by operating the switch 56 at time t1. As a result, the supply of the constant current from the first constant current power supply 24a to the gate is started, so that the gate voltage Vge starts to rise.

  Thereafter, when it is determined that the gate voltage Vge has reached the predetermined voltage Vα at time t2, the second constant current power supply 24b and the charging switching element 26 are connected by the operation of the switch 56. Thereby, the supply source of the gate charging current is switched from the first constant current power supply 24a to the second constant current power supply 24b. In this case, although the charging current is halved, the gate voltage Vge can be limited by the clamp voltage Vclamp because the resistance value of the resistor 32 is set to “2R”. After that, at time t3, the soft cutoff switching element 42 is turned on.

  Incidentally, in this embodiment, the process of operating the switch 56 constitutes a “decreasing means” so that the charging current is as shown in FIG.

  According to the present embodiment described above, the following effects can be obtained in addition to the effects (1) and (2) of the first embodiment.

  (3) A constant current supplied to the gate in a period in which the gate voltage Vge is limited by the clamp voltage Vclamp by the clamping process, and a period in which the gate voltage Vge is not limited by the clamp voltage Vclamp in the period in which the charging process is performed The constant current supplied to the gate is “½”. For this reason, the resistance value of the resistor 32 for securing the same clamp voltage Vclamp can be set large. Thereby, it is possible to further increase the effect of preventing the discharge of the gate charge in a situation where magnetic coupling occurs.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  The “impedance part” is not limited to the resistor 32 but may be, for example, a switching element (MOSFET, hereinafter referred to as an adjustment switching element). In this case, for example, in the first embodiment, an adjustment switching element is provided instead of the resistor 32. Then, during the clamp process, the gate voltage of the adjustment switching element may be adjusted so that the ON resistance of the adjustment switching element is “R” depending on the half-on state of the adjustment switching element. In this case, the adjustment switching element is a member having the highest impedance in the clamp path Lclamp when the gate charge is discharged by the clamp path Lclamp.

  The half-on state of the adjustment switching element is a state in which the gate voltage Vgs of the adjustment switching element when the adjustment switching element is turned on is set to a voltage that drives the adjustment switching element in the saturation region. . Here, the saturation region is a region where the drain current Id is substantially constant regardless of the magnitude of the drain-source voltage Vds of the adjustment switching element in the output characteristics. When the adjustment switching element is driven in the saturation region, the on-resistance of the adjustment switching element increases as compared with the full-on state.

  Further, the “impedance portion” is not limited to one using an element such as a resistor or a MOSFET. For example, a portion having a resistance value larger than that of other portions in the clamp path Lclamp by reducing the width of the wiring pattern as the clamp path Lclamp or lengthening the wiring pattern may be used as the impedance portion.

  The “soft blocking means” is not limited to one that decreases the gate charge discharge rate by increasing the resistance value of the gate charge discharge path. For example, it may be described in the following (A) and (B).

  (A) In FIG. 2, the soft cutoff resistor 40, the fourth terminal T4, and the soft cutoff switching element 42 are removed. Then, a power source is connected to a connection point between the third terminal T3 and the discharging switching element 30 via a switching element (for example, a MOSFET). Further, a configuration in which the discharge rate of the gate charge during the soft cutoff process is made lower than the normal discharge rate by turning on the switching element to supply charges to the connection point may be used as the soft cutoff means. Good. This utilizes the fact that the gate charge is prevented from being discharged by supplying a charge from the power source to the connection point.

  (B) In FIG. 2, the soft cutoff resistor 40, the fourth terminal T4, and the soft cutoff switching element 42 are removed. Then, the source of the discharge switching element 30 is selectively selected from either the emitter of the switching element S ¥ # or a part having a higher potential than the emitter (for example, a power source having an output potential higher than the emitter potential). Are connected by an energizing operation type switching element (for example, MOSFET). Then, the soft shut-off means has a configuration in which the discharge rate of the gate charge is lowered by connecting the source of the discharge switching element 30 and the portion having the high potential by the energization operation of the switching element during the soft shut-off process It may be used.

  The “charge storage element” is not limited to a capacitor, and may be another charge storage element as long as it has a function of storing charges.

  In the third embodiment, the resistance value of the resistor 32 may be set to “3R”, for example. In this case, the constant current output from the second constant current power supply 24b may be set to “1/3” of the constant current output from the first constant current power supply 24a.

  The “driven switching element” is not limited to a single IGBT, and may be a parallel connection body of a plurality of IGBTs. Specifically, the collectors of the plurality of IGBTs may be connected to each other, and the emitters may be connected to each other. Such a configuration is employed in order to increase the maximum value of the collector current Ic that can flow through the drive target switching element. Further, the “driven switching element” is not limited to the IGBT but may be a MOSFET, for example. In this case, the first terminal is the drain and the second terminal is the source.

  24 ... constant current power source, 32 ... resistor, 34 ... switching element for clamping, Lclamp ... path for clamping, S ¥ # ... switching element.

Claims (8)

Constant current supply means (24; 24a, 24b) for supplying a constant current to the open / close control terminal of the drive target switching element in order to switch the drive target switching element (S ¥ #) to the ON state;
A clamp path (Lclamp) connected to the open / close control terminal;
An impedance portion (32) provided in the clamp path, and having the highest impedance in the clamp path when the charge of the switching control terminal is discharged by the clamp path;
In a situation where a constant current is supplied to the open / close control terminal by the constant current supply means, the output current of the constant current supply means is supplied for a predetermined time before the voltage of the open / close control terminal reaches the upper limit voltage. Limiting means (34, 36, 38) for limiting the voltage of the open / close control terminal with a clamp voltage lower than the upper limit voltage due to a voltage drop in the impedance unit accompanying the circulation to the impedance unit;
With
A current flow path connected to the open / close control terminal so that the charge of the open / close control terminal moves to the clamping path side in a situation where the current flowing between the pair of main terminals of the drive target switching element decreases. A drive circuit for a drive target switching element, wherein a main current flow path connected to at least one of the pair of main terminals is magnetically coupled.   The drive circuit of the drive target switching element according to claim 1, wherein the impedance unit is a resistor. The limiting means limits the voltage of the open / close control terminal with the clamp voltage over the predetermined time after the voltage of the open / close control terminal becomes a predetermined voltage,
3. The drive target according to claim 1, wherein the predetermined voltage is set to a voltage that is equal to or higher than a threshold voltage at which the drive target switching element switches from an off state to an on state and is less than the clamp voltage. Switching element drive circuit.   The drive circuit of the drive target switching element according to claim 1, further comprising a charge storage element connected in parallel to the impedance unit and having a function of storing charge.   5. The drive circuit for a drive target switching element according to claim 4, wherein the charge storage element is a capacitor. The constant current supply means (24a, 24b) has a function of making the constant current supplied to the open / close control terminal variable,
A constant current supplied from the constant current supply means to the open / close control terminal during a period in which the voltage of the open / close control terminal is restricted by the limiting means, and a constant current from the constant current supply means to the open / close control terminal. 2. The apparatus according to claim 1, further comprising a lowering unit configured to reduce a constant current supplied to the open / close control terminal during a period during which the voltage of the open / close control terminal is not limited by the limiting unit. The drive circuit of the drive object switching element of any one of -5. The pair of main terminals as a first main terminal and a second main terminal,
The driving target switching element is switched to an on state by setting the voltage of the switching control terminal to be equal to or higher than a threshold voltage by increasing the potential of the switching control terminal with respect to the potential of the second main terminal,
The second main terminal is connected to the opposite side of the both ends of the clamp path to the side to which the open / close control terminal is connected,
The limiting means is
A clamping switching element (34) provided in the clamping path and opened and closed to open and close the clamping path;
Clamping operation means (36, 38) for turning on and off the clamping switching element so that the voltage drop amount in the impedance section is the clamping voltage;
The drive circuit for the drive target switching element according to claim 1, comprising:   The driving is performed when the overcurrent does not flow under the condition that the state where the overcurrent flows between the pair of main terminals is continued for a specified time under the situation where the voltage of the switching control terminal is limited by the limiting means. Soft shut-off means (40) forcibly switching the drive target switching element to the OFF state by discharging the charge at a discharge rate lower than the charge discharge rate of the switching control terminal when the target switching element is switched to the OFF state. , 42, 46, 48), the drive circuit for the drive target switching element according to claim 1.

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