JP2014079031A - Motor controller and refrigerator using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent erroneous detection of position information during a commutation overlapping period in a system where a current flowing in a motor changes due to a fluctuation of load torque and the length of a commutation overlapping period varies, thereby realizing a sensorless drive scheme capable of high efficiency drive from an extremely low speed region near a zero speed.SOLUTION: Provided is a motor controller 100 for operating a permanent magnet motor 2 by 120-degree conduction, wherein the motor controller is equipped with a primary delay filter device for attenuating an input value, which is a motor current flowing in a phase set to be the non-conducting phase of 120-degree conduction assumed as an initial input value for the primary delay filter, to zero or to near zero by a time constant equivalent to an electrical time constant of the entire inverter circuit including the permanent magnet motor 2. The end of the commutation overlapping period of the inverter is determined from the output value of the primary delay filter device, and position estimation is performed by determining whether a terminal voltage in the non-conducting phase of the permanent magnet motor 2 has exceeded a conduction mode switching threshold after the end of the commutation overlapping period. In this way, the non-conducting phase of the inverter 1 is set from the result of the position estimation.

Description

  The present invention relates to a motor control device that drives a permanent magnet motor and a refrigeration apparatus using the motor control device.

  Compact and highly efficient permanent magnet motors (synchronous motors) are widely used in motor control devices for compressors used in refrigerators and air conditioners.

  However, in order to drive the permanent magnet motor, position information of the rotor of the motor is required, and a position sensor for that purpose is required. In recent years, sensorless control that eliminates this position sensor and performs rotation speed and torque control of a permanent magnet motor has become widespread.

  The practical use of sensorless control has the advantage that the cost of the position sensor can be reduced and the size of the apparatus can be reduced. Further, in a compressor, it is difficult to attach a position sensor due to its structure, and it is necessary to drive a permanent magnet motor by sensorless control.

  Currently, in sensorless control of a permanent magnet motor, a method of directly detecting an induced voltage (speed induced voltage) generated by rotation of the rotor of the permanent magnet motor and driving the permanent magnet motor as position information of the rotor, A position estimation technique that estimates and calculates the rotor position from a mathematical model of a permanent magnet motor to be controlled is employed.

  However, since these sensorless controls are based on the induced voltage generated by the permanent magnet motor, there is a problem that the detection sensitivity decreases when the motor is stopped or in a low speed region where the induced voltage is small. Various solutions have been proposed.

  In the method described in Japanese Patent Laid-Open No. 2009-189176 (Patent Document 1), in controlling the energization of a permanent magnet motor by 120 degrees, the magnetic saturation state changes depending on the rotor position in the non-energized phase of the three-phase stator winding. The rotor position information is obtained by observing the change in the electromotive voltage that accompanies (hereinafter referred to as the open phase electromotive voltage), and by switching the energized phase, 120-degree energization from the extremely low speed region near the zero speed is performed. Control is realized.

JP 2009-189176 A

  In the method described in Patent Document 1, a commutation spike voltage is generated at a non-conduction phase terminal after switching an energized phase, and the commutation spike voltage is generated (hereinafter referred to as a commutation overlap period). The position information cannot be obtained from the open phase electromotive voltage of the non-energized phase. Therefore, it is necessary to observe the open phase electromotive voltage after the commutation overlap period.

  As a method of determining whether this commutation overlap period has ended, a method of providing a certain waiting time after switching the energized phase and observing the terminal voltage of the non-energized phase after the elapse of the waiting time is common. Since the commutation overlap period is determined by the magnitude of the current flowing through the inverter and the electrical time constant of the entire inverter circuit including the motor, if these factors change over time, the commutation overlap period It was difficult to uniquely determine the waiting time.

  For example, when a certain waiting time is provided for a system in which the generation time of the commutation overlap period changes, the position information cannot be correctly detected in a state where the generation time of the commutation overlap period is longer than the waiting time. Control may become unstable and the motor may stop.

  Therefore, the present invention estimates the length of the commutation overlap period that occurs during 120-degree conduction control, and observes the change in the open-phase electromotive voltage that occurs in the non-conduction phase of the three-phase stator winding at an appropriate timing. Accordingly, an object of the present invention is to realize a sensorless driving system capable of high-efficiency driving from an extremely low speed region near zero speed even in a system such as a compressor in which the commutation overlap period changes.

In order to solve the above problems, for example, the configuration described in the claims is adopted. One feature of the present invention is that, in a motor control device that operates a permanent magnet motor by energization at 120 degrees, the motor current flowing in a phase set as a non-energized phase at 120 degrees energization is used as an initial input value of the first-order lag filter. A first-order lag filter that attenuates the input value to zero or near zero with a time constant equivalent to the electrical time constant of the entire inverter circuit including the magnet motor, and commutation overlap of the inverter from the output value of the first-order lag filter Determining the end of the period, performing position estimation by determining whether the value of the terminal voltage of the non-energized phase of the permanent magnet motor exceeds the energization mode switching threshold after the commutation overlapping period ends, and the position estimation From this result, the non-energized phase of the inverter is set.
The other features of the present invention are as described in the claims of the present application.

  According to the present invention, the length of the commutation overlap period that occurs when conducting 120-degree conduction control is estimated, and the change in the open-phase electromotive voltage that occurs in the non-conduction phase of the three-phase stator winding is observed at an appropriate timing. By doing so, it is possible to realize a sensorless driving method capable of high-efficiency driving from an extremely low speed region near zero speed even in a system such as a compressor in which the commutation overlap period changes.

It is an example of the circuit block diagram of the motor control apparatus in Example 1 of this invention. It is an example of the simplified diagram for demonstrating the switching system of the 120 degree | times energization control system in this invention. It is an example of the simplified diagram for demonstrating the electricity supply pulse with respect to the motor at the time of 120 degree | times electricity supply in this invention. An example of a simplified diagram for explaining the relationship between an electromotive voltage (open phase electromotive voltage) generated in a non-energized phase when the energization pulse shown in FIG. 3 in the present invention is applied to the motor and the rotational angle position of the rotor. It is. FIG. 3 is an example of a simplified diagram for explaining each relationship among an open-phase electromotive voltage characteristic with respect to a rotation angle position of a rotor, a gate signal of a switching element that constitutes an inverter 2, a conduction mode, and a switching phase. A simplified diagram for explaining the relationship between the energization mode, the non-energization phase, the open phase electromotive voltage of the non-energization phase corresponding to the energization mode, and the reference voltage and the switching trigger generation timing of the energization mode with respect to the rotation angle position of the rotor. It is an example. For explaining the relationship between the energization mode, the non-energized phase, the open phase electromotive voltage of the non-energized phase corresponding to the energized mode, and the reference voltage with respect to the rotation angle position of the rotor when the influence of the commutation overlap period is taken into consideration It is an example of a simplified diagram. It is an example of the simplified diagram for demonstrating the relationship between the load torque fluctuation | variation which arises by the suction of a compressor, and a compression process, and the length of a commutation overlap period. It is an example of the control block block diagram of the motor control apparatus in Example 1 of this invention. It is an example of the control block diagram of the commutation overlap period estimator used with the motor control apparatus in Example 1 of this invention. It is an example of the simplified diagram for demonstrating the characteristic at the time of driving a permanent magnet motor in Example 1 of this invention. It is an example of the simplified diagram for demonstrating the characteristic at the time of driving a permanent magnet motor in Example 1 of this invention. It is an example of the circuit block diagram of the motor control apparatus in Example 2 of this invention. It is an example of the control block block diagram of the motor control apparatus in Example 2 of this invention. It is an example of the control block diagram of the commutation overlap period estimator used with the motor control apparatus in Example 2 of this invention. It is an example of the simplified diagram for demonstrating the method to determine the completion | finish of the commutation overlap period in Example 3 of this invention. It is an example of the control block block diagram of the motor control apparatus in Example 3 of this invention. It is an example of the control structure block diagram of the commutation overlap period estimator used with the motor control apparatus in Example 3 of this invention. It is an example of the circuit structure for performing operation | movement verification of this invention. It is an example of the simplified diagram for demonstrating the operation | movement verification method of this invention. It is an example of the simplified diagram for demonstrating the operation | movement verification method of this invention.

  Embodiments of the present invention will be described below with reference to the drawings.

  An example of the circuit configuration in this embodiment is shown in FIG. In this embodiment, as shown in FIG. 1, an inverter 1 that outputs an alternating voltage, a motor 2 connected to the inverter 1, and a controller that controls the inverter 1 by outputting a pulse width modulation signal to the inverter 1. 3. The controller 3 receives the inverter DC current Idc detected using the shunt resistor 4 and the phase voltages Vu_in, Vv_in, Vw_in of the motor 2 detected by the open phase voltage detecting means 5 as inputs. Each switching element of the inverter 1 is driven by performing 120-degree energization control according to the above.

  First, in describing an embodiment of the present invention, a 120-degree energization control method used in the present embodiment will be described with reference to each drawing.

  In the 120-degree energization control method, the switching operation is performed for two phases among the three-phase upper and lower arms of the inverter 1 in FIG. Since switching is performed for a period of 120 degrees in a phase of 180 degrees in terms of electrical angle, this is called a 120-degree conduction control method.

  There are several switching methods. For example, one of the methods shown in FIG. 2 may be used. FIG. 2 conceptually shows the gate signals of the upper and lower arms in one electrical angle cycle. In the drawing, Gp means an upper arm gate signal, and Gn means a lower arm gate signal. These drive signals for the upper and lower arms are output from the controller 3 in FIG. In order to determine the voltage to be applied to the motor 2, it is necessary to consider three points: the magnitude of the voltage, the waveform of the voltage, and the phase of the voltage with respect to the rotor position of the motor 2.

  When driving the motor 2 with 120-degree energization, two phases to be energized are selected from the three-phase windings of the motor 2 and a pulse voltage is applied to generate torque. There are six possible combinations of two phases to be energized, and these are defined as energization mode 1 to energization mode 6, respectively.

  FIG. 3A shows an energization mode (corresponding to energization mode 3 to be described later) in the state of energizing from the V phase to the W phase, and FIG. 3B on the contrary, energizing from the W phase to the V phase. It is a figure which shows the electricity supply mode of a state.

  On the other hand, the electromotive voltage that appears in the non-energized phase (the U phase in FIG. 4) when the rotational angle position of the rotor is changed by one electrical angle cycle is as shown in FIG. It can be seen that the U-phase electromotive voltage (U-phase terminal voltage) varies depending on the rotation angle position.

  This electromotive voltage is a difference in the rate of change of magnetic flux generated in the V phase and the W phase, which is observed as a voltage in the U phase, which is a non-energized phase, and is different from the speed induced voltage. This is distinguished from the speed induced voltage and is called an open phase induced voltage.

  In FIG. 4, the open phase electromotive voltage when applying a positive pulse indicated by a solid line and the open phase electromotive voltage when applying a negative pulse indicated by a broken line are both higher than the speed induced voltage Emu. The speed-induced voltage is an electromotive voltage that changes in proportion to the rotational speed of the rotor as the name suggests. Therefore, the magnitude relationship between the speed induced voltage in the low speed region and the electromotive voltage of the non-energized phase is as shown in FIG.

  Therefore, if this open phase electromotive voltage is detected, a relatively large rotor position signal can be obtained when the rotational speed of the motor 2 ranges from near zero speed to a low speed range.

  When detecting the open phase voltage of the motor 2, the open phase voltage detection means 5 is used. In many application examples, a voltage dividing resistor is used because the terminal voltage of the motor exceeds the power supply voltage of the control unit. Thereafter, it is amplified by an operational amplifier or a buffer circuit is inserted for the purpose of protecting the control unit.

  FIG. 5 shows an open-phase electromotive force characteristic with respect to the rotational angle position θd of the rotor when the U phase, V phase, and W phase are non-energized phases, the gate signal of the switching element constituting the inverter 1, and the rotation of the motor 2 The rotation angle position θd of the child, the energization mode, and the switching phase relationship are shown.

  As can be seen from FIG. 5, the voltage pulses shown in FIGS. 3A and 3B are applied during the normal operation of the 120-degree conduction method. In the energization mode 3, the state shown in FIG. Two phases to be energized are switched every 60 degrees of electrical angle according to the mode rotation angle position θd. That is, the non-energized phase is also switched sequentially.

  In FIG. 5, in the states of FIGS. 3A and 3B, the energization mode corresponds to the energization mode 3 or the energization mode 6. In the energization mode 3 or the energization mode 6, since the U phase is a non-energization phase, the open-phase electromotive voltage can be detected as indicated by the thick line shown in the U-phase electromotive voltage waveform. That is, as the rotation angle position θd increases, it is possible to detect an open phase electromotive voltage that decreases in the minus direction in the energization mode 3 and increases in the plus direction in the energization mode 6.

  Similarly, in energization mode 2 and energization mode 5, a V-phase electromotive voltage waveform can be detected, and in energization mode 1 and energization mode 4, a W-phase electromotive voltage waveform can be detected.

  FIG. 6 shows the relationship among the energization mode, the non-energization phase, the open phase electromotive voltage of the non-energization phase corresponding to the energization mode, and the reference voltage with respect to the rotation angle position θd. Each time the energization mode is switched, the open phase electromotive voltage of the non-energized phase has a waveform that repeats an increase and decrease in positive and negative, respectively. Therefore, reference voltages (Vhp, Vhn) serving as threshold values are set on the positive side and the negative side, respectively, and the rotational angle position θd can be estimated from the magnitude relationship between the reference voltage and the open-phase electromotive voltage of the non-conduction phase. A trigger signal for switching the energization mode is generated.

  That is, the reference voltage is used as a threshold value indicating a predetermined phase for switching the energization mode (hereinafter referred to as an energization mode switching threshold value). When the detected open phase electromotive voltage of the non-energized phase exceeds this threshold value, At the time, a mode switching trigger signal is generated to switch the energization mode in order. As described above, the open phase electromotive voltage is different from the speed induced voltage and can be detected even when the motor is stopped or rotating at an extremely low speed. Therefore, position sensorless driving is possible even when the rotational speed of the motor 2 is in the low speed range from near zero speed. Thus, by detecting the open phase electromotive voltage of the non-energized phase, the rotor position can be accurately detected even when the motor 2 is stopped or at extremely low speed. Based on this, the rotational speed is also obtained.

  However, when detecting the open-phase electromotive voltage, it is necessary to consider the period during which the spike voltage is generated by the return current during the switching operation of the inverter 1 (commutation overlap period). FIG. 7 shows the relationship of the open phase electromotive voltage of the energized mode, the non-energized phase, and the non-energized phase with respect to the rotation angle position θd when the commutation overlap period is taken into consideration. A period indicated by A in FIG. 7 is a spike voltage generation period, that is, a commutation overlap period. This spike voltage is immediately after switching to the energization mode (modes 1, 3, 5) in which the open-phase electromotive voltage decreases. It occurs in the negative direction and occurs in the positive direction immediately after switching to the energization mode (modes 2, 4, and 6) in which the open phase electromotive force increases.

  For example, when the energization mode is changed from 6 to 1, the spike voltage is generated in the negative direction and becomes equal to or lower than the negative reference voltage. Therefore, the condition for generating the mode switching trigger signal is satisfied at this time, and originally The energization mode to be switched at point B in FIG. 7 is switched at point C. In this case, the energization mode is not set correctly with respect to the rotation angle position θd, and the phase of the voltage applied to the motor 2 is shifted, so that the motor 2 cannot be operated normally.

  Therefore, when detecting the open phase electromotive voltage, the open phase electromotive voltage is not detected or compared with the energization mode switching threshold during the time from when the energization mode is switched to when the commutation overlap period ends. For example, it is necessary to prevent the energization mode from being erroneously switched.

  As a general method of waiting for the time until the commutation overlap period ends, a timer is prepared in the controller, the timer is started when the energization mode is switched, and the commutation is performed when a certain time has elapsed. Although it is conceivable to consider that the overlap period has ended, this may be difficult depending on the load of the inverter.

  This is because the commutation overlap period is determined by the magnitude of the current flowing through the inverter and the magnitude of the electrical time constant of the entire inverter circuit including the motor. The greater the current flowing through the inverter or the greater the electrical time constant, This is because the flow overlap period becomes longer.

  For example, in a compressor used in a refrigerator, an air conditioner, and the like, the load torque varies depending on the refrigerant suction and compression process, so that the current flowing through the compressor driving permanent magnet motor and the inverter also varies according to the load torque variation. . Therefore, in the compressor, as shown in FIG. 8, the length of the commutation overlap period changes due to the load torque fluctuation caused by the refrigerant suction and compression process, so the waiting time until the commutation overlap period ends is uniquely determined. Is difficult.

  In order to solve such a problem, it is necessary to estimate the length of the commutation overlap period and automatically adjust the waiting time. In this embodiment, paying attention to the fact that the length of the commutation overlap period varies depending on the current flowing in the inverter and the electric time constant of the entire inverter circuit including the motor, the current value flowing in the inverter is detected, and the detected current value A method of reproducing the return current flowing through the inverter during the commutation overlap period by performing a filtering process equivalent to an electric time constant on the inverter and determining the end of the commutation overlap period from the reproduced return current state will be described below. .

  FIG. 9 shows a control block diagram of the controller 3 in the present embodiment.

As shown in FIG. 9, the controller 3 receives the inverter DC current Idc detected by the shunt resistor 4 as an input, reproduces the current values Iu, Iv, Iw flowing through the UVW phases, and outputs the current reproducer 6; The commutation overlap period estimator 7 that estimates the commutation overlap period that occurs when the inverter energization mode is switched using the two phase voltages Vu_in, Vv_in, Vw_in and the output values Iu, Iv, Iw of the current reproducer 6 as inputs. A position estimator 8 for estimating the position of the rotor by using as input the respective phase voltages Vu_in, Vv_in, Vw_in of the motor 2 and the output of the commutation overlap period estimator 7 and the output of the energized phase selector 9; 8 includes an energized phase selector 9 that receives an output of 8 as an input and determines an energization mode of the inverter according to the position of the rotor, and includes a voltage command calculator and dq reverse conversion The drive signal of the inverter 1 is output from the PWM generator in accordance with the energization mode determined by the energization phase selector 9 so that the voltage command value of each UVW phase obtained by the generator is applied to the motor 2. In the inverter control component block, there are various calculation methods such as a case where a speed controller or a current controller is used in the process until the voltage command value is obtained. In this embodiment, a method for obtaining the voltage command value is provided. You can use any method. For example, the simplest configuration of the voltage command calculator 22 in this embodiment is such that the d-axis voltage command value Vd * is zero, the q-axis voltage command value Vq * is the motor speed command value ω * and the motor induced voltage. An example is a configuration obtained from the constant Ke by the arithmetic expression described in Expression 1.
Vq * = ω * × Ke (Formula 1)

  An example of a control block diagram of the commutation overlap period estimator 7 is shown in FIG. As shown in FIG. 10, the current selector 10 receives the output (energization mode) of the energized phase selector 9 and the current values Iu, Iv, and Iw flowing through the UVW phases, and the energized phase is changed from energized to de-energized. And output the current of the switched phase.

  An operation example of the commutation overlap period estimator 7 will be described with reference to FIGS. 10 and 11. As shown in FIG. 10, when the energized state of the U phase is switched to the non-energized state, the switch of the current selector 10 is set to the A side, and the U phase current value Iu is output. The output of the current selector 10 is input to the first-order lag filter 11 and is set as a filter input initial setting value.

  FIG. 11 shows the U-phase motor current, the U-phase return current estimation value, and the commutation overlap period estimation result with respect to the load torque fluctuation of the compressor. When the load torque of the compressor is large, the current amplitude of the U phase increases, so that the initial value of the estimated U-phase return current is also set to a large value. On the other hand, when the load torque of the compressor is small, the U-phase current amplitude is small and the initial value of the estimated value of the U-phase return current is also small. Further, as shown in part B, the first-order lag filter 11 attenuates the filter input initial set value to the filter output target value. At this time, the output target value of the first-order lag filter is generated during the commutation overlap period by setting the output target value of the first-order lag filter to zero and setting the filter time constant to the electric time constant of the inverter circuit including the motor. This corresponds to the return current of the inverter 1. Therefore, as shown in FIG. 12, the value of the return current at the end of the commutation overlap period (zero or near zero) is used as a commutation overlap period end determination threshold value, and is compared with the output value of the first-order lag filter by the comparator 12. This makes it possible to determine whether the commutation overlap period has ended.

  According to the present embodiment, even in a system such as a compressor in which the commutation overlap period changes, the current flowing through each phase of the motor 2 is passed through the first-order lag filter to reproduce the return current and the commutation overlap. Since the end of the period can be determined, it is possible to realize a sensorless driving method capable of high-efficiency driving from an extremely low speed region near zero speed.

  In refrigerators and air-conditioning equipment, rotor position information near the zero speed of the compressor can be detected, improving compressor startup performance and reducing power consumption through highly efficient sensorless drive from extremely low speed regions. Can be expected.

  In the present embodiment, in the motor control device described in the first embodiment, a solution to the case where the electrical time constant of the entire inverter circuit including the motor changes depending on the characteristics of the motor to be controlled and the temperature condition of the place of use. Will be described.

  When the current dependency of the motor inductance is high, a change in the inductance of the motor increases due to a change in current caused by a load applied to the motor. Since the electrical time constant changes due to this change in inductance, in the commutation overlap period estimator 7 described in the first embodiment, the accuracy of estimation of the return current estimated value by the first-order lag filter deteriorates. Further, when the ambient temperature of the motor changes, the resistance value of the motor winding changes, so that the electrical time constant also changes. Therefore, the estimation accuracy of the return current estimation value is deteriorated as in the case where the inductance is changed.

  In order to solve such a problem, the motor inductance and the winding resistance value are estimated from the current value flowing through the motor and the motor ambient temperature, and the commutation overlap period estimator 7 is used based on the estimation result. The filter time constant set for the first-order lag filter may be changed.

  An example of the circuit configuration in this embodiment is shown in FIG. In addition, description is abbreviate | omitted about the part which has the same code | symbol shown by Example 1 already demonstrated, and the part which has the same function.

  The present embodiment is different from the first embodiment in that a motor ambient temperature detection value Tm by a motor ambient temperature detector 13 configured by a general temperature sensor such as a thermistor or a thermocouple is input to the controller 3.

  FIG. 14 shows a control block diagram of the controller 14 in the present embodiment.

  As shown in FIG. 14, the controller 14 detects the d-axis current detection value Idc and the q-axis obtained by performing dq coordinate conversion on the current values Iu, Iv, and Iw flowing through the UVW phases outputted from the current reproducer 6. An electric time constant estimator 15 for outputting an estimated value Ts * of an electric time constant of the entire inverter circuit including the motor, using the current detection value Iqc and the motor ambient temperature detection value Tm output from the motor ambient temperature detector 13 as inputs. It is characterized by providing.

  As an example of a method for estimating the electrical time constant by the electrical time constant estimator 15, a dq axis inductance estimated value obtained by linearly approximating each inductance component of the dq axis of the motor from the amount of change in each of the detected current values Idc and Iqc of the dq axis. And a method for calculating the electrical time constant from the estimated motor winding resistance estimated using the detected motor ambient temperature Tm and the motor temperature coefficient, and the electrical time constant that varies depending on the ambient temperature of the motor and the magnitude of the current value. Can be set in advance as table data, and an appropriate value can be selected according to the input value of the electrical time constant estimator 15.

  The electrical time constant estimated value Ts * obtained as described above is used as an input to the commutation overlap period estimator 16.

  As shown in FIG. 15, the commutation overlap period estimator 16 sets the electrical time constant estimated value Ts * as the filter time constant of the first-order lag filter 17 and reproduces the return current of the inverter as in the first embodiment. Thus, the end of the commutation overlap period is determined.

  According to this embodiment, the end of the commutation overlap period can be determined even in a system in which the electrical time constant of the entire inverter circuit including the motor changes due to changes in the current flowing through the motor and the ambient temperature of the motor. It is possible to realize a sensorless driving method capable of high-efficiency driving from an extremely low speed region near the speed.

  In the present embodiment, different means for solving the same problem as in the first embodiment will be described.

  In this embodiment, in order to estimate the length of the commutation overlap period and automatically adjust the waiting time, in this embodiment, paying attention to the change in the open phase voltage before and after the commutation overlap period ends, the magnitude of the open phase voltage is changed. A method for determining the end of the flow overlap period will be described below.

  As shown in FIG. 16, the open-phase electromotive voltage during the commutation overlap period jumps positively and negatively with respect to the reference voltage due to the spike voltage. When the commutation overlap period ends, the spike voltage disappears, and the original open phase voltage can be observed. Therefore, when the commutation overlap period end determination threshold is set on the reference voltage side with respect to the open phase voltage during the commutation overlap period, and the open phase electromotive voltage is within this threshold range, the commutation overlap is determined. It can be assumed that the period has ended. For example, when the energization mode is changed from 6 to 1, the spike voltage is generated on the negative side. Therefore, while energization is performed in mode 1, it may be determined that the commutation overlap period has ended when the open phase electromotive voltage becomes greater than the commutation overlap period end determination threshold.

  FIG. 17 shows a control block diagram of the controller 3 in this embodiment. In addition, description is abbreviate | omitted about the part which has the same code | symbol shown by Example 1 already demonstrated, and the part which has the same function.

  The controller 3 is different from the first embodiment in a commutation overlap period estimator 18.

  FIG. 18 shows a control block diagram of the commutation overlap period estimator 18. The commutation overlap period estimator 18 receives the open phase electromotive voltages Vu_in, Vv_in, Vw_in of each UVW phase and the energization mode. According to the input energization mode, the open phase electromotive voltage selector 19 and the threshold selector 20 switch the open phase electromotive voltage and the commutation overlap period end determination threshold input to the comparator 21. The comparator 21 determines whether the open phase electromotive voltage has exceeded the commutation overlap period end determination threshold value, and outputs the commutation overlap period end determination result.

  For example, when the energization mode changes from 6 to 1, the non-energized phase is switched from the U phase to the W phase, so the switch of the open phase electromotive voltage selector 19 is set to the C side, and the open phase electromotive voltage Vw_in of the W phase is compared. To the device 21. Since the spike voltage is generated on the negative side, the switch of the threshold selector 20 is set to the B side, and the negative commutation overlap period end determination threshold is input to the comparator 21. When the open phase electromotive voltage Vw_in of the W phase exceeds the threshold value, the comparator 21 determines that the commutation overlap period has ended.

  According to the present embodiment, even in a system such as a compressor in which the commutation overlap period changes, the end of the commutation overlap period can be determined, so that high-efficiency driving from a very low speed region near zero speed is possible. A sensorless driving method can be realized.

  Further, since the end of the commutation overlap period can be determined only by the value of the open phase electromotive voltage of each phase of the motor 2, it is not necessary to take in current information from the shunt resistance or the like into the controller 3, and the number of inputs can be reduced. Therefore, the cost applied to the controller 3 can be suppressed.

  Many of the motor control devices related to each embodiment, and the drive devices, refrigerators, and air conditioner controllers 3 and controllers 14 using the semiconductor control circuits such as microcomputers and DSPs (arithmetic control means) ) And is often realized by software. Therefore, there is a problem that it is difficult to verify whether the controller is configured correctly. Therefore, in the present embodiment, a method for verifying whether the configuration related to each embodiment is operating correctly will be described based on the control configuration of the controller 3.

  In addition, description is abbreviate | omitted about the part which has the same function as the structure to which the same code | symbol shown in Example 1-Example 3 was attached | subjected.

  As shown in FIG. 19, the values that need to be measured for verification are the drive signal of each UVW phase of the inverter 1, the three-phase currents Iu, Iv, Iw of the AC output of the motor or inverter 1, and the UVW phase. Open-phase electromotive voltages Vu_in, Vv_in, and Vw_in.

  As an example of a method for detecting each measurement value, first, the drive signal can be measured by a potential difference from the reference potential of the controller 3. Further, the three-phase current can be measured by the current transformer 22. Further, the open phase electromotive voltage of each UVW phase can be measured by measuring the value input to the controller 3 and converting the voltage dividing resistance ratio.

  The DC voltage source 24 is connected to the input part of the open phase electromotive voltages Vu_in, Vv_in, Vw_in of the controller. The DC voltage source has terminals A, B, and C that independently output a DC voltage ranging from 0 V to the power supply voltage of the controller, and the open-phase electromotive voltage detection value input to the controller can be arbitrarily varied. can do.

  As a verification method, first, the drive signal of the inverter 1 is measured to determine which energization mode is in the state. The relationship between the drive signal and the energization mode is as shown in FIG.

  When the inverter is driven by energization at 120 degrees, a reflux current as shown in FIG. 20 can be observed immediately after switching from the energized phase to the non-energized phase in the three-phase current.

  In the configuration of the controller 3 described in the first embodiment, the return current is estimated, and the commutation overlap period ends when the estimated return current value becomes zero or near zero. During this time, the energization mode is not switched.

  Therefore, the open phase electromotive voltage detection value of the controller 3 is clamped as shown in FIG. In FIG. 20, ½ of the inverter DC power supply voltage is used as the 0V reference point. In the energization modes 1, 3, and 5, clamping is performed on the −Edc / 2 side, and in the energization modes 2, 4, and 6, clamping is performed on the Edc / 2 side.

  When clamping the open phase electromotive voltage detection value, the DC voltage source 24 is used. Since the open-phase induced voltage detection value input to the controller 3 is actually the output value of the open-phase voltage detection means 5 (for example, voltage dividing resistor), the value output from the DC voltage source 24 is Edc / The voltage value of 2 or −Edc / 2 may be the same value as the result obtained through the open phase voltage detection means 5.

  As shown in FIG. 20, when the open phase electromotive voltage detection value is clamped, the open phase electromotive voltage detection value exceeds the switching threshold value for the next energization mode when the energization mode is switched. . Therefore, if the invention of the present application is operating correctly, for example, after the energization mode is switched from 2 to 3, the energization mode is not switched until the commutation overlap period ends, that is, until the return current becomes zero or near zero. . Conversely, if the invention of the present application is not operating correctly, the energization mode is switched from 2 to 3, and then immediately switched to the next energization mode 4.

  In the configuration of the controller 3 described in Example 2, the return current is not estimated, and the commutation overlap period ends when the magnitude of the open-phase electromotive voltage detection value falls within the set threshold value range. Therefore, the energization mode is not switched while the open-phase electromotive voltage detection value is outside the set threshold range.

  Therefore, the open-phase electromotive voltage detection value of the controller 3 is varied as shown in FIG. In FIG. 21, 1/2 of the inverter DC power supply voltage is set as the 0V reference point. When clamping the open-phase electromotive voltage detection value, the energization modes 1, 3, and 5 are clamped to the -Edc / 2 side, and the energization modes 2, 4, and 6 are clamped to the Edc / 2 side.

  As an example of the verification method, when the energization mode is changed from 2 to 3, the open phase electromotive voltage detection value is set to −Edc until the return current becomes zero after the energization mode is switched, that is, until the commutation overlap period ends. The DC voltage source 24 is clamped to the / 2 side, and the value of the DC voltage source 24 is changed from the end of the commutation overlap period until the commutation overlap period end determination threshold is gradually exceeded.

  When the invention of the present application is operating correctly, the energization mode is not switched until the value of the DC voltage source 24 exceeds the commutation overlap period end determination threshold, and when the invention of the present application is not operating properly, After switching from 2 to 3, the current mode 4 is immediately switched to.

  As described above, whether or not the invention of the present application is operating correctly can be determined from the switching timing of the energization mode. Although the case where the energization mode is changed from 2 to 3 is described here as an example, the same verification can be made when each energization mode is switched.

  In addition, this invention is not limited to each above-mentioned Example, Various modifications are included. For example, each of the above-described embodiments has been described in detail for easy understanding of the present invention, and is not necessarily limited to one having all the configurations described. Further, a part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment. Further, it is possible to add, delete, and replace other configurations for a part of the configuration of each embodiment.

  Each of the above-described configurations, functions, processing units, processing procedures, and the like may be realized by hardware by designing a part or all of them with, for example, an integrated circuit. Each of the above-described configurations, functions, and the like may be realized by software by interpreting and executing a program that realizes each function by the processor.

  Although the motor has been described as a permanent magnet motor, other electric motors (for example, induction machines, synchronous machines, switched reluctance motors, synchronous reluctance motors, etc.) may be used. At that time, the calculation method in the voltage command value generator varies depending on the electric motor, but other methods can be applied in the same manner, and the object of the present invention can be achieved.

1. Inverter, 2. Permanent magnet motor, 3. Controller, 4. Shunt resistor, 5. Open-circuit voltage detection means, 6. Current reproduction device, 7. Commutation overlap period estimator, 8. Position estimator, 9. Energized phase selector, 10. Current selector, 11. first-order lag filter, 12. comparator, 13. 14. Motor ambient temperature detector, 14 controller, Electric time constant estimator, 16. 16. Commutation overlap period estimator; First-order lag filter, 18. 18. Commutation overlap period estimator, Open phase electromotive voltage selector, 20. Threshold selector, 21. Comparator, 22. Current transformer, 23. Verification means, 24. DC voltage source, 25. Energization mode measuring instrument, 26. Open phase electromotive force measuring instrument, 27. Three-phase current measuring device, 100. Motor control device

Claims (6)

  In a motor control device that operates a permanent magnet motor by energization at 120 degrees, the entire inverter circuit including the permanent magnet motor is obtained by setting the motor current flowing in the phase set to the non-energized phase of 120 degrees energization as the initial input value of the primary delay filter. A first-order lag filter that attenuates the input value to zero or near zero with a time constant equivalent to the electrical time constant of the inverter, determining the end of the commutation overlap period of the inverter from the output value of the first-order lag filter, From the end of the commutation overlap period, position estimation is performed by determining whether the value of the terminal voltage of the non-energized phase of the permanent magnet motor exceeds the energization mode switching threshold, and the inverter is de-energized from the position estimation result. A motor control device characterized by setting a phase.   A function of estimating an inductance and a resistance value of a winding from a current value flowing through the permanent magnet motor and an ambient temperature of the permanent magnet motor, and changing a filter time constant set in the first-order lag filter based on the estimation result The motor control device according to claim 1,   In a motor control device that operates a permanent magnet motor by energization at 120 degrees, the inverter commutation overlap period ends when the magnitude of the terminal voltage of the non-energized phase at 120 degrees energization falls within a certain range. Determining the position using the terminal voltage value of the non-energized phase of the permanent magnet motor from the end of the commutation overlap period, and setting the non-energized phase of the inverter from the result of the position estimation. A motor control device. An energization mode measuring device that detects a drive signal of each UVW phase of the inverter and determines a state of an energization mode; a current detector that measures a three-phase current of the AC output of the permanent magnet motor or the inverter; An open phase electromotive voltage measuring device that measures the open phase electromotive voltage of the motor, and a DC voltage source that varies the detection voltage of the open phase electromotive voltage detection unit of the motor control device,
When the voltage value of the open-phase electromotive voltage detection unit of the phase switched from the energized phase to the non-energized phase is changed by the DC voltage source during 120-degree energization driving, the magnitude of the return current of the inverter is zero or zero 3. The motor control device according to claim 1, wherein the non-energized phase of the inverter is switched when the voltage value in the vicinity and the open phase electromotive voltage detection unit exceeds the energization mode switching threshold.   An energization mode measuring device that detects a drive signal of each UVW phase of the inverter and determines a state of an energization mode; a current detector that measures a three-phase current of the AC output of the permanent magnet motor or the inverter; An open-phase electromotive force measuring device that measures the open-phase electromotive voltage of the motor and a DC voltage source that varies the detection voltage of the open-phase electromotive voltage detection unit of the motor control device. When the voltage value of the open phase electromotive voltage detection unit of the phase switched to the energized phase is changed by the DC voltage source, the magnitude of the return current of the inverter is zero or near zero, and the open phase electromotive voltage detection unit When the voltage value of the inverter exceeds the commutation overlap period end determination threshold value and exceeds the energization mode switching threshold value, the non-energized phase of the inverter is switched. The motor control device according to claim 3.   A refrigeration apparatus comprising the motor control device according to any one of claims 1 to 5.