JP2011188428A - Transmitting and receiving circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To mitigate receiving sensitivity deterioration caused by a transmitting leak signal leaking into a receiving circuit. <P>SOLUTION: In a transmitting and receiving circuit including an Ich mixer 11 and a Qch mixer 12 for performing quadrature demodulation, a signal of the same signal generating source as a transmitting leak signal is used as a receiving local signal of the Ich mixer 11 and the Qch mixer 12, and phases of the transmitting leak signal and the receiving local signal are set into the relationship of an odd-number multiple of 45°, thereby uniformly distributing an amplitude noise component that the transmitting leak signal has to the Ich mixer 11 and the Qch mixer 12. Thus, amplitude noise is prevented from extremely eccentrically appearing at a side selected for data demodulation between the Ich mixer 11 and the Qch mixer 12, and receiving sensitivity deterioration can be prevented. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a transmission / reception circuit, and more particularly to a communication device that performs transmission and reception at the same carrier frequency at the same time, such as a reader / writer in an RFID (Radio Frequency Identification) system. The present invention relates to a transmission / reception circuit having wraparound.

In recent years, there is a communication system called RFID. This is a system in which a reader / writer device as a parent device communicates with a small child device generally called a tag using RF (Radio Frequency), that is, high-frequency radio. In particular, an RFID system using a frequency band around 830 MHz to 960 MHz as an RF frequency is called a UHF band RFID.
FIG. 14 is a diagram illustrating an example of a UHF band RFID reader / writer transceiver using the direct conversion method. The transceiver shown in the figure has a configuration of a general ASK (Amplitude shift keying) modulation type transceiver.

  In the figure, the transmitter / receiver amplifies the local frequency signal generator 1 that generates a local signal, a TX mixer 2 that multiplies (multiplies) the transmission signal and the transmission local signal TXLO, and an output signal of the TX mixer 2. Power amplifier 3, a transmission / reception separator 4 such as a coupler or a circulator, an antenna 5, a low-noise amplifier (hereinafter referred to as LNA) 6 that amplifies a received signal from the antenna 5, and reception amplified by the LNA 6. An Ich mixer 11 that multiplies (multiplies) signal RFIN with Ich local signal IchRXLO, and a Qch mixer 12 that multiplies (multiplies) reception signal RFIN amplified by LNA 6 with Qch local signal QchRXLO.

  In such a configuration, the transmission signal is upconverted to the RF frequency by multiplying the transmission local signal TXLO generated by the local frequency signal generation unit 1 by the TX mixer 2 which is an analog multiplication circuit, amplified by the power amplifier 3, The transmission signal TXRF is a high-frequency signal. The transmission signal TXRF is output to the air at the antenna 5 via the transmission / reception separator 4.

  On the other hand, the high-frequency received signal RXRF is received from the air by the antenna 5 and input to the receiving side of the transceiver via the transmission / reception separator 4. Received signal RXRF is amplified by LNA 6, converted to a baseband frequency by a quadrature demodulator (Ich mixer 11 and Qch mixer 12), passed to a baseband signal processing unit (not shown) in the subsequent stage, and data is extracted. Is done. The local signals used in the quadrature demodulator are an Ich local signal IchRXLO and a Qch local signal QchRXLO whose phases are different from each other by 90 degrees. The Ich local signal IchRXLO and the Qch local signal QchRXLO are generated by the local frequency signal generator 1.

Here, the received signal RXRF is represented as P _AM (t) · sin (2πft + θ), the Ich local signal IchRXLO is represented as sin (2πft + φ), and the Qch local signal QchRXLO is represented as cos (2πft + φ) as an amplitude-modulated high-frequency signal. . Where f is the frequency of the RF signal, t is the time, θ is the phase of the received RF signal, P _AM (t) is the amplitude-modulated data component carried on RXRF, and φ is the phase of the LO signal.
At this time, the received signal P _Ich that appears at the output of the Ich mixer and the received signal P _Qch that appears at the output of the Qch mixer can be expressed by Expression (1).

  As shown in equation (1), the power of the received signal appearing at Ich and Qch differs depending on the phase relationship between the input to Ich mixer and Qch mixer (received signal RFIN) and the local signal. For this reason, in a system in which the phase of the high-frequency received signal RXRF cannot be predicted, there are cases where reception is not possible with a receiving circuit having only one of the Ich mixer and the Qch mixer. Therefore, the IQ receiver having the Ich mixer and the Qch mixer Must be used.

  In such an RF transceiver, for example, in a system in which a transmission RF carrier frequency and a reception RF carrier frequency, such as RFID, use the same frequency and a transmission carrier is simultaneously output during reception, as shown in FIG. Thus, the transmission signal TXRF is not sufficiently separated by the transmission / reception separator or is reflected by the antenna and leaks into the reception circuit. Here, the transmission leak signal leaking into the receiving circuit is referred to as TXRF leak. In the RFID system, this TXRF leak becomes noise (that is, a disturbing wave signal) and affects reception sensitivity.

  When the TXRF leak is considered as noise to the receiving side, this is mainly divided into a phase noise component due to temporal variations in phase and frequency, and an amplitude noise component in which the amplitude of the RF signal varies with time. Most of the phase noise component is generated in the transmission local frequency signal generation unit, and the amplitude noise component is often generated in the transmission baseband signal generation unit and an amplification stage such as a power amplifier.

The phase noise can be canceled by the correlation by using the signal generated by the same local frequency signal generation unit as the transmission side as the local for down-conversion of the reception mixer (Ich mixer and Qch mixer). This is described in Patent Document 1, for example. This fact is also expressed as follows.
When the phase noise component of the TXRF leak included in the reception signal RFIN is N _RF (t) and the phase noise component of the IchRXLO is N _LO (t), and only the phase noise component of the TXRF leak is considered, the TXRF leak is expressed by the equation (2). ) And IchRXLO can be expressed by equation (3).

At this time, the TXRF leak component P _Ich-TX after frequency conversion appearing in _Ich is expressed by Equation (4).

Since the TXRF leak and the local signal are signals generated by the same circuit, there is a time difference between the TXRF leak and the local signal, that is, a path time difference from the generation by the local frequency signal generation unit to the input to the mixer. When it is extremely small, it can be approximated as N _RF (t) = N _LO (t), so the TXRF leak component P _Ich-TX is expressed by equation (5), and the phase noise component is canceled.

Although the above has been described using an equation with Ich as an example, the same applies to Qch.
On the other hand, when the noise of TXRF leak is amplitude noise, the signal generated by the same local frequency signal generator as the transmission side is used as the local signal for down-conversion of the reception mixer (Ich mixer and Qch mixer) But it won't be canceled during down-conversion.

JP 2003-174388 A

As described above, in the case of a system that uses the same carrier frequency for transmission and reception as in RFID, for example, the phase noise component of the TXRF leak that has leaked into the receiver is removed by the above-described cancellation (hereinafter simply referred to as cancellation). It is possible. However, since the amplitude noise component is not canceled by this method and is superimposed on the received signal, the reception sensitivity is affected. Therefore, in order to improve the reception sensitivity, it is necessary to reduce the amplitude noise component of the TXRF leak, that is, the amplitude noise component of the transmission signal TXRF. Generally, in order to reduce the noise of the transmitter, it is necessary to increase the power, and naturally there is a limit to the improvement.
The present invention has been made in view of this point, and an object thereof is a transmission / reception circuit capable of reducing sensitivity deterioration due to an amplitude noise component of a TXRF leak and improving the sensitivity of the receiver by a device on the receiver side. Is to provide.

  The transmission / reception circuit according to the present invention is a transmission / reception circuit having a reception circuit including an I mixer and a Q mixer whose phase of a local signal multiplied by 90 ° differs from that of the reception signal. The reception circuit includes an interference wave having the same frequency as the local signal. Phase adjusting means for adjusting a phase of any one of the jamming signal input to the I mixer and the Q mixer and the local signal input to the I mixer and the Q mixer; The interference wave signal is uniformly distributed to the I mixer and the Q mixer. Thereby, it is possible to prevent the amplitude noise from appearing on the side selected for data demodulation in the I mixer and the Q mixer, and to prevent the reception sensitivity from deteriorating.

  Further, the phase adjusting means has an odd multiple of 45 ° between the interfering wave signal input to the I mixer and the Q mixer and the local signal input to the I mixer and the Q mixer. As described above, the phase of any one of the interference wave signal and the local signal is adjusted in a range of 90 ° or more. By adjusting the phase difference to be an odd multiple of 45 °, the interference signal is evenly distributed to the I mixer and Q mixer, minimizing the effects of TXRF leak amplitude noise, and improving reception sensitivity. Can be made.

  Then, the phase adjusting means is configured to cause the difference between the phase of the interfering wave signal input to the I mixer and the Q mixer and the phase of the local signal to be an odd multiple of 45 °. You may adjust the phase of the local signal each input into Q mixer simultaneously. By simultaneously adjusting the phases of the local signals input to the I mixer and the Q mixer, the phase difference between the jamming signal and the local signals input to the I mixer and the Q mixer is an odd multiple of 45 °. , The phase can be adjusted.

  In addition, the phase adjustment unit may be configured to output the interference signal so that a difference between the phase of the interference signal input to the I mixer and the Q mixer and the phase of the local signal is an odd multiple of 45 °. The phase may be adjusted. By adjusting the phase of the interfering wave signal, the phase can be adjusted so that the phase difference between the interfering wave signal and the local signal input to the I mixer and the Q mixer is an odd multiple of 45 °.

  By the way, a DC detection unit for detecting a DC component of a signal down-converted by the I mixer and the Q mixer, and a phase adjustment control signal for generating a phase adjustment control signal corresponding to the DC component detected by the DC detection unit And a generation unit, and the phase adjustment unit may adjust the phase of the local signal by the phase adjustment control signal generated by the phase adjustment control signal generation unit. If comprised in this way, the production | generation of the phase adjustment control signal for implement | achieving said phase state can be performed easily.

  A DC detection unit for detecting a DC component of the signal down-converted by the I mixer and the Q mixer; and a phase adjustment control signal for generating a phase adjustment control signal corresponding to the DC component detected by the DC detection unit. And a generation unit, and the phase adjustment unit may adjust the phase of the interference wave signal by the phase adjustment control signal generated by the phase adjustment control signal generation unit. If comprised in this way, the production | generation of the phase adjustment control signal for implement | achieving said phase state can be performed easily.

  The DC detection unit includes a first low-pass filter that receives the output of the I mixer, a second low-pass filter that receives the output of the Q mixer, the first low-pass filter, and the second low-pass filter. An adder that adds the two signals that have passed through, a subtracter that subtracts the two signals that have passed through the first low-pass filter and the second low-pass filter, an addition result of the adder, and a subtraction result of the subtractor First and second comparators for respectively determining whether the phase is positive or negative, and the phase adjustment control signal generation unit generates the phase adjustment control signal according to a determination result of the first and second comparators. May be. With this configuration, the phase adjustment amount when one of the determination results is inverted after sweeping the phase adjustment amount can be selected as the optimum adjustment amount, and the absolute value of the DC value of the Ich mixer output and the Qch mixer output Can be set easily.

  The DC detection unit includes a first low-pass filter that receives the output of the I mixer, a second low-pass filter that receives the output of the Q mixer, the first low-pass filter, and the second low-pass filter. First and second analog-to-digital converters for converting the analog voltage values of the two signals passed through the digital signal to digital values, respectively, and the phase adjustment control signal generation unit includes the first and second analog-digital converters The phase adjustment control signal may be generated according to the digital value converted by the converter. With this configuration, the phase adjustment amount when one of the determination results is inverted after sweeping the phase adjustment amount can be selected as the optimum adjustment amount, and the absolute value of the DC value of the Ich mixer output and the Qch mixer output Can be set easily.

  According to the present invention, it is possible to minimize the influence of the amplitude noise of the TXRF leak, which cannot be canceled in the prior art, and to improve the reception sensitivity.

It is a figure which shows an example of the transmitter / receiver which concerns on the Example of this invention. It is a figure which shows an example of the receiving circuit which concerns on Example 1 of this invention. It is a figure which shows the relationship between the phase of a reception RF signal, and the amplitude noise power after down-conversion. It is a figure which shows an example of the receiving circuit which concerns on Example 2 of this invention. It is a figure which shows an example of the receiving circuit which concerns on Example 3 of this invention. It is a figure which shows an example of the receiving circuit which concerns on Example 4 of this invention. It is a figure which shows an example of the phase adjustment circuit which concerns on Example 4 and Example 5 of this invention. It is a figure which shows the relationship between the phase of the received RF signal which concerns on Example 1 of this invention, and the DC value input into a DC detection part. It is a figure which shows an example of the receiving circuit which concerns on Example 5 of this invention. It is a figure which shows an example of the receiving circuit which concerns on Example 6 of this invention. It is a figure which shows an example of the receiving circuit which concerns on Example 6 of this invention. It is a figure which shows an example of the DC detection part which concerns on Example 6 of this invention. It is a figure which shows an example of the DC detection part which concerns on Example 7 of this invention. It is a figure which shows an example of the transmitter / receiver of a general ASK modulation system.

Hereinafter, embodiments of the receiving circuit will be described with reference to the drawings. In the drawings referred to in the following description, the same parts as those in the other drawings are denoted by the same reference numerals.
(Configuration of entire transmission / reception circuit)
First, FIG. 1 shows a configuration example of the whole transceiver in the embodiment of the present invention. In the figure, unlike the configuration of FIG. 14, the transceiver of this example is provided with a distributor 7 and uses the output signal (ie, transmission signal) of the power amplifier 3 as the reception local signal RXLOIN to the reception circuit 100. The configuration of the receiving circuit 100 will be described in each embodiment described later.

  In the configuration of FIG. 1, a received signal is received from the air by an antenna 5 and input to a receiving circuit via a transmission / reception separator 4 such as a coupler or a circulator. The transmission signal is output to the air by the antenna 5 via the mixer 2, the power amplifier 3, and the transmission / reception separator 4, and simultaneously input to the reception circuit as the reception signal RFIN via the transmission / reception separator 4. At the same time, the transmission signal is input to the reception circuit 100 as a reception local signal RXLOIN via the distributor. According to this configuration, the phase noise component in the TXRF leak can be suppressed sufficiently low by cancellation. Here, as an example, in order to reduce the path time difference between the TXRF leak and the reception local signal RXLOIN, the reception local signal RXLOIN is transmitted from the transmission signal through the distributor 7, but the reception mixer 100 in the reception circuit 100 is down-converted. As long as the signal generated by the same local frequency signal generator 1 as that on the transmission side is used as a local signal, the same effect can be obtained even with another configuration.

  FIG. 2 is a diagram showing the configuration of the first embodiment of the present invention. The reception local signal RXLOIN in FIG. 2 is the same as the reception local signal RXLOIN shown in FIG. 2 is the same as the reception signal RFIN shown in FIG. The received local signal RXLOIN is converted into two local signals IchRXLO and QchRXLO having a phase difference of 90 ° by a phase shift circuit that distributes a signal such as a polyphase filter to signals having different phase states of 0 ° and 90 °. The The reception signal RXIN including the transmission leak signal is input to the I mixer and the Q mixer, and frequency-converted by being multiplied by the local signals IchRXLO and QchRXLO, respectively, and the signal RXBBI is output from the I mixer. A signal RXBBQ is output from the mixer.

Here, the amplitude noise of the TXRF leak is not canceled at the time of down-conversion even when a signal generated by the same local frequency signal generation unit as that used for transmission is used for down-conversion of the reception mixer. This situation can be understood by the following equations.
When the amplitude noise component of the TXRF leak included in the received signal RFIN is P _RF (t) and the amplitude noise component of the IchRXLO is P _LO (t), the TXRF leak included in the received signal RFIN can be expressed by Equation (6).

  In addition, the Ich local signal IchRXLO can be expressed by Expression (7).

At this time, the TXRF leak component P _Ich-TX after frequency conversion appearing in Ich is expressed by Equation (8).

Even when P _RF (t) = P _LO (t), Equation (9) is obtained, and the amplitude noise component does not disappear even when the time difference between the TXRF leak and the local signal is extremely small.

  However, the amplitude noise component of the TXRF leak is determined by the down-converted noise level according to the equation (8) according to the phase relationship with the local signal. FIG. 3 shows the relationship between the phase of the Ich signal and the Qch signal and the noise power. The horizontal axis in the figure is the phase difference between the TXRF leak and the Ich local signal IchRXLO, and the vertical axis in the figure is the IQ received signal when the amplitude noise component of the TXRF leak is normalized to “0 dB”. The noise power that appears. Since the Ich local signal IchRXLO and the Qch local signal QchRXLO used in the quadrature demodulator have a phase difference of 90 °, for example, when the phase of the TXRF leak matches the Qch local signal QchRXLO, as shown in “A” in FIG. All noise power appears in the Ich reception signal, ie, the signal RXBBI, and at the same time no noise appears in the Qch reception signal. Conversely, when the phase of the TXRF leak matches the Ich local signal IchRXLO, all noise appears in the Qch reception signal, that is, the signal RXBBQ, as indicated by “C” in FIG. Here, a case is considered in which the phase of the transmission RF leak signal is input with a phase different by an odd multiple of 45 ° with respect to the Ich local signal IchRXLO and the Qch local signal QchRXLO. For example, as indicated by “B” in FIG. 3, when the input is performed with a phase different by 45 ° × 3 = 135 °, the noise appearing in each of Ich and Qch is half that in the case where the phase matches either. Appears in both Ich and Qch at a level that is 3 dB smaller.

The relationship between the phase state and power can be applied to the received signal in the same manner as in the case of noise. However, since the phase state of the received RF signal changes depending on the communication distance, it is impossible to predict in which phase state it will be received. Therefore, the power distribution of the received signal component appearing in the received signal RXBBI and the received signal RXBBQ is different for each reception. Different.
Since the signals appearing in the reception signal RXBBI and the reception signal RXBBQ have the same data, it is sufficient to obtain only one of the information as the function of the receiver. However, as described above, since the power distribution of the received signal components appearing in the received signal RXBBI and the received signal RXBBQ is different for each reception, the receiver compares the signal power of both the received signal RXBBI and the received signal RXBBQ, and receives It is necessary to select the one where more signal components are allocated and perform data demodulation in the baseband signal processing unit.

  In such a situation, for example, the received signal is received in the phase state of “B” in FIG. 3 at the same level on Ich and Qch, and at the same time, the phase state of “A” in FIG. In this case, the Ich having a higher power is selected for data demodulation considering the received signal + noise. At this time, since the noise power is received at the maximum level biased to Ich, the S / N ratio (hereinafter referred to as S / N) is the lowest level.

  Here, for example, when the phase of the TXRF leak is set to the phase state of “B” in FIG. 3, even in the case of “B” in FIG. 3dB is good. Therefore, it can be said that the receiving sensitivity is improved by 3 dB as a receiving system. In other words, in the first embodiment of the present invention, the phase difference between the two local signals IchRXLO and QchRXLO and the received signal RXIN is an odd multiple of 45 °, so that each of the Ich and Qch is as described above. Distribution of the amplitude noise power of the appearing transmission leak signal is equally distributed between Ich and Qch, and reception sensitivity can be improved as compared with a state where noise is biased.

  FIG. 4 is a diagram showing the configuration of the second embodiment of the present invention. In the second embodiment, as shown in FIG. 4, the received local signal RXLOIN is input to the phase adjustment circuit 30 and its phase is adjusted. The phase adjustment circuit 30 has a phase adjustment width of 90 ° or more. The local signal RXLO whose phase is adjusted by the phase adjustment circuit 30 is input to the phase shift circuit 20, and two local signals IchRXLO and QchRXLO having a phase difference of 90 ° are generated. By using these as the Ich local signal IchRXLO to the Ich mixer 11 and the Qch local signal QchRXLO to the Qch mixer 12, the phase difference between the Ich local signal IchRXLO and the Qch local signal QchRXLO and the reception signal RXIN can be easily set to 45 °. The relationship can be an odd multiple.

  FIG. 5 is a diagram showing the configuration of the third embodiment of the present invention. In the second embodiment described above, RXLOIN is input to the phase adjustment circuit 30 and the phase thereof is adjusted. On the other hand, the third embodiment has a configuration in which the received signal RXIN is input to the phase adjustment circuit 30 and the phase thereof is adjusted. The reception signal RXIN2 whose phase is adjusted by the phase adjustment circuit 30 is input to the Ich mixer 11 and the Qch mixer 12, so that the phase difference between the Ich local signal IchRXLO and the Qch local signal QchRXLO and the reception signal RXIN2 can be easily obtained. The relationship can be an odd multiple of 45 °. Note that the phase adjustment circuit 30 used in this example also has a phase adjustment width of 90 ° or more.

  FIG. 6 is a diagram showing a configuration of the fourth embodiment of the present invention. The fourth embodiment provides an effective adjustment means for the phase adjustment circuit 30 of the second embodiment. The signal RXBBI that is the output of the I mixer 11 and the signal RXBBQ that is the output of the Q mixer 12 are input to the DC detection unit 40, and the DC component of the signal RXBBI and the signal RXBBQ is detected by the DC detection unit 40. The detection result of the DC component is input to the phase adjustment control signal generation unit 50, and the phase adjustment control signal generation unit 50 outputs the phase adjustment control signal for changing to the next phase state based on the detection result of the DC component. Output to the adjustment circuit 30.

  FIG. 7 shows a circuit configuration example of the phase adjustment circuit 30. In the figure, a phase adjustment circuit 30 includes an input port A and an input port B, a resistor R1 having one end connected to the input port A, a variable capacitor VC having one end connected to the input port A, and an input port B. A resistor R2 connected at one end, a resistor R3 connected at one end to the input port B, a positive input connected to the other end of the resistor R1 and the other end of the variable capacitor VC, and the other end of the resistor R2 and the resistor R3 And a differential buffer 31 having a negative input connected to the other end. The variable capacitor VC can be changed in capacity by a phase adjustment control signal (not shown).

When the input signal is a differential input, the differential signal is input to the input port A and the input port B. When the input signal is a single input, the input signal is input to port A, and a constant voltage bias such as a power supply voltage is applied to port B.
The port A is connected to one end of the resistor R1 and one end of the variable capacitor VC. The input port B is connected to one end of each of the resistor R2 and the resistor R3. The voltage appearing at the other ends of the resistors R1 and R2 is applied to the positive input of the differential buffer 31 as the differential signal P. The voltage appearing at the other end of the variable capacitor VC and the other end of the resistor R3 is applied as a differential signal N to the negative input of the differential buffer 31. By using the output of the differential buffer 31 as a differential signal, a signal with a phase change corresponding to the amount of adjustment of the capacity of the variable capacitor VC can be obtained.

The specific operation of phase adjustment in the present invention will be described below. It is assumed that a reception signal RXIN including TXRF leakage is input to the reception circuit in a certain phase state. At this time, since the TXRF leak uses the same signal generation source as the received local signal RXLOIN and has the same frequency, only the DC component and the amplitude noise component remain in the signal RXBBI and the signal RXBBQ.
FIG. 8 is a diagram of relative values of DC values appearing in the Ich reception signal (solid line in the drawing) and the Qch reception signal (broken line in the drawing) with respect to the phase state of the local signal. The horizontal axis in the figure is the phase of the TXRF leak included in the received signal RFIN when the phase of the Ich local signal is 0 °, and corresponds to the value of θ−φ in equation (4).

Also, the vertical axis in the figure is the DC value when the maximum DC value is normalized to “1”. From the equation (4), this DC value, that is, P _AM · cos (θ−φ) in the equation (4), and the amplitude noise amount of the TXRF leak, that is, P _RF (t) · cos (θ−) in the equation (4). (φ) is proportional. For this reason, if the phase of the local signal is adjusted so that the absolute value of the DC value is equal between Ich and Qch, the amount of noise also becomes equal between Ich and Qch. That is, the DC detection unit 40 appropriately adjusts the phase adjustment amount of the phase adjustment circuit 30 so that the DC components of the signal RXBBI and the signal RXBBQ are equal to each other, whereby the Ich local signal IchRXLO and the Qch local signal QchRXLO and the received signal are adjusted. The phase difference from RXIN can be easily set to an odd multiple of 45 °.

  FIG. 9 is a diagram showing the configuration of the fifth embodiment of the present invention. In the fourth embodiment, RXLOIN is input to the phase adjustment circuit to adjust the phase. However, in the fifth embodiment, RXIN is input to the phase adjustment circuit as shown in FIG. Further, by appropriately adjusting the phase adjustment amount of the phase adjustment circuit 30 so that the DC components of the signal RXBBI and the signal RXBBQ are equal, the phase difference between the Ich local signal IchRXLO and the Qch local signal QchRXLO and the reception signal RXIN2 is It is possible to easily achieve an odd multiple of 45 °.

10 and 11 are diagrams illustrating the configuration of the sixth embodiment. In the sixth embodiment, a specific circuit configuration for generating the phase adjustment control signal in the fourth or fifth embodiment is provided.
In FIG. 10, the DC detection part 40 in Example 4 is shown with the broken-line part in the figure. The DC detection unit 40 includes an Ich low-pass filter 41, a Qch low-pass filter 42, and a DC determination circuit 43.

  Moreover, in FIG. 11, the DC detection part 40 in Example 5 is shown with the broken-line part in the figure. The DC detection unit 40 includes an Ich low-pass filter 41, a Qch low-pass filter 42, and a DC determination circuit 43. The signal RXBBI and the signal RXBBQ are input to the DC determination circuit 43 as the signal RXDCI and the signal RXDCQ from which the harmonic component is removed by the Ich low-pass filter 41 and the Qch low-pass filter 42 and the DC component due to the TXRF leakage remains. .

  FIG. 12 is a diagram illustrating a configuration example of the DC determination circuit 43. As shown in the figure, the DC determination circuit 43 includes an adder 431 that adds the signal RXDCI and the signal RXDCQ, a subtracter 432 that subtracts the signal RXDCI and the signal RXDCQ, and an adder 431. A comparator 433 for performing positive / negative determination on the addition result and a comparator 434 for performing positive / negative determination on the subtraction result of the subtractor 432 are provided.

In this configuration, for each DC value of the signal RXDCI and the signal RXDCQ, a signal (I + Q) generated by the adder 431 and a signal (I−Q) generated by the subtractor 432 are prepared. Then, for each of the signal (I + Q) and the signal (I−Q), the binary comparators 433 and 434 perform positive / negative determination.
When the phase adjustment amount of the phase adjustment circuit 30 is swept by the phase adjustment control signal generation unit 50, the phase state changes and the determination result of one of the binary comparators 433 and 434 is inverted. That is, when the DC values of the signal RXDCI and the signal RXDCQ match, the determination result of one of the binary comparators 433 and 434 is inverted. Therefore, the phase adjustment control signal generation unit 50 sweeps the phase adjustment amount of the phase adjustment circuit 30 and selects the phase adjustment amount when one of the determination results is inverted as the optimum adjustment amount. Thereby, it is possible to easily set a phase where the absolute values of the DC values of the Ich mixer output and the Qch mixer output substantially coincide.

  In the sixth embodiment, the DC determination circuit 43 in FIG. 12 is used. On the other hand, in the seventh embodiment, two analog-digital converters for Ich and Qch (hereinafter referred to as ADC) are used instead of the DC determination circuit 43 in FIG. That is, as shown in FIG. 13, by measuring the DC value of the signal RXDCI and the DC value of the signal RXDCQ by the ADC 435 receiving the output of the adder 431 and the ADC 436 receiving the output of the subtractor 432, respectively. As in the case of the sixth embodiment, it is possible to easily set the phase at which the absolute values of the DC values of the Ich mixer output and the Qch mixer output substantially coincide.

(Summary)
In the present invention, in a transmission / reception circuit including an I mixer and a Q mixer that perform quadrature demodulation, the signal of the same signal generation source as the transmission leak signal is used as the I mixer and Q mixer reception local signal, and the transmission leak signal and the reception local signal are used. And the amplitude noise component of the transmission leak signal is uniformly distributed to the I mixer and the Q mixer. Thereby, it is possible to prevent the amplitude noise from appearing on the side selected for data demodulation in the I mixer and the Q mixer, and to prevent the reception sensitivity from deteriorating.
According to the present invention, even when the phase of a local or transmission leak signal changes due to a change in temperature or time due to an external factor, the phase state can always be monitored. Can be realized.

1 Local frequency signal generator 2 Mixer 3 Power amplifier 4 Transmitter / receiver separator 5 Antenna 6 LNA
7 Divider 11 Ich mixer 12 Qch mixer 20 Phase shift circuit 30 Phase adjustment circuit 31 Differential buffer 40 DC detection unit 41, 42 Low pass filter 43 DC determination circuit 50 Phase adjustment control signal generation unit 100 Reception circuit 431 Adder 432 Subtractor 433, 434 Comparator A, B Input ports R1-R3 Resistor VC Variable capacitance

Claims (8)

  In a transmission / reception circuit having a reception circuit including an I mixer and a Q mixer whose phase of a local signal multiplied by 90 ° is multiplied by a reception signal, an interference wave signal having the same frequency as that of the local signal is input to the reception circuit. A phase adjusting means for adjusting a phase of any one of the interference signal input to the mixer and the Q mixer and the local signal input to the I mixer and the Q mixer; A transmission / reception circuit characterized by being uniformly distributed to the I mixer and the Q mixer.   The phase adjusting unit is configured so that a phase difference between the interference signal input to the I mixer and the Q mixer and the local signal input to the I mixer and the Q mixer is an odd multiple of 45 °. The transmitter / receiver circuit according to claim 1, wherein the phase of one of the jamming wave signal and the local signal is adjusted in a range of 90 ° or more.   The phase adjusting means includes the I mixer and the Q mixer so that a difference between the phase of the interfering wave signal input to the I mixer and the Q mixer and the phase of the local signal is an odd multiple of 45 °. The transmitter / receiver circuit according to claim 1 or 2, wherein the phase of each of the local signals input to each is adjusted simultaneously.   The phase adjusting means adjusts the phase of the jamming signal so that the difference between the phase of the jamming signal input to the I mixer and the Q mixer and the phase of the local signal is an odd multiple of 45 °. The transmission / reception circuit according to claim 1, wherein the transmission / reception circuit is adjusted.   A DC detection unit for detecting a DC component of a signal down-converted by the I mixer and the Q mixer, and a phase adjustment control signal generation unit for generating a phase adjustment control signal corresponding to the DC component detected by the DC detection unit The transmission / reception circuit according to claim 3, wherein the phase adjustment unit adjusts the phase of the local signal by the phase adjustment control signal generated by the phase adjustment control signal generation unit.   A DC detection unit for detecting a DC component of a signal down-converted by the I mixer and the Q mixer, and a phase adjustment control signal generation unit for generating a phase adjustment control signal corresponding to the DC component detected by the DC detection unit 5. The transmission / reception circuit according to claim 4, wherein the phase adjustment unit adjusts the phase of the interference wave signal by the phase adjustment control signal generated by the phase adjustment control signal generation unit. .   The DC detection unit includes a first low-pass filter that receives the output of the I mixer, a second low-pass filter that receives the output of the Q mixer, the first low-pass filter, and the second low-pass filter. An adder that adds the two signals that have passed through, a subtracter that subtracts the two signals that have passed through the first low-pass filter and the second low-pass filter, an addition result of the adder, and a subtraction result of the subtractor First and second comparators for respectively determining the positive and negative values of the first and second comparators, wherein the phase adjustment control signal generation unit generates the phase adjustment control signal according to the determination results of the first and second comparators. The transmission / reception circuit according to claim 5 or 6.   The DC detection unit includes a first low-pass filter that receives the output of the I mixer, a second low-pass filter that receives the output of the Q mixer, the first low-pass filter, and the second low-pass filter. First and second analog-to-digital converters for converting the analog voltage values of the two signals passed through the digital signal to digital values, respectively, and the phase adjustment control signal generation unit includes the first and second analog-digital converters The transmission / reception circuit according to claim 5 or 6, wherein the phase adjustment control signal is generated according to a digital value converted by a converter.

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