JP2006352907A - Tuner - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a tuner capable of maintaining and improving reception performance over wide frequency bands, in details, to provide a tuner which ensures isolation of input/output of a SAW filter on an intermediate frequency amplification stage and comprises the SAW filter with high intra-band characteristics and cutoff characteristics. <P>SOLUTION: An intermediate frequency signal from a frequency converting means is supplied to an input terminal 61, and the intermediate frequency signal is amplified by an amplifier 62, then limited into a predetermined intermediate frequency band by a SAW filter 63 (or 70), further subjected to intermediate frequency amplification in an amplifier 65, and extracted in an output terminal 66 as a tuner output. A SAW filter 63 (or 70) of a metal package (or plastic package) is surrounded with a shield wall 64. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a tuner that receives digitally modulated signals (QPSK, 64QAM, OFDM, 8VSB, etc.) in CATV, satellite broadcasting, terrestrial broadcasting, and the like.

  With the recent digitalization and multimediaization, not only wireless broadcasting such as the current TV but also wired broadcasting such as CATV in which broadcasting and communication are integrated has attracted attention in the broadcasting field.

  Conventionally, a bidirectional data transmission system using a CATV transmission line has been developed and applied to services such as video-on-demand.

  On the other hand, since the CATV line has a larger capacity for data transmission than a telephone line, a network is formed using the CATV line, and the communication device (for example, a personal computer) of a subscriber terminal is connected to the Internet or the like at high speed. An accessible data communication service has been developed. In this service, an interface with a CATV line called a cable modem is installed in a subscriber's house, and a personal computer as a data communication device is connected to the cable modem. This enables a service in which a user connects to an external network such as the Internet via a CATV broadcasting center.

FIG. 15 is a block diagram showing a schematic configuration of a CATV broadcast system including a cable modem.

  In FIG. 15, the CATV broadcast center 201 includes a video supply device 202, a server device 204 connected to the external network 203, a signal conversion circuit 205 connected to the server device 204, and a modulated video from the video supply device 202. A signal is mixed with the modulated data signal from the signal conversion circuit 205, or a mixing / demultiplexing circuit 206 is provided for demultiplexing the mixed video signal and the data signal from the CATV line 207.

  In the CATV broadcasting center 201, a video signal accompanying a normal broadcasting service such as terrestrial broadcasting or satellite broadcasting wave is modulated from the video supply device 202 and supplied to the mixed demultiplexing circuit 206, and then sent to the CATV line 207. The The CATV line 207 is composed of a hybrid of an optical cable and a coaxial cable, and is connected to a plurality of (two in the illustrated case) subscriber terminals 208. The hybrid system of optical cable and coaxial cable is a system that uses an optical cable for a trunk line extending from the center and draws it into a subscriber's house using a coaxial cable.

  The CATV broadcasting center 201 is provided with a server device 204 connected to an external network 203 such as the Internet. The server device 204 performs management of subscribers, external security, network management, and the like. ing.

  Regarding the downstream of the signal conversion circuit 205 connected to the server device 204, data generation, modulation processing and frequency conversion of this CATV communication system are performed, supplied to the mixing / demultiplexing circuit 206, and output to the CATV line 207.

  The uplink signal transmitted from the subscriber terminal 208 is demodulated in the uplink data and converted into the format data for the server by the signal conversion circuit 205 via the mixing / demultiplexing circuit 206 from the CATV line 207. Supplied to the device 204.

  On the other hand, the subscriber terminal 28 receives the broadcast signal transmitted from the CATV line 27 by the set top box 211, performs the channel selection process and the demodulation process, and then the television receiver 212 as the display device. Reproduce.

  Further, the downstream signal sent from the broadcasting center 21 via the CATV line 27 is demodulated by the cable modem 209, then the control data for the cable modem is taken out, and the connected data communication device The necessary data is supplied to the personal computer 210.

  On the other hand, the uplink data sent from the personal computer 210 is modulated by the cable modem 209 and sent as an uplink signal to the CATV broadcast center 201 via the CATV line 207.

  As described above, this CATV broadcasting system is different from conventional subscriber terminals in that it not only receives broadcast signals from the broadcast center 201 by the CATV set-top box 211 but also actively transmits from the cable modem 209 of each terminal. Transmission to 201 can be performed.

  In the communication between the broadcast center 201 and the subscriber terminal 208, uplink / downlink data is transmitted / received by QPSK modulation having uplink and downlink frequency bands of 6 MHz and 1.5 MHz, respectively.

In the case of CATV broadcasting, a digitally modulated high-frequency signal, typically about 90 to 860 MHz, is distributed from the center station to each home via a cable, and this signal is input to the tuner in the set top box, and once to three times by the tuner. After frequency conversion to an intermediate frequency by frequency conversion, digital demodulation is performed.
In the case of a bidirectional CATV such as a cable modem, in addition to the above-described high-frequency signal of about 90 to 860 MHz (downstream signal), the digitally-modulated high-frequency signal such as QPSK or 16QAM of about 5 to 50 MHz ( Transmission is performed from each household to the center station using an upstream signal (Upstream). In this way, the band of 5 to 50 MHz is allocated in the direction from the subscriber side to the center station (that is, the direction opposite to the broadcast), and images such as subscriber homes and event venues are transmitted to the center, repeaters and CATV sets. It is used to notify the center of abnormalities in the top box.

The downlink signal and the uplink signal are frequency-separated by a demultiplexer. The duplexer needs to ensure sufficient isolation so that the downstream signal and upstream signal do not interfere with each other.
Alternatively, after these media are drawn into the home by a single coaxial cable, they are distributed by the distributor inside and outside the receiving device (terminal) in the home, and the RF of the tuner corresponding to each media in the set top box Is input to the department. The distributor needs to secure sufficient isolation between tuners.

FIG. 16 is a block diagram showing a conventional tuner.
In FIG. 16, a high-frequency signal is input from an input / output terminal 100, passes through a high-pass filter (hereinafter referred to as HPF) 101h of a demultiplexer 101, an attenuator 102 and an amplifier 103, and is supplied to a UHF / VHF changeover switch 104. In the UHF / VHF changeover switch 104, one signal path (105u to 109u path or 105v to 109v path) is selected depending on whether the reception frequency band is the UHF band or the VHF band.

  When the UHF band is selected by the UHF / VHF changeover switch 104, the high frequency signal is mixed through the frequency variable BPF 105u tuned to the reception frequency, the amplifier 106u, and the frequency variable BPF 107u tuned to the reception frequency. It is input to one input terminal of 108u. A local oscillation signal from the local oscillator 109u is input to the other input terminal of the mixer 108u.

When the VHF band is selected by the UHF / VHF changeover switch 104, the high-frequency signal is mixed through the frequency variable BPF 105v tuned to the reception frequency, the amplifier 106v, and the frequency variable BPF 107v tuned to the reception frequency. It is input to one input terminal of 108v. A local oscillation signal from the local oscillator 109v is input to the other input terminal of the mixer 108v. (VHF band frequency variable BPFs 105v and 107v are often divided into two frequency bands (VHF low band and VHF high band), but are omitted here as VHF bands.)
The local oscillators 109u and 109v are constituted by voltage-controlled high-frequency oscillators, and are controlled so that a signal having a local oscillation frequency corresponding to the reception frequency is obtained by the tuning voltage Vt from the input terminal 116. Further, the frequency variable BPFs 105u, 107u, 105v, and 107v are controlled by the tuning voltage Vt, so that a pass band tuned to the reception frequency is obtained.

  The mixer 108u and the local oscillator 109u constitute UHF band side frequency converting means for converting a UHF band high frequency signal into an intermediate frequency signal. The mixer 108v and the local oscillator 109v convert the VHF band high frequency signal into an intermediate frequency signal. The frequency conversion means on the VHF band side for converting to V is configured.

  An intermediate frequency signal from the mixer 108u or 108v is supplied to an intermediate frequency amplification stage constituting an amplifier 111, a surface acoustic wave filter (hereinafter referred to as SAW filter) 112, and an amplifier 113 via a UHF / VHF changeover switch 110. Here, amplification of the intermediate frequency signal and band limitation to the intermediate frequency band are performed, and the result is output from the output terminal 114.

  An upstream signal (home → center direction) from a cable modem (not shown) is input to the input terminal 115 and is output from the input / output terminal 100 through the low-pass filter (LPF) 101 l of the duplexer 101.

FIG. 17 shows the input end reflection loss (return loss) of the frequency variable BPF 105u. The horizontal axis represents frequency (MHz) and the vertical axis represents reflection loss (dB). The input end reflection loss of the frequency variable BPF 105u represents the attenuation of the power of the signal reflected at the input end of the BPF 105u, and is large at the reception frequency and small above and below the reception frequency. (The VHF band variable frequency BPF 105v exhibits similar characteristics.)
When the duplexer 101 is placed in front of a filter circuit having such characteristics, it is known that sufficient isolation between the terminals of the duplexer 101 cannot be obtained. Therefore, as shown in FIG. An attenuator 102 and an amplifier 103 are provided in front of the variable BPFs 105u and 105v so as to secure a reflection loss other than the tuning frequency so as to reduce the frequency component returned to the preceding demultiplexer 101 and the demultiplexer 101 and Connected.
FIG. 18 is a block diagram showing another conventional tuner. An example is shown in which a distributor 121 is connected to the input section of the tuner.
In FIG. 18, the high-frequency signal is input from the input terminal 120 to the distributor 121, the first output terminal of the distributor 121 is connected to the circuit after the amplifier 123, and the second output terminal 122 of the distributor 121 is different from that of the amplifier 121. It is connected to a tuner (not shown) corresponding to the media.
The output of the amplifier 123 passes through an attenuator 124 and is supplied to a UHF / VHF changeover switch 125. The UHF / VHF selector switch 125 selects one signal path (126u to 130u path or 126v to 130v path) depending on whether the reception frequency band is the UHF band or the VHF band.

The frequency variable BPF 126u, amplifier 127u, frequency variable BPF 128u, mixer 129u, and local oscillator 130u in the UHF band path are the frequency variable BPF 105u, amplifier 106u, frequency variable BPF 107u, mixer 108u, and local in FIG. This is the same as the oscillator 109u. Further, the frequency variable BPF 126v, amplifier 127v, frequency variable BPF 128v, mixer 129v, and local oscillator 130v in the VHF band path are the frequency variable BPF 105v, amplifier 106v, frequency variable BPF 107v, mixer 108v, The same as the local oscillator 109v. The input terminal 136 of the tuning voltage Vt is the same as the input terminal 116 of FIG. 16, and the local oscillation frequency of the local oscillators 130u and 130v is controlled according to the reception frequency by the tuning voltage Vt from the input terminal 136. At the same time, the passbands of the frequency variable BPFs 126u, 128u, 126v, and 128v are controlled. (VHF band variable frequency BPF 126v and 128v are often divided into two frequency bands (VHF low band and VHF high band), but are omitted here as VHF bands.)
The intermediate frequency signal from the mixer 129u or 129v is supplied to an intermediate frequency amplification stage composed of an amplifier 132, a SAW filter 133, and an amplifier 134 via a UHF / VHF switch 131, where the intermediate frequency signal is supplied. Is output from the output terminal 135 after being amplified and limited to the intermediate frequency band.

The input end reflection loss of the frequency variable BPF 126u is the same as that shown in FIG. Since it is known that when the distributor 121 is placed in front of the filter circuit having such an input end reflection characteristic, the inter-terminal isolation of the distributor 121 cannot be obtained sufficiently, as shown in FIG. An amplifier 123 and an attenuator 124 are provided in front of the variable frequency type BPFs 126u and 126v to ensure reflection loss other than the tuning frequency and are connected to the distributor 121 in the previous stage.
In FIGS. 16 and 18, the attenuators 102 and 124 and the amplifiers 103 and 123 are shown together, but either an attenuator or an amplifier may be used.
By the way, in the tuner of FIG. 16 or FIG. 18, isolation of the branching filter and distributor cannot be secured unless an attenuator or amplifier is provided before the frequency variable BPF, but an attenuator is provided before the frequency variable BPF. If provided, the reception C / N deteriorates, and if an amplifier is provided, the input level increases, resulting in a problem that nonlinear distortion of nonlinear circuits (amplifiers and frequency converters) after the frequency variable BPF deteriorates.

  In general, a duplexer (corresponding to reference numeral 101 in FIG. 16) for separating an upstream signal and a downstream signal is configured as shown in FIGS.

  FIG. 19 is a circuit diagram of the duplexer, and FIG. 20 shows a state in which the duplexer of FIG. 19 is disposed at the input portion of the tuner including the shield frame 140. FIG. 21 is a cross-sectional view taken along the line AA of FIG. 20. A wiring board 141 on which a duplexer 101 is mounted is housed in a shield frame 140, and metal lids are placed above and below the shield frame 140. The bodies 143 and 142 are disposed.

  In FIG. 19, the duplexer 101 includes a low-pass filter (LPF unit) that passes an upstream signal and blocks a downstream band, and a high-pass filter (HPF unit) that blocks an upstream band and passes a downstream signal. In a terminal including a cable modem, as shown in FIG. 16, it is configured in an input section inside the tuner. The LPF section includes inductors LL1, LL2,..., LLn connected between the input / output terminal 100 and the input terminal 115, the connection points between the inductors LL1, LL2,. , CLn connected between reference potential points, and capacitors CL12, C13,..., CLn connected in parallel to the inductors LL2, LL3,. The HPF unit includes capacitors CH1, CH2,..., CHn + 1 connected in series to the input / output terminal 100, and a capacitor CH11 connected between the connection points of the capacitors CH1, CH2,. It comprises an inductor LH1, a capacitor CH12, an inductor LH2,..., And a series circuit of capacitors CH1n and LHn.

  In a terminal including a cable modem, the level of the input downstream signal is normally −15 to +15 dBmV, whereas the upstream signal transmitted to the center needs to compensate for the loss caused by the tap or cable until reaching the center. Since the maximum output level is +60 dBmV, the following performance is required for the duplexer.

  (B) Since the high band end of the upstream band is covered by the intermediate frequency band of the tuner, when the upstream signal is transmitted at the high band edge of the upstream band, the signal after the conversion of the downstream signal to the intermediate frequency signal is disturbed. Therefore, the high-pass filter of the duplexer needs a sufficient attenuation characteristic in the intermediate frequency band.

  (B) The upstream signal generated by the upstream signal modulator in the cable modem in the terminal includes harmonic components outside the upstream band due to distortion of the amplifier or the like, or the digital synthesizer type modulator is out of the upstream band. Modulation component is generated. Since these interfere with downstream signals, the low pass filter of the duplexer requires sufficient attenuation characteristics in the cutoff region.

  By the way, even if each of the high-pass filter and the low-pass filter of the duplexer is optimally designed, as described above (FIGS. 19 to 21), when configuring the duplexer inside the tuner, Due to the downsizing of the tuner, the distance between the filters cannot be secured sufficiently, and the inductors and the like constituting each filter are coupled to each other. In FIG. 19 to FIG. 21, as indicated by the dotted line in FIG. 19, the inductor LL1 of the LPF part and the inductor LH1 of the HPF part are easily spatially coupled to each other by electromagnetic induction. As a result, isolation between the high-pass filter and the low-pass filter cannot be achieved, and attenuation occurs due to the rise of each other's inductor resonance in the cut-off region of each filter and the apparent decrease in the number of filter stages due to direct signal jumping. There was a problem that the characteristics deteriorated.

Since a digitally modulated signal has phase information, a tuner that receives the phase information needs to reduce the phase noise of a high-frequency oscillator used for a local oscillator or the like so as not to deteriorate the purity of the phase information.
When receiving a CATV or terrestrial broadcast, the oscillator in the tuner needs to oscillate in the range of about 120 to 910 MHz, which is higher than the input signal by an intermediate frequency (about 30 to 60 MHz). In a normally used variable capacitance diode, the ratio between the upper limit and the lower limit of the oscillation frequency is about 1.5 to 2.5 times in the control voltage range (approximately 1 to 25 V). For this reason, the entire frequency range (120 to 910 MHz) is covered by dividing into a plurality of frequency bands. Generally, it is divided into three frequency bands of approximately 120 to 250 MHz, 250 to 500 MHz, and 500 to 910 MHz, and each is covered with a continuous control voltage range of about 1 to 25V.
FIG. 22 is a circuit diagram of a high frequency oscillator in a conventional tuner. The high-frequency oscillator shown in this figure is a circuit called a grounded collector type.

  In FIG. 22, the high-frequency oscillator has capacitors C1 and C2 connected between the base of the transistor Q1 and the reference potential point, and the emitter is connected to the connection point of the capacitors C1 and C2, while being connected to the reference potential point via the resistor R1. The collector is connected to the power supply terminal 150 of the DC voltage Vcc, while being connected to the reference potential point via the capacitor C3, and the voltage obtained by dividing the DC voltage Vcc of the power supply terminal 150 by the resistors R2 and R3 is supplied to the base of Q1. It is like that. The base of Q1 is connected to the cathodes of variable capacitance diodes Cv1 and Cv2 via a capacitor C4, and the anode of Cv1 is connected to a reference potential point via a parallel circuit of a capacitor C5 and a resistor R4, and the tuning voltage Vt The input terminal 151 is connected to the reference potential point via the capacitor C6, while being connected to the cathode connection point of the variable capacitance diodes Cv1 and Cv2 via the resistor R5, and the tuning voltage Vt is connected to each of the variable capacitance diodes Cv1 and Cv2. It is supplied to the cathode. The anode of Cv2 is connected to one end of a resonance circuit in which an inductor L1, a capacitor C7, and an inductor L2 are connected in series, and the other end of the resonance circuit is connected to a power supply terminal 153 of a DC voltage Vcc via a resistor R11. ing. The connection point between the inductor L1 and the capacitor C7 is connected to the reference potential point via the resistor R7, the connection point between the capacitor C7 and the inductor L2 is connected to the cathode of the switching diode D1, and the anode thereof is connected via the capacitor C8. One of the terminals connected to the reference potential point is connected to the input terminal 152 of the frequency band switching control signal SW through the resistor R8. The input terminal 152 is connected to a reference potential point via a parallel circuit of a resistor R9 and a capacitor C9, and a connection point of the inductor L2 and the resistor R11 is connected to a reference potential point via a parallel circuit of the capacitor C10 and the resistor R10. The power supply terminal 153 is connected to the reference potential point via the capacitor C11.

  In the circuit shown in FIG. 22, a common transistor Q1 is used for the first frequency band (120 to 250 MHz) and the second frequency band (250 to 500 MHz), and the inductors L1 and L2 of the resonant circuit are used. It is switched by the diode D1 to cover two oscillation frequency ranges. In this example, when the logic of the band switching control voltage SW is "Low", the diode D1 becomes an open impedance and oscillates at 120 to 250 MHz by changing the applied voltage of the tuning voltage Vt. On the other hand, when the logic of the band switching control voltage SW is “High”, the diode D1 becomes a short-circuit impedance and oscillates at 250 to 500 MHz by changing the applied voltage of the tuning voltage Vt. For this reason, it is necessary to generate a negative countermeasure in the oscillation frequency range of 120 to 500 MHz by the transistor Q1, the capacitor C1, and the capacitor C2.

  FIG. 23 shows the negative resistance when the base side of the transistor Q1 is viewed from the capacitor C3 in the circuit of FIG. A sufficient negative resistance can be obtained in the vicinity of 250 MHz, but a sufficient negative resistance cannot be obtained in the vicinity of 120 MHz and 500 MHz.

Therefore, in the conventional high-frequency oscillator of FIG. 22, it is difficult to generate a negative resistance in the range of 120 to 500 MHz and oscillate with a sufficient margin in the desired oscillation range, resulting in deterioration of phase noise. There is a problem.
In digital CATV broadcasting, signals transmitted in the downstream band are digitally modulated waves such as QPSK and QAM, and are generally transmitted at a level lower by about 10 dB than analog waves. Therefore, when demodulating, even if there is an analog wave in the adjacent channel, it is necessary to sufficiently reduce the level of the adjacent wave. Therefore, a surface acoustic wave filter (hereinafter referred to as a SAW filter) is used at the intermediate frequency stage as a band limiting filter. ing. However, since the SAW filter has a large insertion loss, in order to obtain a desired noise figure and signal level, amplifiers 161 and 163 are generally used before and after the SAW filter 162 as shown in FIG. Further, the amplifier 163 disposed in the subsequent stage often requires a relatively high signal level for demodulation, and thus is often a high gain amplifier. As a method of mounting the SAW filter, there are (i) a case where the SAW filter 162 is arranged outside the tuner and (b) a case where the SAW filter 162 is built in the tuner as shown in FIG.

  By the way, the following problems occur depending on the method of mounting the SAW filter in the intermediate frequency band.

  (B) When the SAW filter is mounted outside the tuner, that is, on the main board of the set on which the tuner is mounted, interference in the intermediate frequency band that travels on the main board (for example, digital noise or upstream signal) There is a problem that jumps directly.

  (B) When it is built in the tuner, it is covered with the tuner chassis that has a shielding effect, so it is an advantageous configuration against disturbance on the main board and is easy to handle, but in a small area. Since the SAW filter and the high gain amplifier are mounted, it is difficult to achieve isolation between the input and output of the SAW filter, the signal component transmitted outside the SAW filter increases, and the band generated by interference with the signal transmitted through the SAW filter There was a problem that the inner ripple increased.

  As described above, the conventional tuner has the following problems (1) to (4).

  (1) If an attenuator or amplifier is not provided in front of the frequency variable BPF, isolation of the duplexer or distributor cannot be ensured. However, if an attenuator is provided in front of the frequency variable BPF, the reception C / N deteriorates. However, when an amplifier is provided, there is a problem that nonlinear distortion of nonlinear circuits (amplifiers and frequency converters) after the frequency variable BPF deteriorates due to an increase in input level.

(2) Even if the high-pass filter and low-pass filter of the duplexer are optimally designed, as described above, if the duplexer is configured inside the tuner, the tuner must be downsized. Therefore, the distance between the filters cannot be sufficiently secured, and the inductors and the like constituting each filter are coupled to each other. As a result, the high-pass filter and the low-pass filter cannot be isolated, and the attenuation characteristics deteriorate due to the rise of the mutual resonance of the inductors in the cut-off region and the apparent decrease in the number of filter stages due to direct signal jumping. There was a problem to do.

(3) A negative resistance is generated in the range of 120 to 500 MHz, and it is difficult to oscillate with a sufficient margin in a desired oscillation range, resulting in a problem of deteriorating phase noise.
(4) The following problems occur depending on the method of mounting the SAW filter in the intermediate frequency band.

  B) When the SAW filter is mounted outside the tuner, that is, on the main board of the set on which the tuner is mounted, interference in the intermediate frequency band (for example, digital noise or upstream signal) transmitted on the main board is generated. There is a problem of jumping directly.

  B) When built in a tuner, it is covered with a tuner chassis that has a shielding effect, so it is an advantageous configuration against disturbance on the main board, and it is easy to handle, but it can be used in a small area. Since the filter and the high gain amplifier are mounted, it is difficult to achieve isolation between the input and output of the SAW filter, so that the signal component transmitted outside the SAW filter increases and is generated by interference with the signal transmitted through the SAW filter. There was a problem that the in-band ripple increased.

  In view of the above problems, an object of the present invention is to provide a tuner capable of maintaining and improving reception performance over a wide frequency band.

  More specifically, in view of the above problem (4), the present invention provides a tuner equipped with a SAW filter that secures input / output isolation of the SAW filter of the intermediate frequency amplification stage and has good in-band characteristics and cutoff characteristics. For the purpose.

The tuner according to the first aspect of the present invention is arranged in a stage subsequent to the frequency conversion means, a frequency conversion means for generating an intermediate frequency signal by mixing an input high frequency signal and a local oscillation signal from the local oscillation means, A first amplifying means for amplifying the intermediate frequency signal; a surface acoustic wave filter for band-limiting the intermediate frequency signal amplified by the first amplifying means; and a second stage disposed after the surface acoustic wave filter. In a tuner equipped with an intermediate frequency amplification stage comprising
The surface acoustic wave filter is covered with a package, and the surface acoustic wave filter covered with the package is mounted in a region surrounded by a shield wall.

  In the first aspect of the invention, the surface acoustic wave filter covered by the package is mounted in a dedicated area surrounded by the shield wall, thereby improving the input / output isolation of the surface acoustic wave filter. Thus, an intermediate frequency amplification stage can be obtained in which the tail is cut off, the in-band ripple is small, and the passband characteristics of the surface acoustic wave filter can be finely adjusted.

  As described above, according to the present invention, reception performance can be maintained and improved over a wide frequency band. In particular, since the input / output isolation of the SAW filter of the intermediate frequency amplification stage can be ensured and a SAW filter having good in-band characteristics and cutoff characteristics can be obtained, a tuner having good characteristics can be realized.

Embodiments of the invention will be described with reference to the drawings.
FIG. 1 is a block diagram showing a tuner according to the first embodiment of the present invention.

  In FIG. 1, a high frequency signal is input from an input / output terminal 10, passes through a high pass filter (HPF) 11 h of a branching filter 11, and is supplied to a UHF / VHF changeover switch 12. In the UHF / VHF changeover switch 12, one signal path (13u to 18u path or 13v to 18v path) is selected depending on whether the reception frequency band is the UHF band or the VHF band.

  When the UHF band is selected by the UHF / VHF changeover switch 12, the UHF band is input to the amplifier 15u through the frequency variable type terminator 13u and the frequency variable type BPF 14u tuned to the reception frequency.

  The output of the amplifier 15u is input to one input terminal of the mixer 17u via the frequency variable BPF 16u. The oscillation signal from the local oscillator 18u is input to the other input terminal of the mixer 17u.

When the VHF band is selected by the UHF / VHF selector switch 12, the signal passes through the frequency variable type terminator 13v and is input to the amplifier 15v through the frequency variable type BPF 14v tuned to the reception frequency.
The output of the amplifier 15v is input to one input terminal of the mixer 17v via the frequency variable BPF 16v. An oscillation signal from the local oscillator 18v is input to the other input terminal of the mixer 17v.

  The configuration of the frequency variable terminators 13u and 13v will be described later.

  The local oscillators 18u and 18v are constituted by voltage-controlled high-frequency oscillators, and are controlled so that a signal having a local oscillation frequency corresponding to the reception frequency is obtained by the tuning voltage Vt from the input terminal 26. Further, the frequency variable BPFs 14u, 16u, 14v, and 16v are controlled by the tuning voltage Vt, so that a pass frequency band tuned to the reception frequency is obtained.

  The mixer 17u and the local oscillator 18u constitute UHF band-side frequency converting means for converting a UHF band high-frequency signal into an intermediate frequency signal, and the mixer 17v and the local oscillator 18v convert a VHF band high-frequency signal into an intermediate frequency signal. Frequency conversion means on the VHF band side.

  The intermediate frequency signal from the mixer 17u or 17v is supplied via the UHF / VHF switch 19 to the intermediate frequency amplification stage 20 constituting the amplifier 21, the SAW filter 22, and the amplifier 23, where the intermediate frequency signal is supplied. Are amplified and limited to the intermediate frequency band, and output from the output terminal 24.

  An input signal from an unillustrated cable modem (home → center direction) is input to the input terminal 25, and is output from the input / output terminal 10 through the LPF 11 l of the duplexer 11.

  The frequency variable terminators 13u and 13v are in the form of a bridge T-type BPF. The configuration of the frequency variable terminator 13u will be described below. The inductor Lu1 and the capacitor Cu1 constitute a series resonance circuit 13u-1. The inductor Lu2 and the capacitor Cu2 constitute a parallel resonance circuit 13u-2. The capacitors Cu1 and Cu2 use variable capacitance diodes, and their capacitance values change according to the channel selection voltage Vt from the input terminal 26 as in the frequency variable type BPF 14u, whereby the series resonance circuit 13u-1 and the parallel resonance circuit 13u. -2 resonance frequency changes. By making the two resonance frequencies of the resonance circuits 13u-1 and 13u-2 coincide with the reception frequency, the series resonance circuit 13u-1 exhibits a short-circuit impedance and the parallel resonance circuit 13u-2 is open at the reception frequency. Since it exhibits an impedance, the insertion loss is almost zero and is input to the frequency variable BPF 14u. As the frequency deviates from the reception frequency, the series resonant circuit 13u-1 gradually exhibits an open impedance, while the parallel resonant circuit 13u-2 exhibits a short-circuit impedance, so that the input of the frequency variable termination unit 13u The output terminal is terminated with a termination resistor Ru1 and a termination resistor Ru2. The termination resistors Ru1 and Ru2 are selected to be about 75Ω in the case of CATV or the like. The configuration of the frequency variable termination unit 13v is the same as described above.

  FIG. 2 shows the reflection loss at the input end after the frequency variable terminator 13u. The horizontal axis represents frequency (MHz) and the vertical axis represents reflection loss (dB).

  As can be seen from FIG. 2, the reflection loss at the input end of the frequency variable terminator 13u is good in frequency bands other than the reception frequency regardless of the impedance of the subsequent frequency variable BPF 14u.

  As described above, a circuit having a substantially constant impedance (for example, 76Ω) is connected to the output terminal of the HPF 11h of the duplexer 11 in FIG. 1 regardless of the reception frequency. The upstream signal input from the upstream signal input terminal 25 is output from the input / output terminal 10 through the LPF 11 l of the duplexer 11. In this case, since the impedance after the frequency variable terminator 13u is substantially constant, the HPF 11h and the LPF 11l of the duplexer 11 can obtain excellent characteristics, and the output terminal of the HPF 11h and the input terminal of the LPF 11l are sufficient. Secure isolation.

  FIG. 3 is a block diagram showing a tuner according to the second embodiment of the present invention.

In the embodiment of FIG. 3, the high frequency signal is input from the input terminal 30 to the distributor 31, and the first output terminal of the distributor 31 is connected to the circuit after the UHF / VHF changeover switch 32. The second output terminal 36 is connected to a tuner (not shown) corresponding to another medium.
In the UHF / VHF changeover switch 31, one signal path (33u to 38u path or 33v to 38v path) is selected depending on whether the band of the reception frequency is the UHF band or the VHF band. The configuration of frequency variable terminators 33u and 33v is mainly different from FIG.

The frequency variable type terminator 33u, frequency variable type BPF 34u, amplifier 35u, frequency variable type BPF 36u, mixer 37u, and local oscillator 38u in the UHF band path are the frequency variable type terminator 13u, frequency variable type BPF 14u, This is the same as the amplifier 15u, the variable frequency BPF 16u, the mixer 17u, and the local oscillator 18u. Further, the frequency variable type termination unit 33v, the frequency variable type BPF 34v, the amplifier 35v, the frequency variable type BPF 36v, the mixer 37v, and the local oscillator 38v in the VHF band path are the frequency variable type termination unit 13v, frequency variable type in FIG. This is the same as the BPF 14v, the amplifier 15v, the frequency variable BPF 16v, the mixer 17v, and the local oscillator 18v. The input terminal 37 for the tuning voltage Vt is the same as the input terminal 26 in FIG. 1, and the local oscillation frequencies of the local oscillators 38u and 38v are set according to the reception frequency at the tuning voltage Vt from the input terminal 37. In addition to controlling, the passbands of the variable frequency BPFs 34u, 36u, 34v, 36v are controlled. (VHF band variable frequency BPFs 34v and 36v are often divided into two frequency bands (VHF low band and VHF high band), but are omitted here as VHF bands.)
The intermediate frequency signal from the mixer 37u or 37v is supplied to an intermediate frequency amplification stage 40 including an amplifier 41, a SAW filter 42, and an amplifier 43 via a UHF / VHF changeover switch 39. The signal is amplified and limited to the intermediate frequency band and output from the output terminal 44.

  The frequency variable type terminator 33u includes a parallel resonant circuit 33u-1 including an inductor Lu3 and a capacitor Cu3 and a resistor Ru3 connected in series. The series connected circuit is connected to the UHF output terminal of the changeover switch 39 and a reference potential point. It is connected and configured. The capacitor Cu3 uses a variable capacitance diode in the same manner as the capacitor Cv3 described later, and its capacitance value is varied according to the reception frequency by the tuning voltage Vt. The parallel resonance circuit 33u-1 configured by the inductor Lu3 and the capacitor Cu3 exhibits an open impedance at the reception frequency, and the parallel resonance circuit 33u-1 gradually exhibits a short-circuit impedance as the frequency is separated from the reception frequency. Accordingly, in the case of a circuit in which the outside of the reception frequency band of the frequency variable type BPF 34u exhibits an open impedance, the input end reflection loss after the frequency variable terminator 33u is reduced by being terminated by the frequency variable type terminator Ru3. 2 can be obtained.

  The VHF band frequency variable type terminator 33v connects the inductors Lv3, Lv3 ′ and the parallel resonant circuit 33v-1 of the capacitor Cv3 and the resistor Ru3 in series, and opens or shorts both ends of the inductor Lv3 ′. A switch SW is provided, and this series connection circuit is connected between the UHF output terminal of the changeover switch 39 and a reference potential point. The switch SW is used to switch between two frequency bands (VHF low band and VHF high band) of the VHF band.

  As the capacitors Cu3 and Cv3, variable capacitance diodes whose capacitance values change depending on the tuning voltage Vt are used.

  The frequency variable terminators 33u and 33v described above are connected to one output terminal of the distributor 31, and a circuit having a good input end reflection loss is connected to the other output terminal 36, whereby the distributor 31 is connected. The two output terminals can be sufficiently isolated from each other.

  FIG. 4 is a circuit diagram showing a duplexer used in the tuner according to the third embodiment of the present invention, FIG. 5 is a perspective view showing a state where the duplexer of FIG. 4 is disposed in the tuner, and FIG. 5 is a cross-sectional view taken along line AA of FIG. 5, in which the wiring board 42 on which the duplexer 11 is mounted is accommodated in the shield frame 40, and metal lids 44 and 43 are arranged on the upper and lower portions of the shield frame 40. It is installed.

  In FIG. 4, the duplexer 11 includes a low-pass filter (LPF unit) that transmits an upstream signal and blocks a downstream band, and a high-pass filter (HPF unit) that blocks an upstream band and transmits a downstream signal. A terminal including a cable modem is configured in an input unit inside the tuner as shown in FIG. The LPF section includes inductors LL1, LL2,..., LLn connected between the input / output terminal 10 and the input terminal 25, the connection points between the inductors LL1, LL2,. , CLn connected between reference potential points, and capacitors CL12, C13,..., CLn connected in parallel to the inductors LL2, LL3,. The HPF unit includes capacitors CH1, CH2,..., CHn + 1 connected in series to the input / output terminal 10, and a capacitor CH11 connected between the connection point of each capacitor CH1, CH2,. It comprises an inductor LH1, a capacitor CH12, an inductor LH2,..., A series circuit of capacitors CH1n and a capacitor LHn.

  5 and 6, a shield plate 41 is provided between the HPF part and the LPF part so as to separate the inductors LH1 to LHn of the HPF part and the inductors LL1 to LLn of the LPF part. In addition, it is assumed that the inductors in the HPF part and the LPF part are arranged in directions that are difficult to couple with each other.

  With the above configuration, spatial coupling due to mutual induction between the inductor of the HPF unit and the inductor of the LPF unit is eliminated, and deterioration of the attenuation characteristics of the attenuation region of each filter can be eliminated.

  FIG. 7 is a cross-sectional view of the duplexer disposed in the tuner according to the fourth embodiment of the present invention. In the shield frame 40, the wiring board 42 on which the duplexer 11 is mounted is housed. Metal lids 44 and 43 are disposed on the upper and lower sides of the frame.

  In FIG. 7, LH1 is the inductor closest to the coupling portion with the LPF portion among the inductors LH1 to LHn of the HPF portion, and LL1 is the closest to the coupling portion with the HPF portion among the inductors LL1 to LLn of the LPF portion. Indicates a close inductor. The inductor LH1 of the HPF portion is disposed on the front surface of the substrate 42, and the inductor LL1 of the LPF portion is disposed on the back surface of the substrate 42. In addition, it is assumed that the inductors in the HPF part and the LPF part are coupled to each other and arranged away from each other.

  With the above configuration, it becomes difficult to couple the inductors closest to the coupling portion in a positional relationship that facilitates coupling, and thus it is possible to suppress deterioration of the attenuation characteristics in the cutoff region. In the embodiment of FIG. 7, the inductor LH1 of the HPF part is arranged on the front surface of the substrate 42, and the inductor LL1 of the LPF part is arranged on the back surface of the substrate 42. The same effect can be obtained even if LL1 is arranged on the surface of the substrate 42. Further, in FIG. 7, the inductor LL1 on the back surface of the substrate is shown as an air-core coil like the substrate surface.

  It is to be noted that the combination of both the arrangement of the shield plate shown in FIGS. 4 to 6 and the arrangement of the inductors on the front and back surfaces of the substrate 42 as shown in FIG. 7 is another embodiment according to the present invention. This is an effective embodiment for isolating HPF and LPF.

  FIG. 8 is a circuit diagram showing a local oscillator in the tuner according to the fifth embodiment of the present invention. The high-frequency oscillator shown in this figure is a circuit called a grounded collector type. The same parts as those in FIG. 22 are denoted by the same reference numerals.

  In FIG. 8, the high-frequency oscillator has capacitors C1 and C2 connected between the base of the transistor Q1 and a reference potential point, and an emitter connected to the connection point of the capacitors C1 and C2, while being connected to the reference potential point via a resistor R1. The collector is connected to the power supply terminal 50 of the DC voltage Vcc, and connected to the reference potential point via the capacitor C3, and the voltage obtained by dividing the DC voltage Vcc of the power supply terminal 50 by the resistors R2 and R3 is supplied to the base of Q1. I am doing so.

  A series circuit comprising a capacitor C21, a first variable capacitance diode Cv11 and a capacitor C22 is connected between the base and the emitter of Q1, and a capacitor C23 and a second variable capacitance diode are connected between the emitter of Q1 and the reference potential point. A series circuit consisting of Cv12 is connected.

  The base of Q1 is connected to the cathodes of the third and fourth variable capacitance diodes Cv1 and Cv2 through a capacitor C4, and the anode of Cv1 is connected to a reference potential point through a parallel circuit of a capacitor C5 and a resistor R4. The input terminal 51 of the tuning voltage Vt is connected to the reference potential point via the capacitor C6, while being connected to the connection point of the cathodes of the third and fourth variable capacitance diodes Cv1 and Cv2 via the resistor R5. The local voltage Vt is supplied to the cathodes of the variable capacitance diodes Cv1 and Cv2. The node between the capacitor C21 and the anode of the first variable capacitance diode Cv11 is connected to the reference potential point via the resistor R21, and the input terminal 51 for the tuning voltage Vt is connected to the first potential via the resistor R22. The variable voltage diode Cv11 is connected to the cathode of the second variable capacitance diode Cv12 through the resistor R23, and the tuning voltage Vt is applied to the cathodes of the first and second variable capacitance diodes Cv11 and Cv12. To be supplied.

  The anode of the fourth variable capacitance diode Cv2 is connected to one end of a resonance circuit formed by connecting an inductor L1, a capacitor C7, and an inductor L2 in series, and the other end of the resonance circuit is connected to a DC voltage Vcc via a resistor R11. The power supply terminal 53 is connected. The connection point between the inductor L1 and the capacitor C7 is connected to the reference potential point via the resistor R7, the connection point between the capacitor C7 and the inductor L2 is connected to the cathode of the switching diode D1, and the anode thereof is connected via the capacitor C8. One of the terminals connected to the reference potential point is connected to the input terminal 52 of the frequency band switching control signal SW through the resistor R8. The input terminal 52 is connected to a reference potential point via a parallel circuit of a resistor R9 and a capacitor C9, and a connection point of the inductor L2 and the resistor R11 is connected to a reference potential point via a parallel circuit of the capacitor C10 and the resistor R10. The power supply terminal 53 is connected to the reference potential point via the capacitor C11.

  In the circuit shown in FIG. 8, the first variable capacitance diode Cv11 is connected so as to vary the capacitance value between the base and the emitter of the transistor Q1, and the second variable capacitance diode Cv12 is connected to the transistor Q1. The capacitance value between the emitter and the reference potential point is connected to be variable. The third and fourth variable capacitance diodes Cv1, Cv2 are the same as Cv1, Cv2 in the conventional example of FIG.

  The inductors L1 and L2 of the resonance circuit are switched by a diode D1, and two oscillation frequency ranges of approximately 120 to 250 MHz and 250 to 500 MHz are covered with a common transistor Q1.

  When the logic of the band switching control voltage SW is "Low", the diode D1 becomes an open impedance and oscillates at 120 to 250 MHz by changing the applied voltage of the tuning voltage Vt. On the other hand, when the logic of the band switching control voltage SW is “High”, the diode D1 becomes a short-circuit impedance and oscillates at 250 to 500 MHz by changing the applied voltage of the tuning voltage Vt.

  In other words, when the tuning voltage Vt is about 1V, if the SW logic is “low”, it oscillates at 120 MHz. If the SW logic is “high”, the tuning voltage Vt oscillates at 250 MHz. When the voltage is 1 V, a negative resistance is required in the frequency range from 120 to 250 MHz. In addition, when the tuning voltage Vt is about 25 V, if the SW logic is “low”, oscillation occurs at 250 MHz, and if the SW logic is “high”, oscillation occurs at 500 MHz. Negative resistance is required in the frequency range from 250 to 500 MHz at 25V.

  According to the circuit of FIG. 8, sufficient negative resistance can be obtained in the desired oscillation range as shown in FIG.

  FIG. 9 shows the negative resistance when the base side of the transistor Q1 is viewed from the capacitor C4 in the circuit of FIG. A solid curve A indicates a negative resistance when the applied voltage of Vt is 1V, and a dotted curve B indicates a negative resistance when the applied voltage of Vt is 25V. When the applied voltage of Vt is gradually increased from 1 V to 25 V, the negative resistance curve gradually changes from the solid curve A in FIG. 9 to the dotted curve B. In this way, the negative resistance when the base side of C4 to Q1 in the conventional tuner of FIG. 22 is seen is a characteristic that changes as shown in FIG. 23 regardless of the change of the tuning voltage Vt. In the tuner of FIG. 8, the negative resistance changes in the frequency range from 120 to 250 MHz when the tuning voltage Vt is 1 V as shown in FIG. When the voltage Vt is 25 V, the negative resistance changes in the frequency range from 250 to 500 MHz. As a result, it is possible to obtain a sufficient negative resistance in the desired oscillation range and suppress deterioration of phase noise.

  FIG. 10 is a circuit diagram showing a local oscillator in the tuner according to the sixth embodiment of the present invention. The circuit shown in this figure is a grounded collector type high frequency oscillator as in the circuit of FIG.

  10 differs from the circuit of FIG. 8 in that the second variable capacitance diode Cv12 is connected so as to vary the capacitance value between the emitter and the collector of the transistor Q1. Further, FIG. 8 shows an example in which a change relating to means for supplying the tuning voltage Vt is made.

  The configuration of FIG. 10 will be described below. In the high frequency oscillator, capacitors C1 and C2 are connected between the base of the transistor Q1 and a reference potential point, an emitter is connected to a connection point of the capacitors C1 and C2, and a collector is connected to a reference potential point via a resistor R1. A voltage Vcc is connected to a power supply terminal 50 and connected to a reference potential point via a capacitor C3 so that a voltage obtained by dividing the DC voltage Vcc of the power supply terminal 50 by resistors R2 and R3 is supplied to the base of Q1. .

  A series circuit consisting of a capacitor C21 and a first variable capacitance diode Cv11 is connected between the base and emitter of Q1, and a series circuit consisting of a second variable capacitance diode Cv12 and a capacitor C23 is connected between the emitter and collector of Q1. Is connected.

  The base of Q1 is connected to the cathode of the third variable capacitance diode Cv1 via capacitors C4 and C5, and the anode of Cv1 is connected to the reference potential point via a parallel circuit of capacitor C5 and resistor R4. The base of Q1 is connected to the cathode of the fourth variable capacitance diode Cv2 through the capacitor C4, and the anode of Cv2 is connected to the reference potential point through the resistor R7, while the inductor L1, capacitor C7, and inductor L2 are connected in series. One end of the connected resonance circuit is connected, and the other end of the resonance circuit is connected to the power supply terminal 53 of the DC voltage Vcc via the resistor R11.

  The input terminal 51 of the tuning voltage Vt is connected to the reference potential point via the capacitor C6, connected to the cathode of the third variable capacitance diode Cv1 via the resistor R4, and the fourth variable via the resistor R5. Connected to the cathode of the capacitive diode Cv2, the tuning voltage Vt is supplied to the cathodes of the variable capacitive diodes Cv1 and Cv2.

  The node between the capacitor C21 and the anode of the first variable capacitance diode Cv11 is connected to a reference potential point via a resistor R21, and further the anode of the capacitor C23 and the second variable capacitance diode Cv12. The connection point is connected to the reference potential point via the resistor R23, and the input terminal 51 of the tuning voltage Vt is connected to the cathodes of the first and second variable capacitance diodes Cv11 and Cv12 via the resistor R22. The tuning voltage Vt is supplied to the cathodes of the first and second variable capacitance diodes Cv11 and Cv12.

  The connection point between the capacitor C7 and the inductor L2 is connected to the cathode of the switching diode D1, and the anode is connected to the reference potential point via the capacitor C8, while the input terminal 52 for the frequency band switching control signal SW via the resistor R8. Connected to. The input terminal 52 is connected to a reference potential point via a parallel circuit of a resistor R9 and a capacitor C9, and a connection point of the inductor L2 and the resistor R11 is connected to a reference potential point via a parallel circuit of the capacitor C10 and the resistor R10. The power supply terminal 53 is connected to a reference potential point via a capacitor C11. The effect of FIG. 10 is the same as that of FIG.

  FIG. 11 is a block diagram showing the configuration of the intermediate frequency amplification stage in the tuner according to the seventh embodiment of the present invention, and FIG. 12 is a perspective view showing the structure thereof.

  This embodiment relates to an improvement of the intermediate frequency amplification stage 20 or 40 in FIG. 1 or FIG.

  The intermediate frequency amplifying stage of FIG. 11 includes an amplifier 62 as a first amplifying means, a SAW filter 63 of a metal package, for example, and an amplifier 65 as a second amplifying means, between an input terminal 61 and an output terminal 66. Are connected in series. Here, the SAW filter 63 is configured to be surrounded by a shield wall 64.

  An intermediate frequency signal from a frequency conversion means (not shown) is supplied to an input terminal 61. The intermediate frequency signal is amplified by an amplifier 62 and then limited to a predetermined intermediate frequency band by a SAW filter 63. Thereafter, the signal is further amplified at an intermediate frequency by an amplifier 65 and taken out as a tuner output to an output terminal 66.

  As shown in FIG. 12, the SAW filter 63 is surrounded by a shield wall 64, and each shield wall 64 is at least partially soldered to the ground pattern on the surface of the wiring board 65. It is assumed that it penetrates the back surface of 65 and is soldered to the ground pattern on the back surface of substrate 65.

  With the above configuration, the periphery of the SAW filter 63 is partitioned by the shield wall 64 having a low impedance, so that a signal that wraps around from the input terminal of the SAW filter 63 to the output terminal via the circuit pattern is reduced. Since the front-stage amplifier 62 and the rear-stage amplifier 65 are separated from each other in the shielded SAW filter 63 region, the signal that flows from the output of the amplifier 65 to the input of the amplifier 62 is also reduced. And it becomes possible to suppress degradation of the in-band ripple characteristic.

  In the embodiment of FIGS. 11 and 12, the SAW filter 63 is a metal package, but the same effect can be obtained with a plastic package SAW filter as shown in FIGS. .

  FIG. 13 is a perspective view showing another configuration example of the intermediate frequency amplification stage in the tuner according to the eighth embodiment of the present invention, and FIG. 14 is a sectional view taken along line AA in FIG. In this example, a plastic package SAW filter 70 is provided instead of the metallic package SAW filter 63 in FIG. 12, and in FIG. 13 and FIG. A vertical plastic package SAW filter 70 extending in a direction perpendicular to the surface is disposed.

  In the case of the SAW filter 70 in the vertical plastic package, the inclination of the SAW filter 70 can be changed as shown in FIG. 14, and therefore, the surface pattern of the SAW filter in the package and the shield are changed depending on the distance between the package and the shield wall 64. The capacitance formed by the wall 64 changes, and the power output impedance of the SAW filter can be finely adjusted. That is, by changing the inclination of the SAW filter 70, the passband characteristics of the SAW filter can be finely adjusted.

1 is a block diagram illustrating a tuner according to a first embodiment of this invention. The figure which shows the input end reflection loss after the frequency variable type | mold termination of FIG. The block diagram which shows the tuner of the 2nd Embodiment of this invention. FIG. 6 is a circuit diagram showing a duplexer in a tuner according to a third embodiment of the present invention. The perspective view of the state which has arrange | positioned the splitter of FIG. 4 in the tuner. AA line sectional view of Drawing 5. Sectional drawing of the duplexer in the tuner of the 4th Embodiment of this invention. The circuit diagram which shows the local oscillator in the tuner of the 5th Embodiment of this invention. The figure which shows the negative resistance at the time of seeing the base side of Q1 from the capacitor | condenser C4 in FIG. The circuit diagram which shows the local oscillator in the tuner of the 6th Embodiment of this invention. The block diagram which shows the structure of the intermediate frequency amplification stage in the tuner of the 7th Embodiment of this invention. The perspective view which shows the structure of FIG. The perspective view which shows the other structural example of the intermediate frequency amplification stage in the tuner of the 8th Embodiment of this invention. AA line sectional view of Drawing 13. 1 is a block diagram showing a general bidirectional CATV transmission / reception system. FIG. The block diagram which shows the tuner of a prior art example. The figure which shows the input end reflection loss of the frequency variable type BPF of FIG. The block diagram which shows the tuner of another prior art example. FIG. 17 is a circuit diagram showing a duplexer used in the tuner of FIG. 16. The perspective view of the state which has arrange | positioned the splitter of FIG. 19 in a tuner. AA sectional view taken on the line AA of FIG. The circuit diagram which shows the local oscillator in the tuner of a prior art example. The figure which shows the negative resistance at the time of seeing the base side of Q1 from the capacitor | condenser C4 in FIG. The block diagram which shows the structure of the intermediate frequency amplification stage in the tuner of a prior art example. The perspective view which shows the structure of FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 ... Input / output terminal 11 ... Demultiplexer 11h ... High pass filter (HPF)
11l ... Low-pass filter (LPF)
12, 32... UHF / VHF changeover switch 13u, l3v, 23u, 33v... Frequency variable type terminator 14u, 14v, 34u, 34v ... Frequency variable type BPF (frequency variable type filter means)
15u, 15v, 35u, 35v ... amplifier 18u, 18V, 38u, 38V ... local oscillator (high frequency oscillator)
20, 40... Intermediate frequency amplification stage 21, 41, 62... Amplifier (first amplification means)
22, 42, 63, 70 ... surface acoustic wave filter (SAW filter)
23, 43, 65 ... amplifier (second amplification means)
25 ... I / O terminal 31 ... Distributor 40 ... Shield frame 41 ... Shield plate 42 ... Substrate 64 ... Shield wall LH1, LH2, ..., LHn ... HPF side inductor LL1, LL2, ..., LLn ... LPF side inductor Q1 ... Transistor Cv11 ... first variable capacitance diode Cv12 ... second variable capacitance diode Cv1 ... third variable capacitance diode Cv2 ... fourth variable capacitance diode

Claims (1)

A frequency converting means for generating an intermediate frequency signal by mixing the input high frequency signal and a local oscillation signal from the local oscillating means; and a first stage for amplifying the intermediate frequency signal, which is arranged after the frequency converting means. An intermediate frequency amplification stage comprising amplification means, a surface acoustic wave filter for band-limiting the intermediate frequency signal amplified by the first amplification means, and second amplification means disposed downstream of the surface acoustic wave filter In a tuner equipped with
A tuner, wherein the surface acoustic wave filter is covered with a package, and the surface acoustic wave filter covered with the package is mounted in a region surrounded by a shield wall.

Images