JP2006337348A - Radio wave guidance seeker apparatus - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a radio wave guidance seeker apparatus which can shorten a blind distance in a short distance by carrying out adjustable of the transmitted pulse width according to a distance with a target. <P>SOLUTION: A migration average process is given with a running average filter 53 to a received signal after receiver 21 receives a reflective signal from the target and carrying out a rectangular detection, gain control processing is performed with a gain regulator 57 to this received signal, the received signal after this gain control processing is re-sampled with a re-sample unit 63 according to any one re-sample clock signal obtained by performing the frequency division of the basic timing signal to a plurality of stages, and obtained it with a frequency division term unit 41 is followed, the received signal after this gain control processing is re-sampled with the re-sample unit 63, received signal after this re-sample is passed by the period of three kinds of gate signals related to the distance to a target with a prediction distance computing unit 29, a pulse width controller 31 forms a pulse width control signal based on an error between differential signals after this passage, with a transmitter 33, according to this pulse width control signal, the width of the surface of the pulse signal is altered and is transmitted. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a radio wave induction seeker device that can shorten a blind distance at a short distance by changing a transmission pulse width according to a distance from a target.

  Conventionally, a radio wave induction seeker device is known as a kind of pulse radar device.

  In this radio wave induction seeker device, transmission spire power, transmission pulse width, wavelength, antenna gain, signal processing loss, etc. are required to satisfy the required maximum detection distance.

At this time, the transmission blind occurs as a time during which the reflected wave cannot be received corresponding to the transmission pulse width, and therefore, there is a problem that the radar cannot observe when the distance to the target is short.
"Maximum detection distance and pulse width" MERRILL I. SKOLNIK, "RADAR HANDBOOK", McGRAW-HILL (1990) pp. 2.1.2.11

  As described above, since transmission blinds corresponding to the transmission pulse width occur, there is a problem that a target in a region shorter than the transmission pulse width cannot be tracked.

  In addition, when the transmission pulse width is varied, the reception band changes, so that a problem arises that a circuit bank becomes large because a filter bank or the like composed of an analog circuit is required.

  The present invention has been made in view of the above, and as an object thereof, a radio wave induction seeker device capable of shortening a blind distance at a short distance by changing a transmission pulse width according to a distance from a target. Is to provide.

  In order to solve the above-mentioned problem, the invention according to claim 1 is a detector for receiving a reflected signal from a target and performing quadrature detection to output a received signal, and moving average processing is performed on the received signal after detection. A moving average means for performing band limitation by applying, a gain adjusting means for performing gain adjustment processing on the received signal after the moving average processing, and any one obtained by dividing a basic clock signal into a plurality of stages Resample means for re-sampling the received signal after gain adjustment processing according to two re-sample clock signals, and passing the re-sampled received signal during the period of the gate signal related to the distance to the target. The signal processing means for generating the pulse width control signal based on the error between the different signals, and the transmission means for changing the width of the pulse signal according to the pulse width control signal and transmitting it. To.

  According to the radio wave induction seeker device of claim 1, the received signal after receiving the reflected signal from the target and performing quadrature detection is subjected to moving average processing to limit the band, and the received signal after the moving average processing In accordance with any one resample clock signal obtained by dividing the basic clock signal into a plurality of stages, the received signal after the gain adjustment process is resampled. The received signal after this re-sampling is passed during the period of the gate signal associated with the distance to the target, and a pulse width control signal is generated based on the error between the different signals after this passage, and the pulse width control signal is Since the pulse signal width is changed and transmitted, the transmission pulse width can be varied according to the distance from the target, and the transmission blind distance can be shortened. As a result, it is possible to output a guidance signal up to near the meeting distance with the target.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  The radio wave induction seeker device according to an embodiment of the present invention uses a plurality of antennas. In the following, for the sake of simplicity, the number of antennas is described as “8”, for example, but the number of antennas is not limited to this and is arbitrary.

  FIG. 1 is a block diagram showing a configuration of a radio wave induction seeker device 11 according to an embodiment of the present invention.

  The radio wave induction seeker device 11 includes an antenna 13, a receiver 21, A / D converters 23-1 to 23-3, a reception processing circuit 25, a signal processor 27, and a transmitter 33.

  The antenna 13 is connected to a distribution comparator 19 via phase shifters 17-1 to 17-8 that perform phase adjustment in order to form beams by a plurality of antennas 15-1 to 15-8. As reception signals output from the device 19, ΔAZ representing the difference signal in the azimuth direction, ΔEL representing the difference signal in the elevation direction, and Σ representing the sum signal are output to the receiver 21. The antenna pattern has outputs of a sum beam and a difference beam, and it can be understood from which direction the radio wave comes from the ratio of the sum and difference.

  Further, a transmitter 33 is connected to the distribution comparator 19 of the antenna 13, and a transmission signal from the transmitter 33 is input to the distribution comparator 19.

  In FIG. 1, the transmission signal output from the transmitter 33 to the distribution comparator 19 of the antenna 13 is signal-distributed by the distribution comparator 19 and output to the phase shifters 17-1 to 17-8 and the phase shifters 17-1 to 17-17. At -8, the phase of the transmission signal is adjusted, beam formation is performed by the antennas 15-1 to 15-8, and the transmission signal is radiated into space. The transmission signal is reflected by the target and input to the antenna 13, and the reception signals ΔAZ, ΔEL, and Σ are output from the distribution comparator 19 to the receiver 21.

  The receiver 21 performs A / D conversion on the intermediate frequency signal IF obtained by mixing the input reception signals ΔAZ, ΔEL, and Σ with a signal having a predetermined frequency oscillated by a local oscillator (not shown). It outputs to the devices 23-1 to 23-3.

  In the A / D converters 23-1 to 23-3, the reception signals ΔAZ, ΔEL, and Σ converted to the intermediate frequency signals are A / D converted into digital signals and output to the reception processing circuit 25.

  The reception processing circuit 25 generates orthogonal signals ΔAZ (I / Q), ΔEL (I / Q), Σ obtained by performing orthogonal detection and band limitation on the received signals ΔAZ, ΔEL, Σ converted into digital signals. (I / Q) is output to the signal processor 27.

  The signal processor 27 calculates the predicted distance from the target, and outputs the calculation result to the transmitter 33 and the reception processing circuit 35 as a pulse width control signal. Corresponding to the pulse width control signal output from the signal processor 27, the transmitter 33 controls the transmission pulse width, and at the same time, the reception processing circuit 25 controls the pass band.

  Specifically, the signal processor 27 includes a predicted distance calculator 29 and a pulse width controller 31. The predicted distance calculator 29 receives the received signal Σ (I / Q) from the resampler 63 and obtains the distance error ΔR from the distance error voltage Δr calculated using an arithmetic expression described later. The pulse width controller 31 determines the range of the above-mentioned distance error voltage Δr, adds, for example, an instruction value of 2 bits to the pulse width control signal and outputs it. The signal processor 27 normally performs angle measurement processing based on ΔAZ (I / Q), ΔEL (I / Q), and Σ (I / Q) input from the reception processing circuit 25. Since there is no direct relationship, the description thereof will be omitted.

  Thereby, the transmission blind distance is shortened as the target is approached. For example, FIG. 8 shows the relationship between the transmission pulse width and the blind distance.

  Next, FIG. 2 is a detailed block diagram from the receiver shown in FIG. 1 to the A / D converter and the reception processing circuit.

  The intermediate frequency received signal that has passed through the analog filter 22 provided in the receiver 21 is sampled and converted into a digital signal by an A / D converter 23-1, for example, to which an 80 MHz clock signal is input.

  For example, the 80 MHz clock signal is input to a frequency divider 41 and an NCO (Numerically Controlled Oscillator) 43. , 55, and 10 MHz, 5 MHz, 2.5 MHz, and 1.25 MHz clock signals divided by 1/8 to 1/64 by the frequency divider 41 are output to the selector 61.

  The NCO 43 outputs a COS signal consisting of a plurality of bits and forming a COS waveform to the multiplier 45 as time elapses, and outputs a SIN signal having a phase difference of 90 ° from the COS signal to the multiplier 47.

  The multiplier 45 receives the reception signal converted by the A / D converter 23-1 and the COS signal output from the NCO 43, and the Q signal is detected by the digital filter 49 as a detection signal represented by the multiplication result of both. Is output. On the other hand, to the multiplier 47, the reception signal converted by the A / D converter 23-1 and the SIN signal output from the NCO 43 are input, and the I signal is digitally converted as a detection signal represented by the multiplication result of both. It is output to the filter 51. As a result, the multipliers 45 and 47 perform I / Q quadrature detection.

  Thereafter, the digital filters 49 and 51 remove the harmonic image from the I and Q signals resulting from the multiplication and output them to the moving average filters 53 and 55. The moving average filters 53 and 55 perform band limitation on the passing I and Q signals and output band-limited I and Q signals to the gain adjusters 57 and 59, respectively.

  The gain adjusters 57 and 59 are composed of a shift register that reduces the input signal in steps of, for example, 1/2, 1/4, and 1/8, and controls the pulse width output from the signal processor 27. The amount of decrease is determined according to the 2-bit selection signal indicated by the signal, and the I signal and Q signal as a result of gain adjustment are output to the resampler 63.

  On the other hand, the pulse width control signal output from the signal processor 27 is input to the selector 61, which selects 10 MHz, 5 MHz, 2.5 MHz, 1... According to the 2-bit selection signal indicated by this signal. Any one of the 25 MHz clock signals is selected and output to the resampler 63 as a resample CLK.

  The resampler 63 resamples the I and Q signals whose gains have been adjusted by the gain adjusters 57 and 59 by the resample CLK output from the selector 61, and the resulting I and Q signals are converted into the signal processor 27. Is output.

  Hereinafter, the operation of the radio wave induction seeker apparatus will be described in detail with reference to the drawings.

  The radio wave induction seeker device shown in the present embodiment is configured to switch the transmission pulse width in accordance with the distance from the target. In addition, the reception band changes when the transmission pulse width is switched. The reception band is also made variable by digital processing at the same time. With this configuration, even if there is a pulse width necessary for maximum detection, if the distance to the target is shortened, the transmission pulse width can be reduced in terms of power, and the distance to the target can be accommodated. If the pulse width is narrowed, the blind distance is shortened according to the distance.

  First, when the IF center frequency of the received signal input from the receiver 21 to the A / D converter 23-1 is selected to be, for example, 180 MHz, the A / D converter 23-1 underlines at, for example, 80 MHz as shown in FIG. Sampled. The analog filter provided in the receiver 21 in the front stage of the A / D converter 23-1 is a filter for aliasing measures, and is a filter whose pass band is determined by the shortest pulse width.

  Since the center frequency after digital conversion by the A / D converter 23-1 is 20 MHz, a 20 MHz SIN signal and COS signal are generated by the NCO 43, and the result of multiplication by the multipliers 45 and 47 is multiplied by the digital filter 49. 51, I / Q quadrature detection is performed by removing the image.

Here, Expressions 1 and 2 show multiplication expressions applied by the multipliers 45 and 47, respectively. The input received signal S = COS (ω ₁ t) is multiplied by the COS signal (COS (ω ₂ )) output from the NCO 43 and the SIN signal (SIN (ω ₂ )) as follows.

  The digital filters 49 and 51 perform processing for removing this image. The digital filters 49 and 51 are configured as shown in FIG. 3, for example. For simplification of description, only the configuration of the digital filter 49 will be described, but the same applies to the digital filter 51.

  A multiplication result composed of a plurality of bits output from the multiplier 45 is input to the digital filter 49. This input signal has a function of a low-pass filter composed of multipliers 71-1 to 71-7, latches 73-1 to 73-6, and adders 75-1 to 75-6 shown in FIG. The result is input to the seven-element filter, and the final calculation result is output from the adder 75-6.

  Here, the frequency response of the digital filters 49 and 51 is shown in FIG. As shown in FIG. 4, it can be seen that by passing through the digital filters 49 and 51, it gradually attenuates to, for example, about −55 dB between 0 MHz and 20 MHz, and an image of 20 MHz or more is removed to, for example, about −55 dB. .

  Next, the I signal and the Q signal output from the digital filters 49 and 51 are band-limited by moving average processing through the moving average filters 53 and 55, respectively, and this moving average result is input to the gain adjusters 57 and 59. The amount of decrease is determined according to the pulse width control signal, and the I signal and Q signal resulting from the gain adjustment are output to the resampler 63.

  Next, the resampler 63 resamples the I signal and Q signal gain-adjusted by the gain adjusters 57 and 59 with the resample CLK divided by the same frequency clock as the reception band corresponding to the transmission pulse. Is done.

  As a result, as shown in FIGS. 5 to 7, the reception band can be limited to the transmission pulse width. FIG. 9 shows the relationship between the transmission pulse width and the reception band.

  As shown in FIG. 10, the moving average numbers are 4, 8, and 16 corresponding to the transmission pulse widths of 0.2 μs, 0.4 μs, and 0.8 μs, respectively. In FIGS. 5 to 7, the horizontal axis represents the normalized frequency, and therefore, when the numerical value on the horizontal axis is multiplied by 80 MHz, it corresponds to the frequency shown as an example.

  As can be seen from FIG. 5 to FIG. 7, as a result of the moving average processing performed by the moving average filters 53 and 55, the gain differs in each case of the moving average numbers 4, 8 and 16. , 59 to perform gain adjustment processing.

  The gains of the signals output from the moving average filters 53 and 55 are 12 dB, 18 dB, and 24 dB as shown in FIGS. 5 to 7, so that the gain adjusters 57 and 59 can adjust the gain by bit shift. .

  Next, prediction distance calculation by the prediction distance calculator 29 provided in the signal processor 6 shown in FIG. 1 will be described.

  Here, the radio wave induction seeker device waits by opening the gate with the gate signal at the time when the received signal will return.

  At this time, as shown in FIG. 11, for example, an M (Medium) gate for the center distance (position) with respect to the received pulse width, and a distance (position) ahead of the pulse width by half of the pulse width. The target E (Eariy) gate, as well as the L (Late) gate for the distance (position) behind the pulse width by a half of the pulse width, overlap each other by 50%, Sample the data in the range direction.

  Here, the gate widths of E, M, and L described above depend on the transmission pulse width, and correspond to the range of the moving average processing by the moving average filter provided in the reception processing circuit 25. As a result of this processing, three signals E, M, and L are obtained.

  Note that distance tracking can be performed by using the signals E, M, and L described above. As shown in FIG. 12, when the signal amplitude of the reflected wave reflected from the target is modeled by a triangular wave as shown in FIG. 12, the distance tracking principle is obtained by subtracting the L signal from the E signal and the distance error voltage Δr ( Δr = E−L) occurs. The predicted distance calculator 29 receives the sum signal Σ (I / Q) from the resampler 63, and when there is a reflected signal (received signal) from the target within the range of the M gate, the distance error voltage Δr is Vref1˜. When it is between Vref2 and within the range of the E gate, the distance error voltage Δr is a positive voltage of Vref2 or more, and when within the range of the L gate, the distance error voltage Δr is a negative voltage of Vref2 or less.

Here, the distance error voltage Δr is

It becomes.

  The predicted distance calculator 29 obtains a distance error ΔR from the distance error voltage Δr calculated in (Equation 3) using generally known α and β filters. The predicted distance calculator 29 controls a gate signal waiting for the next time.

  As shown in FIG. 8, when the distance predicted by the α and β filters is 120 m or more, the distance less than 120 m is 60 m or more, and the distance less than 60 m and 30 m or more is converted into a combination of indicated values of 2 bits. The pulse width controller 31 outputs the 2-bit indication value combination (00, 01, 10) as a pulse width control signal to the selector 61 and the gain adjusters 57, 59 provided in the reception processing circuit 25 and transmits them. Output to the device 33.

  In the selector 61 provided in the reception processing circuit 25 to which the pulse width control signal output from the pulse width controller 31 is input, 10 MHz, 5 MHz, 2.5 MHz, and the like according to the 2-bit instruction value included in the pulse width control signal. A resample clock signal is selected from 1.25 MHz and output to the resampler 63.

  At the same time, in the gain adjusters 57 and 59, the output gain of the gain adjusters 57 and 59 is set to any one of 12 dB, 18 dB, and 24 dB according to the 2-bit instruction value included in the pulse width control signal. Adjust the gain.

  At the same time, the transmitter 33 sets the width of the pulse signal to be transmitted according to the 2-bit instruction value included in the pulse width control signal, and transmits this pulse signal.

  In this way, the received signal after receiving the reflected signal from the target and performing quadrature detection is subjected to moving average processing to limit the band, and the received signal after this moving average processing is subjected to gain adjustment processing. Then, according to any one of the resample clock signals obtained by dividing the basic clock signal into a plurality of stages, the received signal after the gain adjustment processing is resampled, and the resampled received signal is set as a target. The pulse width control signal is generated based on the error between the different signals after the passage, and the width of the pulse signal is set in accordance with the pulse width control signal. Since transmission is performed after changing, the transmission pulse width can be varied according to the distance from the target, and the transmission blind distance can be shortened. As a result, it is possible to output a guidance signal up to near the meeting distance with the target.

  In addition, by controlling the gain according to the generated pulse width control signal and selecting the resample clock signal, the reception band can be varied in response to the variation of the transmission pulse width. .

It is a figure which shows the structure of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is a figure which shows the reception processing circuit of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is a figure which shows the digital filter structure of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is a figure which shows the frequency response of the digital filter of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is explanatory drawing of the zone | band limitation by 4 moving averages of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is explanatory drawing of the band restriction | limiting by 8 moving averages of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is explanatory drawing of the zone | band limitation by 16 moving averages of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is a table | surface which shows the relationship between the pulse width and blind distance of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is a table | surface which shows the relationship between the pulse width of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention, and a receiving band. It is a table | surface which shows the relationship between the pulse width of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention, and a moving average number. It is explanatory drawing of E, M, and L gate of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention. It is explanatory drawing of the distance error voltage using E, M, and L gate of the electromagnetic wave induction seeker apparatus which concerns on one embodiment of this invention.

Explanation of symbols

13 Antenna 21 Receiver 22 Analog Filter 23 A / D Converter 25 Reception Processing Circuit 27 Signal Processor 29 Predicted Distance Calculator 31 Pulse Width Controller 33 Transmitter 41 Divider 43 NCO
45, 47 Multiplier 49, 51 Digital filter 53, 55 Moving average filter 57, 59 Gain adjuster 61 Selector 63 Resampler

Claims (2)

Detecting means for receiving a reflected signal from a target, performing quadrature detection and outputting a received signal;
Moving average means for performing band averaging on the received signal after detection to limit the bandwidth;
Gain adjusting means for performing gain adjustment processing on the received signal after the moving average processing;
Re-sampling means for re-sampling the received signal after the gain adjustment processing according to any one re-sampling clock signal obtained by dividing the basic clock signal into a plurality of stages;
A signal processing means for passing the resampled received signal during a period of the gate signal associated with the distance to the target, and generating a pulse width control signal based on an error between the different signals after the passage;
A radio wave induction seeker device comprising: transmission means for changing the width of the pulse signal according to the pulse width control signal and transmitting the pulse signal.   2. The radio wave according to claim 1, wherein a gain of the gain adjusting unit is controlled in accordance with a pulse width control signal generated by the signal processing unit, and a resample clock signal of the resample unit is selected. Induction seeker device.

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