JP2006271172A - Switching power circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To improve power conversion efficiency and reduce circuit components as a power circuit which has a power factor for improving the functions, capable of coping with a wide voltage range of commercial AC power. <P>SOLUTION: A current resonance type converter, which is equipped with a rectifying and smoothing means for switching a full-wave rectifying action and a voltage-doubling and rectifying action, according to the voltage of a commercial power source, is provided with a power regeneration system of power factor improving circuit 10, and is provided with a series resonance circuit on the secondary side also, whereby a coupled resonance circuit by the electromagnetic coupling of a converter PIT is made, and to obtain unimodal characteristics as to this coupled resonance circuit, approximately 0.7 is set as to the coupling coefficient of the converter transformer PIT itself. Hereby, an active filter can be dispensed with, thereby improving the AC→DC power conversion efficiency. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit used as a power supply for various electronic devices.

JP-A-6-327246

  In recent years, the development of switching elements that can withstand relatively high currents and voltages at high frequencies has led to most switching power supply circuits as power supply circuits that rectify commercial power and obtain a desired DC voltage. . The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

  By the way, in general, when a commercial power supply is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the power supply utilization efficiency is impaired. In addition, a countermeasure for suppressing harmonics generated by such a distorted current waveform is required. As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V, for example, to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V AC region such as Europe. A so-called wide-range power supply circuit that is configured so as to be able to be used is known.

  Here, as the above-mentioned resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known. As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched and driven by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, fs = 50 KHz to 250 KHz. It is about.

  In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 300 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve each of these problems, a method using a so-called active filter is known as a conventional technique for realizing a power factor improvement and a wide-range configuration (see, for example, Patent Document 1).

  The basic configuration of such an active filter is, for example, as shown in FIG. In FIG. 15, a bridge rectifier circuit Di is connected to a commercial AC power source AC. An output capacitor Cout is connected in parallel to the positive / negative line of the bridge rectifier circuit Di. By supplying the rectified output of the bridge rectifier circuit Di to the output capacitor Cout, a DC voltage Vout is obtained as a voltage across the output capacitor Cout. This DC voltage Vout is supplied as an input voltage to a load 50 such as a subsequent DC-DC converter.

  As shown in the figure, the power factor improvement includes an inductor L, a fast recovery diode D, a resistor Ri, a switching element Q, and a multiplier 11. The inductor L and the diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout. The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the output capacitor Cout.

  In this case, the switching element Q1 is a MOS-FET, and is inserted between the connection point of the inductor L and the diode D and the primary side ground as shown in the figure.

  To the multiplier 11, a current detection line LI and a waveform input line Lw are connected as a feedforward circuit, and a voltage detection line LV is connected as a feedback circuit. The multiplier 11 detects the level of the rectified current that flows from the current detection line LI and flows to the negative output terminal of the bridge rectifier circuit Di.

  Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.

  Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV.

  The multiplier 11 outputs a drive signal for driving the switching element Q.

  The multiplier 11 first multiplies the rectified current level detected from the current detection line LI as described above and the fluctuation difference of the DC input voltage detected from the voltage detection line LV. Then, a current command value having the same waveform as the AC input voltage VAC is generated based on the multiplication result and the waveform of the AC input voltage detected from the waveform input line Lw.

  Furthermore, the multiplier 11 in this case compares the current command value with the actual AC input current level (detected based on the input from the current detection line LI), and performs PWM control on the PWM signal according to this difference. And generate a drive signal based on the PWM signal. The switching element Q is switched by this drive signal. As a result, the AC input current is controlled to have the same waveform as the AC input voltage, and the power factor is improved so that the power factor approaches one. In this case, the current command value generated by the multiplier is controlled such that the amplitude changes according to the fluctuation difference of the rectified voltage, so that the fluctuation of the rectified voltage is also suppressed.

  FIG. 16A shows the input voltage Vin and the input current Iin input to the active filter circuit shown in FIG. The voltage Vin corresponds to the voltage waveform as the rectified output of the bridge rectifier circuit Di, and the current Iin corresponds to the current waveform as the rectified output of the bridge rectifier circuit Di. Here, the waveform of the current Iin has the same conduction angle as the rectified output voltage (voltage Vin) of the bridge rectifier circuit Di. This is the waveform of the AC input current flowing from the commercial AC power supply AC to the bridge rectifier circuit Di. Also indicates that the conduction angle is the same as that of the current Iin. That is, a power factor close to 1 is obtained.

  FIG. 16B shows a change in energy (power) Pchg input / output to / from the output capacitor Cout. The output capacitor Cout stores energy when the input voltage Vin is high, and releases energy when the input voltage Vin is low to maintain the flow of output power.

  FIG. 16C shows a waveform of the charge / discharge current Ichg with respect to the output capacitor Cout. This charge / discharge current Ichg flows in correspondence with the energy Pchg accumulation / discharge operation in the output capacitor Cout, as can be seen from the fact that it is in phase with the waveform of the input / output energy Pchg in FIG. It is.

  Unlike the input current Iin, the charge / discharge current Ichg has substantially the same waveform as the second harmonic of the AC line voltage (commercial AC power supply AC). In the AC line voltage, a ripple voltage Vd is generated in the second harmonic component as shown in FIG. 16D due to the flow of energy with the output capacitor Cout. The ripple voltage Vd has a phase difference of 90 ° with respect to the charge / discharge current Ichg shown in FIG. The rating of the output capacitor Cout is determined in consideration of processing the second harmonic ripple current and the high frequency ripple current from the boost converter switch that modulates the current.

  FIG. 17 shows a configuration example of an active filter having a basic control circuit system based on the circuit configuration of FIG. In addition, about the part made the same as FIG. 15, the same code | symbol is attached | subjected and description is abbreviate | omitted.

  A switching pre-regulator 51 is provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout. The switching preregulator 51 is a part formed by the switching element Q, the inductor L, the diode D, and the like in FIG.

  The control circuit system including the multiplier 11 includes an error voltage amplifier 12, a divider 13, and a squarer 14 in addition. In the error voltage amplifier 12, the DC voltage Vout of the output capacitor Cout is divided by a voltage dividing resistor Rvo-Rvd and input to the non-inverting input of the operational amplifier 15. A reference voltage Vref is input to the inverting input of the operational amplifier 15. The operational amplifier 15 amplifies a voltage of a level corresponding to the error of the divided DC voltage Vout with respect to the reference voltage Vref by an amplification factor determined by the feedback resistor Rvl and the capacitor Cvl, and outputs the error output voltage Vvea to the divider 13. Output to.

  Further, a so-called feedforward voltage Vff is input to the squarer 14. The feedforward voltage Vff is an output (average input voltage) obtained by averaging the input voltage Vin by the averaging circuit 16 (Rf11, Rf12, Rf13, Cf11, Cf12). The squarer 14 squares the feedforward voltage Vff and outputs it to the divider 13.

  The divider 13 divides the error output voltage Vvea from the error voltage amplifier 12 by the square value of the average input voltage output from the squarer 14, and outputs a signal as a result of the division to the multiplier 11. That is, the voltage loop is made up of a system of a squarer 14, a divider 13, and a multiplier 11. The error output voltage Vvea output from the error voltage amplifier 12 is divided by the square of the average input voltage (Vff) before being multiplied by the rectified input signal Ivac in the multiplier 11. With this circuit, the gain of the voltage loop is kept constant without changing as the square of the average input voltage (Vff). The average input voltage (Vff) has a function of open loop correction that is sent forward in the voltage loop.

  The multiplier 11 receives the output obtained by dividing the error output voltage Vvea by the divider 13 and the rectified output (Iac) of the positive output terminal (rectified output line) of the bridge rectifier circuit Di via the resistor Rvac. Here, the rectified output is shown not as a voltage but as a current (Iac). The multiplier 11 multiplies these inputs to generate and output a current programming signal (multiplier output signal) Imo. This corresponds to the current command value described with reference to FIG. The output voltage Vout is controlled by varying the average amplitude of this current programming signal. That is, a PWM signal corresponding to a change in the average amplitude of the current programming signal is generated, and switching drive is performed by a drive signal based on the PWM signal, thereby controlling the level of the output voltage Vout.

  Thus, the current programming signal has an average amplitude waveform that controls the input and output voltages. Note that the active filter controls not only the output voltage Vout but also the input current Vin. Since the current loop in the feedforward circuit can be said to be programmed by the rectified line voltage, the input to the subsequent converter (load 50) becomes resistive.

  FIG. 18 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG. In the power supply circuit shown in this figure, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

  First, in the power supply circuit shown in FIG. 18, two sets of line filter transformers LFT and LFT and three sets of across capacitors CL are connected to the commercial AC power supply AC according to the illustrated connection mode. The bridge rectifier circuit Di is connected through the inrush current limiting circuit 22 shown in the figure. The inrush current limiting circuit 22 includes a parallel connection circuit including an inrush current limiting resistor Ri and a switch SW. For example, when the switch SW is turned off by an external signal, the commercial AC power supply side at the time of starting the power supply circuit Inrush current flow from is restricted.

  The rectified output line of the bridge rectifier circuit Di is connected to a normal mode noise filter 24 formed by connecting one set of choke coils LN and two sets of filter capacitors (film capacitors) CN as shown in the figure. Yes.

  The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci via a series connection of the choke coil LN, the inductor Lpc of the power choke coil PCC, and the fast recovery type rectifier diode D10. The smoothing capacitor Ci corresponds to the output capacitor Cout in FIGS. Further, the inductor Lpc and the rectifier diode D10 of the power choke coil PCC correspond to the inductor L and the diode D shown in FIG. In addition, an RC snubber circuit including a capacitor Csn and a resistor Rsn is connected in parallel to the rectifier diode D10 in this figure.

  The switching element Q6 corresponds to the switching element Q in FIG. That is, in actually mounting the switching element of the active filter, in this case, the switching element Q6 is connected to the connection point between the power choke coil Lpc and the fast recovery type rectifier diode D10, and the primary side ground (negative rectified output line). It is inserted between. In this case, a MOS-FET is selected as the switching element Q6. A gate-source resistor R52 is connected between the gate and source of the switching element Q6.

  In this case, the active filter control circuit 20 controls the operation of the active filter for improving the power factor so that the power factor is close to 1, and is, for example, a single stone integrated circuit (IC). In this case, the active filter control circuit 20 includes a multiplier, a divider, an error voltage amplifier, a PWM control circuit, a drive circuit that outputs a drive signal for switching the switching element, and the like. Circuit sections corresponding to the multiplier 11, error voltage amplifier 12, divider 13, and squarer 14 shown in FIG. 17 are mounted in the active filter control circuit 20.

  In this case, the feedback circuit is formed so that the voltage value obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistors R56 and R57 is input to the terminal T1 of the active filter control circuit 20. .

  As the feedforward circuit, first, the rectified output is input to the terminal T3 via the resistor R58. This forms an AC input voltage waveform detection and corresponding feedforward circuit for the averaging circuit.

  Further, the rectified current level is input to the terminal T6 through the resistor R60 from the connection point between the negative output terminal of the bridge rectifier circuit Di and the resistor R61 inserted between the primary side ground. That is, a feedforward circuit is formed as a line corresponding to the current detection line LI in FIG.

  Further, the positive rectified output of the bridge rectifier circuit Di via the starting resistor Rs is input to the terminal T4 as the starting voltage. The active filter control circuit 20 is activated by the activation voltage input to the terminal T4 when the power supply is activated.

  In the power choke coil PCC, a winding N5 that is transformer-coupled with the inductor Lpc is wound. The alternating voltage excited in the winding N5 is converted into a predetermined low-voltage DC voltage by a half-wave rectifier circuit comprising a diode D11 and a capacitor C11. This low-voltage DC voltage is also input to the terminal T4. . After being activated by the activation voltage, the active filter control circuit 20 is operated by inputting this low-voltage DC voltage as a power source. The terminal T5 is connected to the primary side ground via a resistor R59.

  A drive signal for driving the switching element is output from the terminal T2. The drive signal output from the terminal T2 is output to the gate of the switching element Q6 via the resistor R51.

  In the switching element Q6, a gate voltage is generated at both ends of the gate-source resistor R52 in accordance with an applied drive signal. Then, the switching operation is performed so that the gate voltage is turned on when the gate voltage becomes equal to or higher than the threshold value, and turned off when the gate voltage becomes lower than the threshold value.

  The switching drive of the switching element Q6 is a drive based on PWM control so that the conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform as described with reference to FIGS. Done by signal. The conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform. That is, the conduction angle of the AC input current flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC. As a result, the power factor is controlled to be approximately 1. That is, power factor improvement is achieved. In practice, a characteristic of power factor PF = 0.99 to 0.98 is obtained.

  Further, depending on the active filter control circuit 20 shown in FIG. 18, the average value of the rectified and smoothed voltage Ei (corresponding to Vout in FIG. 17) = 375 V is constant in the range of AC input voltage VAC = 85 V to 288 V. Also works. That is, a DC input voltage stabilized at 375 V is supplied to the subsequent-stage current resonance type converter regardless of the fluctuation range of the AC input voltage VAC = 85 V to 288 V.

  The range of the AC input voltage VAC = 85V to 288V continuously covers the commercial AC power supply AC100V system and the 200V system. Therefore, the switching converter at the subsequent stage includes the commercial AC power supply AC100V system and the 200V system, The DC input voltage (Ei) stabilized at the same level is supplied. That is, the power supply circuit shown in FIG. 18 is configured as a wide-range power supply circuit by including an active filter.

  As shown in the figure, the current resonance converter at the latter stage of the active filter includes two switching elements Q1 and Q2. In this case, a half-bridge connection is made such that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and the switching element Q1 is connected in parallel with the rectified and smoothed voltage Ei (DC input voltage). That is, a current resonance type converter by a half bridge coupling method is formed.

  In this case, the current resonance type converter is a separately excited type, and corresponding to this, MOS-FETs are used for the switching elements Q1 and Q2. Damper diodes DD1 and DD2 are connected in parallel to these switching elements Q1 and Q2, respectively, thereby forming a switching circuit. These damper diodes DD1, DD2 form a path through which a reverse current flows when the switching elements Q1, Q2 are turned off.

  The switching elements Q1 and Q2 are driven to be switched by the drive circuit 21 at a required switching frequency at the timing when they are alternately turned on / off. The drive circuit 21 variably controls the switching frequency in accordance with the level of a secondary side DC output voltage Eo described later, thereby stabilizing the secondary side DC output voltage Eo.

  Converter transformer PIT is provided to transmit the switching outputs of switching elements Q1, Q2 from the primary side to the secondary side. One end of the primary winding N1 of the converter transformer PIT is connected to a connection point (switching output point) of the switching elements Q1 and Q2, and the other end is primary through the primary side series resonance capacitor C1. Connected to side ground. Here, the primary side series resonance capacitor C1 forms a series resonance circuit by its own capacitance and the leakage inductance (L1) of the primary winding N1. This series resonance circuit causes a resonance operation when the switching outputs of the switching elements Q1 and Q2 are supplied. By this, the operation of the switching circuit composed of the switching elements Q1 and Q2 is made a current resonance type.

  Although the description by illustration here is abbreviate | omitted, as a structure of the converter transformer PIT mentioned above, the EE type | mold core which combined the E type | mold core by the ferrite material is provided, for example. Then, after the winding part is divided between the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the inner magnetic leg of the EE core. Further, a gap of about 1.0 mm or less is formed with respect to the inner magnetic leg of the EE type core of the converter transformer PIT, and the primary winding N1 and the secondary winding N2 are about 0.80 to 0.90. The coupling coefficient is obtained.

  The secondary winding N2 of the converter transformer PIT is provided with a center tap as shown in the figure and connected to the secondary side ground, and then comprises rectifier diodes Do1 and Do2 and a smoothing capacitor Co as shown in the figure. Both wave rectifier circuits are connected. As a result, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the drive circuit 21. As described above, the drive circuit 21 switches the switching frequency so that the secondary side DC output voltage Eo is stabilized based on the level of the input secondary side DC output voltage Eo. Elements Q1 and Q2 are driven. That is, stabilization by the switching frequency control method is performed.

  As can be seen from the above description, the power supply circuit shown in FIG. 18 is configured by mounting the conventionally known active filter shown in FIGS. 15 and 17. By adopting such a configuration, the power factor is improved. However, the power supply circuit having the configuration shown in FIG. 18 has the following problems.

  As a result of actual experiments on the power supply circuit of FIG. 18, the AC → DC power conversion efficiency in the active filter (ηAC → DC is about ηAC → DC = 92% when the AC input voltage VAC = 100V, AC input voltage Under the condition of VAC = 230V, ηAC → DC = 95%.

  Further, the power conversion efficiency of the current resonance type converter when the average value of the DC input voltage Ei generated by the active filter is set to Ei = 375 V and the load power Po = 300 W is ηDC → DC = 94 %.

  For this reason, the total efficiency of the circuit shown in FIG. 18 is represented by the product of the power conversion efficiency of the active filter in the previous stage and the power conversion efficiency of the current resonant converter, and ηAC → DC = 86 at VAC = 100V. About 5%. Further, the overall efficiency at VAC = 230 V is about ηAC → DC = 89.3%, and an efficiency of 90% or more cannot be obtained.

  In addition, since the active filter circuit is a hard switching operation, the level of noise generation is very high, and thus a relatively severe noise suppression measure is required. For this reason, in the circuit shown in FIG. 18, a noise filter composed of two sets of line filter transformers LFT and three sets of across capacitors is formed on the line of the commercial AC power supply AC in order to suppress common mode noise. Yes. That is, two or more stages of line noise filters are required.

  Further, a normal mode noise filter including one set of choke coils LN and two sets of filter capacitors CN is provided for the rectified output line. Further, an RC snubber circuit is provided for the rectifying fast recovery type rectifying diode D10. Thus, an actual circuit requires countermeasures against noise due to a very large number of parts, resulting in a complicated circuit configuration and an increase in cost, and an increase in the mounting area of the power supply circuit board.

  Furthermore, while the switching frequency of the switching element Q6 operated by the active filter control circuit 20 as a general-purpose IC is fixed, the switching frequency of the subsequent current resonance type converter varies within a range of about 80 kHz to 200 kHz, for example. . Since the switching timings of the two are independently performed in this way, the primary side ground potential interferes and becomes unstable due to the switching operation of the two, and abnormal oscillation is likely to occur, for example. As a result, for example, circuit design becomes difficult, and problems such as deterioration in reliability are caused.

  Therefore, in the present invention, in view of the various problems described above, the switching power supply circuit is configured as follows.

  That is, a commercial AC power supply is input to generate a rectified and smoothed voltage, and the rectified and smoothed voltage having a level corresponding to the commercial AC power supply level is generated according to the level of the input commercial AC power supply. A rectifying / smoothing means that is switched between an equal voltage rectifying operation and a voltage rectifying operation that generates the rectified and smoothed voltage at a level corresponding to a predetermined multiple of the commercial AC power supply level is provided.

  In addition, there are provided switching means having a switching element for switching the rectified voltage supplied from the rectifying / smoothing means, and switching drive means for generating a drive signal for switching the switching element.

  A converter formed by winding at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the primary winding. A transformer is provided.

  A primary side series resonant circuit formed by at least a leakage inductance component of the primary winding of the converter transformer and a capacitance of a primary side series resonant capacitor connected in series to the primary winding; and at least the converter transformer. And a secondary side series resonance circuit formed by a leakage inductance component of the secondary winding and a capacitance of a secondary side series resonance capacitor connected in series to the secondary winding.

  Further, a secondary side DC output voltage generating means for generating a secondary side DC output voltage by performing a rectifying operation by inputting a resonance output obtained in the secondary side series resonance circuit, and a secondary side DC output voltage of the secondary side DC output voltage Voltage control means for changing the frequency of the drive signal according to the level and setting the secondary side DC output voltage to a predetermined voltage value.

  And a power factor correction circuit for improving a power factor by supplying a current corresponding to a current flowing through the primary side series resonant circuit or the secondary side series resonant circuit to the rectifying and smoothing means.

  Then, with respect to the electromagnetic coupling type resonance circuit formed by including the primary side series resonance circuit and the secondary side series resonance circuit, the output characteristic of the secondary side DC output voltage with respect to the frequency of the drive signal becomes a single peak characteristic. It was characterized by that.

The switching power supply circuit having the above configuration includes the rectifying / smoothing means for switching between the equal voltage rectifying operation and the voltage doubler rectifying operation according to the level of the commercial AC power input. The range of the voltage supplied to can be made narrow, and it is possible to cope with a wide range of commercial AC power supply levels. Furthermore, the output characteristics of the electromagnetically coupled resonant circuit formed by the primary winding, secondary winding, primary side series resonant capacitor, and secondary side series resonant capacitor of the converter transformer are unimodal. Therefore, the secondary side DC output voltage obtained from the secondary side DC output voltage generating means can be set to a predetermined voltage value in a narrow frequency range of the drive signal of the switching element.
And since it has the power factor improvement circuit which supplies the electric current according to the electric current which flows into a primary side series resonance circuit or a secondary side series resonance circuit to a rectification smoothing means and improves a power factor, it can make a power factor favorable. .

  In this way, according to the present invention, the equal voltage rectification operation for generating the rectified and smoothed voltage at a level corresponding to the commercial AC power supply level and the rectification at the level corresponding to the predetermined multiple of the commercial AC power supply level. Since the rectifying / smoothing means that is switched between the voltage doubler rectifying operation for generating the smoothing voltage is provided, it is possible to deal with a commercial AC power supply in a wide power supply voltage range while maintaining high efficiency.

  In addition, since the output characteristic of the secondary side DC output voltage with respect to the frequency of the drive signal has become a single peak characteristic, the variable control range (necessary control range) of the switching frequency necessary for constant voltage control is reduced, and noise is reduced. In order to suppress, the filter band may be narrow, and the noise filter can be simplified.

  In addition, as a resonant converter having a power factor improving function, a configuration without an active filter can be adopted. For example, power conversion efficiency is improved as compared with a power supply circuit that improves power factor by an active filter.

  In addition, the power supply circuit of the present invention does not require a large number of component elements for constituting an active filter. Further, the current resonance type converter and the power factor correction circuit constituting the power supply circuit are soft switching operations, and the switching noise is greatly reduced. Therefore, it is not necessary to strengthen the noise filter. For this reason, compared with the prior art, the number of parts is greatly reduced, and the power circuit size can be reduced / lightened. In addition, the cost can be reduced accordingly. Further, since the active filter is omitted, the interference of the primary side ground potential is eliminated, so that the primary side ground potential is also stabilized and the reliability is improved.

  FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit as a first embodiment in the best mode for carrying out the present invention (hereinafter also referred to as an embodiment). The power supply circuit shown in this figure employs a configuration in which a power factor correction circuit 10 is combined with a separately excited current resonance converter using a half-bridge coupling method as a basic configuration on the primary side.

  Further, the switching power supply circuit shown in FIG. 1 has a so-called wide-range configuration capable of operating in accordance with both AC100V system and AC200V system inputs. In this case, the power supply circuit is assumed to be provided as a power supply for a printer device which is a peripheral device of a personal computer, for example, and corresponds to a load fluctuation of 0 W (no load) to 300 W as the load power Po. It is said.

  FIG. 2 is a circuit in which the power factor correction circuit 10 is not provided in the switching power supply circuit shown in FIG. First, referring to FIG. 2, a circuit configuration and a function of a portion that obtains the rectified and smoothed voltage Ei from the commercial AC power supply AC will be described, and a circuit configuration of a separately excited current resonance type converter portion by a half-bridge coupling method will be described. The operation will be described, and finally, the switching power supply circuit shown in FIG. 1 including the circuit configuration of the power factor correction circuit 10 and its operation will be described with reference to FIG.

  In FIG. 2, power from a commercial AC power supply AC is input to a common mode noise filter formed by an across capacitor CL and a common mode choke coil CMC.

  The input side of the bridge rectifier circuit Di formed by the rectifier diode Da or the rectifier diode Dd is connected to the output side of the common mode filter, and two smoothing connected in series to the output side of the bridge rectifier circuit Di. A capacitor Ci1 and a smoothing capacitor Ci2 are connected. Then, according to the level of the commercial AC power input, the same voltage rectification operation for generating the rectified smoothing voltage having a level corresponding to the commercial AC power level equal to the same level, and a predetermined multiple of the commercial AC power level Switching is performed between the voltage doubler rectification operation for generating the level of the rectified and smoothed voltage.

  Here, the bridge rectifier circuit Di, the smoothing capacitor Ci1, and the smoothing capacitor Ci2 constitute an example of the rectifying and smoothing means. The rectified and smoothed voltage Ei is obtained as a voltage across the series connection circuit of the smoothing capacitors Ci1 and Ci2.

  The specific connection mode of the rectifying / smoothing means in the present embodiment is different between the equal voltage rectification operation and the voltage double rectification operation. In the equal voltage rectification operation, the cathode which is the positive polarity end of the rectification diode Da and the rectification diode Db is connected to the positive polarity end which is the side end of the smoothing capacitor Ci1 connected in series, and the rectification diode Dc and the rectification diode are connected. The anode which is the negative terminal of Dd is connected to the negative terminal which is the negative terminal of the smoothing capacitor Ci2, the connection point between the negative terminal of the rectifier diode Da and the positive terminal of the rectifier diode Dc, The electric power from the commercial AC power supply AC is input between the connection point between the negative polarity end of the diode Dc and the positive polarity end of the rectifier diode Dd.

  On the other hand, in the voltage doubler rectification operation, in addition to the connection mode described above, the connection point between the negative polarity end of the rectification diode Db and the positive polarity end of the rectification diode Dd, and the smoothing capacitor Ci1 and the smoothing capacitor Ci2. The connection points are connected to each other. Here, the connection between the connection point of the smoothing capacitor Ci1 and the smoothing capacitor Ci2 is made via the relay switch S.

  The relay switch S is turned on / off according to the driving state of the relay RL connected to the rectifier circuit switching module 5. That is, the rectifier circuit switching module 5 drives the relay RL to switch the operation of the rectifying / smoothing means depending on whether the input of the commercial AC power supply AC is a 100V system or a 200V system.

  For this reason, the detection voltage output from the voltage detection circuit comprising the diode D-1, the peak hold capacitor C-1, the resistor R1, and the resistor R2 is input to the rectifier circuit switching module 5. Yes. In this detection circuit, first, a commercial AC power supply AC is rectified and smoothed by a half-wave rectifier circuit including a diode D-1, a rectifier diode Dd, and a peak hold capacitor C-1. As described above, a DC voltage of a level corresponding to the commercial AC power supply AC is obtained. This DC voltage is divided by a series connection circuit of a resistor R1 and a resistor R2 connected in parallel to the peak hold capacitor C-1 (hereinafter, this divided voltage is referred to as a divided voltage). Called). The divided voltage is input to the rectifier circuit switching module 5 as a detection voltage.

  This detection voltage is obtained by dividing the DC voltage corresponding to the level of the commercial AC power supply AC as described above according to the resistance values of the resistors R1 and R2, but the peak hold capacitor before the voltage is divided. At both ends of C-1, a voltage approximately 1.4 times the effective value of the commercial AC power supply AC is obtained. That is, the level of the detection voltage indicates the level of the commercial AC power supply AC.

  Further, a relay RL is connected to the rectifier circuit switching module 5, and the rectifier circuit switching module 5 operates the relay RL to control on / off of the relay switch S. Here, it is assumed that no current flows through the relay RL when the level of the detection voltage is smaller than a predetermined value, and in this case, the relay switch S is turned off. In this way, the relay switch S is turned on at the moment when the power from the commercial AC power supply AC is applied, and the voltage doubler rectification operation is performed. As a result, the smoothing capacitor Ci1 and the smoothing capacitor Ci2 connected in series are It is possible to prevent a situation in which a large voltage exceeding a specified value is generated at both ends of the.

  The switching operation of the rectifier circuit configured as described above is as follows. For example, when the AC input voltage VAC is 100 V, a voltage of approximately 141 V is generated at both ends of the peak hold capacitor C-1, and when the AC input voltage VAC is 220 V, the voltage is generated at both ends of the peak hold capacitor C-1. Produces a voltage of approximately 310V.

  When the AC input voltage VAC is 150 V, a voltage of approximately 210 V is generated at both ends of the peak hold capacitor C-1, and this voltage corresponds to a divided voltage obtained by dividing the voltage by the resistor R1 and the resistor R2. The rectifier circuit switching module 5 detects whether the reference voltage is equal to or higher than this reference voltage, and if it is equal to or higher than this reference voltage, if it is equal to or higher than this voltage, a current is passed through the relay RL. First, the relay switch S is turned off.

  Therefore, when the divided voltage divided by the resistor R1 and the resistor R2 is larger than the reference voltage, that is, when the AC input voltage VAC is 150 V or more, the relay switch S is turned off, and the rectifier diode Da and A full-wave rectifying and smoothing circuit in which a smoothing capacitor Ci1 and a smoothing capacitor Ci2 connected in series are connected between the cathode of the rectifying diode Db, the rectifying diode Dc, and the rectifying diode Dd anode is configured. As a voltage across the −Ci2 series connection circuit, a rectified and smoothed voltage Ei corresponding to the AC input voltage VAC is obtained.

  On the other hand, when the divided voltage divided by the resistor R1 and the resistor R2 is smaller than the reference voltage, that is, when the AC input voltage VAC is less than 150V, the relay switch S is turned on and the AC During a period when the input voltage VAC is positive, a rectification current path through which a charging current flows is formed via the rectifier diode Da and the smoothing capacitor Ci1, and thereby, both ends of the smoothing capacitor Ci1 are equal to the AC input voltage VAC. A corresponding rectified and smoothed voltage is obtained. On the other hand, when the AC input voltage VAC is negative, a rectification current path through which the charging current flows is formed via the rectifier diode Da and the smoothing capacitor Ci1, and thereby the AC input voltage VAC is connected to both ends of the smoothing capacitor Ci2. A rectified and smoothed voltage corresponding to the same magnification is obtained. Here, since the smoothing capacitor Ci1 and the smoothing capacitor Ci2 are connected in series, the rectified smoothing voltage Ei which is the voltage across the series connection circuit of the smoothing capacitors Ci1-Ci2 corresponds to twice the AC input voltage VAC. Level to be obtained. That is, a so-called voltage doubler rectifier circuit is formed.

  In this way, in the circuit shown in FIG. 2, in the case of the commercial AC power supply AC100V system, the rectified smoothing voltage Ei corresponding to twice the AC input voltage VAC is generated by the voltage doubler rectification operation, and the commercial AC power supply AC200V In the case of the system, a rectified and smoothed voltage Ei corresponding to an equal magnification of the AC input voltage VAC is generated by a normal full-wave rectification operation. That is, as a result, the same level of rectified and smoothed voltage Ei is obtained when the commercial power input is an AC 100V system and an AC 200V system. This rectified and smoothed voltage Ei is input as a DC input voltage to the subsequent current resonance type converter.

  Next, a description will be given of the circuit configuration and operation of a separately-excited current-resonant converter using a half-bridge coupling method.

  As a current resonance type converter for switching (intermittently) by inputting a DC input voltage, as shown in FIG. 2, a switching circuit in which two switching elements Q1 and Q2 by MOS-FETs are connected by a half bridge connection is used. Prepare. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes included in the switching element Q1 and the switching element Q2, respectively.

  A primary side partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the primary side partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In addition, an oscillation / drive circuit 2 is provided for switching the switching elements Q1, Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and for example, a general-purpose IC can be used. Then, a drive signal (gate voltage) having a required frequency is applied to the gates of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit in the oscillation / drive circuit 2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

  The converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side.

  Here, the converter transformer PIT has a structure as shown in the sectional view of FIG. As shown in this figure, the converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other. And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding (N1) and the secondary side winding (N2) are wound in this way to the EE cores (CR1, CR2), the primary side winding and the secondary side winding are provided. The wire is wound around the inner magnetic leg of the EE core by the different winding regions. In this way, the overall structure of the converter transformer PIT is obtained.

  In addition, a gap G is formed as shown in the figure for the inner magnetic leg of the EE type core. A predetermined width is set as the gap G, whereby a predetermined value is obtained as the value of the coupling coefficient k between the primary side and the secondary side of the converter transformer PIT itself. Note that k = 0.8 or less is obtained as the value of the coupling coefficient k between the secondary side and the secondary side of the actual converter transformer PIT in the switching power supply circuit of the present embodiment. Note that the value of the gap G of the converter transformer PIT is expanded to 2 mm or more (see FIG. 3).

  Returning to FIG. The converter transformer PIT generates a predetermined leakage inductance L1 in the primary winding N1 with the structure described with reference to FIG. A primary side series resonance circuit is formed by at least the capacitance of the primary side series resonance capacitor C1 and the leakage inductance L1.

  According to the connection mode described above, the switching outputs of the switching elements Q1, Q2 are transmitted to the above-described primary side series resonant circuit. Then, the primary side series resonance circuit performs a resonance operation by the transmitted switching output, so that the operation of the primary side switching converter becomes a current resonance type.

  Here, according to the description so far, the primary side switching converter shown in this figure includes the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the above-described primary side partial voltage resonance circuit (Cp). // Partial voltage resonance operation by L1) is obtained.

  That is, on the primary side of the power supply circuit shown in this figure, a configuration in which the resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit is adopted. Here, a switching converter in which two resonance circuits are combined in this way is referred to as a “composite resonance type converter”.

  An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited (induced) in the secondary winding N2 of the converter transformer PIT. In this case, a secondary side series resonant capacitor C2 is connected in series to the winding end end side of the secondary winding N2. As a result, on the secondary side of the converter transformer PIT, a secondary side series resonance circuit is formed by at least the capacitance of the secondary side series resonance capacitor C2 and the leakage inductance L2 of the secondary winding N2. That is, in the present embodiment, a series resonant circuit is formed on each of the primary side and the secondary side of the converter transformer PIT.

  The secondary winding N2 is connected to the secondary side series resonant capacitor C2 as described above, and is connected to four rectifier diodes Do1 to Do4 as shown in the figure. A full-wave rectifier circuit formed by the bridge rectifier circuit and the smoothing capacitor Co is connected.

  In this full-wave rectifier circuit, in one half cycle of the alternating voltage excited (induced) in the secondary winding N2, a pair of rectifier diodes [Do1, Do2] is conducted to rectify the smoothing capacitor Co. An operation of charging current is obtained. Further, in the other half cycle of the alternating voltage excited in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining a set of rectifier diodes [Do3, Do4] conductive.

  As a result, the secondary side DC output voltage Eo having a level corresponding to the same level as the level of the alternating voltage excited in the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo obtained in this way is supplied to a load (not shown) and is also branched and input as a detection voltage for the control circuit 1 described later.

  In addition, since the above-described full-wave rectifier circuit performs a rectifying and smoothing operation on the resonant output of the secondary side series resonant circuit, the secondary side rectifying operation by the full-wave rectifier circuit is also a current resonance type. That is, the rectified current waveform includes a sine waveform based on the resonance frequency of the secondary side series resonance circuit.

  According to the description so far, the switching power supply circuit of the present embodiment includes the primary side series resonance circuit (L1-C1) and the primary side partial voltage resonance circuit (L1 // Cp) on the primary side, and the secondary side Is provided with a secondary side series resonance circuit (L2-C2). As described above, for example, as in the case where only the primary side of FIG. 1 is seen, a switching converter in which two resonance circuits of a series resonance circuit and a partial voltage resonance circuit are combined is a composite resonance type converter. However, a switching converter in which three or more resonance circuits are combined as in the present embodiment is referred to as a multiple resonance type converter.

  The control circuit 1 is provided to stabilize the secondary side DC output voltage Eo by the switching frequency control method. In this case, the control circuit 1 supplies the oscillation / drive circuit 2 with a detection output corresponding to the level change of the secondary side DC output voltage Eo as a detection input. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. For this purpose, the frequency of the oscillation signal generated by the internal oscillation circuit is varied.

  By changing the switching frequency of the switching elements Q1, Q2, the resonance impedance of the primary side series resonance circuit changes, and the amount of power transmitted from the primary winding N1 of the converter transformer PIT to the secondary winding N2 changes. However, this operates so as to stabilize the level of the secondary side DC output voltage Eo.

  Although the details will be described later, the switching frequency control method in the power supply circuit of the present embodiment is based on the resonance frequency fo1 of the primary side series resonance circuit and the intermediate resonance frequency fo determined by the resonance frequency fo2 of the secondary side series resonance circuit. The higher frequency range is set as the variable range of the switching frequency. That is, a so-called upper side control method is adopted.

  As a general matter, the series resonant circuit has the lowest resonance impedance at the resonance frequency. For this reason, when the upper side control method based on the resonance frequency of the series resonance circuit is adopted as in this embodiment, the resonance impedance is increased as the switching frequency fs increases. become.

  Therefore, for example, when the secondary side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. This lowers the resonance impedance and increases the amount of power transmission from the primary side to the secondary side, so that the secondary side DC output voltage Eo increases.

  On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the switching frequency is controlled to be higher. As a result, the resonance impedance is increased and the power transmission amount is reduced, so that the secondary side DC output voltage Eo is lowered. Thus, the secondary side DC output voltage Eo is stabilized by changing the switching frequency.

  Next, how the secondary side DC output voltage Eo is stabilized in the switching power supply circuit shown in FIG. 2 will be described.

  First, in order to explain step by step, the secondary side series resonance capacitor C2 is omitted from the configuration of the switching power supply circuit of the embodiment shown in FIG. 2, and no secondary side series resonance circuit is formed. A description will be given of how the secondary side DC output voltage Eo is stabilized in the composite resonance type converter as shown in FIG.

  Such a composite resonance type converter includes a primary side series resonance circuit (and a primary side partial voltage resonance circuit), but does not include a secondary side series resonance circuit. Therefore, in stabilizing the secondary side DC output voltage Eo by the switching frequency control method of the upper side control, the switching frequency is variably controlled in a frequency range higher than the resonance frequency fo1 of the primary side series resonance circuit, A change in resonance impedance caused by this is used.

  This will be described with reference to FIG. FIG. 19 shows a constant voltage control characteristic for the secondary side DC output voltage Eo by the composite resonance type converter. In this figure, the horizontal axis represents the switching frequency fs, and the vertical axis represents the secondary side DC output voltage Eo.

  Here, the series resonance circuit has the smallest resonance impedance at the resonance frequency. As a result, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs in the upper side control, the level of the secondary side DC output voltage Eo is equal to the switching frequency fs with respect to the resonance frequency fo1 of the primary side series resonance circuit. It increases as it gets closer, and decreases as it gets away from the resonance frequency fo1.

  Accordingly, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs under the condition that the load power Po is constant, the switching frequency fs is the same as the resonance frequency fo1 of the primary side series resonance circuit as shown in FIG. It sometimes becomes a peak, and shows a quadratic curve change that decreases as it moves away from the resonance frequency fo1.

  The level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs has a characteristic of shifting so as to decrease by a predetermined amount at the maximum load power (Pomax) than at the minimum load power (Pomin). can get. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  Under such characteristics, when the secondary side DC output voltage Eo is stabilized by upper side control so that Eo = tg, the variable range of the switching frequency required in the power supply circuit is obtained. (Necessary control range) is a range indicated as Δfs in the figure.

  For example, as an actual example of this composite resonance type converter, an AC input voltage VAC = 85V to 120V as an AC 100V system, a maximum load power (Pomax) of the secondary side DC output voltage Eo = 150 W, a minimum load power (Pomin) ) = 0. Assuming that it corresponds to the load condition of 0 W (no load), it is assumed that the specification for stabilization at the secondary side DC output voltage Eo = 135 V is set by the switching frequency control method.

  In this case, the variable range of the switching frequency fs that can be varied for constant voltage control by the composite resonance type converter is fs = 80 kHz to 200 kHz or more, and Δfs, which is a necessary control range amount, is 120 kHz or more and a correspondingly wide range. It will be a thing.

  With this in mind, let's consider configuring this composite resonant converter to support a wide range.

  In order to make it compatible with a wide range, for example, it should correspond to an AC input voltage range of AC85V to 288V. Therefore, for example, the level fluctuation range of the secondary side DC output voltage Eo is larger than that corresponding to a single range of only the AC 100 V system or only the AC 200 V system. In order to perform the constant voltage control operation against the level fluctuation of the secondary side DC output voltage Eo expanded according to such an AC input voltage range, a wider switching frequency control range is required. For example, the control range of the switching frequency fs needs to be expanded to about 80 kHz to 500 kHz.

  However, for an IC (oscillation / drive circuit 2) for driving a current switching element, the upper limit of the drive frequency that can be handled is preferably about 200 kHz. Further, even if a switching drive IC capable of driving at a high frequency as described above is configured and mounted, the power conversion efficiency in the switching elements Q1 and Q2 decreases, so that an actual power supply circuit can be obtained. It is no longer practical. In consideration of such points, for example, the upper limit of the AC input voltage VAC level that can be efficiently stabilized by the composite resonance converter is about 100V.

  Therefore, conventionally, as one of the configurations for making the switching power supply circuit that is stabilized by the switching frequency control system compatible with a wide range, as shown in FIG. It is known that That is, control is performed so that the value of the DC input voltage (Ei) generated on the output side of the active filter becomes a substantially constant value.

  However, as described above, since the active filter circuit is a hard switching operation, the noise generation level is very high. For this reason, a relatively severe noise suppression measure is required for both common mode noise and normal mode noise.

  Furthermore, since the switching frequency of the active filter circuit and the composite resonance type converter are different, it is necessary to reduce mutual interference of the ground potential and mutual magnetic interference generated from each component. The placement of each component on the board is limited, and the ground pattern must be designed in consideration of the path of the noise current flowing out from each component, which makes it difficult to route the pattern on the board. there were.

  However, as described above with reference to FIG. 19, when the switching frequency control range has a wide characteristic, the secondary side DC output voltage is set to a required level in response to a load fluctuation such as a switching load. It takes a relatively long time to change the frequency to the switching frequency. That is, an unsatisfactory result is obtained as a response characteristic of constant voltage control.

  For such a wide range, the switching power supply circuit of the present embodiment shown in FIG. 1 or 2 is a full-wave rectifier circuit or a double voltage rectifier circuit depending on the voltage of the commercial AC power supply AC. By selecting this, it is possible to support a wide range basically.

  In addition, the primary side and the secondary side are provided with series resonance circuits (primary side series resonance circuit, secondary side series resonance circuit), respectively. Thereby, as a power supply circuit based on a current resonance type converter, by constant voltage control of switching frequency control, even in a 100V system, for example, in a range of 85V to 150V, in a 200V system, for example, 150V ~ A wide range can be supported in the range of 288V. Hereinafter, this point will be described.

  The circuit diagram of FIG. 4 shows an equivalent circuit when the power supply circuit of the present embodiment shown in FIG. 2 is viewed from the relationship between the primary side series resonant circuit and the secondary side series resonant circuit. In the equivalent circuit diagram, the same parts as those in FIG.

  In FIG. 4, the converter transformer PIT is indicated by a broken line, and the ideal transformer IT is indicated by a one-dot chain line. Here, the ideal transformer IT converts the voltage according to a winding ratio of 1: n (1: number of turns of the secondary winding N2 / number of turns of the primary winding N1). Further, in FIG. 4, reference numerals L1 and L1e denote a leakage (leakage) inductance of the primary winding N1 and an excitation inductance of the primary winding N1, respectively. Symbols L2 and L2e indicate the leakage (leakage) inductance of the secondary winding N2 and the excitation inductance of the secondary winding N2, respectively.

  In the equivalent circuit diagram shown in FIG. 4, an alternating current (frequency signal) at the switching frequency fs is input on the primary side of the converter transformer PIT. That is, the switching output of the primary side switching converter (switching elements Q1, Q2) is an input.

  Then, on the primary side of the converter transformer PIT, an alternating current input with this switching frequency fs is supplied to the primary side series resonance circuit. The value of the resonance frequency fo1 of the primary side series resonance circuit is obtained based on the equivalent circuit shown in FIG. Similarly, the value of the resonance frequency fo2 of the secondary side series resonance circuit is obtained based on the equivalent circuit shown in FIG. In the present embodiment, the value of the resonance frequency fo2 is selected to be 1.6 times or more the value of the resonance frequency fo1.

In the equivalent circuit of FIG. 4, Equation 1 is established for the coupling coefficient k of the converter transformer PIT, the self-inductance of the primary winding N1 L1s, and the leakage inductance L1 of the primary winding N1.

Further, Equation 2 is established for the excitation inductance L1e of the primary winding N1.

Similarly, (Equation 3) and (Equation 4) are established for the leakage inductance L2 and the excitation inductance L2e of the secondary winding N2, respectively. Here, L2s is the self-inductance of the secondary winding N2.

  Here, it is shown that the equivalent circuit shown in FIG. 4 includes a primary side series resonance circuit on the primary side and a secondary side series resonance circuit on the secondary side via electromagnetic induction of the converter transformer PIT. Has been. Therefore, the circuit shown in this figure can be regarded as forming a coupled resonance circuit by electromagnetic coupling. For this reason, the constant voltage control characteristic for the secondary side DC output voltage Eo in the power supply circuit shown in FIG. 2 differs depending on the coupling coefficient k of the converter transformer PIT. This point will be described with reference to FIG.

  FIG. 5 shows the output characteristics with respect to the input (switching frequency signal) for the equivalent circuit of FIG. That is, the control characteristic for the secondary side DC output voltage Eo is shown by the relationship with the switching frequency fs. In this figure, the horizontal axis represents the switching frequency, and the vertical axis represents the level of the secondary side DC output voltage Eo.

  The description of FIG. 5 is a general one that applies regardless of the frequency relationship between the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit. In the figure, reference numerals fo1 and fo2 are shown in parentheses, indicating that the characteristics shown in FIG. 5 are established regardless of the vertical relationship between the resonance frequency fo1 and the resonance frequency fo2. In FIG. 6, the meanings of the symbols fo1 and fo2 shown in parentheses are the same.

Here, it is assumed that a tight coupling state with a coupling coefficient k = 1 is set. In this case, the leakage inductance L1 of the primary winding N1 and the leakage inductance L2 of the secondary winding N2 are obtained by substituting k = 1 into (Equation 1) and (Equation 3), respectively (Equation 5). Can be obtained.

  That is, it is shown that the leakage inductance of the primary winding N1 and the secondary winding N2 does not exist because the converter transformer PIT is tightly coupled.

In this way, as a constant voltage control characteristic in a state where the primary side and the secondary side of the converter transformer PIT are tightly coupled, the resonance of the primary side series resonance circuit as shown by the characteristic curve 1 in FIG. A so-called bimodal characteristic is obtained in which the secondary side DC output voltage Eo peaks at frequencies f1 and f2 different from the frequency fo1 and the resonance frequency fo2 of the secondary side series resonance circuit. Here, the frequency f1 is represented by (Equation 6), and the frequency f2 is represented by (Equation 7).

Further, fo, which is one of the terms in (Expression 6) and (Expression 7), is an intermediate resonance frequency that exists between the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit. The frequency is determined by the impedance on the primary side, the impedance on the secondary side, and the impedance (mutual coupling inductance M) common to the primary side and the secondary side. The mutual coupling inductance M is expressed by (Equation 8).

  Further, assuming that the coupling coefficient k is gradually decreased from the state of k = 1, that is, when the degree of loose coupling is gradually increased from the tightly coupled state, it is shown in FIG. The characteristic curve 1 shows a change in which the bimodal tendency gradually diminishes and becomes flat in the vicinity of the intermediate resonance frequency fo. Then, when the total coupling coefficient Kt is reduced to a certain level, a so-called critical coupling state is obtained. In this critical coupling state, as shown by the characteristic curve 2, there is no tendency as a bimodal characteristic, and the curve shape becomes flat with the intermediate resonance frequency fo as the center.

  Further, if the coupling coefficient k is decreased from the critical coupling state and the loose coupling state is increased, the peak and peak are obtained only at the intermediate frequency fo as shown by the characteristic curve 3 in FIG. A unimodal characteristic is obtained. Further, comparing this characteristic curve 3 with the characteristic curves 1 and 2, the characteristic curve 3 has a peak level itself lower than that of the characteristic curves 1 and 2, but as a quadratic curve shape thereof, It can be seen that it has a steeper slope.

  As described above, the relationship between the coupling coefficient k and various characteristics has been described in consideration of only the transformer coupling coefficient k which is the coupling coefficient of the converter transformer PIT itself. However, when an inductance is added to the outside of the converter transformer PIT, The same explanation can be made by using the total coupling coefficient Kt between the primary side and the secondary side instead of the coupling coefficient k used in (Equation 6) to (Equation 8).

  The total coupling coefficient Kt is an inductance component connected to the primary side, and an inductance that is not magnetically coupled to the secondary side is equivalently considered as a leakage inductance on the primary side. Inductance component connected to the secondary side, which is not magnetically coupled to the primary side, and is equivalent to the secondary side leakage inductance. This indicates the degree of proper coupling. Therefore, when no inductance is added outside the converter transformer PIT, the coupling coefficient k (in this case, the transformer coupling coefficient k of the converter transformer PIT itself) and the total coupling coefficient Kt are equal.

  Further, when the value of the coupling coefficient k is reduced, even if the value of the coupling coefficient k is reduced in the converter transformer PIT itself, an inductance is added to the primary side or the secondary side of the converter transformer PIT so that the coupling coefficient k is reduced. Even when the value is changed, the characteristics shown in FIG. 5 can be applied. Therefore, when the coupling coefficient between the primary side and the secondary side is a problem regardless of the cause, the total coupling coefficient Kt The terminology is used.

  When the unimodal characteristic shown in FIG. 5 and the constant voltage control characteristic of the composite resonance type converter shown in FIG. 19 are actually compared, the characteristic shown in FIG. 19 with respect to FIG. In particular, the slope is considerably gentler.

  Since the characteristic shown in FIG. 19 is gentle as described above, the necessary control range of the switching frequency for performing the constant voltage control on the secondary side DC output voltage Eo is, for example, a condition corresponding to a single range. Even so, since fs = 80 kHz to 200 kHz or more and Δfs = 120 kHz or more, it is very difficult to adapt to a wide range only by constant voltage control by switching frequency control, as described above. It is.

  In addition, when changing the switching frequency over a wide range as described above, both the common mode noise filter for common mode noise and the normal mode filter for normal mode noise have wide bandwidths corresponding to such a wide range of switching frequencies. It must be done, and it will be difficult to realize it. Furthermore, the power switching elements Q1 and Q2 must be operated at a higher frequency, which is not preferable from the viewpoint of increasing the switching loss.

  On the other hand, the constant voltage control characteristic of the present embodiment is a single peak characteristic indicated by the characteristic curve 3 in FIG. 5, and the constant voltage control operation is as shown in FIG. In FIG. 6, each characteristic curve for the maximum load power (Pomax) and the minimum load power (Pomin) at the AC input voltage VAC = 100 V (AC 100 V system) for the power supply circuit of the present embodiment shown in FIG. Four characteristic curves are shown: A and B, and characteristic curves C and D at the maximum load power (Pomax) and at the minimum load power (Pomin) when the AC input voltage VAC = 230 V (AC 200 V system). Yes.

  As can be seen from FIG. 6, first, when the AC input voltage VAC = 100 V corresponding to the AC 100 V system input, the switching required to make the secondary side DC output voltage Eo constant at the required rated level tg. The frequency variable control range (necessary control range) is represented by Δfs1. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve A to the switching frequency fs at level tg in the characteristic curve B.

  In addition, when the AC input voltage VAC = 230 V corresponding to the AC 200 V system input, the variable control range (necessary control) of the switching frequency required to make the secondary side DC output voltage Eo constant at the required rated level tg. (Range) is indicated by Δfs2. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve C to the switching frequency fs at level tg in the characteristic curve D.

  As described above, the unimodal characteristic that is the control characteristic of the secondary side DC output voltage Eo in the present embodiment is considerably steep in a quadratic function curve as compared with the control characteristic shown in FIG. It is. Therefore, Δfs1 and Δfs2, which are the necessary control ranges when the AC input voltage VAC = 100V and VAC = 230V, are considerably reduced as compared with Δfs shown in FIG. .

  In addition, the frequency variable range (ΔfsA) from the lowest switching frequency at Δfs1 (switching frequency fs at level tg in characteristic curve A) to the highest switching frequency at Δfs2 (switching frequency fs at level tg in characteristic curve A) is also used. , It is correspondingly narrow.

  Here, the actual frequency variable range ΔfsA in the power supply circuit of the present embodiment shown in FIG. 1 is well within the variable range of the switching frequency corresponding to the current switching drive IC (oscillation / drive circuit 2). It has become. Furthermore, the switching element Q1 and the switching element Q2 are also in a frequency range in which the loss is reduced and the operation is performed.

  That is, in the power supply circuit shown in FIG. 1, the switching frequency can be actually variably controlled within the frequency variable range ΔfsA. This means that the power supply circuit shown in FIG. 1 can stabilize the secondary side DC output voltage Eo in accordance with any commercial AC power supply input of AC100V system and AC200V system. In other words, the power supply circuit shown in FIG. 1 can support a wide range only by switching frequency control. In addition, by reducing the necessary control range in this way, the high-speed response when stabilizing the secondary side DC output voltage Eo is improved. For example, the load referred to as the above-described switching load Corresponding to the fluctuation, good constant voltage control performance can be obtained.

  Incidentally, a coupled resonance circuit using electromagnetic coupling is already known as a technique for expanding the amplification bandwidth of an amplifier circuit using a transistor, for example, as an intermediate frequency transformer amplifier. However, in such a field, the bimodal characteristic in the tight coupling or the flat characteristic in the critical coupling is used, and the single peak characteristic in the loose coupling is not used. In the present embodiment, in such a coupled resonant circuit technology using electromagnetic coupling, a single-peak characteristic with loose coupling that has not been employed in the field of communication technology is actively used in the field of resonant switching converters. It can be said that. Thereby, as described above, the variable range (necessary control range) of the switching frequency necessary for stabilizing the secondary side DC output voltage Eo is reduced, and the wide range only by constant voltage control in the switching frequency control is performed. This is possible.

  By the way, if the loosely coupled state in the present embodiment is to be obtained by the structure of the converter transformer PIT, for example, the gap G of the inner magnetic leg of the EE type core of the converter transformer PIT is enlarged to convert the converter transformer. The PIT itself is configured as a loosely coupled transformer.

  By adopting such a configuration, the unimodal characteristics described in FIG. 5 can be obtained, so that the necessary control range of the switching frequency is reduced as described in FIG. Switching between equal voltage rectification operation that generates a rectified and smoothed voltage at a level corresponding to the commercial AC power supply level and double voltage rectification operation that generates a rectified and smoothed voltage at a level corresponding to twice the commercial AC power supply level The secondary side DC voltage can be stabilized in response to a wide range of commercial AC power supply inputs in a range that is greater than or equal to the range that can be accommodated.

  In the embodiment shown in FIG. 2, the value of the coupling coefficient k is set to 0.8 in order to obtain a single peak characteristic. The smaller the value of the coupling coefficient k, the narrower the required control range of the switching frequency can be for a wide range of commercial AC power supply input voltages. From this point, the value of the coupling coefficient k is 0.7 or less. More desirable is.

  However, in the case of the converter transformer PIT structure in which such a gap G is enlarged, an increase in eddy current loss in the vicinity of the gap G of the core of the converter transformer PIT becomes dominant. The power conversion efficiency (ηAC → DC) will decrease. From such a viewpoint, the value of the gap G cannot be increased too much.

  FIG. 7 and FIG. 8 show the operation waveforms of the respective parts, and FIG. 9 shows the characteristic diagrams. In obtaining these, constants and members related to the main parts of the circuit shown in FIG. 2 were selected as follows. .

First, the gap length of the converter transformer PIT was set to 2 mm. Further, as described above, 0.8 is obtained as the value of the coupling coefficient k of the converter transformer PIT itself. The induced voltage level per turn (T) in the secondary winding N2 was set to a predetermined value of 5 V / T or more.

  Using the converter transformer PIT set in this way, the operation waveforms of the main part of the switching power supply circuit obtained by the circuit shown in FIG. 2 are shown in FIGS. 7 (a) to 7 (h) and FIGS. FIG. 9 shows the characteristics of the switching power supply circuit shown in FIG.

  Here, FIGS. 7 (a) to 7 (h) and FIGS. 8 (a) to 8 (d) are waveform diagrams showing the switching period of the switching converter. FIG. 7 (a) to FIG. FIG. 8 (a) and FIG. 8 (b) show operation waveforms when the AC input voltage VAC = 100V.

  Specifically, the rectangular wave voltage V1 shown in FIG. 7A is the voltage across the drain and source of the switching element Q2, and the switching current IQ2 shown in FIG. 7B is the load power Po = 300W. Is the drain current of the switching element Q2.

  7C is a voltage across the secondary winding N2 of the converter transformer PIT, and the current I2 shown in FIG. 7D is obtained when the load power Po = 300 W. This shows the current flowing through the secondary winding N2 of the converter transformer PIT.

  8 (a) and 8 (b) show waveforms when the load power Po = 0W, and FIG. 8 (a) shows the voltage across the drain and source of the switching element Q2. A voltage V1 is shown, and FIG. 8B shows a switching current IQ2 which is a drain current of the switching element Q2.

  On the other hand, FIGS. 7 (e) to 7 (h), 8 (c) and 8 (d) show operation waveforms when the AC input voltage VAC = 230V. Specifically, FIG. 7E shows the voltage V1 that is the voltage across the drain and source of the switching element Q2, and FIG. 7F shows the switching current IQ2 that is the drain current of the switching element Q2. These are waveforms when the load power Po = 300 W.

  Further, a rectangular wave voltage V2 shown in FIG. 7E is a voltage across the secondary winding N2 of the converter transformer PIT, and a current I2 shown in FIG. 7F is a secondary winding of the converter transformer PIT. This shows the current flowing in the line N2. These are waveforms when the load power Po = 300 W.

  8 (c) and 8 (d) each show a waveform when the load power Po = 0W, and FIG. 8 (c) shows the voltage across the drain and source of the switching element Q2. The voltage V1 is shown, and FIG. 8D shows the switching current IQ2 which is the drain current of the switching element Q2.

  The rectangular wave voltage V1 shown in FIGS. 7A and 7E is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2. This voltage V1 has a waveform clamped at the level of the rectified and smoothed voltage Ei during an on period in which the switching element Q2 is conductive and turned on, and in an off period in which the switching element Q2 is nonconductive.

  During the ON period of the switching element Q2, a switching current IQ2 having a waveform shown in the figure flows through the switching circuit system including the switching element Q2 and the damper diode DD2. Further, the switching current IQ2 becomes 0 level during the OFF period of the switching element Q2.

  Although not shown, the voltage across the other switching element Q1 and the switching current flowing through the switching circuit (Q1, DD1) are obtained as a waveform obtained by shifting the phases of the voltage V1 and the switching current IQ2 by 180 °. . That is, the switching element Q1 and the switching element Q2 perform the switching operation at the same cycle timing so as to be alternately turned on / off.

  Further, the switching current IQ2 shown in FIG. 7B and FIG. 7F shows the current flowing through the primary side series resonance circuit (L1-C1) in the half cycle of the switching cycle. In the (on period of the switching element Q1), a switching current IQ1 (not shown) flows through the switching circuit system including the switching element Q1 and the damper diode DD1. A component (not shown) in which switching currents flowing through the switching circuits (Q1, DD1) (Q2, DD2) are combined flows through the primary side series resonance circuit (L1-C1).

  In this way, a voltage is induced in the secondary winding N2 of the converter transformer PIT by the switching operation on the primary side, and the secondary side series resonance circuit operates. An alternating voltage shown in FIGS. 7D and 7F is generated in the secondary winding N2 with a period corresponding to the resonance operation of the secondary side series resonance circuit.

  Here, the secondary side series resonance circuit including the secondary winding N2 of the converter transformer PIT (electrically the leakage inductance component (L2) of the secondary winding N2) and the secondary side series resonance capacitor C2 includes a bridge. One input end of the bridge rectifier circuit composed of the connected rectifier diodes Do1 to Do4 (the connection point between the anode of the rectifier diode Do1 and the rectifier diode Do4) and the other input end of the bridge rectifier circuit (rectifier diode) The anode of Do3 and the node of the rectifier diode Do2 are connected. Also, one output end of the bridge rectifier circuit (a connection point between the cathode of the rectifier diode Do1 and the rectifier diode Do3) and the other output end of the bridge rectifier circuit (the anode of the rectifier diode Do2 and the anode of the rectifier diode Do4). The smoothing capacitor Co is connected to the connection point), the negative terminal of the smoothing capacitor Co is set to the secondary side ground potential, and the secondary side DC output voltage is obtained from both ends of the positive and negative terminals of the smoothing capacitor Co. Has been made.

  In a secondary rectifier circuit that operates by inputting an alternating voltage (output of the secondary side series resonant circuit) from the secondary side series resonant circuit, the rectifier diodes [Do1, Do2 in a half cycle in which the alternating voltage is positive. And the smoothing capacitor Co is charged with the rectified current, and the rectifier diodes [Do3, Do4] are turned on and the smoothing capacitor Co is charged with the rectified current in the half cycle of negative polarity. .

  By performing such an operation, as shown in FIGS. 7D and 7F, the secondary side rectified current I2 flowing through the secondary side rectified current path has a period of flowing due to the positive polarity. Due to the negative polarity, the waveform flows alternately and periodically. 7 (d) and 7 (f) show the maximum load power (Pomax), that is, at 300 W, and this secondary side rectified current I2 is the minimum load power (Pomin), That is, it becomes 0 level at 0W.

  Here, comparing FIG. 7 (a) and FIG. 7 (f), first, the waveform at the time of AC input voltage VAC = 230V and maximum load power (Pomax) = 300W shown in FIG. 7 (f). In the figure, the conduction period of the secondary side rectified current I2 falls within the conduction period of the primary side switching current IQ2. This state can be regarded as a state in which the phase of the primary side series resonant current flowing through the primary side series resonant circuit as the switching output of the switching elements Q1 and Q2 and the secondary side rectified current I2 coincide.

  On the other hand, when the AC input voltage VAC = 100V and the maximum load power (Pomax) = 300 W shown in FIG. 7A, the conduction period of the secondary side rectified current I2 is within the conduction period of the switching current IQ2. It is in a state where it is flowing periodically and not. That is, under the conditions of the AC input voltage VAC = 100 V and the maximum load power (Pomax) = 300 W, the secondary side rectified current I2 has a constant phase difference with respect to the primary side series resonance current. It can be said that.

  As described above, such a phase difference has a relationship of fo2≈fo1 × 1.65 with respect to the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit. It is caused by setting. In other words, the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit are the primary side series resonance under the condition of the AC input voltage VAC below a certain level and the heavy load above the certain level. It is set so that a constant phase difference is obtained between the resonance current and the secondary side rectified current I2.

  In this way, according to the present embodiment, the switching current IQ2 and the secondary side rectified current are set under the conditions of the AC input voltage VAC = 100 V and the load power Pomax = 300 W by setting the resonance frequencies fo1 and fo2. A certain phase difference is generated with respect to I2. As a result, an experimental result was obtained that the peak level of the switching current IQ2 was effectively suppressed under the conditions of the maximum input power (Pomax) (300 W) when the AC input voltage VAC = 230 V (AC 200 V system).

  The suppression of the peak level of the switching current IQ2 when the AC input voltage VAC = 230V is achieved in this way is that the peak level of the switching current IQ2 at the maximum load power (Pomax) in FIG. And the minimum level of the load current (Pomin) = 0W, the peak level and minimum level of the switching current IQ2 at the maximum load power (Pomax) in FIG. 7 (b) with respect to the ratio (Lv1 / Lv2) of the peak level of the switching current IQ2 This is also indicated by the fact that the ratio (Lv3 / Lv4) of the peak level of the switching current IQ2 when the load power (Pomin) = 0 W is smaller.

  By suppressing the peak level of the switching current IQ2 in this way, the peak level of the primary side series resonance current (Io) flowing through the primary side series resonance circuit can be suppressed. By suppressing the peak level of the primary side series resonance current, the current level flowing through each switching element Q1, Q2 is suppressed, so that the switching loss in the switching elements Q1, Q2 is also reduced. It will be.

  That is, in the power supply circuit of FIG. 2, the power loss is reduced compared with the conventional power conversion efficiency characteristics, particularly under the condition that the AC input voltage level is high and the load tends to be heavy. It can be said that the improvement is achieved.

  When the resonance frequencies fo1 and fo2 are set as described above, as shown in FIG. 7, the switching current IQ2 (primary side series) when the AC input voltage VAC = 100 V / maximum load power (Pomax) = 300 W is set. The waveform of (resonance current) is substantially sinusoidal. Further, the peak portion of the waveform of the switching current IQ2 (primary side series resonance current) when VAC = 230 V and the maximum load power (Pomax) = 300 W is substantially M-shaped.

Further, in the present embodiment, an example is given in which the resonance frequencies fo1 and fo2 are set so as to obtain a relationship of fo2≈fo1 × 1.65. However, the resonance frequency to be set to obtain the above-described effect The relationship between the values of fo1 and the resonance frequency fo2 is not limited to this, and may actually be appropriately changed depending on, for example, the corresponding load condition.

  That is, for example, in order to obtain a state in which the peak level of the primary side series resonance current is suppressed until it is actually required, an AC input voltage range of a certain level or less and a heavy load of a certain level or more are set. It is only necessary to set the values of the resonance frequency fo1 and the resonance frequency fo2 so that a constant phase difference is obtained for the primary side series resonance current and the secondary side rectified current under the conditions.

  In the above-described embodiment shown in FIG. 1 or FIG. 2, the source of the switching element Q1 is connected to one end of the primary winding N1 of the converter transformer PIT via the series connection of the primary series resonant capacitor C1. Is connected to a connection point (switching output point) between the drain of the switching element Q2 and a switching output is transmitted. However, in order to obtain the same effect, one end of the primary winding N1 of the converter transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, and the converter The other end of the primary winding N1 of the transformer PIT may be connected to one side end of the primary side series resonant capacitor C1, and the other side end of the primary side series resonant capacitor C1 may be connected to the primary side ground. .

  Furthermore, although not shown, the switching means may be provided with four switching elements and may be a full bridge configuration formed by combining two so-called half bridges. In this case, the primary side series resonance In the series connection of the capacitor C1 and the primary winding N1, the other end of the primary side series resonant capacitor C1 on the side that is not the connection point between one end of the primary side series resonant capacitor C1 and one end of the primary winding N1. The part may be connected to the switching output point of one set of half bridges, and the other end of the primary winding N1 may be connected to the switching output point of the other set of half bridges. .

  As for the secondary side DC output voltage generating means, not only the full-wave rectifier circuit for rectifying the power from one winding shown in FIG. 1 or FIG. Any of a wave rectifier circuit and a quadruple voltage rectifier circuit (both not shown) may be used.

  However, by using the full-wave rectifier circuit shown in FIG. 1 or FIG. 2 in which the rectification current flows in the entire secondary winding in each positive and negative period of the induced voltage of the secondary winding, in particular, the following The problem is also solved.

  First, in the case of a double-wave rectifier circuit, as shown also as the secondary side of the current resonance type converter of FIG. 18, the secondary winding N2 is provided with a center tap, and the two secondary winding portions N2A and N2B are provided. Will be formed. In these two secondary winding portions, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding portion N2A → rectifier diode Do1 → smoothing capacitor Co → It flows through the path of the secondary winding portion N2A]. Further, in the other half cycle of the above alternating voltage, the rectified current flows through [secondary winding portion N2B → rectifier diode Do2 → smoothing capacitor Co → secondary winding portion N2B]. That is, in the two-wave rectification, the two secondary winding portions are in a state in which current flows only in one half cycle and does not flow in the other half cycle.

  According to such a double-wave rectification operation, a required capacitance exists between the secondary winding portion N2A and the secondary winding portion N2B wound around the bobbin of the converter transformer PIT. It will be. Since the inter-line capacitance is present in this way, the capacitor Cv is equivalently connected in parallel with the secondary winding N2 on the secondary side of the converter transformer PIT in this case. It becomes a state.

  By connecting the capacitor Cv in parallel to the secondary winding N2, in this case, a parallel resonance circuit is formed on the secondary side by the leakage inductance of the secondary winding N2 and the capacitance of the capacitor Cv. The equivalent state is obtained.

  Incidentally, the capacitance of the capacitor Cv is determined by the number of bundles of litz wires used as the secondary winding N2 and the window area of the bobbin around which the secondary winding N2 is wound. In the design specification of the current resonance type converter shown, it is a very small amount of about 100 pF to 500 pF.

  As a result of the parallel resonance circuit being formed on the secondary side as described above, the actual constant voltage control characteristics are changed from those shown in FIG. 19 to those shown in FIG. In FIG. 20, first, as described above, a parallel resonant circuit is formed also on the secondary side, so that when the resonance frequency of the primary side series resonant circuit is fo1, the secondary side parallel resonant circuit of FIG. A resonance frequency fov2 exists.

  Since two different resonance points exist in this way, the characteristic curve particularly at Pomin (light load) shows that the resonance between the peak and the secondary side depends on the resonance frequency fo1 on the primary side. A bimodal curve as shown in the figure having two peaks corresponding to the frequency fov2 is obtained.

  In this case, as the capacitance of the capacitor Cv is relatively small as described above, when the level of the secondary side DC output voltage Eo tends to be relatively low under heavy load conditions, The secondary resonance point does not become apparent (characteristic curve at Pomax). However, if the secondary-side DC output voltage Eo tends to rise sharply by approaching a no-load state due to a light load trend, does the secondary-side resonance point become obvious? Thus, a bimodal characteristic curve such as a characteristic curve at Po = 0 in the figure is obtained.

  When comparing the characteristic curve of this bimodal with the characteristic curve at the same Po = 0W in FIG. 19, the bimodal curve shown in FIG. 20 has no load compared to the case where it is a unimodal curve. It can be seen that the switching frequency at the time tends to be higher. According to this, as can be seen by comparing Δfs in each figure, the required control range Δfs of the switching frequency becomes wider in the case of the bimodal in FIG.

  In this way, when a double-wave rectifier circuit is provided as a secondary side rectifier circuit, the required control range is caused by the presence of a resonance point due to the resonant circuit of the parallel resonant circuit formed equivalently on the secondary side. Δfs expands. The above-described capacitor Cv is not formed in the power supply circuit of the present embodiment that adopts the secondary-side configuration described so far. Therefore, there is an expansion factor of the necessary control range Δfs due to the formation of the secondary-side parallel resonance. It has been excluded.

  FIG. 9 shows characteristics obtained by the switching power supply shown in FIG. The solid line shows the characteristics at the AC input voltage VAC = 100V, and the broken line shows the characteristics at the AC input voltage VAC = 230V.

  In FIG. 9, the vertical axis indicates the value of AC → DC power conversion efficiency (ηAC → DC) and the value of the switching frequency fs that is the frequency of the drive signal, and the horizontal axis indicates the value of the load power Po. . As shown in FIG. 9, the AC → DC power conversion efficiency (ηAC → DC) at the maximum load power (Pomax) = 300 W is 92.8% when the AC input voltage VAC = 100 V, and the AC input voltage VAC = 230 V, 95, 3% was obtained.

  At this time, the value of the switching frequency fs was 109 kHz when the minimum load power (Pomain) = 0 W and 80 kHz when the maximum load power (Pomax) = 300 W when the AC input voltage VAC = 100 V. Further, the value of the switching frequency fs was 112 kHz when the minimum load power (Pomain) = 0 W at the AC input voltage VAC = 230 V, and 96.5 kHz when the maximum load power (Pomax) = 300 W.

  Next, a switching power supply circuit further including a power factor correction circuit 10 will be described with reference to FIG. In addition, a power supply circuit provided with a power factor improvement means like the power factor improvement circuit 10 is named generically as a power regeneration system power factor improvement power source.

  The power factor correction circuit 10 includes a filter capacitor CN, a high frequency inductor L10, a high speed diode D1, and a high speed diode D2.

  One side end of the primary winding N1 of the converter transformer PIT is connected to the anode of the high speed diode D1, the cathode of the high speed diode D2, and the one side end of the high frequency inductor L10. The anode of the high speed diode D2 is connected to the primary side ground. The cathode of the high-speed diode D1 is connected to the drain of the switching element Q1, the cathodes of the rectifier diode Da and the rectifier diode Db, the one end of the filter capacitor CN, and the positive polarity end that is the side end of the smoothing capacitor Ci1. . The other end of the high frequency inductor L10 is connected to the other end of the filter capacitor CN, the anode of the rectifier diode Da, and the cathode of the rectifier diode Db.

  According to the above connection mode, the power factor correction circuit 10 supplies a current corresponding to the magnitude of the current flowing in the primary side series resonant circuit to the rectifying and smoothing means, which is a part of the current flowing in the primary side series resonant circuit. It is designed to be able to return and improve the power factor. At this time, the current fed back to the supply to the rectifying / smoothing means passes through the high speed diode D1.

  In addition, the constant and member which concern on the principal part of the switching power supply circuit shown in FIG. 1 were selected as follows.

  First, an EER-40 type ferrite core was selected for the converter transformer PIT, and the gap length of the gap G was 2 mm. The number of turns of each winding (number of turns: T) was set to primary winding N1 = 38T and secondary winding N2 = 26T. With this structure, 0.7 is obtained as the value of the coupling coefficient k of the converter transformer PIT itself. The induced voltage level per turn (T) in the secondary winding N2 was set to a predetermined value of 5 V / T or more.

  Further, the inductance value of the high frequency inductor L10 is 47 μH (microhenry), and the total coupling coefficient Kt is 0.64. Further, 0.016 μF (microfarad) was used as the value of the primary side series resonance capacitor C1, and 0.018 μF (microfarad) was used as the value of the secondary side series resonance capacitor C2.

  At this time, for each of the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit, 65 kHz is obtained as the value of the resonance frequency fo1, and 107.5 kHz is obtained as the value of the resonance frequency fo2, and fo2≈fo1. The resonance frequency of each resonance circuit was set so that the relationship of × 1.65 was obtained.

  Further, the load condition was set to Eo = 175 V for the secondary side DC output voltage Eo, and the value of the maximum load power (Pomax) that is the maximum value of the load power Po was 300 W. Further, the input condition of the commercial AC power supply was a wide range input of AC input voltage VAC = 85V to 288V.

  The waveform diagram of FIG. 10 shows the operation waveform of the power factor correction circuit 10 by the commercial AC power supply cycle.

  10 (a) to 10 (d) show operation waveforms of each part when the value of the AC input voltage VAC is 100V, and FIGS. 10 (e) to 10 (h) show the value of the AC input voltage VAC of 230V. The operation waveform at the time is shown. Specifically, FIGS. 10A and 10E show the AC input voltage VAC, FIGS. 10B and 10F show the AC input current IAC, and FIGS. 10 (g) shows the voltage V1 at the winding end of the converter transformer PIT, and FIGS. 10 (d) and 10 (h) show the waveform of the current I1 flowing through the high-frequency inductor L10. In addition, the vertical line part of each waveform in FIG.10 (c), FIG.10 (d), FIG.10 (g), and FIG.10 (h) represents switching with the switching frequency fs.

  As shown in FIGS. 10B and 10F, the AC input current IAC obtained as a result of the power factor correction operation of the power factor correction circuit 10 at each of VAC = 100 V / 230 V is shown. This AC input current IAC has a longer conduction period (conduction angle) than, for example, when the power factor correction circuit 10 is not provided. That is, in response to the input of the AC input voltage VAC and the operation of the switching converter, the power factor correction circuit 10 performs the power factor correction operation by performing power regeneration from the primary side series resonance circuit. .

  FIG. 11A shows the ripple voltage ΔEo included in the secondary side DC output voltage Eo when the AC input voltage VAC = 100V. The peak-to-peak voltage of the ripple voltage ΔEo is 100 mV (millivolts) under the load condition of the maximum load power Pomax = 300 W. Note that the value of the secondary side DC output voltage Eo is 175V.

  FIG. 11B shows the ripple voltage ΔEo of the secondary side DC output voltage Eo when the AC input voltage VAC = 230V. The peak-to-peak voltage of the ripple voltage ΔEo of the secondary side DC output voltage Eo under the load condition of the maximum load power Pomax = 300 W is 60 mV (millivolt). The value of the secondary side DC output voltage Eo is 175 V as in the case of the AC input voltage VAC = 100 V.

  FIG. 12 shows the characteristics of the switching power supply circuit shown in FIG. In FIG. 12, the vertical axis represents power factor PF, AC → DC power conversion efficiency (ηAC → DC), and rectified smoothing voltage Ei, and the horizontal axis represents load power Po. Here, the solid line indicates each value when the value of the AC input voltage VAC is 100V, and the broken line indicates each value when the value of the AC input voltage VAC is 230V.

  First, the power factor PF obtained in accordance with the operation of the power factor correction circuit 10 is close to 1 when the AC input voltage VAC = 100 V, up to a power value of about 110 W as the load becomes heavy. Thereafter, the power value decreases to about 260 W, and then increases to 300 W, and the power factor PF is 0.85 when the maximum load power Pomax = 300 W.

  Further, when the value of the load power Po is 10 W to the maximum load power Pomax = 300 W, the value of the power factor PF is 0, 75 or more, and the power supply harmonic distortion regulation value is cleared. Here, the inflection point of the power factor PF described above occurs according to the change of the load power Po because the phase of the primary side series resonance current changes due to the change of the secondary side series resonance current.

  In addition, the power factor PF at the AC input voltage VAC = 230 V tends to be close to 1 until it reaches a power value of about 200 W as the load becomes heavy, and then decreases to a power value of 300 W. However, 0.79 is obtained as the value of the power factor PF at the maximum load power Pomax = 300 W.

  According to such a value of the power factor PF, it can be said that the power harmonic distortion regulation can be cleared with respect to a wide range of load power Po, and a practically sufficient power factor can be obtained.

  As for AC → DC power conversion efficiency (ηAC → DC), when AC input voltage VAC = 100 V, AC → DC power conversion efficiency of 93% is obtained at maximum load power Pomax = 300 W, and AC input voltage VAC = At 230 V, the AC → DC power conversion efficiency of 91.5% is obtained at the maximum load power Pomax = 300 W. At this time, the value of the ripple voltage ΔEi of the rectified and smoothed voltage Ei was 85V, and the value of the ripple voltage ΔEo of the secondary side DC output voltage Eo was 100 mV.

  Further, when the AC input voltage VAC = 230 V, the load power Po becomes higher as the load becomes heavier, and under the load condition of maximum load power (Pomax) = 300 W, ηAC → DC = 91%. Yes. At this time, the value of the ripple voltage ΔEi of the rectified and smoothed voltage Ei was 50V, and the value of the ripple voltage ΔEo of the secondary side DC output voltage Eo was 60 mV.

  That is, when the power supply circuit of the embodiment described so far is compared with the power supply circuit shown in FIG. 18 which is the prior art for achieving the same power factor improvement and wide range compatibility, I can say that.

  First, as can be seen from the characteristic diagram from the experimental results shown in FIG. 12, the power conversion efficiency in the AC100V system and AC200V system is higher in the power supply circuit shown in FIG. 1 than in the power supply circuit in FIG. It will be improved. This is configured to include a rectifying and smoothing means that is switched between equal voltage rectification operation and voltage double rectification operation according to the level of the input commercial AC power supply, and a power factor improvement circuit using a voltage feedback method. This is because the active filter is unnecessary. That is, in the present embodiment, the total efficiency is not lowered by the two power conversion efficiency values of the former stage and the latter stage as in the case where the active filter is provided.

  Further, in the circuit shown in FIG. 1, the active filter is not required, so that the number of circuit components can be reduced. In other words, the active filter constitutes a set of converters. As can be seen from the description of FIG. 19, in practice, there are many switching elements and ICs for driving them. It consists of the number of parts.

  On the other hand, in the power supply circuit shown in FIG. 1, at least a filter capacitor CN, a high-speed diode D1, a high-speed diode D2, and two smoothing capacitors connected in series as additional components necessary for power factor improvement and wide range compatibility. It is only necessary to provide Ci1, smoothing capacitor Ci2, and relay switch S, and the number of parts can be made very small as compared with the active filter. If a short plug is used instead of the relay switch S, the price can be further reduced.

  Further, in addition to the relay switch S described above, if the relay RL connected to the rectifier circuit switching module 5 and the rectifier circuit switching module 5 is provided, automatic switching according to the value of the AC input voltage VAC can be automatically performed. Become.

  As a result, the power supply circuit shown in FIG. 1 can be manufactured at a much lower cost than the circuit shown in FIG. 19 as a wide-range power supply circuit having a power factor correction function. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight.

  In the power supply circuit shown in FIG. 1, since the operation of the resonant converter and the power factor correction circuit 10 is a so-called soft switching operation, the level of switching noise is greatly reduced as compared with the active filter shown in FIG. The

  For this reason, as shown in FIG. 1, if a one-stage noise filter comprising a pair of common mode choke coils CMC and an across capacitor CL is provided, it is possible to sufficiently satisfy the power disturbance standard. The Further, as shown in FIG. 1, the normal mode noise of the rectified output line is taken by only one filter capacitor CN. By reducing the number of components as a noise filter in this way, the cost reduction of the power supply circuit and the reduction in size and weight of the circuit board are promoted.

  In the case of the power supply circuit shown in FIG. 1, the switching elements Q1 and Q2 forming the primary side switching converter perform a switching operation synchronously. Accordingly, the primary-side ground potential can be prevented from interfering between the active filter side and the subsequent switching converter as in the power supply circuit of FIG. Thereby, for example, the problem of abnormal oscillation that becomes a problem in the power supply circuit of FIG. 19 is also solved.

  As described above, the power supply circuit according to the present embodiment shown in FIG. 1 obtains a wide-range power supply circuit having a power factor improvement function after solving various problems of the power supply circuit including the active filter. Is.

  Subsequently, as a second embodiment of the switching power supply circuit according to the present invention, FIG. 13 shows a variation of another power factor improving means as an embodiment. The switching power supply circuit shown in FIG. 13 includes a power factor correction circuit 30 as an example of the power factor improvement means. A power supply circuit including a power factor improving unit such as the power factor improving circuit 30 is collectively referred to as a voltage feedback power factor improving power source.

  In FIG. 13, the same parts as those in FIG. 1 or FIG.

  Further, in the switching power supply circuit shown in FIG. 13, the operation waveform of the AC input voltage VAC, the operation waveform of the AC input current IAC, the operation waveform of the voltage V1, the operation waveform of the current I1, and the operation waveform of the secondary side DC output voltage Eo. Are substantially the same as the waveforms shown in FIGS. 10 (a) to 10 (h) and FIGS. 11 (a) to 11 (d). In which part each of the AC input voltage VAC, the AC input current IAC, the voltage V1, the current I1, and the secondary side DC output voltage Eo indicates a voltage or a current is indicated by a symbol in FIG.

  The power factor correction circuit 30 is formed by a filter capacitor CN, a high speed diode D1, a high speed diode D2, a power factor improvement transformer VFT, and a capacitor C3.

  One end of the filter capacitor CN is connected to the cathode of the high speed diode D1, the drain of the switching element Q1, the positive polarity terminal of the smoothing capacitor Ci1, the cathode of the rectifier diode Da, and the cathode of the rectifier diode Db. The other end of the filter capacitor CN is connected to one end of the capacitor C3 connected in parallel and the secondary winding N12 of the power factor improving transformer VFT.

  Further, the anode of the high speed diode D1 and the cathode of the high speed diode D2 are connected to one end of the parallel connected capacitor C3 and the secondary winding N12 of the power factor improving transformer VFT. The anode of the high speed diode D2 is connected to the primary side ground.

  Further, the primary winding N11 of the power factor improving transformer VFT is further connected in series to the series circuit of the secondary winding N2 of the converter transformer PIT and the secondary side series resonance capacitor C2, and the secondary side series resonance current is obtained. Flows in the primary winding N11 of the power factor improving transformer VFT.

  Since the power factor correction circuit 30 is connected as described above, a current corresponding to the secondary side series resonance current is also supplied to the secondary winding N12 of the power factor improvement transformer VFT. Here, the path of the current flowing through the high speed diode D1 and the path of the current flowing through the high speed diode D2 are different. Then, only the current flowing through the high-speed diode D1 passes through the smoothing rectifier, and the rectified current is also passed during a period in which the level of the AC input voltage VAC is lower than the voltage across the smoothing capacitor Ci. That is, the conduction angle of the rectified current is increased.

  As the conduction angle of the rectified current is increased as described above, the conduction angle of the AC input current IAC is also increased, whereby the average waveform of the AC input current IAC is changed to the AC input voltage VAC. The power factor is improved by approaching the waveform.

  The main part of the switching power supply circuit shown in FIG. 13 is configured as follows.

  First, an EER-40 type ferrite core was selected for the converter transformer PIT, and the gap length of the gap G was 2 mm. The number of turns of each winding (number of turns: T) was set to primary winding N1 = 40T and secondary winding N2 = 26T. With this structure, 0.7 is obtained as the value of the coupling coefficient k of the converter transformer PIT itself. The induced voltage level per turn (T) in the secondary winding N2 was set to a predetermined value of 5 V / T or more.

  For the inductance of the power factor improving transformer VFT, an EER-40 type ferrite core was selected, and the gap length of the gap G was set to 1 mm. At this time, 33 μH (microhenry) was obtained as the primary inductance value, and 105 μH (microhenry) was obtained as the secondary inductance value.

  The value of the primary side series resonance capacitor C1 is 0.022 μF (microfarad), the value of the secondary side series resonance capacitor C2 is 0.018 μF (microfarad), and the value of the secondary side series resonance capacitor C2 is 0.018 μF (microfarad), 1 μF (microfarad) was used as the value of the filter capacitor CN, and 1000 μF (microfarad) was used as the values of the smoothing capacitor Ci1 and smoothing capacitor Ci2.

  At this time, with respect to the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit, the resonance frequency of each resonance circuit was set so as to obtain a relationship of fo2≈fo1 × 1.65.

  Further, the load condition was set to Eo = 175 V for the secondary side DC output voltage Eo, and the value of the maximum load power (Pomax) that is the maximum value of the load power Po was 300 W.

  FIG. 14 shows the characteristics of the switching power supply circuit shown in FIG. In FIG. 14, the vertical axis represents power factor PF, AC → DC power conversion efficiency (ηAC → DC), and rectified smoothing voltage Ei, and the horizontal axis represents load power Po. Here, the solid line indicates each value when the value of the AC input voltage VAC is 100V, and the broken line indicates each value when the value of the AC input voltage VAC is 230V.

  When the AC input voltage VAC is 100 V, the power factor is 0.75 or more in the range of the load power Po from 50 W to 300 W, and when the AC input voltage VAC is 230 V, the value of the load power Po is from 75 W to In the range of 300 W, the power factor is 0.75 or more, and the power supply harmonic distortion regulation value is cleared. Further, by adding the capacitor C3 in the power factor correction circuit 30, the power factor PF is greatly improved with respect to fluctuations in the load power Po, and the rectified and smoothed voltage Ei hardly changes. Yes.

  The switching power supply circuit shown in FIG. 13 can be described as follows when compared with the switching power supply circuit of the background art shown in FIG.

  As can be seen from the experimental results shown in FIG. 14, in the switching power supply circuit shown in FIG. 13, the power conversion efficiency in the AC100V system and AC200V system is improved as compared with the switching power supply circuit in FIG. This is configured to include a rectifying and smoothing means that is switched between equal voltage rectification operation and voltage double rectification operation according to the level of the input commercial AC power supply, and a power factor improvement circuit using a voltage feedback method. This is because the active filter is unnecessary.

  In the circuit shown in FIG. 13, the number of circuit components can be reduced by eliminating the need for an active filter. Further, in the case of the power supply circuit shown in FIG. 1, the switching elements Q1 and Q2 forming the primary side switching converter perform switching operations synchronously. Accordingly, the primary-side ground potential can be prevented from interfering between the active filter side and the subsequent switching converter as in the power supply circuit of FIG. Thereby, for example, the problem of abnormal oscillation that becomes a problem in the power supply circuit of FIG. 19 is also solved.

  As described above, the power supply circuit according to the present embodiment shown in FIG. 13 obtains a wide-range power supply circuit having a power factor improvement function after solving various problems of the power supply circuit including the active filter. Is.

  The present invention should not be limited to the configuration described so far. For example, an element other than a MOS-FET such as an IGBT (Insulated Gate Bipolar Transistor) may be employed as the switching element. Also, the constants of the component elements described above may be changed according to actual conditions. In addition, the wide range compatible and power factor improving configuration according to the present invention can be applied to a self-excited current resonance type converter.

It is a circuit diagram which shows the structure of the switching power supply circuit as embodiment. It is a circuit diagram explaining operation | movement of the converter part of the switching power supply circuit of embodiment. It is sectional drawing which shows the structural example of the converter transformer of embodiment. It is the equivalent circuit diagram which looked at the switching power supply circuit of embodiment as an electromagnetic coupling type resonance circuit. It is a figure which shows the constant voltage control characteristic about the switching power supply circuit of embodiment. It is a figure which shows the switching frequency control range (required control range) according to alternating current input voltage conditions and load fluctuation | variation as constant voltage control operation | movement of the switching power supply circuit of embodiment. It is a wave form diagram which shows operation | movement of the principal part of embodiment by a switching period. It is a wave form diagram which shows operation | movement of the principal part of embodiment by a switching period. It is a figure which shows each characteristic of the AC-> DC power conversion efficiency with respect to load fluctuation | variation, and the frequency of a drive signal about the power supply circuit of embodiment. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the switching power supply circuit of embodiment. It is a wave form diagram of the ripple voltage of the secondary side DC output voltage of an embodiment. It is a figure which shows each characteristic of the power factor with respect to load fluctuation | variation, AC-> DC power conversion efficiency, and a rectification smoothing voltage about the power supply circuit of embodiment. It is a circuit diagram which shows the structure of the switching power supply circuit as embodiment. It is a figure which shows each characteristic of the power factor with respect to load fluctuation | variation, AC-> DC power conversion efficiency, and a rectification smoothing voltage about the power supply circuit of embodiment. It is a circuit diagram which shows the basic circuit structure of an active filter. It is a wave form diagram which shows the operation | movement in the active filter shown in FIG. It is a circuit diagram which shows the structure of the control circuit system of an active filter. It is a circuit diagram which shows the structural example of the power supply circuit as background art. It is the figure which showed the constant voltage control characteristic in the case of setting the coupling coefficient of a primary side and a secondary side as the conventional setting. It is the figure which showed the constant voltage control characteristic at the time of making a secondary side rectifier circuit into a double wave rectifier circuit.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 10, 30 power factor correction circuit, Di bridge rectifier circuit, Da, Db, Dc, Dd rectifier diode, Ci1, Ci2 smoothing capacitor Q1, Q2 switching element, PIT converter transformer, C1 primary Side series resonant capacitor, Cp primary side partial resonant capacitor, C2 secondary side series resonant capacitor, N1 (converter transformer) primary winding, N2 (converter transformer) secondary winding, Do1 to Do4 (secondary side) rectifier diode, Co (secondary side) smoothing capacitor, CN filter capacitor, D1, D2 high speed diode, L10 high frequency inductor VFT power factor improving transformer, N11 (power factor improving transformer) primary winding, N12 (power factor improving transformer) 2 Next winding

Claims (4)

A commercial AC power supply is input to generate a rectified and smoothed voltage. According to the level of the commercial AC power that is input, the same magnification that generates the rectified and smoothed voltage having a level corresponding to the same level as the commercial AC power supply level. Rectifying and smoothing means that is switched between a voltage rectifying operation and a voltage doubler rectifying operation that generates the rectified and smoothed voltage at a level corresponding to a predetermined multiple of the commercial AC power supply level;
Switching means formed with a switching element for switching the rectified voltage supplied from the rectifying and smoothing means;
Switching drive means for generating a drive signal for switching the switching element;
A converter transformer formed by winding at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the primary winding; ,
A primary side series resonance circuit formed by at least a leakage inductance component of the primary winding of the converter transformer and a capacitance of a primary side series resonance capacitor connected in series to the primary winding;
A secondary side series resonant circuit formed by at least a leakage inductance component of the secondary winding of the converter transformer and a capacitance of a secondary side series resonant capacitor connected in series to the secondary winding;
A secondary side DC output voltage generating means for inputting a resonance output obtained by the secondary side series resonance circuit and performing a rectifying operation to generate a secondary side DC output voltage;
Voltage control means for changing the frequency of the drive signal according to the level of the secondary side DC output voltage and setting the secondary side DC output voltage to a predetermined voltage value;
A power factor improving circuit for improving a power factor by supplying a current corresponding to a current flowing through the primary side series resonant circuit or the secondary side series resonant circuit to the rectifying and smoothing means,
With respect to the electromagnetic coupling type resonance circuit formed by including the primary side series resonance circuit and the secondary side series resonance circuit, the output characteristic of the secondary side DC output voltage with respect to the frequency of the drive signal is unimodal. A switching power supply circuit characterized by comprising: The power factor improving means is
Provided with an inductor and a diode element through which at least a part of the current flowing through the primary side series resonant circuit passes, and supplies a current in a direction corresponding to the polarity of the diode element flowing through the primary side series resonant circuit to the rectifying and smoothing means The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed as described above. The power factor improving means is
A voltage feedback transformer having at least an electromagnetically coupled primary winding and a secondary winding, and a first diode element and a second diode forming two different current paths including the secondary winding of the voltage feedback transformer A diode element,
A current flowing through the secondary side series resonance circuit is supplied to the primary winding of the voltage feedback transformer, and a current flowing through the first diode element among the current flowing through the secondary winding of the voltage feedback transformer is 2. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed so as to be supplied to a rectifying and smoothing means. The rectifying and smoothing means is
A first smoothing capacitor and a second smoothing capacitor connected in series; a first rectifier diode element to a fourth rectifier diode element;
Connecting one polar end of the first rectifier diode and the second rectifier diode element to a side end on the first smoothing capacitor side connected in series;
The other polar ends of the third rectifier diode and the fourth rectifier diode element are connected to the side ends on the second smoothing capacitor side connected in series,
A connection point between the other polar end of the first rectifier diode element and the one polar end of the third rectifier diode element, the other polar end of the third rectifier diode element, and the fourth rectifier diode The power from the commercial AC power source is input between the connection point with the one polar end of the element,
In the voltage doubler rectification operation, a connection point between the other polar end of the third rectifier diode element and one polar end of the fourth rectifier diode element, the first smoothing capacitor, and the second The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed to connect a connection point of a smoothing capacitor.

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