JP2006271027A - Switching power circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain high efficiency by realizing coping with a wide range of a voltage resonance type converter. <P>SOLUTION: A voltage resonance type converter is provided with a secondary parallel circuit, whereby it is put in a loose coupling of approximately 0.6 in the total coupling coefficient k of an insulated converter transformer PIT. Hereby, a constant-voltage control characteristic is made a sharp unimodal characteristic, thereby reducing the control range of the switching frequency required for stabilization. Moreover, it is so arranged that favorable characteristics can be obtained as to power conversion efficiency, and primary parallel resonance frequency fo1 and secondary parallel resonance frequency fo2 are set. The peak level of switching voltage is suppressed by the drop in the total coupling coefficient kt. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit including a voltage resonance type converter.

As a so-called soft switching power supply of a resonance type, a current resonance type and a voltage resonance type are widely known. In the present situation, a current resonance type converter using a two-bridge switching element and a half-bridge coupling system is widely used on the background of practical application.
However, with the background of, for example, the improvement of the characteristics of high voltage switching elements, for example, the problem of withstand voltage in the practical use of voltage resonant converters has been cleared. In addition, a single-ended voltage resonant converter with a single switching element is advantageous in terms of input feedback noise and DC output voltage line noise components compared to a single-current resonant forward converter. It is also known that.

FIG. 13 shows a configuration example of a switching power supply circuit including a voltage resonance type converter by a single end system.
In the switching power supply circuit shown in this figure, the commercial AC power supply AC is rectified and smoothed by a rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci, and a rectified and smoothed voltage Ei is generated as a voltage across the smoothing capacitor Ci. Yes.
Note that the line of the commercial AC power supply AC is provided with a noise filter that includes a pair of common mode choke coils CMC and two across capacitors CL and removes common mode noise.

  The rectified and smoothed voltage Ei is input to the voltage resonant converter as a DC input voltage. As described above, this voltage resonance type converter adopts a single-end configuration including one switching element Q1. In this case, the voltage resonance type converter is a separately excited type, and the switching element Q1 of the MOS-FET is switched by the oscillation / drive circuit 2.

  A MOS-FET body diode DD is connected in parallel to the switching element Q1. A primary side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching element Q1.

  The primary side parallel resonant capacitor Cr forms a primary side parallel resonant circuit (voltage resonant circuit) with the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. The primary side parallel resonance circuit can obtain a voltage resonance type operation as the switching operation of the switching element Q1.

  The oscillation / drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1 in order to drive the switching element Q1. Thereby, the switching element Q1 performs a switching operation at a switching frequency corresponding to the cycle of the drive signal.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As the structure of the insulating converter transformer PIT, for example, as shown in FIG. 14, an EE type core in which an E type core CR11 and an E type core CR12 made of a ferrite material are combined is provided. And the bobbin B10 formed with resin etc. by the shape divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side is provided. The primary winding N1 is wound around one winding portion of the bobbin B10. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B10 on which the primary side winding and the secondary side winding are wound in this way to the EE type core (CR11, CR12), the primary side winding and the secondary side winding are different from each other. By the winding region, the EE type core is wound around the central magnetic leg. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, whereby k = 0.80 between the primary side and the secondary side. A coupling coefficient k of about ˜0.85 is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained.

  In FIG. 13, one end of the primary winding N1 of the insulating converter transformer PIT is inserted between the positive terminal of the switching element Q1 and the smoothing capacitor Ci, so that the switching output of the switching element Q1 is transmitted. It is like that. An alternating voltage induced by the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.

In this case, a secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. As a result, a secondary side parallel resonance circuit (voltage resonance circuit) is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2.
In addition, a half-wave rectifier circuit is formed by connecting a rectifier diode Do1 and a smoothing capacitor Co to the secondary side parallel resonant circuit as shown in the figure. This half-wave rectifier circuit uses, as a voltage across the smoothing capacitor Co, the secondary side DC output voltage Eo at a level corresponding to the same voltage as the alternating voltage V2 obtained in the secondary winding N2 (secondary side parallel resonant circuit). Generate. The secondary side DC output voltage Eo is supplied to the load and is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side DC output voltage Eo input as a detection voltage to the oscillation / drive control circuit 2.
The oscillation / drive circuit 2 adjusts the switching element Q1 so that the secondary side DC output voltage Eo becomes constant at a predetermined level in accordance with the level of the secondary side DC output voltage Eo indicated by the input detection output. Controls the switching operation. That is, a drive signal for obtaining a switching operation to be controlled is generated and output. Thereby, stabilization control of the secondary side DC output voltage Eo is performed.

15 and 16 show the experimental results for the power supply circuit having the configuration shown in FIG. In the experiment, as a condition of VAC = 100 V corresponding to the AC 100 V system, the main part of the power supply circuit in FIG. 13 is set as follows.
For the isolated converter transformer PIT, EER-35 is selected for the core, and the gap length of the central magnetic leg is set to 1 mm. The number of turns T (number of turns) of the primary winding N1 and the secondary winding N2 was set to N1 = 43T and N2 = 43T, respectively. For the coupling coefficient k of the insulating converter transformer PIT, k = 0.81 was set.
Further, a primary side parallel resonant capacitor Cr = 6800 pF and a secondary side parallel resonant capacitor C2 = 0.01 μF were selected. Accordingly, the resonance frequency fo1 = 175 kHz of the primary side parallel resonance circuit and the resonance frequency fo2 = 164 kHz of the secondary side parallel resonance circuit are set.
The rated level of the secondary side DC output voltage Eo is 135V, and the corresponding load power is the maximum load power Pomax = 200 W to the minimum load power Pomin = 0 W.

First, FIG. 15 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG. 13 by the switching period of the switching element Q1, and FIG. 15A shows the switching voltage at the maximum load power Pomax = 200 W. V1, switching current IQ1, primary winding current I1, secondary winding voltage V2, secondary winding current I2, and secondary side rectified current ID1 are shown. FIG. 15B shows the switching voltage V1, switching current IQ1, primary winding current I1, secondary winding voltage V2, secondary winding current I2, and secondary side rectified current ID1 when the minimum load power Pomin = 0 W. It is shown.
The switching voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a zero level in the period TON in which the switching element Q1 is turned on and becomes a sinusoidal resonance pulse in the period TOFF in which the switching element Q1 is turned off. The resonance pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is a voltage resonance type.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD). The switching current IQ1 is zero level during the period TOFF, and has a negative polarity by flowing in the forward direction with respect to the body diode DD during the turn-on. Then, it is inverted and then obtained as a waveform that flows between the drain and source of the switching element Q1 and increases until it is turned off. For this reason, a peak level is obtained at the turn-off timing as the switching current IQ1.
The primary winding current I1 flowing in the primary winding N1 is obtained by combining the current component flowing as the switching current IQ1 in the period TON and the current flowing in the primary side parallel resonant capacitor Cr in the period TOFF. It becomes a waveform.

Further, as the operation of the secondary side rectifier circuit, the rectified current ID1 flowing through the rectifier diode Do1 is set to 0 level by the waveform shown after the peak level is obtained when the rectifier diode Do1 is turned on at the maximum load power Pomax = 200 W. The voltage decreases and flows with a waveform that becomes 0 level during the off period of the rectifier diode Do1. Note that, when the minimum load power Pomin = 0 W, the level is 0 even during the on period.
In this case, the secondary winding voltage V2 is a voltage obtained in the parallel circuit of the secondary winding N2 // secondary parallel resonant capacitor C2, and the secondary rectifier diode Do1 is conductive. Corresponding to the ON period, it is clamped by the level of the secondary side DC output voltage Eo, and a sine waveform in the negative polarity direction is obtained in the OFF period of the secondary side rectifier diode Do1. Further, the secondary winding current I2 flowing through the secondary winding N2 is a combination of the rectified current ID1 and the current flowing through the secondary parallel resonant circuit (N2 (L2) // C2). It flows by.

FIG. 16 shows the switching frequency fs with respect to the load fluctuation, the ON period TON, the OFF period TOFF, and the AC → DC power conversion efficiency (ηAC → DC) with respect to the load variation for the power supply circuit shown in FIG.
First, looking at AC → DC power conversion efficiency (ηAC → DC), it is 90% or more in the range of load power Po = 75 W to 200 W. As a voltage resonance type converter, it is known that a single-ended type in which the switching element Q1 is one stone can obtain a good result in terms of power conversion efficiency.

Further, depending on the switching frequency fs, the ON period TON, and the OFF period TOFF shown in FIG. 16, the switching operation as the constant voltage control characteristic with respect to the load fluctuation in the power supply circuit of FIG. 13 is shown. In this case, the switching frequency fs is controlled such that the switching frequency becomes higher as the load tends to be lighter. Further, with respect to the ON period TON and the OFF period TOFF, the OFF period TOFF is almost constant with respect to the load variation, whereas the ON period TON is shortened as the light load tends to be reduced. Yes. In other words, the power supply circuit shown in FIG. 13 variably controls the switching frequency so as to shorten the on-period TON in accordance with, for example, a light load tendency while keeping the off-period TOFF constant. become.
Thus, the inductive impedance obtained by providing the primary side parallel resonant circuit and the secondary side parallel resonant circuit is varied by variably controlling the switching frequency. Depending on the variable inductive impedance, the amount of power transmitted from the primary side to the secondary side and the amount of power transmitted from the secondary side parallel resonant circuit to the load change, and the secondary side DC output voltage Eo The level of is variable. As a result, the secondary side DC output voltage Eo is stabilized.

FIG. 17 schematically shows the constant voltage control characteristics of the power supply circuit shown in FIG. 13 by the relationship between the switching frequency fs (kHz) and the secondary side DC output voltage Eo.
Here, assuming that the resonance frequency of the primary side parallel resonance circuit is fo1, and the resonance frequency of the secondary side parallel resonance circuit is fo2, the circuit of FIG. The secondary parallel resonance frequency fo2 is low.
In addition, assuming constant voltage control characteristics with respect to the switching frequency fs under the condition of a certain AC input voltage VAC, as shown in the figure, the resonance impedance under the resonance impedance corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit is shown. The constant voltage control characteristics at the time of maximum load power Pomax / minimum load power Pomin are shown as characteristic curves A and B, respectively, and are maximum under the resonance impedance corresponding to the resonance frequency fo2 of the secondary side parallel resonance circuit. The constant voltage control characteristics at the time of load power Pomax / minimum load power Pomin are shown by characteristic curves C and D, respectively.
Furthermore, when the primary side parallel resonance frequency and the secondary side parallel resonance circuit are provided as in the circuit of FIG. 13, an intermediate resonance frequency fo exists between the resonance frequencies fo1 and fo2. The resonance impedance characteristic based on the relationship between the intermediate resonance frequency fo and the switching frequency fs is indicated by the characteristic curve E at the maximum load power Pomax, and is indicated by the characteristic curve F at the minimum load power Pomin.
In the voltage resonance type converter including the secondary side parallel resonance circuit, the level of the secondary side DC output voltage Eo is determined by the resonance impedance characteristic with respect to the switching frequency fs of the intermediate resonance frequency fo. Further, the voltage resonance type converter shown in FIG. 13 adopts a so-called lower side control method in which the switching frequency fs is variably controlled in a frequency region lower than the center resonance frequency fo.
Then, under the characteristics shown as characteristic curves E and F corresponding to the intermediate resonance frequency fo in FIG. 17, the rated level of the secondary side DC output voltage Eo (FIG. 13) by switching frequency control corresponding to the lower side control. In the case of the above circuit, when trying to achieve a constant voltage with 135V) as a target value, the variable range (necessary control range) of the switching frequency fs necessary for this is the section indicated by Δfs. In other words, the secondary side DC output voltage Eo becomes the rated level tg by changing the switching frequency so as to be a required value according to the load fluctuation in the frequency range corresponding to the section indicated by Δfs. It is controlled in this way.

JP 2000-152617 A

  By the way, against the background of diversification of various electronic devices, the power supply circuit is also required to be compatible with a so-called wide range that operates in accordance with any commercial AC power input of AC100V system and AC200V system. .

As described above, the power supply circuit having the configuration shown in FIG. 13 operates so as to stabilize the secondary side DC output voltage Eo by switching frequency control, and the switching frequency variable range (for this purpose) Necessary control range) is indicated by Δfs described in FIG.
The power supply circuit shown in FIG. 13 is adapted to handle a relatively wide range of load fluctuations from 200 W to 0 W. In the power supply circuit of FIG. 13, the actual required control range corresponding to this load condition is fs = 117.6 kHz to 208.3 kHz, and Δfs = 96.7 kHz, which is a relatively wide range.

Here, as a matter of course, the level of the secondary side DC output voltage Eo also varies as the level of the AC input voltage VAC changes. That is, the level of the secondary side DC output voltage Eo similarly increases and decreases according to the level increase and decrease of the AC input voltage VAC.
Therefore, in response to fluctuations in the AC input voltage in the wide range from the AC100 system to the AC200V system, for example, in the case of dealing with fluctuations in a single range of only the AC100 system or AC200V system, the secondary side It can be said that the level fluctuation of the DC output voltage Eo1 also increases. In order to perform the constant voltage control operation corresponding to the level fluctuation of the secondary side DC output voltage Eo1 expanded in this way, the above-mentioned range of 117.6 kHz to 208.3 kHz is set in the direction of higher frequency. An expanded, wider range of required control is required.

However, as an IC (oscillation / drive circuit 2) for driving a switching element at present, the upper limit of the drive frequency that can be handled is about 200 kHz. In addition, even if an IC capable of driving at a high frequency as described above is developed, the power conversion efficiency is remarkably lowered when the switching element is driven at a high frequency, so that it can be used as a power supply circuit. Practically impossible.
From this, it can be seen that, for example, it is very difficult to realize the wide range by the configuration of the power supply circuit shown in FIG.

  For this reason, as a switching power supply circuit equipped with a resonant converter, when an operation corresponding to a wide range is to be realized, for example, according to an AC100V system / 200V system commercial AC power supply input, a primary side switching converter The configuration is switched between half bridge and full bridge. Alternatively, the operation of the rectifier circuit that performs the rectification operation for the commercial AC power supply AC is switched by full-wave rectification / double voltage rectification according to the AC 100V / 200V commercial AC power input. .

However, when the circuit configuration is switched between the AC100V system and the AC200V system, the following problems occur.
For example, for switching according to such a commercial AC power supply level, a threshold value (for example, 150V) for the input voltage is set, and circuit switching corresponding to the AC 200V system is performed when the threshold is exceeded, and AC 100V system is coupled when the threshold is decreased. However, if only such switching is performed, the AC 100 system is also supported for a temporary decrease in AC input voltage due to, for example, a momentary power failure at the time of AC 200 V input. There is a risk of switching. In other words, for example, in the switching configuration of the rectifying operation, the AC voltage is switched to the voltage doubler rectifier circuit even though the input is AC200V, and the switching element is overvoltage-resistant. May be destroyed.

Therefore, in order to prevent the above-described malfunction, in practice, not only the DC input voltage of the main switching converter but also the DC input voltage of the converter circuit on the standby power supply side is detected. To be.
However, detecting the converter circuit on the standby power supply side in this way means mounting a comparator IC or the like for comparing the reference voltage and the input voltage, but this increases the number of components. Therefore, an increase in circuit manufacturing cost and an increase in circuit board size are promoted.

  In addition, detecting the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunction in this way means that it can only be actually used if it is an electronic device having a standby power supply in addition to the main power supply. become. That is, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and there is a problem that the use range is narrowed accordingly.

Further, in the configuration for switching between the half bridge and the full bridge, it is necessary to include at least four switching elements in order to enable the full bridge configuration. That is, if switching is not necessary, a half bridge that requires only two switching elements must be added in this case.
Further, even when the rectifying operation is switched, two smoothing capacitors Ci must be provided in order to obtain a double voltage rectifying operation. That is, one smoothing capacitor Ci has to be added as compared with a configuration in which only full-wave rectification is performed.
Also in these respects, the wide-range configuration with circuit switching as described above causes an increase in circuit manufacturing cost and an increase in the size of the power circuit board. In particular, in the configuration of switching the rectifying operation, the smoothing capacitor Ci and the like enter a large category among the components constituting the power supply circuit, so that the increase in the substrate size is further promoted.

Further, as described above, another problem caused by the wide control range of the switching frequency is that the high-speed response characteristic for stabilization of the secondary side DC output voltage Eo is deteriorated. Is mentioned.
Particularly in recent electronic devices, for example, the load power may be a so-called switching load condition in which the load power changes instantaneously between the maximum load and no load according to on / off of various driving units. In response to this, it is necessary for the power supply circuit side to perform constant voltage control of the secondary side DC output voltage Eo in accordance with the load power which fluctuates at such a high speed and widely.
However, if the switching frequency control range is wide as described above, it takes a lot of time to change to the switching frequency necessary for constant voltage control corresponding to the load changing between the maximum value and the minimum value. Will be required. That is, the responsiveness of constant voltage control becomes dull.

  Further, the power supply circuit shown in FIG. 13 has a configuration including a primary side voltage resonance converter, and the power supply circuit having such a configuration has a characteristic that tends to be advantageous in terms of power conversion efficiency. As explained. However, in consideration of recent energy circumstances, environmental circumstances, and the like, electronic devices are required to have higher power conversion efficiency characteristics. In connection with this, the power supply circuit itself mounted in the electronic device is in a situation where further improvement in power conversion efficiency is required.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, a switching means formed by including a switching element that performs switching by inputting a DC input voltage, and a switching drive means that switches the switching element.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the switching output obtained in the primary winding are wound. Insulated converter transformer formed.
Further, a primary side parallel resonance circuit is provided which is formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of the primary side parallel resonance capacitor, and the operation of the switching means is a voltage resonance type.
Also, by connecting a secondary side parallel resonant capacitor in a parallel relationship with the secondary winding of the insulating converter transformer, a leakage inductance component including the secondary winding and the secondary side parallel resonant capacitor And a secondary side parallel resonant circuit formed by the capacitance of the second side.
A secondary-side DC output voltage generator configured to generate a secondary-side DC output voltage by performing a rectifying operation based on an alternating voltage induced in the secondary winding of the insulating converter transformer; Constant voltage control means for controlling the switching drive means according to the level of the secondary side DC output voltage and changing the switching frequency of the switching means to perform constant voltage control on the secondary side DC output voltage; Is provided.
In addition, the insulating converter transformer has an electromagnetic coupling type resonance circuit in which a coupling coefficient between the primary side and the secondary side is formed by including the primary side parallel resonance circuit and the secondary side parallel resonance circuit. A unimodal characteristic is obtained as an output characteristic with respect to the input of a frequency signal having the switching frequency, and the ratio of the ON period of the switching element when an AC input voltage of a predetermined level or higher is input is less than a predetermined value. It is set to a predetermined value that is considered a combination.
Then, the resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set so that power conversion efficiency of a certain level or more is obtained at least under a predetermined load condition. It was decided to.

The power supply circuit having the above configuration adopts a basic configuration as a voltage resonance type converter including a secondary side parallel resonance circuit on the secondary side. That is, the primary side and the secondary side are each provided with a secondary side parallel resonance circuit, thereby forming a coupled resonance circuit by electromagnetic coupling of the insulating converter transformer. Then, by reducing the coupling coefficient of the insulating converter transformer to a predetermined loose coupling value, the output characteristic for the frequency signal (switching output) of the switching frequency that is the input to the coupled resonant circuit is a steep single unit. Peak characteristics can be obtained. As a result, the variable range (necessary control range) of the switching frequency required for stabilizing the secondary side DC output voltage can be reduced.
Further, by reducing the value of the coupling coefficient, the ratio of the ON period, particularly when an AC input voltage exceeding a predetermined level is input, becomes small as a characteristic of the OFF period and the ON period of the switching element with respect to the load fluctuation. Became clear. In other words, the ratio of the ON period can be reduced to a predetermined value or less particularly when an AC input voltage of a predetermined level or higher is input, by reducing the coupling coefficient to a value that is a predetermined loose coupling as described above. This suppresses the peak level of the switching voltage applied to the switching element.
Further, by setting the resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit, a power conversion efficiency characteristic of a certain level or more can be obtained under a predetermined load condition.

Thus, the present invention reduces the variable control range (required control range) of the switching frequency necessary for constant voltage control for the voltage resonance type converter including the secondary side parallel resonance circuit.
As a result, the voltage resonance type switching converter can be easily realized in a wide range only by the switching frequency control.

  In addition, since the coupling coefficient is reduced to a value that is a predetermined loose coupling as described above, the ratio of the ON period of the switching element particularly when an AC input voltage of a predetermined level or higher is input is reduced. The peak level of the switching voltage applied to the switching element can be suppressed.

  Further, according to the present invention, a power conversion efficiency of a certain level or more can be obtained by setting the resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit so as to correspond to the load condition of a predetermined load power. I am trying to do it. The voltage resonance type converter originally has high power conversion efficiency characteristics. However, according to the present invention, a power supply circuit including a voltage resonance type converter can be provided having better power conversion efficiency characteristics.

  As a basic configuration for realizing the wide range, suppression of the peak level of the switching voltage, and high efficiency as described above, an insulating converter transformer is provided for a voltage resonant converter including a secondary parallel resonant circuit. It should be a structure that can provide the required coupling coefficient. Therefore, it can be used for a wide range without increasing costs, increasing the size of the circuit, increasing the weight, etc. I can say.

The circuit diagram of FIG. 1 shows a configuration example of a power supply circuit as the best mode (embodiment) for carrying out the present invention. The power supply circuit shown in this figure employs a basic configuration as a voltage resonance type switching converter by a single end system.
In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by the bridge rectifier circuit Di, and the rectified output is charged to the smoothing capacitor Ci. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter.

  In this figure, the switching converter that performs a switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage is formed as, for example, a single-ended voltage resonant converter including a single switching element Q1. In this case, a high breakdown voltage MOS-FET is selected as the switching element Q1. In this case, the driving method of the voltage resonance type converter is a separately excited type in which the switching element is switched by the oscillation / drive circuit 2.

A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.
The drain of the switching element Q1 is connected to the winding start end of a primary winding N1 of an insulating converter transformer PIT described later. The winding end end of the primary winding N1 is connected to the positive terminal of the smoothing capacitor Ei. Therefore, in this case, the DC input voltage (Ei) is supplied to the switching element Q1 through the serial connection of the primary winding N1. The source of the switching element Q1 is connected to the primary side ground.

  Since the MOS-FET is selected for the switching element Q1 in this case, a body diode DD is incorporated so as to be connected in parallel with the source and drain as shown in the figure. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
The primary side parallel resonance capacitor Cr has a primary side parallel resonance circuit (voltage resonance circuit) for the switching current flowing in the switching element Q1 by its own capacitance and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. Form. The primary side parallel resonance circuit performs a resonance operation, whereby a voltage resonance type operation is obtained as the switching operation of the switching element Q1. Accordingly, a sinusoidal resonance pulse waveform is obtained during the off period as the switching voltage V1, which is the voltage across the switching element Q1 (drain-source voltage).

  The oscillation / drive circuit 2 is a drive that is a gate voltage for switching the MOS-FET based on an oscillation circuit and an oscillation signal obtained by the oscillation circuit in order to drive the switching element Q1 by, for example, separate excitation. A signal is generated and applied to the gate of the switching element Q1. Thereby, the switching element Q1 continuously performs on / off operation according to the switching frequency corresponding to the cycle of the drive signal. That is, a switching operation is performed.

The insulating converter transformer PIT transmits the switching output of the primary side switching converter to the secondary side in a state where the primary side and the secondary side are galvanically insulated.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core in which E type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other. The E-type cores CR1 and CR2 are cores each having an E-shaped cross section as shown in the figure.
And the bobbin B formed with resin etc. is provided by the shape divided | segmented so that it might mutually become independent about the primary side and the secondary side winding part. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound.

  Furthermore, a gap G is formed as shown in the figure with respect to the central magnetic leg of the EE type core. The length of the gap G is an element in determining the coupling coefficient between the primary side and the secondary side. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

In addition, in the insulating converter transformer PIT in this case, the central portion of the two outer magnetic legs of the EE core, that is, the winding position of the primary winding N1 and the winding position of the secondary winding N2 The ferrite sheet cores FSC1 and FSC2 are provided so as to sandwich each of the portions corresponding to the boundary. These ferrite sheet cores FSC1 and FSC2 each have a predetermined thickness Ln2.
Further, as can be seen from the names, the ferrite sheet cores FSC1 and FSC2 in this case are made of the same ferrite as the E-type cores CR1 and CR2, and are provided between the outer magnetic legs. In addition, the ferrite sheet cores FSC1 and FSC2 further have a protruding portion (magnetic path generating portion) with a predetermined length Ln3 from the inner end surface portion of the outer magnetic leg to the inner magnetic leg side of the EE core. Provided. As a result, the end portions of the ferrite sheet cores FSC1 and FSC2 are inserted into the portion between the winding portions of the primary winding N1 and the secondary winding N2 in the bobbin B by a predetermined length. Become.

Here, for example, in the case of a simple EE type core structure in which the ferrite sheet cores FSC1 and FSC2 are not inserted as in the insulating converter transformer PIT shown in FIG. 14, the magnetic flux φ1 in FIG. A magnetic path is formed as shown by φ2. In FIG. 3, the bobbin B is omitted for easy understanding.
As can be seen from this figure, the magnetic paths of the magnetic fluxes φ1 and φ2 pass through the outer magnetic legs so as to straddle the primary winding N1 side and the secondary winding N2 side. For this reason, in the insulating converter transformer PIT shown in FIG. 14, the degree of coupling between the primary winding N1 and the secondary winding N2 is correspondingly high, and the coupling coefficient k set in the circuit of FIG. = 0.80 is obtained by forming a gap G having a gap length of about 1.0 mm on the inner magnetic leg.

On the other hand, when the ferrite sheet cores FSC1 and FSC2 are provided as shown in FIG. 2, the E-type cores CR1 and CR2 are ferrites of the same material as the ferrite sheet cores FSC1 and FSC2. Substantially, as shown in FIG. 3B, the shape of the EE-type core is assumed to have a shape in which the center part of the outer magnetic leg protrudes toward the center part side of the inner magnetic leg. Can do. The shape of such an EE type core is the same as that of the EE type core shape of FIG. 3A by the length Ln3 of the protruding portion (magnetic path generation site) and the central portion of the outer magnetic leg The spatial distance from the center part of the magnetic leg is shortened.
As these portions approach, the insulating converter transformer PIT shown in FIG. 3B generates magnetic fluxes indicated by φ11 and φ12 indicated by broken lines in FIG. 3B. The components of the magnetic fluxes φ11 and φ12 increase as the thickness Ln2 of the protruding portions of the ferrite sheet cores FSC1 and FSC2 increases and the amount of magnetic flux passing through the protruding portions of the ferrite sheet cores FSC1 and FSC2 increases. On the other hand, the component amount of the magnetic fluxes φ1 and φ2 decreases.

Here, the magnetic paths of the magnetic fluxes φ11 and φ12 are formed respectively corresponding to the primary winding N1 side and the secondary winding N2 side. Therefore, while the component amounts of the magnetic fluxes φ1 and φ2 described above decrease, the magnetic fluxes φ11 and φ12 tend to increase, which means that the degree of coupling between the primary winding N1 and the secondary winding N2 decreases. become.
That is, the coupling coefficient can be changed also by this ferrite sheet core (FSC), and the coupling coefficient can be set low according to the thickness Ln2 as described above.

In this way, the fact that the coupling coefficient is lowered by providing the ferrite sheet cores FSC1 and FSC2 means that according to the configuration of this example, a lower value is set as the coupling coefficient without increasing the gap G. be able to.
In the case of the present embodiment, specifically, the length of the gap G is set to about 1.0 mm as in the prior art, and the above-described thickness Ln2 of the ferrite sheet cores FSC1 and FSC2 is set to 1.5 mm. As a total coupling coefficient (hereinafter referred to as a total coupling coefficient kt) in the insulating converter transformer PIT, about kt = 0.6 is obtained.

As can be understood from the above description, the setting of the total coupling coefficient kt is such that a magnetic path generation site similar to the ferrite sheet core (FSC1, FSC2) is formed with respect to the basic shape of the EE type core. However, according to this, for example, an E-type core is molded so that an actual core shape as shown in FIG. 3B can be obtained without using a ferrite sheet core. Thus, by combining these, it is possible to set the total coupling coefficient kt similar to the case where the ferrite sheet core is used.
However, in the present embodiment, the ferrite sheet cores (FSC1, FSC2) are used as outer magnetic legs in consideration of the fact that the manufacturing process is simpler than molding the E-type core as described above. It is the form which is inserted.

In the case of the present embodiment, the insulating converter transformer PIT may have the structure shown in FIG.
In FIG. 4A, the insulating converter transformer PIT in this case is such that the magnetic legs of the E-type core CR3 and E-type core CR4 made of a ferrite material whose cross-sectional shape is E-shaped are opposed to each other. EE type core combined with
However, in this case, each of the two bobbins B5 for independently winding the primary winding (N1) and the secondary winding (N2) is formed on the EE core (CR3 / CR4). By attaching to each of the two outer magnetic legs, the primary side winding is wound around one outer magnetic leg and the secondary side winding is wound around the other outer magnetic leg as shown in the figure. It is supposed to be.
Also in this case, a gap G is formed with respect to the central magnetic leg (inner magnetic leg) of the EE type core. Also in this case, the gap G can be formed by making the central magnetic legs of the E-type cores CR3 and CR4 shorter than the two outer magnetic legs.

In the insulating converter transformer PIT having the structure shown in FIG. 4A, a magnetic path as shown in FIG. 4B is formed. In FIG. 4B, the bobbin B5 is omitted for convenience of explanation.
First, the primary side winding and the secondary side winding are wound around different outer magnetic legs as described above, and as a magnetic path in this case, as shown in FIG. The magnetic path by the magnetic flux φ3 formed so as to straddle the two outer magnetic legs on the core CR3 side and the E-type core CR4 side, and the magnetic flux φ13 and the magnetic flux φ14 passing from the outer magnetic leg side to the inner magnetic leg side, respectively. To form a magnetic path.
As shown in the figure, the magnetic path by the magnetic flux φ3 is formed so as to straddle the primary winding N1 side and the secondary winding N2 side.
The magnetic paths by the magnetic fluxes φ13 and φ14 are formed corresponding to the primary winding N1 side and the secondary winding N2 side, respectively.

As described above, the magnetic paths by the magnetic fluxes φ13 and φ14 formed respectively corresponding to the primary winding N1 side and the secondary winding N2 side pass through the inner magnetic legs as described above. Has been. Therefore, in this case as well, it can be understood that the length of the gap G formed on the inner magnetic leg of the EE core is one element for setting the degree of coupling between the primary side and the secondary side.
However, according to the structure in this case, the degree of coupling between the primary side and the secondary side can be increased by decreasing the magnetic fluxes φ13 and φ14 by increasing the length of the gap G. On the contrary, by shortening the length of the gap G, the magnetic fluxes φ13 and φ14 are increased and lowered. In other words, in the case of the structure shown in FIG. 4, in reducing the coupling coefficient, the length of the gap G is made shorter. It is possible to reduce the value of the coupling coefficient without accompanying.

Thus, according to the structure of the insulating converter transformer PIT of the present embodiment, the coupling coefficient between the primary side and the secondary side can be reduced without increasing the gap G.
For example, when the coupling coefficient is decreased by increasing the gap G, the eddy current loss in the insulating converter transformer PIT increases. In this example, the gap G is increased when the coupling coefficient is decreased. By not having this, it is possible to suppress this eddy current loss and effectively suppress the decrease in efficiency.

  In the following, it is assumed that the configuration shown in FIG. 2 is adopted as the insulating converter transformer PIT, and the coupling coefficient on the primary side and the secondary side is indicated by the total coupling coefficient kt. For confirmation, the length of the gap G is set to be the same when setting the same coupling coefficient value regardless of the configuration of FIGS. 2 and 4. The effect of suppressing the eddy current loss can be obtained equally.

Returning to FIG.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the drain of the switching element Q1 as described above. As a result, the switching output of the switching element Q1 is transmitted to the primary winding N1, and an alternating voltage is generated in the primary winding N1.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2.
A secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. Thus, a secondary side parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2. This secondary side parallel resonance circuit performs a resonance operation in accordance with a rectification operation of a secondary side rectifier circuit described later. That is, the voltage resonance operation can be obtained on the secondary side as well as the primary side.

  In this case, the secondary side rectifier circuit includes one rectifier diode Do1 and one smoothing capacitor Co2 with respect to the secondary winding N2 to which the secondary side parallel resonant capacitor C2 is connected in parallel as described above. Are connected to form a half-wave rectifier circuit. As a connection mode of the half-wave rectifier circuit, first, the anode of the rectifier diode Do1 is connected to the winding end end side of the secondary winding N2. Further, the cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The winding start end of the secondary winding N2 and the negative terminal of the smoothing capacitor Co are connected to the secondary side ground. Note that the rectifier diode Do1 performs an on / off operation at a relatively high frequency corresponding to the switching frequency, and therefore, a high-speed type (high-speed recovery type) is selected.

In the half-wave rectifier circuit formed in this way, with respect to the secondary winding voltage V2, the rectifier diode Do1 conducts during the half-wave period in which the winding end end side of the secondary winding N2 is positive, and the rectified current The operation of charging the smoothing capacitor Co is obtained. As a result, the secondary side DC output voltage Eo at a level corresponding to the same voltage as the induced voltage of the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co.
The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching element Q1 is driven such that the switching frequency is varied in accordance with the input detection output of the control circuit 1. In this embodiment, since the switching frequency is variable, the degree of coupling between the primary side and the secondary side is lowered as compared with the conventional case as described in FIG. The off period TOFF is not completely constant, but mainly the operation of variably controlling the on period TON. Such an operation is a constant voltage control operation for the secondary side DC output voltage.

  By variably controlling the switching frequency of the switching element Q1, the primary and secondary resonance impedances in the power supply circuit are changed and transmitted from the primary winding N1 to the secondary winding N2 side of the insulating converter transformer PIT. The amount of power and the amount of power to be supplied from the secondary side rectifier circuit to the load will change. As a result, an operation for controlling the level of the secondary side DC output voltage Eo is obtained such that the level fluctuation of the secondary side DC output voltage Eo is canceled. That is, the secondary side DC output voltage Eo is stabilized.

Here, as a practical example of the power supply circuit of the circuit configuration shown in FIG. 1, the main part is configured by setting as follows.
First, for the insulating converter transformer PIT, EER-35 was selected for the core, and a gap length of 1.0 mm was set for the gap G. As for the number of turns (number of turns) T of the primary winding N1 and the secondary winding N2, N1 = 63T and N2 = 25T were selected. In this case, the leakage inductance L1 of the primary winding N1 = 403 μH and the leakage inductance L2 of the secondary winding N2 = 6 μH. As described above, kt = 0.57 is set as the total coupling coefficient kt of the insulating converter transformer PIT.
For the capacitance of the primary side parallel resonant capacitor Cr, Cr = 4300 pF was selected. The resonance frequency fo1 = 166.0 kHz of the primary side parallel resonance circuit is set by the capacitance setting for the primary side parallel resonance capacitor Cr and the leakage inductance L1 of the primary winding N1 obtained by the structure of the insulating converter transformer PIT. . Further, C2 = 0.047 μF is selected as the capacitance of the secondary parallel resonant capacitor C2, and the capacitance setting and the leakage inductance L2 of the secondary winding N2 obtained by the structure of the insulating converter transformer PIT The secondary parallel resonance frequency fo2 = 110.5 kHz is set. Relatively, it can be said that the relationship of fo1≈1.5 × fo2 is obtained.
The corresponding load power is the maximum load power Pomax = 200 W, the minimum load power Pomin = 0 W (no load), and the rated level of the secondary side DC output voltage Eo is 135V.

  The waveform diagram of FIG. 5 shows the operation of the main part of the power supply circuit of FIG. 1 configured as described above by the switching cycle of the switching element Q1, and FIG. 5 (a) shows the maximum load power Pomax = 200W. The switching voltage V1, switching current IQ1, primary winding current I1, alternating voltage V2, secondary winding current I2, and secondary side rectified current ID1 when the AC input voltage VAC = 100V is shown. Further, FIG. 5B shows the above waveforms when the AC input voltage VAC = 230 V under the condition of the maximum load power Pomax = 200 W.

In FIG. 5, a switching voltage V1 is a voltage between the drain and source of the switching element Q1, and a switching current IQ1 is a current that flows from the drain side to the switching element Q1 (and the body diode DD). The on / off timing of the switching element Q1 is indicated by the switching voltage V1 and the switching current IQ1. One switching cycle is divided into a period TON in which the switching element Q1 is to be turned on and a period TOFF in which the switching element Q1 is to be turned off. It becomes. The resonance pulse of the switching voltage V1 is obtained as a sinusoidal resonance waveform because the operation of the primary side switching converter is a voltage resonance type.
In this case, when the AC input voltage VAC = 100V shown in FIG. 5A, the peak level (V1-p) of the switching voltage V1 is 660Vp. On the other hand, when the AC input voltage VAC = 230 V shown in FIG. 5B, V1−p = 780 Vp.

The switching current IQ1 is 0 level in the period TOFF. When the period TON ends and the period TON starts and reaches the turn-on timing, first, the switching current IQ1 has a negative waveform by flowing through the body diode DD. The waveform reverses to the positive polarity by flowing from the drain to the source. Such a waveform of the switching current IQ1 indicates that ZVS is properly performed.
When the AC input voltage VAC = 100V in FIG. 5A, the peak level of the switching current IO1 is 3.8 Ap, and when the AC input voltage VAC = 230 V shown in FIG. 5B, IQ1 = 4.0 Ap. .

  The primary winding current I1 is a current that flows through the primary winding N1, and is a combination of the current component that flows through the switching current IQ1 and the current that flows through the primary side parallel resonant capacitor Cr. The waveform of the period TOFF in the primary winding current I1 corresponds to the current waveform flowing in the primary side parallel resonant capacitor Cr.

When the alternating voltage is induced in the secondary winding N2, the rectifier diode Do1 is turned on every half cycle in which the secondary winding voltage V2 is positive and at a certain level or higher. The rectified current ID1 is supplied to Correspondingly, as the secondary winding voltage V2, the induced voltage level of the secondary winding N2 becomes a voltage higher than the secondary side DC output voltage Eo with respect to the rectifying diode Do1, and the rectifying diode Do1 is set. Corresponding to the conducting period, it is clamped by the secondary side DC output voltage Eo, and the period in which the rectifier diode Do1 is non-conducting is a sine wave with an envelope of a level equal to or lower than the secondary side DC output voltage Eo. . The secondary winding current I2 is obtained as a waveform obtained by synthesizing the rectified current ID1 and the current flowing through the secondary side parallel resonant capacitor C2.
As shown in the figure, the peak level of the secondary side rectified current ID1 when the AC input voltage VAC = 100 V is 3.8 Ap, and the peak level of the secondary side rectified current ID1 when the AC input voltage VAC is 230 V is 4.2 Ap. It has become.

FIG. 6 shows, as experimental results for the power supply circuit shown in FIG. 1, AC → DC power conversion efficiency (ηAC → DC) with respect to load fluctuation, switching frequency fs, time length of period TON and period TOFF, and switching voltage V1. The change characteristic of the peak level V1-p is shown.
As will be described later, according to the configuration of the power supply circuit of the present embodiment described with reference to FIG. 1, a wide-range configuration capable of operating in accordance with both the AC100V system and AC200V system inputs is realized. Is done. Accordingly, in FIG. 6, each characteristic when the AC input voltage VAC = 100 V is indicated by a solid line, and each characteristic when the AC input voltage VAC = 230 V is indicated by a broken line.

  In FIG. 6, the AC → DC power conversion efficiency (ηAC → DC) first increases as the load becomes heavy, and decreases as the AC input voltage VAC increases. ing. Specifically, the AC-to-DC power conversion efficiency in this case has a characteristic that increases as the load tends to be heavy as shown in the figure when the AC input voltage VAC = 230 V. However, when the AC input voltage VAC = 100 V, ηAC → DC tends to increase as the load power Po increases to near load power Po = 100 W, but in response to a further increase in load power Po. As a result, a characteristic that slightly decreases is obtained. In other words, when the AC input voltage VAC = 100 V, the AC → DC power conversion efficiency tends to increase as the load power decreases from the maximum load power Pomax = 200 W to the load power Po = 100 W. It is.

Further, the switching frequency fs changes with a tendency to increase as the load becomes lighter. Further, when the AC input voltage VAC is 100V and 230V, the value is higher when VAC = 230V.
In these cases, the constant voltage control operation in this case is a control to increase the switching frequency fs as the secondary side DC output voltage Eo increases in accordance with the light load tendency and the AC input voltage increasing tendency. It is shown that.

In FIG. 6, the time length of the period TON decreases as the light load tendency is reached. In this case, the period TOFF has a characteristic in which the light load tendency tends to decrease in a similar manner to the light load tendency.
Further, comparing the AC input voltage VAC = 100 V and VAC = 230 V, the period TON has a shorter time length when the AC input voltage VAC = 230 V, and conversely, the period TOFF has the AC input voltage VAC = 230 V. The time length is slightly longer.
According to this, in the circuit of FIG. 1, in stabilizing the secondary side DC output voltage Eo, the time length of the period TOFF slightly changes with respect to the load fluctuation and the AC input voltage fluctuation. Basically, it can be understood that the switching frequency control operation is mainly performed in which the time length of the period TON is variable.

  In FIG. 6, the peak level V1-p of the switching voltage V1 tends to increase as the load of the AC input voltage VAC increases at 100V and 230V as the load increases. It can also be seen that under the same load power condition, the value is higher when the AC input voltage VAC = 230V.

Here, in each characteristic shown in FIG. 6, the specific value of the switching frequency fs corresponds to the range of the maximum load power Pomax = 200 W to the minimum load power Pomin = 0 W when the AC input voltage VAC = 100 V. When fs = 100 kHz to 135 kHz, Δfs = 35.0 kHz, and the period TON / TOFF corresponding to the change of the switching frequency is TON = 6.6 μs to 4.7 μs and TOFF = 3.4 μs to 2.9 μs.
Also, when AC input voltage VAC = 230V, corresponding to the range of maximum load power Pomax = 200W to minimum load power Pomin = 0W, Δfs = 21kHz at fs = 138kHz-159kHz, corresponding to this change in switching frequency The period TON / TOFF to be performed is TON = 3.8 μs to 3.3 μs and TOFF = 3.5 μs to 3.0 μs.

As for AC → DC power conversion efficiency (ηAC → DC), the maximum load power Pomax = 200 W when AC → DC power conversion efficiency (ηAC → DC), and when AC input voltage VAC = 100 V, ηAC → DC = 91 When the AC input voltage VAC = 230 V, the measurement result of ηAC → DC = 90.8% was obtained.
Further, in this case, the characteristic particularly when the AC input voltage VAC = 100 V is set to increase as the load power decreases from the maximum load power to around the load power Po = 100 W as described above. Thus, the result that the range where ηAC → DC = 90% or more is from the load power Po = 200 W to about 30 W was obtained.

  Further, as described in FIG. 5, the peak level V1-p of the switching voltage V1 is V1-p = 660Vp when the maximum input power Pomax = 200 W and the AC input voltage VAC = 100V. When the input voltage VAC = 230 V, V1−p = 780 Vp.

Here, in the characteristics of the power supply circuit of FIG. 1 as described above, first, the characteristics regarding the switching frequency fs will be compared with the conventional power supply circuit shown in FIG.
In the power supply circuit of FIG. 13, with an input of AC input voltage VAC = 100 V, fs = 117.6 kHz to 208.3 kHz and Δfs = 96.7 kHz with respect to fluctuations of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W. It had been.
On the other hand, in the power supply circuit of FIG. 1, when the input voltage is AC input voltage VAC = 100 V, fs = 100 kHz to 135 kHz and Δfs = 100 kHz to fluctuations of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W. 35.0 kHz, and it can be seen that the required control range is greatly shortened compared to the characteristics of the power supply circuit of FIG. Further, in the power supply circuit of FIG. 1, with an input of AC input voltage VAC = 230 V, fs = 138 kHz to 159 kHz and Δfs = 21 kHz with respect to fluctuations of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W. Even under this condition, the required control range is significantly shortened compared to the characteristics of the power supply circuit shown in FIG.

  The characteristics of the switching frequency fs of the power supply circuit of FIG. 1 are as follows: AC 100 V system to AC 200 V system range (for example, VAC = 85 V to under the condition of corresponding load power of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W). H.264V) that can be stabilized in response to commercial AC power input, so-called wide-range compatibility is realized. Hereinafter, this point will be described.

First, the power supply circuit shown in FIG. 1 has a basic configuration as a voltage resonance type converter including a secondary side parallel resonance circuit. That is, it can be said that the power supply circuit shown in FIG. 1 includes parallel resonant resonance circuits on the primary side and the secondary side via electromagnetic induction of the insulating converter transformer PIT. If such a configuration is seen from the relationship between the primary side parallel resonance circuit and the secondary side resonance circuit, a coupled resonance circuit by electromagnetic coupling is formed, to which a frequency signal corresponding to the switching frequency fs is input. Can be seen equivalently.
The constant voltage control characteristic for the secondary side DC output voltage Eo of the power supply circuit of FIG. 1 that includes the electromagnetically coupled resonance circuit in this way depends on the degree of coupling (coupling coefficient k) of the insulating converter transformer PIT. Will be different. This point will be described with reference to FIG.

FIG. 7 shows the output characteristics with respect to the input (switching frequency signal) for the above-described electromagnetically coupled resonance circuit. That is, the control characteristic for the secondary side DC output voltage Eo is shown by the relationship with the switching frequency fs. In this figure, the horizontal axis represents the switching frequency, and the vertical axis represents the level of the secondary side DC output voltage Eo.
As described in FIG. 1, in the present embodiment, the resonance frequency fo1 of the primary side series resonance circuit is set to be about 1.5 times the resonance frequency fo2 of the secondary side parallel resonance circuit. Therefore, the resonance frequency fo1 is higher than the resonance frequency fo1.
In FIG. 7, the horizontal axis indicating the switching frequency fs is shown corresponding to the resonance frequencies fo1 and fo2. In this case, the resonance frequency fo1 is set corresponding to the relationship between the resonance frequencies fo1 and fo2. This is shown as being higher than the resonance frequency fo2.

  Here, it is assumed that the coupling degree of the insulating converter transformer PIT is set to a tight coupling state where the total coupling coefficient kt = 1. In this case, the leakage inductance L1 of the primary winding N1 and the leakage inductance L2 of the secondary winding N2 are 0 respectively.

で表され、
周波数ｆ２は、

で表される。
また、上記（数１）（数２）における項の１つであるｆｏは、一次側直列共振回路の共振周波数ｆｏ１と、二次側並列共振回路の共振周波数ｆｏ２との中間に存在する中間共振周波数であり、１次側のインピーダンスと２次側のインピーダンスと、一次側と二次側とで共通となるインピーダンス（相互結合インダクタンスＭ）により決定される周波数である。
なお、相互結合インダクタンスＭについては、

により表される。 As described above, as a constant voltage control characteristic in a state where the primary side and the secondary side of the insulating converter transformer PIT are tightly coupled, as shown as the characteristic curve 1 in FIG. The resonance frequency fo1 and the resonance frequency fo2 of the secondary side parallel resonance circuit have so-called bimodal characteristics in which the secondary side DC output voltage Eo peaks at frequencies f1 and f2.
Here, the frequency f1 is

Represented by
The frequency f2 is

It is represented by
Further, fo, which is one of the terms in the above (Equation 1) and (Equation 2), is an intermediate resonance that exists between the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side parallel resonance circuit. It is a frequency determined by the impedance on the primary side and the impedance on the secondary side, and the impedance (mutual coupling inductance M) that is common to the primary side and the secondary side.
For the mutual coupling inductance M,

It is represented by

  Further, assuming that the total coupling coefficient kt is gradually decreased from the state of kt = 1, that is, when the degree of loose coupling is gradually increased from the tightly coupled state, it is shown in FIG. The characteristic curve 1 shows a change in which the bimodal tendency gradually diminishes and becomes flat near the intermediate resonance frequency fo. Then, when the coupling coefficient kt is lowered to a certain coupling coefficient kt, a so-called critical coupling state is obtained. In this critical coupling state, as shown by the characteristic curve 2, there is no tendency as a bimodal characteristic, and the curve shape becomes flat with the intermediate resonance frequency fo as the center.

Further, if the coupling coefficient kt is decreased from the critical coupling state and the loose coupling state is increased, the peak is obtained only at the intermediate frequency fo as shown by the characteristic curve 3 in FIG. Unimodal characteristics are obtained. Further, when comparing the characteristic curve 3 with the characteristic curves 1 and 2, the characteristic curve 3 has a peak level itself lower than that of the characteristic curves 1 and 2, but as a quadratic curve shape thereof, It can be seen that it has a steeper slope.
Insulating converter transformer PIT of the present embodiment is set in a loosely coupled state in which coupling coefficient kt≈0.6 or less. In the setting of the coupling coefficient kt, the operation is based on the single peak characteristic shown as the characteristic curve 3.

  When the unimodal characteristic shown in FIG. 7 is compared with the constant voltage control characteristic of the conventional power supply circuit (FIG. 13) shown in FIG. 17, the characteristic shown in FIG. For the characteristics, the slope is considerably gentler in terms of a quadratic function.

  As described above, since the characteristic shown in FIG. 17 is gentle in a curve, the necessary control range of the switching frequency for performing constant voltage control on the secondary side DC output voltage Eo is, for example, AC input voltage VAC = 100V. Even under the condition corresponding to a single range by the input of fs = 117.6 kHz to 208.3 kHz, Δfs = 96.7 kHz. For this reason, as described above, it is very difficult to achieve a wide range only by constant voltage control by switching frequency control.

On the other hand, the constant voltage control characteristic of the present embodiment is a single peak characteristic indicated by the characteristic curve 3 in FIG. 7, and the constant voltage control operation is, for example, as shown in FIG. .
In FIG. 8, the characteristic curves A and B at the maximum load power Pomax and the minimum load power Pomin when the AC input voltage VAC = 100 V (AC 100 V system) for the power supply circuit of the present embodiment shown in FIG. Four characteristic curves are shown, which are characteristic curves C and D at the maximum load power Pomax and at the minimum load power Pomin when the AC input voltage VAC = 230 V (AC 200 V system).

As can be seen from FIG. 8, first, when the AC input voltage VAC = 100 V corresponding to the AC 100 V system input, the switching required for making the secondary side DC output voltage Eo constant at the required rated level tg. The frequency variable control range (necessary control range) is indicated by Δfs1. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve A to the switching frequency fs at level tg in the characteristic curve B.
In addition, when the AC input voltage VAC = 230 V corresponding to the AC 200 V system input, the variable control range (necessary control) of the switching frequency necessary for making the secondary side DC output voltage Eo constant at the required rated level tg. (Range) is indicated by Δfs2. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve C to the switching frequency fs at level tg in the characteristic curve D.

As described above, the unimodal characteristic that is the control characteristic of the secondary side DC output voltage Eo in the present embodiment is considerably steep in a quadratic function curve as compared with the control characteristic shown in FIG. It is.
For this reason, Δfs1 and Δfs2 that are required control ranges when the AC input voltage VAC = 100 V and VAC = 230 V are considerably reduced as compared with Δfs shown in FIG.
This also changes the frequency variable range (ΔfsA) from the lowest switching frequency at Δfs1 (switching frequency fs at level tg in characteristic curve A) to the highest switching frequency at Δfs2 (switching frequency fs at level tg in characteristic curve A). ) Will be correspondingly narrow.
For confirmation, the actual Δfs1, Δfs2, and ΔfsA measured in the power supply circuit of FIG.
Δfs1 = 35kHz (= 135kHz-100kHz)
Δfs2 = 21kHz (= 159kHz−138kHz)
ΔfsA ≒ 59 kHz (= 159kHz-100kHz)
It becomes.

The frequency variable range ΔfsA is sufficiently within the variable range of the switching frequency corresponding to the current switching drive IC (oscillation / drive circuit 2). That is, in the power supply circuit shown in FIG. 1, the switching frequency can be actually variably controlled within the frequency variable range ΔfsA.
In this way, the power supply circuit of the present embodiment shown in FIG. 1 is suitable for any AC 100 V system or AC 200 V system commercial AC power input, and the secondary side DC output voltage Eo that is a proper main DC power supply. Can be stabilized. In other words, wide range support is possible only by switching frequency control.

  Incidentally, a coupled resonance circuit using electromagnetic coupling is already known as a technique for expanding the amplification bandwidth of an amplifier circuit using a transistor, for example, as an intermediate frequency transformer amplifier. However, in such a field, the bimodal characteristic in the tight coupling or the flat characteristic in the critical coupling is used, and the single peak characteristic in the loose coupling is not used. In the present embodiment, in such a coupled resonant circuit technology using electromagnetic coupling, a single-peak characteristic with loose coupling that has not been employed in the field of communication technology is actively used in the field of resonant switching converters. It can be said that. As a result, as described above, the variable range (necessary control range) of the switching frequency necessary for stabilizing the secondary side DC output voltage Eo is reduced, and the wide range only by the constant voltage control in the switching frequency control. This is possible.

Generally, as the degree of loose coupling between the primary side and the secondary side of the insulating converter transformer PIT is increased, the power loss (eddy current loss) in the insulating converter transformer PIT tends to increase. As a result, the power conversion efficiency also decreases accordingly. However, in the present embodiment, as will be described later, a practically sufficient power conversion efficiency characteristic is obtained. This is because a resonance circuit (secondary parallel resonance circuit) is also formed on the secondary side.
That is, by providing the secondary side parallel resonance circuit, it is possible to supply power as the secondary side DC output voltage Eo including the increase in energy obtained by the resonance operation, and the coupling is loosely coupled. This compensates for a decrease in power conversion efficiency.

In addition, as described above, the necessary control range (Δfs) of the switching frequency fs for constant voltage control is reduced under the conditions of AC 100 V system and AC 200 V system commercial AC power input. The responsiveness and control sensitivity of voltage control will be greatly improved.
Some electronic devices perform an operation of changing the load power Po so as to be switched (switched) at a relatively high speed between a maximum and no load, which is a so-called switching load. An example of a device that performs such an operation as a switching load is a printer that is a peripheral device of a personal computer.
When a power supply circuit having a relatively wide required control range Δfs as shown in FIG. 13 is mounted on a device that operates as such a switching load, for example, as described above, the device has a steep Following the change of the load power, the switching frequency fs is variably controlled by a correspondingly large amount of change. For this reason, it has been difficult to obtain high-speed constant voltage control response.
On the other hand, in the present embodiment, since the necessary control range Δfs is greatly reduced particularly in the region for each single range, with respect to the steep fluctuation between the maximum load power Po and no load, It is possible to stabilize the secondary side DC voltage Eo in response to high speed. That is, the response performance of the constant voltage control with respect to the switching load is greatly improved.

By the way, in the present embodiment, the total coupling coefficient kt is set to a value lower than the conventional value as described above. However, this suppresses an increase in the peak level of the switching voltage V1. The experimental results were obtained. This point will be described below.
First, the peak level V1-p of the switching voltage V1 is
V1−p = Ei {(1 + π / 2) × (TON / TOFF)}
It is represented by
From this equation, it can be understood that the peak level V1-p increases as the rectified smoothing voltage (DC input voltage) Ei increases (that is, in this case, the AC input voltage VAC increases). In addition, it can be understood that the value increases as the value of “TON / TOFF” increases, that is, as the ratio of the period TON in one switching cycle increases.

  From the characteristic diagram of FIG. 6, it can be seen that the value of “TON / TOFF” increases as the heavy load condition is reached, and becomes maximum at the maximum load power Pomax = 200 W. Since the value of the peak level V1-p increases as the AC input voltage VAC rises as described above, the value of the peak level V1-p depends on the AC input voltage VAC and the load power Po. It can be understood that the maximum value is obtained when becomes the maximum.

Here, if attention is paid to the characteristic that the peak level V1-p increases according to the load power as described above, the characteristic of the period TON with respect to the period TOFF is shown as the characteristic of the period TON and the period TOFF with respect to the load fluctuation shown in FIG. If the time length is further reduced, the above-described “TON / TOFF” value is decreased, and as a result, an increase in the peak level V1-p can be suppressed.
In particular, since the peak level V1-p has a maximum value at the maximum AC input voltage, in order to suppress the peak value V1-p on the maximum value side, on the maximum AC input voltage side as indicated by a broken line in FIG. As a characteristic of the period TON and the period TOFF, it is only necessary to obtain a characteristic in which the ratio of the period TON becomes smaller.

As a result of the experiment, when the value of the total coupling coefficient kt of the insulating converter transformer PIT in the configuration of FIG. In the characteristics obtained on the side (in this case, the AC 200V system is input because of the wide range configuration), the change in the time ratio between the period TOFF and the period TON becomes large, and as a result, the period TON within one switching cycle The result was that the proportion of the share decreased. That is, by setting the value of the total coupling coefficient kt lower in this way, the result that the value of “TON / TOFF” as described above can be further reduced particularly on the maximum AC input voltage side is obtained. It is.
In this way, since the characteristic that the value of “TON / TOFF” becomes smaller especially on the maximum AC input voltage side is obtained, the value of “TON / TOFF” when the maximum AC input voltage is input is also smaller. To be. As the value of “TON / TOFF” at the time of inputting the maximum AC input voltage becomes smaller in this way, the maximum value of the peak level V1-p of the switching voltage V1 is also lowered accordingly. Become.

As the value of the coupling coefficient kt is lowered in this way, the maximum value of the peak level V1-p is suppressed.
If the maximum value of the peak level V1-p of the switching voltage V1 is suppressed, the voltage level applied to the switching element Q1 is also reduced accordingly, so that a part having a lower withstand voltage is selected as the switching element Q1. can do.
If a component with a lower breakdown voltage can be selected, the element size can be reduced, and the circuit area can be reduced. Further, since the component cost is lower, the circuit manufacturing cost can be reduced accordingly.

Here, in the present embodiment, kt = 0.57 is set as the total coupling coefficient kt. However, according to the setting of the total coupling coefficient kt, in this case, the maximum peak level V1-p of the switching voltage V1 is set. The value (at the time of maximum AC input voltage and maximum load power: in this case VAC = 264V · Po = 200 W) can be suppressed to 850 Vp.
With the maximum value of the peak level V1-p, a 900V withstand voltage product can be used as the switching element Q1 in this case.

By the way, regarding the characteristics of the power conversion efficiency (ηAC → DC) in the power supply circuit shown in FIG. 1, in the description of FIG. 6, ηAC → in a wide range from the load power Po = 200 W to about 30 W when the AC input voltage VAC = 100 V. DC = 90% or more. When this characteristic is compared with the characteristic in the case of the circuit of FIG. 13 at the same AC input voltage VAC = 100 V shown in FIG. 16, the range of ηAC → DC = 90% or more in the case of the circuit of FIG. The load power Po = 200 W to 75 W.
That is, according to this, in the power supply circuit of FIG. 1, the range where ηAC → DC = 90% or more is expanded as compared with the prior art. This means that better power conversion efficiency characteristics than the conventional ones are obtained.

  As described above, the reason why a good power conversion efficiency is obtained in this way is that the reduction of the power conversion efficiency is compensated by providing the secondary side resonance circuit as described above. However, in this embodiment, the setting of the resonance frequencies fo1 and fo2 of the primary side parallel resonance circuit and the secondary side parallel resonance circuit is a major factor. The power conversion efficiency characteristics with respect to the load conditions of the present embodiment as described above are finally obtained by adjusting the resonance frequencies fo1 and fo2. That is, the characteristics are finally obtained by performing various settings for the resonance frequencies fo1 and fo2 and performing experiments and setting fo1 = 166.0 kHz and fo2 = 110.5 kHz as described above. Comparing the resonance frequencies fo1 and fo2 with the prior art, in the power supply circuit shown in FIG. 13, fo1 = 175.0 kHz and fo2 = 164.0 kHz, and the magnitude relationship is the same when fo1> fo2. The value of each frequency is different from the frequency difference associated therewith. Respective frequency values of the resonance frequencies fo1 and fo2 are reduced as compared with the conventional case, and the frequency difference is greatly expanded.

One of the reasons why the power conversion efficiency is improved by setting the resonance frequencies fo1 and fo2 as described above is as follows. As can be seen by comparing the switching current IQ1 at the same AC input voltage VAC = 100 V in FIGS. 5A and 15A, the switching current IQ1 of FIG. 5A corresponding to this embodiment is shown. The waveform is such that a peak level of 3.8 Ap can be obtained at the timing before the turn-off when the ON period TON of the switching element Q1 ends and transitions to the OFF period TOFF. When the turn-off timing is reached, the level is lower than the peak level.
The waveform of the switching current IQ1 is influenced by the waveform of the secondary winding current I2. That is, it has a waveform component corresponding to the current flowing in the secondary parallel resonant circuit. The waveform of the secondary winding current I2 is determined by the setting of the resonance frequency fo2 with respect to the resonance frequency fo1.
Therefore, the waveform of the switching current IQ1 of the power supply circuit shown in FIG. 1 is obtained by appropriate setting of the resonance frequencies fo1 and fo2 of the primary side parallel resonance circuit and the secondary parallel resonance circuit. It will be.
The waveform of the switching current IQ1 shown in FIG. 5 (a) means that the level of the switching current IQ1 at the time of turn-off is suppressed. If the level of the switching current IQ1 at the time of turn-off is suppressed, the switching loss and the conduction loss at the time of turn-off are reduced correspondingly.
Further, in the power supply circuit of the present embodiment, as shown in FIG. 5A, the secondary side rectified current corresponds to the fact that the waveform of the switching current IQ1 falls below the peak level at the time of turn-off as described above. Similarly, ID1 has a waveform in which the level at turn-off is suppressed. In the conventional power supply circuit, as shown in FIG. 15A, a peak level is obtained at the time of turn-off. In this way, the switching loss and conduction loss in the rectifier diode Do1 are also reduced.
Such a reduction in switching loss and conduction loss of the switching element and the rectifying element is a main factor in obtaining high power conversion efficiency characteristics for the power supply circuit of the present embodiment.

  Here, the waveform diagram at the time of the load power Po = 200 W shown in FIG. 5 is illustrated and described, but actually, the peak level suppression effect similar to that described above is particularly the load power. It is thought that the efficiency improvement in this load range is aimed at by having appeared more prominently around Po = 25W-100W. As a result, as described above, the range of ηAC → DC = 90% or more is expanded in comparison with the circuit of FIG.

As described above, in the switching power supply circuit according to the present embodiment, the overall coupling coefficient kt of the insulating converter transformer PIT is loosely coupled as a voltage resonant converter in which the secondary parallel resonant circuit is combined with the primary parallel resonant circuit. As a result, it is possible to achieve a wide range only by variable control of the switching frequency.
Further, as described above, the power conversion efficiency is further improved by setting the resonance frequencies fo1 and fo2 of the primary side parallel resonance circuit and the secondary side parallel resonance circuit.

  Further, in the present embodiment, as the total coupling coefficient kt is reduced to a predetermined value that is loosely coupled as described above, the ratio of the ON period TON particularly on the maximum AC input voltage side is reduced. As a result, the peak level V1-p of the switching voltage V1 is suppressed from increasing, and the breakdown voltage level of the switching element Q1 is thereby reduced. At this time, the maximum value of the peak level V1-p is suppressed to about 850 Vp by reducing the total coupling coefficient kt to the above-described kt = 0.57, whereby a 900 V withstand voltage product is selected as the switching element Q1. Can do.

  As a basic configuration for realizing such wide-range compatibility, high efficiency, and the effect of suppressing the peak level V1-p of the switching voltage V1, a voltage resonance type converter including a secondary side parallel resonance circuit ( For example, in contrast to the conventional configuration shown in FIG. 13, it is sufficient that at least the insulating converter transformer PIT has a structure capable of obtaining a total coupling coefficient kt below the required level. Therefore, the cost increases due to an increase in the number of parts and the circuit becomes large. There is also a merit that this can be realized without increasing the weight or increasing the weight.

  In order to reduce the coupling coefficient to a relatively low value as described above, in this embodiment, the insulating converter transformer PIT has the structure shown in FIG. This can be achieved. In other words, it is not necessary to enlarge the gap G when setting a coupling coefficient lower than that of the prior art, and it is possible to effectively prevent a decrease in efficiency due to eddy current loss accompanying the enlargement of the gap G.

Next, in FIGS. 9 to 12 below, variations of the secondary side rectifier circuit will be described as modifications of the power supply circuit of the present embodiment.
First, FIG. 9 shows a configuration of a power supply circuit as a first modification.
9 to the following FIG. 12, only the insulating converter transformer PIT and the secondary side rectifier circuit are shown. Other parts other than these are the same as those in FIG. 1 and are not shown again. 9 to 12, the same parts as those in FIG.

Also in the power supply circuit shown in FIG. 9, a secondary side parallel resonant circuit is formed by connecting a secondary side parallel resonant capacitor C2 in parallel to the secondary winding N2. In addition, a bridge full-wave rectifier circuit is provided as a secondary rectifier circuit. This bridge full-wave rectifier circuit is formed by forming a bridge rectifier circuit by four rectifier diodes Do1, Do2, Do3, and Do4 and then connecting the bridge rectifier circuit and the smoothing capacitor Co as shown in the figure. The
In the bridge full-wave rectifier circuit formed in this way, the rectifier diodes Do1 and Do4 are turned on to charge the smoothing capacitor Co every half cycle of the induced voltage of the secondary winding N2, and the rectifier diodes Do2 and The operation of charging Do3 with the smoothing capacitor Co is alternately obtained. As a result, as the voltage across the smoothing capacitor Co, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the induced voltage level of the secondary winding N2 is obtained.

Also in the power supply circuit as the first modified example, kt≈0.6 is set as the total coupling coefficient kt for the insulating converter transformer PIT by the same configuration as in FIG. In addition, for the primary side parallel resonance frequency fo1 and the secondary side parallel resonance frequency fo2, the relationship of fo1 ≒ 1.5 x fo2 is obtained so that an AC to DC power conversion efficiency value of a certain level or higher can be obtained even under the predetermined load conditions Each resonance frequency is set so as to obtain the above.
As a result, the power supply circuit as the first modification can be adapted to a wide range only by the switching frequency control, and the responsiveness of the constant voltage control operation to the switching load can be improved. Further, the AC → DC power conversion efficiency can be improved as in the case of FIG.
Further, as in the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

FIG. 10 shows a configuration as a second modification.
In the second modification, a double-wave rectifier circuit is provided as a secondary-side rectifier circuit.
In order to form a double-wave rectifier circuit, first, a center tap is applied to the secondary winding N2, so that the secondary winding portions N2A and N2B are divided with the center tap as a boundary. The center tap is connected to the secondary side ground.
In this case, the secondary side parallel resonant capacitor C2 for forming the secondary side parallel resonant circuit is connected in parallel to the entire secondary winding N2.
In addition, in this case, two rectifier diodes Do1 and Do2 and one smoothing capacitor Co are provided as component elements forming the secondary side rectifier circuit. The anode of the rectifier diode Do1 is connected to the end of the secondary winding N2 on the secondary winding portion N2A side, and the anode of the rectifier diode Do2 is connected to the end of the secondary winding N2 on the secondary winding portion N2B side To do. The cathodes of the rectifier diodes Do1 and Do2 are both connected to the positive terminal of the smoothing capacitor Co, and the negative terminal of the smoothing capacitor Co is connected to the secondary side ground.

  In the secondary-side double-wave rectifier circuit formed in this way, the secondary winding portion NA corresponds to the half cycle of one polarity of the secondary winding voltage V2 induced in the secondary winding N2. The rectification current ID1 flows through the path of the rectifier diode Do1 → smoothing capacitor Co to charge the smoothing capacitor Co. Further, in response to the other half cycle of the polarity of the secondary winding V2, a rectified current ID2 flows through the path of the secondary winding NB → rectifier diode Do2 → smoothing capacitor Co to charge the smoothing capacitor Co. . In this way, the full-wave rectification operation for charging the rectification current to the smoothing capacitor Co is performed corresponding to the positive and negative half-cycle periods of the secondary winding voltage V2. As a result, as the voltage across the smoothing capacitor Co, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the induced voltage level of the secondary winding N2 is obtained.

Also in the power supply circuit as the second modified example, the insulation converter transformer PIT is set to kt≈0.6 as the total coupling coefficient kt by the same configuration as in FIG. 1, and the primary side parallel resonance frequency fo1 and the secondary side parallel are set. With respect to the resonance frequency fo2, a relationship of fo1≈1.5 × fo2 is set so that even in this case, AC → DC power conversion efficiency exceeding a certain level can be obtained.
As a result, the power supply circuit as the second modified example can be compatible with a wide range only by switching frequency control, and the responsiveness of the constant voltage control operation with respect to the switching load is improved, and further AC → DC power conversion The efficiency is improved as in the case of FIG.
Similarly to the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

FIG. 11 shows a configuration of a power supply circuit as a third modification.
The third modification includes a voltage doubler rectifier circuit as the secondary side rectifier circuit. This voltage doubler rectifier circuit illustrates two rectifier diodes Do1 and Do2 and two smoothing capacitors Co1 and Co2 with respect to the parallel connection circuit of the secondary winding N2 // secondary parallel resonant circuit C2. In this way, they are connected.
In this case, the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 are connected to the winding end of the secondary winding N2. The cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co1.
The smoothing capacitors Co1 and Co2 are connected in series so that the negative electrode terminal of the smoothing capacitor Co1 and the positive electrode terminal of the smoothing capacitor Co2 are connected, and the winding of the secondary winding N2 is connected to the connection point of the smoothing capacitors Co1 and Co2. Connect the start end.
The negative terminal of the smoothing capacitor Co2 and the anode of the rectifier diode Do2 are connected to the secondary side ground.

  In the secondary voltage doubler rectifier circuit formed in this way, the path of [secondary winding N2 → rectifier diode Do1 → smoothing capacitor Co1] in the half cycle of one polarity of the secondary winding voltage V2. The rectified current flows through and the smoothing capacitor Co1 is charged. Further, in the half cycle of the other polarity of the secondary winding voltage V2, a rectification current flows through a path of [secondary winding N2 → rectifier diode Do2 → smoothing capacitor Co2] to charge the smoothing capacitor Co2. In this way, the charging of the smoothing capacitor Co1 and the charging of the smoothing capacitor Co2 are alternately performed every half cycle when the secondary winding voltage V2 is positive and negative, and each of the smoothing capacitors Co1 and Co2 has a secondary winding. A potential corresponding to an equal magnification of the induced voltage level of the line N2 is obtained. As a result, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the induced voltage level of the secondary winding N2 is obtained as the voltage across the series connection circuit of the smoothing capacitors Co1-Co2.

Also in the power supply circuit of the third modified example, the insulation converter transformer PIT is set to kt≈0.6 as the total coupling coefficient kt by the same configuration as in FIG. 1, and the primary side parallel resonance frequency fo1 and the secondary side parallel resonance are set. Regarding the frequency fo2, the relationship of fo1≈1.5 × fo2 is set so that even in this case, AC / DC power conversion efficiency exceeding a certain level can be obtained.
As a result, the power supply circuit as the third modified example can also be compatible with a wide range only by switching frequency control, improve the responsiveness of the constant voltage control operation to the switching load, and further convert AC to DC power. The efficiency is improved as in the case of FIG.
Similarly to the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

FIG. 12 shows a configuration of a power supply circuit as a fourth modification.
In the fourth modification, the secondary side rectifier circuit is a quadruple voltage rectifier circuit.
Even in this case, first, a parallel resonant capacitor C2 is connected in parallel to the secondary winding N2.
As shown in the figure, the quadruple voltage rectifier circuit includes four rectifier diodes including rectifier diodes Do1 to Do4, a capacitor Cc1, a capacitor Cc2, and smoothing capacitors Co1 and Co2.
In this case, the positive electrode of the smoothing capacitor Co1 is connected to one end (winding end) of the secondary winding N2 through a series connection of a capacitor Cc1 → rectifier diode Do1 (anode → cathode) as shown in the figure. Terminal is connected. The negative terminal of the smoothing capacitor Co1 is connected to the other end (winding end) of the secondary winding N2.
Further, the positive terminal of the smoothing capacitor Co2 is connected to the connection point between the negative terminal of the smoothing capacitor Co1 and the other end of the secondary winding N2, and the negative terminal of the smoothing capacitor Co2 is connected to the secondary side ground. It is connected.

Further, a series connection circuit of a capacitor Cc2 → rectifier diode Do4 (cathode → anode) is inserted between the one end of the secondary winding N2 and the secondary side ground.
The anode of the rectifier diode Do3 is connected to the connection point between the capacitor Cc2 and the rectifier diode Do4 as shown. The cathode of the rectifier diode Do3 is connected to the connection point between the connection point of the smoothing capacitors Co1 and Co2 and the other end of the secondary winding N2.
Further, the anode of the rectifier diode Do2 is connected to the connection point between the cathode of the rectifier diode Do3 and the other end of the secondary winding N2. The cathode of the rectifier diode Do2 is connected to the connection point between the capacitor Cc1 and the rectifier diode Do1.

In the quadruple voltage rectifier circuit having the above configuration, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding N2 → rectifier diode Do2 → capacitor Cc1 → secondary winding N2. ] By the circulation route. Similarly, in the other half cycle of the alternating voltage, the rectified current flows through [secondary winding N2 → capacitor Cc2 → rectifier diode Do3 → secondary winding N2] through the circulation path.
That is, in this case, a DC voltage having a level corresponding to the same multiple of the alternating voltage level excited by the secondary winding N2 is obtained at both ends of the capacitors Cc1 and Cc2 in a corresponding half cycle.

In this case, in each half cycle, the rectified current branches from the circulation path and also flows through the following paths.
First, in one half cycle of the alternating voltage described above, the rectified current branches and flows through the path [smoothing capacitor Co2 → rectifier diode Do4 → capacitor Cc2 → secondary winding N2]. At this time, the capacitor Cc2 is charged during this period due to the previous circulation path. For this reason, depending on the rectification current path as described above, the smoothing capacitor Co2 is charged with a voltage level due to the superposition of the alternating voltage obtained at the secondary winding N2 and the voltage across the capacitor Cc2. Become.
That is, a smoothing capacitor Co2 generates a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2.

Further, in the other half cycle of the above alternating voltage, the rectified current branches and flows through a path [capacitor Cc1 → rectifier diode Do1 → smoothing capacitor Co1 → secondary winding N2]. The smoothing capacitor Co1 is charged according to the voltage level of the alternating voltage of the secondary winding N2 that has received the superimposed voltage across the capacitor Cc1.
In other words, even with the smoothing capacitor Co1, a voltage corresponding to twice the alternating voltage level obtained at the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co1.

  In this manner, a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is generated at both ends of the smoothing capacitor Co1 and the smoothing capacitor Co2. As a result, a secondary side DC output voltage Eo having a level corresponding to four times the alternating voltage level excited in the secondary winding N2 is obtained at both ends of the series connection of the smoothing capacitor Co1 and the smoothing capacitor Co2. It will be.

Also in the power supply circuit of the fourth modification shown in FIG. 12, with respect to the insulating converter transformer PIT, kt≈0.6 is set as the total coupling coefficient kt by the same configuration as in FIG. 1, and the primary side parallel resonant frequency fo1 As for the secondary side parallel resonance frequency fo2, the relationship of fo1≈1.5 × fo2 is set so that the AC → DC power conversion efficiency exceeding a certain level can be obtained even in this case.
As a result, the power supply circuit as the fourth modified example can be compatible with a wide range only by switching frequency control, and the responsiveness of the constant voltage control operation with respect to the switching load is improved. The efficiency is improved as in the case of FIG.
Similarly to the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

In addition, as this invention, it is not limited to the structure shown as said each embodiment. For example, the detailed circuit configuration of the primary side voltage resonance type converter and the configuration of the secondary side rectifier circuit formed including the secondary side parallel resonance circuit are also conceivable.
As the switching element, it may be possible to select an element other than the MOS-FET. In each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

It is a circuit diagram which shows the structural example of the power supply circuit as embodiment of this invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a figure for demonstrating the magnetic path formed in the insulated converter transformer by the structure shown in FIG. It is a figure shown about the other structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped, and its magnetic path. It is a wave form diagram which shows operation | movement of the principal part of the power supply circuit of embodiment by a switching period. It is a figure which shows the fluctuation | variation characteristics of the AC-> DC power conversion efficiency with respect to load fluctuation | variation, switching frequency, the ON / OFF period of a switching element, and the peak level of a switching voltage about the power supply circuit of embodiment. It is a figure which shows the constant voltage control characteristic about the power supply circuit of embodiment. It is a figure which shows the switching frequency control range (required control range) according to alternating current input voltage conditions and load fluctuation | variation as constant voltage control operation | movement of the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as a 1st modification. It is a circuit diagram which shows the structural example of the power supply circuit as a 2nd modification. It is a circuit diagram which shows the structural example of the power supply circuit as a 3rd modification. It is a circuit diagram which shows the structural example of the power supply circuit as a 4th modification. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art example. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit shown in FIG. 13 is provided. FIG. 14 is a waveform diagram showing an operation of a main part of the power supply circuit shown in FIG. 13. It is a figure which shows the fluctuation characteristic of the AC-> DC power conversion efficiency with respect to load fluctuation | variation, the switching frequency, and the ON / OFF period of a switching element about the power supply circuit shown in FIG. It is a figure which shows notionally the constant voltage control characteristic about the conventional power supply circuit.

Explanation of symbols

  1 Control circuit, 2 Oscillation / drive circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, CR1, CR2, CR3, CR4 E type core, FSC1, FSC2 ferrite sheet core, Cr primary side parallel Resonant capacitor, N1 primary winding, N2 secondary winding, C2 secondary side parallel resonant capacitor, Do1, Do2, Do3, Do4 rectifier diode, Co, Co1, Co2 (secondary side) smoothing capacitor, N2A, N2B secondary Winding part, Cc1, Cc2 capacitor

Claims (3)

Switching means formed with a switching element for switching by inputting a DC input voltage;
Switching driving means for switching and driving the switching element;
Formed by winding at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the switching output obtained in the primary winding An isolated converter transformer,
A primary side parallel resonant circuit formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of the primary side parallel resonant capacitor, and the operation of the switching means is a voltage resonant type;
By connecting a secondary side parallel resonant capacitor in parallel with the secondary winding of the insulating converter transformer, a leakage inductance component including the secondary winding and a capacitance of the secondary side parallel resonant capacitor A secondary parallel resonant circuit formed by
A secondary side DC output voltage generating means configured to perform a rectifying operation based on an alternating voltage induced in the secondary winding of the insulating converter transformer to generate a secondary side DC output voltage;
Constant voltage control means for performing constant voltage control on the secondary side DC output voltage by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. And
The insulation converter transformer has a coupling coefficient between the primary side and the secondary side,
With respect to the electromagnetic coupling type resonance circuit formed by including the primary side parallel resonance circuit and the secondary side parallel resonance circuit, the output characteristic with respect to the input of the frequency signal having the switching frequency is a single peak characteristic and has a predetermined level. The ratio of the ON period of the switching element at the time of the above AC input voltage input is set to a predetermined value that is regarded as loosely coupled, so as to be a predetermined value or less,
Furthermore, the resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set so that a power conversion efficiency of a certain level or more can be obtained at least under a predetermined load condition.
A switching power supply circuit. The insulation converter transformer is
The primary winding and the secondary winding are wound around the central magnetic leg of the EE type core, a gap is formed with respect to the central magnetic leg, and the outer magnet of the EE type core is formed. A ferrite sheet core is provided so as to protrude toward the central magnetic leg at a predetermined position of the leg,
The switching power supply circuit according to claim 1. The insulation converter transformer is
The primary winding is wound around one outer magnetic leg of the EE core, and the secondary winding is wound around the other outer magnetic leg, and a gap is formed with respect to the central magnetic leg of the EE core. Formed,
The switching power supply circuit according to claim 1.

Images