JP2006197073A - Code diversity communication method and system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a code diversity communication method and system in which interference suppression performance can be enhanced without increasing a burden on the output stage of a transmitter by an arrangement of constant amplitude composite sequence for keeping the amplitude of a transmission signal at a constant level. <P>SOLUTION: A transmitter 10 comprises a modulator 11 for multiplying the sum of orthogonal cyclic shift M series sets by a spread code sequence to create a sequence of constant amplitude and multiplying the sequence thus created by transmission data before being transmitted. A receiver 20 comprises a demodulator 21 for despreading the signal transmitted from the transmitter 10 by the sum sequence of subset of the spread code sequence and the orthogonal cyclic shift M series; an integrator 22 for integrating the despread signal and outputting correlation of branch; a multiplier 23 for multiplying the despread signal by a weighting factor corresponding to variance of correlation, and a combiner 24 for combining the multiplied signals. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a code diversity communication method and a code diversity communication system in a direct sequence / spread spectrum (DS / SS) system, and in particular, a code diversity communication method and a code capable of improving interference suppression performance. The present invention relates to a diversity communication system.

  In recent years, demand for mobile communication such as mobile phones and wireless local area networks (LANs) has increased rapidly, and various communication methods are being studied. Among them, a spread spectrum (SS) communication method capable of various communication forms is attracting attention. One of the SS systems is a direct sequence / SS (DS / SS) system that directly spreads the spectrum. In this DS / SS system, multiple access communication (DS / SSMA) can be performed by assigning a code to each station.

  In DS / SSMA, a plurality of users simultaneously use a common communication path by using a spreading sequence assigned in advance. However, at this time, the other station signal due to multiple access becomes an interference component with respect to the own station signal. Normally, since the influence of this interference is small, the receiver can correctly detect the correlation of the local station signal. However, if the power of the interference signal is larger than that of the local station signal, the communication quality is significantly degraded. I will. This problem is generally called a perspective problem (see Non-Patent Documents 1 and 2).

In order to solve this near / far problem, an inter-station interference cancellation method (see Non-Patent Document 1) has been proposed. It is necessary to know the PN (Pseudo Noise) sequence of the station.
In addition, a method of reducing interference by orthogonalizing interference sequences using an orthogonal filter has been proposed (see Non-Patent Documents 3 and 4). However, with this method, it takes a convergence time to make them orthogonal.

  In addition to this, a partial correlation weight control method (see Non-Patent Document 5) that estimates a cross-correlation value in a partial portion of a received signal and weights the interval, and LMS (Least Mean Square: minimum) is used as the partial correlation weight control method. A method of introducing a root mean square algorithm (see Non-Patent Document 6) has been proposed. However, these methods are difficult to control and complicate the system.

  Therefore, a code diversity method (see Non-Patent Document 7) based on the application of an existing system has been proposed as one method for suppressing interference without knowing the PN sequence of the other station and without estimating the sequence of the other station. ing. In this system, several code sequence sums are used as spreading sequences, and diversity reception is performed on individual sequences before summation, and the system configuration is simple.

  Note that a spread spectrum communication apparatus that converges an interference canceller by orthogonalization when performing spread spectrum communication by the direct spread code division multiple access method is disclosed in Japanese Patent Laid-Open No. 9-200188 published on July 31, 1997. (See Patent Document 1).

Japanese Patent Laid-Open No. 9-200188 R. Kohno, Pseudo-noise sequence and interference cancellation techniques for spread spectrum system-spread spectrum theory and techniques in Japan, IEICE, Trans., Vol.E74, no.5, pp.1083-1092, May 1991. M.Okubo and M.Nakagawa, Countermeasure against SSMA near-far problem using unknown sequence estimator, IEEE ISSSTA94, vol.1PP.167-170, July 1994. Yoshimasa Naomasa, Gohkawa Akihisa, Yanagi Shuzo, Furuya Nozuna, `` DS / CDMA Adaptive Interference Canceller Suitable for Mobile Communication Environments '', IEICE Technical Report, RCS93-76, pp.47-54 (1993.11). Yoshimasa Naomasa, Gohkawa Akihisa, Yanagi Shuzo, Furuya Nozuna, "Delay Detection Waveform DS / CDMA Adaptive Interference Canceller Suitable for High-Speed Fading Transmission Lines", IEICE, vol.J77-B-II, no.11, pp .618-627 (1994.11). ] Hideki Asato, Shinichi Tachikawa, Hajime Marubayashi, "Suppression of interference between other stations of DS / SSMA by partial correlation weight control method", IEICE theory, vol.J79-A, no.3, pp.785-794 (1996) -03). Kazuya Umeka and Shinichi Tachikawa, "Application of DS / CDMA Partial Correlation Weight Control Method and LMS Algorithm", IEICE Technical Report, SST96-99, pp.7-12 (1997-03). T. Seki, M. Hamamura, S. Tachikawa, Suppression Effects of Multiple Access Interference in DS / CDMA with Code-Diversity, IEICE, Trans., Vol.E82-A, no.12, pp.2720-2727, December 1999 .

  However, in the conventional code diversity system, since the transmission output becomes multilevel, a large amplifier that maintains linearity in the transmitter output stage is required, which increases the burden on the transmitter output stage. In addition, when an amplitude limiter for noise removal is used, there is a problem that characteristics deteriorate.

  The present invention has been made in view of the above circumstances, and improves the interference suppression performance without increasing the burden on the transmitter output stage by configuring the constant amplitude synthesis sequence to keep the amplitude of the transmission signal constant. It is an object to provide a code diversity communication method and a code diversity communication system that can be used.

  Another object of the present invention is to provide a code diversity communication method and a code diversity communication system in which characteristics under a perspective problem are evaluated and a branch configuration method is devised in order to improve the characteristics. .

  The present invention for solving the problems of the above-described conventional example is a code diversity communication method in the direct spread spectrum spread system, in which a transmitter code sequence is added to a sum of a set of orthogonalized cyclic shift M sequences on the transmitter side. Is used to generate a series having a constant amplitude, and the generated series is multiplied by transmission data for transmission.

  In the above code diversity communication method, the transmission power is normalized by dividing the value of each chip of the orthogonal cyclic shift M sequence by the value of the combined amplitude of all sequences of the orthogonal cyclic shift M sequence. It is characterized by that.

  The present invention relates to a code diversity communication method in the direct spread spectrum spread system, and a receiving side that receives a signal transmitted by the code diversity communication method includes a subset of a code sequence for spreading and a set of orthogonalized cyclic shift M sequences. The branch sequence is despread by the sum series, the correlation of the branch is output, the corresponding weighting factor is calculated from the variance of the correlation, and demodulated by the sum of the despread signal multiplied by the corresponding weighting factor.

  The present invention is characterized in that, in the above-described code diversity communication method, a set of orthogonalized cyclic shift M sequences is divided by the number of diversity branches and assigned to the branches.

  The present invention is characterized in that, in the code diversity communication method described above, a subset of orthogonalized cyclic shift M-sequences is assigned to each branch in order from one having a small correlation variance.

  The present invention is characterized in that, in the code diversity communication method described above, a subset of orthogonalized cyclic shift M-sequences is intensively assigned to one branch from one having a small correlation variance.

  The present invention is characterized in that, in the code diversity communication method, a branch having the largest correlation variance is not used.

  The present invention is a code diversity communication system in a direct spread spectrum spread system, in which a transmitter generates a sequence having a constant amplitude by multiplying a set of orthogonalized cyclic shift M sequences by a spreading code sequence, It has a modulator for multiplying the generated sequence by transmission data and transmitting it.

  The present invention is a code diversity communication system in a direct spread spectrum spread system, in which a receiver transmits a signal transmitted from the transmitter by a sum sequence of a subset of a spreading code sequence and an orthogonalized cyclic shift M sequence. A demodulator that despreads, an integrator that integrates the despread signal and outputs the correlation of the branch, a multiplier that multiplies the despread signal by a weighting factor corresponding to the variance of the correlation, and And a synthesizer for synthesizing signals.

  According to the present invention, in the receiver, a rearrangement circuit is provided between the integrator and the multiplier, and the correlation output from the integrator is assigned to each branch in order from one having a small correlation variance. It is characterized by.

  According to the present invention, in the receiver, a rearrangement circuit that intensively assigns a correlation output from the integrator to one branch from a component having a small correlation variance is provided between the integrator and the multiplier. It is characterized by.

  The present invention is characterized in that, in the receiver, the rearrangement circuit does not use a branch having the largest correlation variance.

  According to the present invention, there is provided a code diversity communication method in the direct spread spectrum spreading method, wherein a sequence having a constant amplitude is obtained by multiplying a sum of a set of orthogonalized cyclic shift M sequences by a spreading code sequence on the transmitter side. Since the code diversity communication method of generating and multiplying the generated sequence by transmission data is used, there is an effect that the burden is not increased at the output stage of the transmitter. Further, even if an amplitude limiter is provided at the front stage of the receiver, there is an effect that the signal waveform is hardly deteriorated.

  According to the present invention, the code diversity communication method for normalizing transmission power by dividing the value of each chip of the orthogonal cyclic shift M sequence by the value of the combined amplitude of all sequences of the orthogonal cyclic shift M sequence Therefore, there is an effect that transmission power can be suppressed.

  According to the present invention, there is provided a code diversity communication method in the direct spread spectrum system, and a receiving side that receives a signal transmitted by the code diversity communication method includes a spreading code sequence and an orthogonal cyclic shift M sequence portion. Code diversity communication method that despreads by sum series of set, outputs branch correlation, calculates corresponding weighting factor from variance of correlation, and demodulates by sum of weighted factor corresponding to despread signal Therefore, there is an effect that the interference suppression performance can be improved without degrading the performance.

  According to the present invention, there is provided a code diversity communication system in the direct spread spectrum spreading method, which generates a sequence having constant amplitude by multiplying a sum of a set of orthogonalized cyclic shift M sequences by a spreading code sequence. Since the transmitter has a modulator that multiplies transmission data by the transmission data and transmits the result, there is an effect that the burden is not increased in the output stage of the transmitter.

  According to the present invention, there is provided a code diversity communication system in a direct spread spectrum spread system, wherein a signal transmitted from the transmitter is despread by a sum sequence of a subset of a spreading code sequence and an orthogonal cyclic shift M sequence. A demodulator that integrates the despread signal and outputs a branch correlation, a multiplier that multiplies the despread signal by a weighting factor corresponding to the variance of the correlation, and a multiplied signal. Since the receiver has a combiner for combining, there is an effect that the interference suppression performance can be improved without degrading the performance.

  According to the present invention, the receiver is provided with a rearrangement circuit that assigns one sequence at a time to each branch from the one having a small correlation variance between the integrator and the multiplier, with respect to the correlation output from the integrator. Therefore, there is an effect that the characteristics can be improved even if there is a difference in the initial phase between the own station series and the other station series.

  According to the present invention, the receiver is provided with a rearrangement circuit that allocates a correlation output from the integrator in a concentrated manner to one branch from those having a small correlation variance between the integrator and the multiplier. Therefore, there is an effect that the characteristics can be improved even if there is a difference in the initial phase between the own station series and the other station series.

[Outline of Embodiment of the Present Invention]
Embodiments of the present invention will be described with reference to the drawings.
In the code diversity communication method and code diversity communication system in the direct spread spectrum spread system according to the embodiment of the present invention, the transmitter side multiplies the sum of the set of orthogonalized cyclic shift M sequences by the spreading PN sequence ± 1 creates a sequence with constant amplitude, multiplies the data, and transmits it. On the receiving side, the spread PN sequence and a part of the orthogonal cyclic shift M sequence are despread and correlated to obtain a correlation variance. The weighting factor is obtained from the above and multiplied by the weighting factor to obtain the addition of each diversity branch. As a result, it is possible to improve the interference suppression performance without increasing the burden at the output stage of the transmitter and without degrading the performance even if the amplitude limiter of the receiver is provided.

[Configuration of constant amplitude composite series]
First, the principle and configuration method of a code diversity communication method in the direct spread spectrum spreading method according to the embodiment of the present invention will be described with reference to FIG. 1 and FIG. FIG. 1 is an explanatory diagram showing the sum of sets of orthogonalized cyclic shift M sequences in the code diversity system according to the embodiment of the present invention, and FIG. 2 shows 3 in the code diversity system according to the embodiment of the present invention. It is explanatory drawing which shows the sum of two signal series. In comparison with FIG. 2, FIG. 11 shows the sum of three conventional signal sequences.

In the code diversity system according to the embodiment of the present invention, two sequences of OM and PN ₁ are required in order to construct a constant amplitude composite sequence used for spreading (in practice, only PN ₁ is sufficient. This will be described later). Here, OM is the sum of orthogonally-shifted cyclic shift M sequences, and PN ₁ is another M sequence. FIG. 1 shows the configuration of this orthogonalized cyclic shift M sequence.

The sequence of each row of Ortho M in FIG. 1 is orthogonalized by adding an orthogonalization coefficient α to all chips of sequence length N. Therefore, a in the determinant is a value such as a ₁ = 1 + α obtained by adding α to the normal chip value. The orthogonalization coefficient α is shown in the following [Equation 1]. However, ± in the formula takes only one of + and −.

  The combined amplitude (sum of each column) by all the sequences of the orthogonalized cyclic shift M sequences expressed in this way becomes a certain constant value (c in FIG. 1). This is apparent from the fact that each column has one −1 due to the nature of the M-sequence and that the orthogonalization coefficient α is added to the total number of chips N. The relationship is expressed by the following equation [Equation 2].

Since the orthogonalized cyclic shift M-sequence sum (OM) and the M-sequence, each chip having a constant value, are integrated with each chip, a composite sequence having a constant amplitude can be realized. However, as it is, the transmission power becomes larger than that in communication with the normal DS / SS system. Therefore, by multiplying each chip by the coefficient k expressed by the following equation [Equation 3], power can be obtained. Perform normalization.
In FIG. 1, since the sum c of each column becomes a constant value, normalization is performed in principle by multiplying by 1 / c.

At this time, s _i represents the amplitude component of each chip of the transmission signal. However, power normalization in the code diversity system according to the embodiment of the present invention is synonymous with setting all the values of each chip of the orthogonalized cyclic shift M sequence to 1.
Therefore, in the code diversity system according to the embodiment of the present invention, the value of the chip at the time of data transmission has a constant amplitude of ± as shown in FIG. 4, which is equivalent to the DS / SS system using only PN _1. . FIG. 4 is a graph of comparison of chip values during transmission in the conventional code diversity and the proposed code diversity (according to the embodiment of the present invention).
As shown in FIG. 4, in the conventional code diversity, the interior of the bar is indicated by diagonal lines, and in the proposed code diversity, the interior of the bar is indicated by dotted lines. In the case of the proposed code diversity, the chip value has a constant amplitude of ± 1.

  This means that, as shown in FIG. 2, the spread sequence (the sum of three sequences) has a constant amplitude of ± 1. On the other hand, in the conventional code diversity, as shown in FIG. 11, the spreading sequence (sum of three sequences) is multilevel.

[System configuration]
Next, the code diversity communication system according to the embodiment of the present invention will be described with reference to FIG. FIG. 3 is a block diagram showing the configuration of the code diversity communication system according to the embodiment of the present invention.
The code diversity communication system (this system) according to the embodiment of the present invention basically includes a transmitter 10 and a receiver 20 as shown in FIG.
In FIG. 3, an adder is provided in the transmission path between the transmitter 10 and the receiver 20, but this is due to artificially generated interference (Interference) and noise (AWGN [Additive White Gaussian Noise: additive white Gaussian noise]) is included in the transmission signal.
Although not shown, when an impulsive burst noise or the like is included in the transmission path, an amplitude limiter is generally provided at the front stage of the receiver 20.

[Transmitter]
The transmitter 10 of this system includes a modulator 11 that spreads and outputs data (DATA) to be transmitted by a sequence OM × PN ₁ constituting a constant amplitude composite sequence. Here, OM is the sum of orthogonalized cyclic shift M sequences, and PN ₁ is another M sequence. The modulator 11 can be realized by a DSP (Digital Signal Processor) or the like.

In the transmitter 10, although not shown, in principle, in order to form an orthogonal cyclic shift M sequence, a cyclic shift register that cyclically shifts the values a _{1 to} a _N of each chip, and the shift A storage unit that cyclically shifts in a register and stores the contents so as to form a determinant as shown in FIG. 1, an arithmetic means for each column sum that calculates the sum of the columns (vertical direction) of the determinant in FIG. It is obtained by normalization coefficient calculation means for obtaining a coefficient k for power normalization from the sum (constant value) and multiplication means for multiplying each chip by the coefficient k.
However, as a result, OM becomes a constant value of +1 (or -1). Therefore, in practice, the OM generation process can be omitted only for the transmitter, and the conventional DS / SS system using only PN ₁ may be used.

[Receiving machine]
The receiver 20 assigns individual sequences constituting the combined sequence OM used on the transmission side to the branches, and performs a demodulation operation on each branch in order to perform a correlation operation in each branch. M number of integrators 22 for integrating the despread signal, M number of multipliers 23 for weighting the integration results, a synthesizer 24 for synthesizing each weighted signal, and data for determining ± 1 information And a determiner 25.

At the receiver 20, a set of orthogonal cyclic shift M series in FIG. 1 the number (N) is equally divided at substantially the number of diversity branches, OM ₁ equally divided aggregate thereof OM ₁ ～OM _M becomes, in the demodulator 21 × PN _{1 to} OM _M × PN ₁ are respectively despread.
Each integrator 22 obtains a correlation by integrating the despread signal.

Generation of OM _{1 to} OM _M is not shown, but in order to form an orthogonal cyclic shift M sequence, a cyclic shift register that cyclically shifts the values a _{1 to} a _N of each chip, and a cyclic shift by the shift register A storage unit that stores the contents that are shifted to store a determinant as shown in FIG. 1 and a column sum calculator that calculates the sum of a subset of each column (vertical direction) of the determinant in FIG. Yes. The above means are realized by a DSP or the like collectively.
Each partial sequence (OM ₁ × PN ₁ ,... OM _M × PN ₁ ) is calculated by a device external to the receiver 20, for example, a calculation device such as a computer, and provided to the demodulator 21 of the receiver 20. You may make it do.
Specifically, the cyclic shift register is used to cycle through the values of each chip, and the cyclic result is stored in the storage unit. Then, OM _1, ..., thereby calculating the sum of each subset required to configure the OM _M. Note that the receiver 20 does not need to normalize the entire power if the calculation is performed with the amplitude of each series of the cyclic shift register being constant.

When the correlation value in the integrator 22 is close to the value of only the own station, the influence of the cross-correlation due to the interference with the other station series is small, so that the multiplier 23 performs large weighting using the coefficient g. For example, the most stable data demodulation is performed.
When the correlation value in the integrator 22 is significantly different from the value of only the own station, the influence of cross-correlation due to interference with other station sequences is large, so that the multiplier 23 performs small weighting using the coefficient g. To do.
That is, the coefficient g becomes a small value when there is much interference, and the coefficient g becomes a large value when there is little interference.

Then, the synthesizer 24 synthesizes the weighted correlation output of each branch and outputs it via the data decision unit 25.
Although the weighting factor g in the multiplier 23 will be described below, the weighting factor g is determined by estimating the variance of cross-correlation and the variance of noise from the correlation output of each branch. Although not shown in the figure, a control unit for inputting a correlation output of each branch and calculating a weighting factor g is provided, and the weighting factor g is output to each multiplier 23 from the control unit. . The calculation of the weighting factor g is obtained by the equation [Equation 9], and the control unit includes means for calculating the weighting factor g using the equation.

[Derivation of weighting factor expression]
In code diversity, for a composite sequence used on the transmission side, individual sequences constituting the composite sequence are assigned to branches on the reception side, and correlation calculation is performed in each branch. At this time, since the sequence assigned to each branch has a different pattern, the influence of cross-correlation due to interference also differs in each branch.

Therefore, the most stable data demodulation can be performed if a large weight is applied to the correlation output having the least influence of the cross-correlation. For this purpose, a method for estimating the variance of cross-correlation and the variance of noise from the correlation output of each branch and a method for determining the weighting factor g _M are shown.

  First, focus on one branch k among several branches. The output after integration in this branch is defined as the following equation [Formula 4].

S _k is a local signal component in each branch, I _k is an interference component, and N _k is a noise component. Here, if E [•] represents the average of •, the data of the own station and other stations are random and the average is 0, it is expressed by the following equation [Equation 5].

Further, since S _k , I _k , and N _k are independent of each other, the average of the products can be considered as an independent average product. Therefore, it is expressed by the following formula [Formula 6].

  Therefore, the root mean square of the equation [Equation 4] is expressed by the following equation [Equation 7].

Here, assuming that the root mean square E [S ² ] of the integral output of only the local station signal is known, the equation [Equation 7] is modified to change the interference component as shown in the following Equation [Equation 8]. The sum of the variance and the variance of the noise component, that is, the magnitude of the variance of the disturbing component can be estimated.

At this time, since the magnitude of noise input to each branch is the same, the variance value of noise is the same for each branch. Therefore, the magnitude of the influence of the cross-correlation between the sequence assigned to each branch and the interference sequence can be known from the equation [Equation 8]. As a result, the coefficient g _M corresponding to the integrated output including the influence of the cross-correlation with the other station series in each branch is defined as the following equation [Equation 9]. Here, k is a branch number (1, 2,... M), and μ is an integrated output of the signal component.
The coefficient g _M decreases as the interference increases.

[Branch configuration method]
In code diversity, for a composite sequence used on the transmission side (each sequence has an orthogonal relationship with each other), an individual sequence constituting the sequence is assigned on the reception side. In conventional code diversity, a spread sequence is configured by adding several code sequences to each chip, and a branch is provided for each ± 1 constant amplitude sequence used for the spread sequence on the receiving side.

  However, in this method, since the code sequence to be used is fixed, it is considered that there is a large difference in the characteristic improvement effect depending on the initial phase shift between the own station sequence and the other station sequence. At this time, if the characteristics are poor, it is necessary to change or increase the spreading sequence to be used, so that sequence control at the transmitter is required.

On the other hand, in the case of the code diversity system according to the embodiment of the present invention, a constant amplitude composite sequence using orthogonalized cyclic shift M sequences is used. A series can be used.
If these N sequences can be combined for each branch so as to reduce the influence of the cross-correlation due to the interference sequence, it is expected that the characteristics will be further improved regardless of the initial phase of the other station sequence. it can.

Hereinafter, five branch configuration methods studied in the embodiment of the present invention are shown ((a) to (d)). Here, (b) to (d) are methods in which all sequences (N) used for despreading are arranged in order of increasing cross-correlation for each phase shift, and configured from them.
FIG. 5 shows an example of the branch configuration shown in FIG. FIG. 5 is a schematic diagram showing an example of the branch configuration method (a). FIG. 5 shows a case where the sequence length N = 31 and the number of branches M = 3. Β in the figure is obtained by multiplying the sequence represented by the determinant in FIG. 1 by PN ₁ . (B) Thereafter, the series of elements to be synthesized changes.

  The branch configuration method (a) is a method in which a set (N pieces) of orthogonalized cyclic shift M sequences is substantially equally divided by the number of diversity branches and assigned to branches.

  The branch configuration method (b) is a method (similar to the allocation method of (a)) in which one sequence is allocated to each branch from the one with a small variance.

  The branch configuration method (c) is a method of allocating intensively to one branch from one having a small variance (a certain branch is a collection of those having a small variance, and a certain branch is a collection of those having a large variance).

  The branch configuration method (d) is a method in which the branch having the largest variance in (c) is not used.

The branch configuration methods (b) to (d) will be described more specifically. For example, each row (horizontal direction) in the OM matrix of FIG. 1 is called a row sequence. A variance value of values (hereinafter referred to as variance) is obtained.
If the branch is 3, the row having the smallest variance among the row series is assigned to the first branch, the row sequence having the next smallest variance is assigned to the second branch, and the row sequence having the next smallest variance is assigned to the third branch. In the branch configuration method (b), the row series having the next smallest variance is assigned to the first branch, and the assignment is repeated in the same manner.

In the branch configuration method (c), a plurality of row series having a small variance among each row series are intensively assigned to the first branch, and a plurality of row series having the next smallest variance are assigned to the second branch. is there.
Note that the branch configuration method (d) is such that the branch having the largest variance, that is, the third branch is not used after assigning each row series to the first to third branches in (c).

[Another system configuration]
The branch configuration method (a) can be realized with the system configuration of FIG. 3, but the branch configuration methods (b) to (d) are special and cannot be realized with the system configuration of FIG. A configuration of a code diversity communication system (another system) according to another embodiment of the present invention that realizes the branch configuration methods (b) to (d) will be described with reference to FIG. FIG. 6 is a block diagram showing the configuration of a code diversity communication system according to another embodiment of the present invention.

  As shown in FIG. 6, the other system is basically the same as that shown in FIG. 3, except that the receiver 20 includes a demodulator 21 and an integrator 22 for each row series (N). In addition, a rearrangement circuit 26 is provided between the integrator 22 and the multiplier 23.

  The rearrangement circuit 26 receives the correlation value of each row series by the demodulator 21 and the integrator 22, determines the magnitude of the variance based on the correlation value, and supplies the branch configuration method (b) to the multiplier 23 corresponding to the number of branches. ... (D) are assigned, the multiplier 23 multiplies the weighting coefficient, the synthesizer 24 synthesizes the data, and the data is output via the data determiner 25.

  Therefore, the rearrangement circuit 26 includes a determination unit that determines the magnitude of variance from the correlation value, and a unit that assigns a row series according to the algorithm of the branch configuration methods (b) to (d) and calculates a correlation value based on the combination. Have.

[simulation]
The characteristics of the proposed code diversity by the constant amplitude composite sequence and the conventional code diversity are compared. And the change of the characteristic by the difference in the branch structure method in a receiver is shown.

[Characteristics of code diversity]
The characteristics of the code diversity based on the constant amplitude composite sequence and the conventional code diversity will be compared. The simulation specifications are shown in FIG. 7, and the obtained BER (Bit Error Rate) characteristics are shown in FIG. FIG. 7 is a diagram showing simulation specifications, and FIG. 8 is a graph showing a comparison of the BER characteristics of the proposed method (embodiment of the present invention) and the conventional method. At this time, the number of branches in the conventional method is 5, and the branch configuration method in the proposed method is (a).
In addition, an estimation time of E [I _i ² ] + E [N _i ² ] of 1000 [bits] is provided for setting the weighting coefficient in the formula [Equation 9].

  From FIG. 8, the characteristics of the code diversity system (proposed system) according to the embodiment of the present invention are shown in SIR (Signal Interference Rate) = − 10 [dB] (the other station signal power is 10 times the local station signal power). It can be seen that the same characteristic tendency as that of the conventional code diversity method is exhibited, and interference can be suppressed.

Specifically, when the Eb / No is 10 [dB], the BER by the DS / SS method is 3.8 × 10 ⁻² , whereas the proposed method has a maximum of 8.5 × 10 ^−3. It was possible to improve.

  Further, when the characteristics of the number of branches are 2, 3, and 5, the effect of improving the characteristics increases as the number of branches increases. However, when the number of branches is 8, the characteristics are slightly worse. I understand that. Regarding this, it is known that there is an optimum value for the number of branches from the trade-off relationship between the SNR (Signal Noise Rate) after synthesis and the SIR of each branch.

  In this way, if the proposed scheme has a performance equivalent to that of the conventional code diversity, the proposed scheme that can be transmitted by one M-sequence or the like as in the conventional DS / SS scheme is more suitable for hardware. There is an advantage that the circuit scale can be reduced in view of the configuration.

[Characteristic change by branch configuration method]
Here, the difference in characteristics due to the difference in the branch configuration method in the receiver is compared. The branch configuration patterns to be verified are (b) to (d), and the simulation environment is the same as in FIG.

  However, the number of branches to be targeted is 3 and 5 branches in (b) to (d). Moreover, the result by (b)-(d) is shown in FIG. 9, FIG. FIG. 9 is a graph showing the characteristics of the branch configuration methods (b) to (d) when the number of branches is 3, and FIG. 10 is a graph showing the characteristics of the branch configuration methods (b) to (d) when the number of branches is 5. is there.

  From the results of FIGS. 9 and 10, it can be seen that the branch configuration method (c) in which concentrated distribution is assigned to a branch from a sequence with small variance has the best BER characteristics. The reason for this is that the weighting factor of a branch configured with a sequence having a small cross-correlation variance with an interference sequence is large, and the weighting factor of a branch configured with a sequence having a large cross-correlation variance is small. The branch for which the correct determination is made is emphasized well, and it is considered that errors are reduced.

  On the other hand, even in the same branch configuration method, the reason why the method of not using the worst branch as shown in (d) deteriorates compared to (c) is considered as follows. First, in the method (d), since one of the branches is discarded, the number of series used in each branch is smaller than that in (c). Therefore, since the signal component in each branch is larger in (c), it is considered that the characteristics of (d) are degraded than in (c) (the contribution of the signal component is somewhat greater than the degradation of the interference interference component). It ’s better to leave the branch).

  Thus, in the proposed method, the code sequence is not changed on the transmitter side as in the conventional code diversity method, but the characteristics can be improved by changing the configuration of the branch in the receiver. is there.

[Summary]
In the embodiment of the present invention, a code diversity method using a composite sequence having a constant amplitude has been proposed, and its principle and configuration method have been described. In addition, it was confirmed that the proposed method has an interference suppression performance equivalent to that of the conventional code diversity method, and it was confirmed that the interference suppression performance can be further improved depending on the branch configuration method.

  For these reasons, the proposed method may be a conventional DS / SS system with respect to the transmitter configuration, and the receiver side changes the receiver branch configuration and performs diversity reception in order to improve interference suppression performance. There is an advantage that only the ingenuity is required.

[Effect of the embodiment]
According to the code diversity communication method and the code diversity communication system according to the embodiment of the present invention, on the transmitter side, the sum of all orthogonal cyclic shift M sequences is multiplied by the spreading PN sequence to obtain a constant amplitude of ± 1. And the data is multiplied by the data and transmitted. On the receiving side, the spreading PN sequence is correlated with a partial sequence of the orthogonalized cyclic shift M sequence, the weighting factor is obtained from the variance of the correlation, and the weighting factor is calculated. Multiplication is added to obtain the addition of each diversity branch, so that the burden on the output stage of the transmitter is not increased, and the performance is not degraded by the amplitude limiter of the receiver, so that the interference suppression performance can be improved. effective.

  Moreover, according to the code diversity communication method and the code diversity communication system according to another embodiment of the present invention, by devising the branch configuration method and adopting the branch configuration methods (b) to (d), Even if there is a difference in the initial phase between the own station series and the other station series, there is an effect that the characteristics can be improved.

  INDUSTRIAL APPLICABILITY The present invention is suitable for a code diversity communication method and a code diversity communication system that can improve interference suppression performance by adopting a constant amplitude composite sequence configuration that keeps the amplitude of a transmission signal constant.

It is explanatory drawing which shows the sum of the set of the orthogonalization cyclic shift M series in the code diversity system which concerns on embodiment of this invention. It is explanatory drawing which shows the sum of three signal series in the code diversity system which concerns on embodiment of this invention. 1 is a configuration block diagram of a code diversity communication system according to an embodiment of the present invention. It is a graph of the comparison of the chip value at the time of transmission by the conventional code diversity and the proposed code diversity (according to the embodiment of the present invention). It is the schematic which shows an example of the branch structure method (a). It is a block diagram of the configuration of a code diversity communication system according to another embodiment of the present invention. It is a figure which shows a simulation specification. It is a graph which shows the comparison of the BER characteristic of the proposed method (embodiment of this invention) and the conventional method. It is a graph which shows the characteristic of the branch structure method (b)-(d) in the number of branches 3. It is a graph which shows the characteristic of the branch structure method (b)-(d) in the number of branches 5. It is explanatory drawing which shows the sum of the three signal series in the conventional code | symbol diversity system.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 10 ... Transmitter, 11 ... Modulator, 20 ... Receiver, 21 ... Demodulator, 22 ... Integrator, 23 ... Multiplier, 24 ... Synthesizer, 25 ... Data judgment device, 26 ... Rearrangement circuit

Claims (13)

A code diversity communication method in a direct spread spread spectrum system,
The transmitter side generates a sequence having a constant amplitude by multiplying the sum of the set of orthogonalized cyclic shift M sequences by a spreading code sequence, and transmits the generated sequence by multiplying the generated sequence by transmission data. Code diversity communication method.   The transmission power is normalized by dividing the value of each chip of the orthogonalized cyclic shift M sequence by the value of the combined amplitude of all sequences of the orthogonalized cyclic shift M sequence. Code diversity communication method. A code diversity communication method in a direct spread spread spectrum system,
3. A receiving side that receives a signal transmitted by the code diversity communication method according to claim 1 or 2 outputs a correlation of a branch by despreading with a sum sequence of a subset of a spreading code sequence and an orthogonal cyclic shift M sequence. A code diversity communication method characterized in that a corresponding weighting factor is calculated from the variance of the correlation and demodulated by the sum of the despread signal multiplied by the corresponding weighting factor.   4. The code diversity communication method according to claim 3, wherein a sum of variance of cross-correlation due to interference and variance of noise is estimated from a correlation output of the branch, and a weighting factor is calculated using a code length.   5. The code diversity communication method according to claim 3, wherein a set of orthogonalized cyclic shift M sequences is divided by the number of diversity branches and assigned to the branches.   5. The code diversity communication method according to claim 3, wherein a subset of orthogonal cyclic shift M-sequences is assigned to each branch in order from one having a small correlation variance.   5. The code diversity communication method according to claim 3, wherein a subset of orthogonalized cyclic shift M-sequences is intensively assigned to one branch from one having a small correlation variance.   8. The code diversity communication method according to claim 7, wherein the branch having the largest correlation variance is not used. A code diversity communication system in a direct spread spectrum spread system,
A transmission having a modulator that multiplies the sum of sets of orthogonalized cyclic shift M sequences by a spreading code sequence to generate a sequence having a constant amplitude, and multiplies the generated sequence by transmission data to transmit the sequence. Machine. A code diversity communication system in a direct spread spectrum spread system,
A demodulator for despreading a signal transmitted from the transmitter according to claim 10 by a sum sequence of a subset of a spreading code sequence and an orthogonal cyclic shift M sequence, and integrating the despread signal A receiver comprising: an integrator that outputs a correlation; a multiplier that multiplies the despread signal by a weighting factor corresponding to a variance of the correlation; and a combiner that combines the multiplied signals.   11. A rearrangement circuit is provided between the integrator and the multiplier, and the correlation output from the integrator is assigned to each branch one by one in order from the one having a small correlation variance. Receiving machine.   11. The rearrangement circuit for allocating a correlation output from the integrator intensively to one branch from the one having a small correlation variance is provided between the integrator and the multiplier. Receiving machine.   The receiver according to claim 12, wherein the rearrangement circuit does not use a branch having the largest correlation variance.

Images