JP2002108465A - Temperature detection circuit, heating protection circuit and various electronic equipment including these circuits - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a temperature detection circuit having a small occupancy area and small power consumption, a heating protection circuit and various pieces of electronic equipment including these circuits. SOLUTION: This heating protection circuit comprises a circuit obtained by serially connecting a diode 61 having temperature dependence and a 1st current source 62 provided between a 1st voltage power supply Vdd and a 2nd voltage power supply Vss, a circuit obtained by serially connecting a 1st resistor 63 and a 2nd current source 64 provided between a 1st voltage power supply Vdd and a 2nd voltage power supply vss, and a comparator 65 for making the voltage of the junction of the diode 61 and the current source 62 to be a 1st input and the voltage of the junction of the resistor 63 and the current source 64 to be a 2nd input and outputting a comparison result signal. The 2nd current source 64 is configured by connecting MOS transistors 641, 642 and 643 as illustrated.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a temperature detecting circuit, a heating protection circuit, and an I-type circuit incorporating these circuits.
C, portable electronic devices such as mobile phones, voltage regulators, DC-DC converters, battery packs,
Related to in-vehicle electrical components and various home appliances.

[0002]

2. Description of the Related Art FIG. 9 is a diagram showing a circuit configuration of a conventional general voltage regulator (for example, see Japanese Patent Application Laid-Open No. 8-272461). In the figure, 71 is a reference voltage source, 72 is an error amplifier circuit (differential amplifier circuit), 7
3 is an output transistor, 74 is an output terminal, and R1 and R2 are resistors. The voltage Vref output from the reference voltage source 71
And the voltage detected from the output transistor 73 and the resistors R1 and R2 are compared by the error amplifier circuit 72, and the output transistor 73 is controlled based on the comparison result to stabilize the output voltage Vout to the output terminal 74. ing.

In the above-described voltage regulator configuration, when a large current flows through the output transistor 73 or when the voltage between the source and drain of the output transistor 73 is increased, the power consumption of the output transistor 73 increases and heat is generated. The voltage
If the temperature of the regulator becomes too high, the IC may be destroyed. Therefore, the temperature of the IC is detected,
It is necessary to protect the IC from overheating.

[0004] As a voltage regulator,
A device using a band gap circuit using a bipolar transistor is known (for example, Japanese Patent Application Laid-Open No. 7-136).
No. 43, FIG. 7 and Japanese Utility Model Publication No. 7-51620, No. 3
See figure). If the voltage regulator is composed of bipolar transistors, the bipolar transistor and the diode are structurally similar, so not only the advantage that the manufacturing process can be the same process as the diode for temperature detection, but also the accuracy of the detected temperature And there is an advantage that variation can be reduced.

[0005]

However, when a voltage regulator is formed using bipolar transistors as described above, there is a problem that the occupied area is large and the power consumption is large.

The object of the present invention is to solve these problems. Specifically, the inventions according to claims 1 to 5 have a small occupied area and a low temperature consumption temperature detection circuit, and in particular, the invention according to claim 3 can precisely adjust the resistance value. The invention described in claim 4 is capable of adjusting the size of the diode. The invention described in claim 5 is directed to providing a temperature detection circuit capable of simplifying a constant current source. I have.

According to the present invention, a heating protection circuit occupying a small area and consuming less power is provided.
In particular, the invention according to claim 8 is capable of adjusting the resistance value with high accuracy, the invention according to claim 10 is capable of adjusting the size of the diode, and the invention according to claim 11 is capable of adjusting the constant current. It is an object of the invention according to claims 12 and 13 to provide a temperature detection circuit capable of efficiently and accurately detecting a temperature.

A further object of the present invention is to provide various devices incorporating a temperature detecting circuit or a heating protection circuit which occupies a small area and consumes less power.

[0009]

According to the present invention, in order to achieve the above object, a MOS transistor is used in place of a conventional band gap circuit using a bipolar transistor. The specific structure of each claim is shown below.

(1) A temperature detecting circuit according to claim 1, wherein a diode having a temperature dependency provided between the first voltage power supply and the second voltage power supply and the first current source are connected in series. Circuit, a circuit in which a first resistor and a second current source provided between a first voltage power source and a second voltage power source are connected in series, and a connection between the diode and the first current source. A point voltage is used as a first input, a voltage at a connection point between a first resistor and the second current source is used as a second input, and both inputs are compared in magnitude. A comparison result signal is output as a temperature detection signal. A temperature detection circuit characterized in that the second current source has a MOS transistor circuit configuration having no temperature dependence.

(2) A temperature detecting circuit according to a second aspect, wherein the second current source according to the first aspect has a drain connected to the first resistor and a source connected to the first power source. MO
A drain connected to the S transistor and the first power supply,
A depletion type MOS transistor having a gate and a source connected to the gate of the first MOS transistor; a drain connected to the gate of the first MOS transistor and the gate and source of the depletion type MOS transistor; A second MOS transistor having a gate connected to the source and a source connected to the second power supply, wherein the depressin MOS transistor and the second MOS transistor have a temperature dependency in the second current source; It is characterized by having a conductivity coefficient that does not exist.

(3) The temperature detecting circuit according to claim 3, further comprising the step of processing the first resistor or the second resistor to an arbitrary one of a plurality of resistors provided in advance by a laser trimming technique. It is specified as the one with the adjusted resistance value. By adjusting the resistance value and controlling the reference voltage, it is possible to control the detected temperature and improve the accuracy.

(4) In the temperature detecting circuit according to a fourth aspect, the diode is further specified to be a diode whose size has been adjusted by a laser trimming technique. As a result, the forward bias of the diode can be controlled, and the detection temperature can be controlled and the accuracy can be improved.

(5) The invention according to claim 5 further specifies that the first current source is constituted by a depletion type transistor connected in saturation. Thus, a constant current source having no temperature dependency can be easily configured.

(6) A heating protection circuit according to claim 6, wherein a diode having a temperature dependency provided between the first voltage power supply and the second voltage power supply and the first current source are connected in series. Circuit, a circuit in which a first resistor and a second current source provided between a first voltage power source and a second voltage power source are connected in series, and a connection between the diode and the first current source. A point voltage is used as a first input, and a voltage at a connection point between a first resistor and a second current source is used as a second input, and the two inputs are compared in magnitude. , And the second current source has a MOS transistor circuit configuration having no temperature dependency. The invention according to claim 7 is characterized in that the comparator has substantially hysteresis.

(7) The heating protection circuit according to claim 8, wherein the second current source has a drain connected to a first resistor,
A first MOS transistor having a source connected to the first power source, a depletion type MOS transistor having a drain connected to the first power source, and a gate and a source connected to the gate of the first MOS transistor; M of 1
The gate of the OS transistor and the depletion type M
A drain connected to the gate and source of the OS transistor, a gate connected to the source of the first MOS transistor, and a second MOS transistor having a source connected to the second power supply;
The transistor and the second MOS transistor are characterized in that the second current source has a conductivity coefficient that does not cause temperature dependency.

(8) In the heating protection circuit according to the ninth aspect, the first resistor or the second resistor is further processed by a laser trimming method for an arbitrary one of a plurality of resistors provided in advance. It is specified that the resistance value is adjusted. As a result, it is possible to control the detected temperature and improve the accuracy.

(9) The overheat protection circuit according to claim 10 is
Further, the diode is specified as one whose size is adjusted by a laser trimming technique. As a result, the forward bias of the diode can be controlled, and the detection temperature can be controlled and the accuracy can be improved.

(10) In the overheat protection circuit according to the eleventh aspect, the first current source is further specified to include a depletion-type transistor connected in saturation. Thus, a constant current source having no temperature dependency can be easily configured.

(11) In the overheat protection circuit according to the twelfth aspect, the diode is arranged near an output transistor for supplying a power supply voltage to the IC.
3. The overheat protection circuit according to claim 3, further comprising an output transistor disposed so as to surround the diode.

(12) The invention according to the fourteenth to nineteenth aspects is directed to an IC circuit, a portable electronic device, a voltage regulator, a DC-DC converter, and a battery pack incorporating the above-described temperature detection circuit or overheat protection circuit. ,
It is a vehicle electrical component.

[0022]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a diagram for explaining an embodiment in which a heating protection circuit according to the present invention is applied to a voltage regulator. In the figure, 1 is a reference voltage source, 2 is an error amplifier circuit (differential amplifier circuit), 3 is an output transistor (output driver), and 4 and 5 are resistors.

A reference voltage source 1, an error amplifier (differential amplifier) 2, an output transistor 3, and resistors 4 and 5 are a reference voltage source 21, an error amplifier (differential amplifier) 22 in FIG. , The output transistor 23, the output terminal 24, and the resistors R1 and R2. Reference numeral 6 denotes a heating protection circuit according to the present invention for detecting a temperature rise and turning off the output transistor 3 to prevent the output transistor 3 and ICs connected thereto from being destroyed by temperature.

FIG. 2 is a structural example of the heating protection circuit 6 in FIG. In the figure, 61 is a diode having temperature-dependent characteristics, 62 is a constant current source 1, 63 is a resistor, 64
Is a constant current source 2, 65 is a comparator, 66 is a p-channel MOS transistor, Vdd is a high voltage side power supply voltage, V
ss is the power supply voltage (or ground voltage) on the low voltage side.

The voltage (VF) at the connection point between the diode 61 and the constant current source 1 (62) is input to the (+) input terminal of the comparator 65, and the connection point B between the resistor 63 and the constant current source 2 (64)
(Vref) is input to the (−) input terminal of the comparator 65, and the output of the comparator 65 is
The voltage is applied to the gate of the S transistor 66, and the p-channel M
The output from the drain of the OS transistor 66 is applied to the gate of the output transistor 3 to control on / off of the output transistor 3. That is, when the voltage (VF) is lower than the voltage (Vref), the output transistor 3 is turned on. When the temperature rises, the output of the comparator 65 is inverted by the temperature-dependent characteristic of the diode 61 to output the output transistor (driver transistor). Turn off 3 to suppress heat generation.

The operation of the heating protection circuit 6 will be described in more detail with reference to FIG. A constant current flows through the diode 61 by the constant current source 1 (62). The constant current source 1 (62) is preferably constituted by a depletion type MOS transistor which is connected in saturation as shown in FIG. 3 as an embodiment. Diode 6 when a constant current flows
The forward bias temperature dependence of 1 is approximately-
2 mV / ° C. The voltage (VF) on the cathode side of the diode 61 is a voltage obtained by subtracting a forward bias voltage depending on the temperature of the diode 61 from the power supply Vdd. In a normal state, the voltage (VF) <the reference voltage (Vref), and the output of the output transistor 3
Turns on, raising the temperature of the IC.

As the temperature of the IC rises, the voltage (VF) changes at a rate of about -2 mV / .degree.
5 becomes equal to the other input voltage (Vref), the output of the comparator 65 is inverted, and the output transistor 3 is turned off. As a result, the output transistor 3 does not generate heat, and the temperature of the IC decreases. When the voltage (VF) becomes lower than the reference voltage (Vref), the comparator 6
5 is inverted and the output transistor 3 is turned on.

Next, a specific example of a circuit comprising a resistor 63 for generating a reference voltage (Vref) and a constant current circuit source 64 constituting the overheat protection circuit 6 in this embodiment will be described. In order to solve the problem in the case where the circuit for generating the reference voltage (Vref) is configured using bipolar transistors, that is, the problem that the occupied area is large and the power consumption is large, in this embodiment, the reference is made by using the MOS technology. A circuit for generating a voltage (Vref) is configured.

FIG. 4 shows an example in which the heating protection circuit 6 of FIG. 2 is constructed by using the MOS technology, and specifically shows a specific example of the constant current circuit 64. In the figure, an enhancement type n-channel MOS transistor 641, a depressin type n-channel MOS transistor 642, an enhancement type n-channel MOS transistor 643, and a resistor 644 constitute the constant current circuit 64 of FIG. Reference numerals 3, 61, 62, 63, 65, and 66
Is the same as FIG.

The temperature dependence can be eliminated by adjusting the conductivity coefficients of the depressin type n-channel MOS transistor 642 and the enhancement type n-channel MOS transistor 643. Hereinafter, the reason will be described (see JP-A-8-30345). In FIG. 4, when the power supply voltage is larger than the absolute sum of the threshold voltages of the transistors 642, 643, and 641, each transistor operates in a saturation region higher than the drain-source voltage. The conductivity coefficient of the transistor 642 is K1,
Assuming that the threshold value is VT1 and the current flowing through the source of the transistor 642 is I1, I1 = K1 × | VT1 | ² (1) The conductivity coefficient of the transistor 643 is K2. Threshold VT
2. The current flowing through the source of the transistor 643 is represented by I2,
The voltage at the connection point between the transistor 641 and the resistor 644 is Vo
ut1, and the voltage at the connection point between the transistor 642 and the transistor 643 is Vout2. I2 = K2 × (Vout1-VT2) ² (2) where the conductivity coefficient of the transistor 641 is K3 and the threshold is Value VT
3. Assuming that the current flowing through the source of the transistor 641 is I3, I3 = K3 × (Vout2-Vout1-VT3) ² ... (3) Assuming that the current flowing through the resistor 644 is IR, IR = Vout1 / R I1 From I = I2, Vout1 = (K1 / K2) ^1/2 × | VT1 | + VT2 (4) The threshold values of the transistors 642, 643, and 641 at room temperature are VT10, VT20, and VT30. The amount of change per 1 ° C. is △ VT1, △ VT2, respectively.
Assuming that △ VT3, VT1 = VT10 + △ T × △ VT1 (5) VT2 = VT20 + △ T × △ VT2 (6) VT3 = VT30 + △ T × △ VT3 Equation (7) Let K10, K20, and K30 be the conductivity coefficients of transistors 642, 643, and 641 at room temperature, and set the conductivity coefficient to 1 ° C.
Assuming that the change rates per hit are △ K1, △ K2, and △ K3, respectively, K1 = (1 + △ T × △ K1) × K10 (8) K2 = (1 + △ T × ΔK2) × K20 (Equation (9)) K3 = (1 + ΔT × △ K3) × K30 (Equation (10)) Let the resistance value be R0,
Assuming that the amount of change per ° C is ΔR, the resistance value R of the resistor 644 is given by: R = R0 + ΔT × ΔR Equation (11) Equation (4) ) And (9). On the same substrate, △ K1 = △ K2, so Vout1 = (K10 / K20) ^1/2 × | VT10 | + VT20 + ΔT ((K10 / K20) ^1/2 ) × | △ VT1 | + △ VT2) Equation (12) From this, the voltage Vout1 and the temperature change rate thereof are determined by the threshold values and the conductivity coefficient of the transistors 642, 643, 641. Since the conductivity coefficients K1 and K2 can be adjusted by the channel length and channel width of the transistor, the amount of change in the voltage Vout1 per 1 ° C. can be controlled, and K10 = K2
If it is set to 0, the temperature change rate of Vout1 will be zero because it is canceled by the depletion type MOS transistor and the enhancement MOS transistor. As described above, it can be understood that the temperature dependence can be eliminated by adjusting the conductivity coefficients of the depressin n-channel MOS transistor 642 and the enhancement n-channel MOS transistor 643.

As shown in FIG. 4, a depressin type n-channel MOS transistor 642 whose gate and source are connected operates as a constant current source, and a depressin type n-channel MOS transistor 642 and an enhancement type n-channel MOS transistor 643 are connected. A constant current flows. Therefore, the drain voltage and the gate voltage of the enhancement-type n-channel MOS transistor 643 are uniquely determined values.

The drain of the enhancement n-channel MOS transistor 643 is connected to the gate of the enhancement n-channel MOS transistor 641, and the gate of the enhancement n-channel MOS transistor 643 is connected to the enhancement n-channel MOS transistor 6.
Since they are connected to the sources of the N-channel MOS transistors 41, respectively, the gate voltage and the source voltage (point A) of the enhancement type n-channel MOS transistor 641 also have univocally determined values.

Enhancement type n-channel MOS transistor
The current value passing through the transistor 641 is represented by I₆₄₁Of the resistor 63
Set the resistance value to R₆₃Then the enhancement n-channel
Voltage of drain (point B) of MOS transistor 641
(VB) is VB = Vdd-I ₆₄₁× R₆₃Becomes here
By changing the resistance value of the resistor 644, I₆₄₁Change
Change the resistance values of the resistors 63 and 644
Arbitrarily adjust the reference voltage (VB = Vref) at point B
can do.

As described above, the reference potential at point B (VB = V
ref) can be adjusted by changing the resistance values of the resistors 63 and 644. As a method of changing the resistance values of the resistors 63 and 644, for example, a plurality of resistors made of polysilicon or a metal thin film are provided. In advance,
There is a method of selectively trimming them by laser trimming, which makes it possible to set the detected temperature at which the output transistor 3 is turned off to a desired value.

The temperature at which the output transistor 3 is turned off can be changed by controlling the size of the diode 61 to change the forward bias of the diode. In this case, it is needless to say that the laser trimming method as described above can be applied.

It is desirable that the temperature detecting section (the diode 61 having a temperature dependency) constituting the heating protection circuit 6 be provided near the output transistor 3 efficiently and accurately. FIG. 5 is a diagram showing an embodiment of an in-chip layout of a heating protection circuit, in which output transistors (drive transistors and power transistors) are arranged in the periphery, and a temperature detection unit (a temperature-dependent diode) 6
This is an example in which 1) is provided.

In the above embodiment, the voltage (V
While F) is lower than the voltage (Vref), the comparator 65
The output transistor 3 is turned on by the output of the comparator 65. However, when the temperature rises and the voltage (VF) becomes equal to the voltage (Vref), the output of the comparator 65 turns off the output transistor 3 and the power consumption is eliminated, thereby lowering the temperature. Although the output transistor 3 is turned on again by the output of the comparator 65 when the voltage (VF) becomes lower than the voltage (Vref) due to a decrease in the temperature, the comparator 65 has a circuit having no such hysteresis characteristics. In such a case, the following problem occurs.

The temperature rises and the output transistor 3 turns off → the temperature drops → the output transistor 3 turns on → the temperature rises due to power consumption → the output transistor 3 turns off → the temperature falls →. A phenomenon called thermal oscillation occurs. Further, the voltage (VF) is changed to the voltage (V
In the vicinity of (ref), an oscillation state in which the output transistor 3 is repeatedly turned on and off may occur due to noise or the like.

In order to eliminate such an oscillation state and to stabilize the output of the comparator 65, the comparator 6
The hysteresis may be provided to the magnitude determination level of the two input voltages (VF) and (Vref).

FIG. 6 is a diagram showing an example of the comparator 65 having hysteresis. In the figure, 651,
652 and 655 are n-channel MOS transistors;
Reference numerals 3, 654, 656, and 657 denote p-channel MOS transistors. The n-channel MOS transistor 651 and the n-channel MOS transistor 652 have the same gate size (gate width / gate length).
Transistor 653 and p-channel MOS transistor 6
Gate size of 54 (gate width / gate length (W / L))
And the p-channel MOS transistors 656 and p
The gate size (gate width / gate length) of the channel MOS transistor 657 is made the same. Also, the p-channel M
The current gain β of the OS transistors 653 and 654 is set smaller than the current gain β of the p-channel MOS transistors 656 and 657, or the p-channel MOS transistor 6
53, 654, 656 current amplification factor β
The current amplification factor β of the S transistor 657 is smaller than β.

In this configuration, the temperature is low and the voltage (V) applied to the gate of n-channel MOS transistor 652 is low.
F: While the input terminal (+) is lower than the voltage (Vref: input terminal (-)) applied to the gate of the n-channel MOS transistor 651, the n-channel MOS transistor 651 and the p-channel MOS transistors 653, 6
56 is on, n-channel MOS transistor 652, p
The channel MOS transistors 654 and 657 are turned off. At this time, the drain voltage of the n-channel MOS transistor 652 is turned on via the inverter and the output transistor 3 formed of a p-channel MOS transistor.

When the temperature rises and the gate voltage (VF: input terminal (+)) of the n-channel MOS transistor 652 rises and becomes equal to the gate voltage of the n-channel MOS transistor 651, the n-channel MOS transistor 652 turns on. Turns on, and p-channel MOS transistor 65
4,657 on, n-channel MOS transistor 65
1. Turn off the p-channel MOS transistors 653 and 654. When the n-channel MOS transistor 652 is turned on, its drain voltage is reduced, and the output transistor 3 formed of a p-channel MOS transistor is turned off via the inverter.

At this time, as described above, the p-channel M
The current amplification factor β of the OS transistors 653 and 654 is made smaller than the current amplification factor β of the p-channel MOS transistors 656 and 657, or the p-channel MOS transistor 6
53, 654, 656 current amplification factor β
By making the current amplification factor β smaller than that of the S transistor 657, after the gate voltage (VF: input terminal (+)) of the n-channel MOS transistor 652 once becomes high, the voltage of the input terminal (+) Even if the voltage decreases, p-channel MOS transistor 652 keeps on, and output transistor 3 formed of a p-channel MOS transistor
Keep off. In this way, by providing the determination level between the input terminal (+) and the input terminal (-) with hysteresis, the operation of the output transistor 3 composed of a p-channel MOS transistor can be stabilized.

In the above example, when the voltage (VF) at the input terminal (+) rises to the same voltage as the voltage (Vref) at the input terminal (-), the output is switched and the p-channel MO is switched.
Although the output transistor 3 composed of the S transistor is turned off, the voltage difference between the input terminal (+) and the input terminal (-) when the output of the comparator 65 is switched can be freely set.

For example, in FIG. 6, the channel size W / L (width / length) of the gate of the n-channel MOS transistor 651 and the gate of the n-channel MOS transistor 652 constituting the comparator is made different so that the input terminal (+) The voltage (VF) is set to V1 and the potential (Vre) of the input terminal (-).
When f) is V2, the output of the comparator 65 can be switched when V2-V1 reaches a predetermined value.

As an example, when the voltage (Vref) at the input terminal (-) is 3 V and the predetermined value is 0.2 V, the voltage (VF) at the input terminal (+) increases with temperature. And rises to 2.8V when the comparator 65
Is switched, and the output transistor 3 is turned off. With this configuration, the voltage difference between the two input terminals that switch off the output transistor 3 can be made a desired value, and the degree of freedom in designing the heating protection circuit of the present invention can be given.

Next, another configuration for providing hysteresis will be described. FIG. 7 is a circuit configuration diagram for making the output of the comparator 65 substantially have hysteresis.

In the same figure, in the normal state, VF <Vr
ef, the p-channel MOS transistor M1 (corresponding to the p-channel MOS transistor 66 in FIGS. 2 and 4),
The n-channel MOS transistor M7 is off (Vref at this time is set to Vref1). Temperature rises and V
When F> Vref, the p-channel MOS transistor M1 and the n-channel MOS transistor M7 are turned on. When the p-channel MOS transistor M1 is turned on, the p-channel MOS transistor M0 (FIGS. 2 and 4)
Output transistor 3) is turned off.

On the other hand, n-channel MOS transistor M7
Is turned on, the potential at the point C becomes the GND (ground) potential, and the resistance 1, the n-channel MOS transistor M
6, since the current flowing through the path of the resistor 2-1 increases (the potential at the point A is constant), the potential of Vref decreases (assuming that the Vref potential at this time is Vref2, Vref1).
> Vref2).

As the temperature decreases, VF also decreases,
In order for the comparator 65 to invert again, VF becomes Vref1.
Not to Vref2. As a result, hysteresis can be provided.

FIG. 2 shows that the diode 61 and the resistor 63 are connected to the high-voltage power supply voltage Vdd side and the low-voltage power supply voltage Vs
In this example, the constant current sources 62 and 64 are provided on the s side. On the other hand, as shown in FIG. Needless to say, a configuration in which a diode 61 ′ and a resistor 63 ′ are provided on the Vss side may be used. In this case, the constant current source 64 'may have the same configuration (the configuration in which Vdd and Vss are reversed) as the constant current source 64 shown in FIG.

In the above embodiments, the heating protection circuit has been described. However, this configuration is useful not only for protection against heating by controlling the ON / OFF of the output transistor, but also as a temperature detection circuit for simply detecting the temperature of an IC or the like. Needless to say, there is.

The heating protection circuit (or simply as a temperature detection circuit) having the configuration of the above-described embodiment has a small occupied area,
In addition, since power consumption is small and temperature detection efficiency and detection accuracy can be improved, various devices, for example, various mobile devices such as mobile phones, DC-DC converters, various IC circuits, and voltage regulators. , Battery packs, and various in-vehicle electrical components.

[0054]

The present invention has the following effects. According to the first and sixth aspects of the present invention, since the reference voltage circuit is formed by using the MOS technology, a temperature detection circuit and a heat protection circuit having a small occupation area and a small power consumption can be realized by using a conventional process. Can be.

According to the second and eighth aspects of the present invention, M
A constant current circuit having no temperature dependency can be realized by using the OS transistor. According to the third, fourth, ninth, and tenth aspects of the present invention, it is possible to control the detected temperature and improve the accuracy by the laser trimming technique. Further, the detection temperature can be determined in a post-process.

According to the fifth and eleventh aspects of the present invention, a constant current source having no temperature dependency can be realized with a simple configuration. According to the seventh aspect of the present invention, by providing the comparator with substantially hysteresis, it is possible to realize a stable overheat protection circuit which prevents thermal oscillation and is stable. According to the twelfth and thirteenth aspects, the temperature can be detected efficiently and accurately. According to the fourteenth to nineteenth aspects of the present invention, it is possible to realize various devices having a small occupied area, a small power consumption, an accurate control of a detected temperature and an overheat protection function.

[Brief description of the drawings]

FIG. 1 is a diagram for explaining an embodiment when a heating protection circuit according to the present invention is applied to a voltage regulator.

FIG. 2 is a configuration example of a heating protection circuit 6 in FIG.

FIG. 3 is an embodiment of a constant current source composed of a depletion type MOS transistor connected in saturation.

FIG. 4 is an example in which the heating protection circuit 6 of FIG. 2 is configured using MOS technology.

FIG. 5 is a diagram showing one embodiment of an in-chip layout of a heating protection circuit.

FIG. 6 is a diagram illustrating an example of a comparator having hysteresis.

FIG. 7 is a diagram showing another embodiment in which a comparator has substantially hysteresis.

FIG. 8 shows a high power supply voltage Vdd and a low power supply voltage V
FIG. 3 is a diagram illustrating a configuration in which ss is opposite to that in FIG. 2.

FIG. 9 is a diagram showing a circuit configuration of a conventional general voltage regulator.

[Explanation of symbols]

1, 21: Reference voltage source, 2, 22: Error amplifier circuit (differential amplifier circuit), 3, 23: Output transistor (output driver), 4, 5, R1, R2: Resistance, 6: Heat protection circuit, 61 , 61 ′: diode having temperature dependency, 62, 62 ′: constant current source, 63, 63 ′: resistor, 64, 64 ′: constant current source, 641: enhancement type n-channel MOS transistor, 642: depressin type n-channel MOS transistor, 643: enhancement type n-channel MOS transistor, 644: resistance, 65: comparator, 651, 652, 655: n-channel MOS transistor, 653, 654, 656, 657: p-channel MOS transistor, 66: p-channel MOS transistor, Vdd: power supply voltage on the high voltage side, Vss: power supply voltage on the low voltage side Power supply voltage (or ground voltage).

──────────────────────────────────────────────────続 き Continued on the front page (51) Int.Cl. ⁷ Identification code FI Theme coat ゛ (Reference) H03K 17/687 F-term (Reference) 5F038 AV03 BB08 BH02 BH04 BH05 BH07 BH16 DF07 EZ20 5H430 BB01 BB05 BB09 BB11 EE04 FF02 FF02 FF13 GG09 HH03 HH07 LA10 LA26 5J055 AX12 AX15 AX32 AX44 AX64 BX16 CX19 DX14 DX22 EX02 EY01 EY02 EY12 EY21 EY23 EY24 EZ03 EZ07 EZ08 EZ10 EZ51 FX06 FX18 FX33 GX01 GX08

Claims (19)

[Claims] A circuit in which a temperature-dependent diode provided between a first voltage power supply and a second voltage power supply and a first current source are connected in series; A circuit in which a first resistor and a second current source provided in series with a second voltage power supply are connected in series; and a voltage at a connection point between the diode and the first current source is supplied to a first input; A temperature detection circuit for comparing the magnitude of both inputs with a voltage at a connection point between the first resistor and the second current source as a second input, and outputting a comparison result signal as a temperature detection signal; A temperature detection circuit, wherein the second current source has a MOS transistor circuit configuration having no temperature dependency. 2. The second current source has a drain connected to the first resistor, a first MOS transistor having a source connected to the first power supply, and a drain connected to the first power supply. And the gate and the source are connected to the first MOS.
A depletion-type MOS transistor connected to the gate of the transistor; a drain connected to the gate of the first MOS transistor and a gate and a source of the depletion-type MOS transistor;
A second MOS transistor having a gate connected to the source of the OS transistor and a source connected to the second power supply, wherein the depressin MOS transistor and the second MOS transistor are connected to the second current source; A temperature detecting circuit having a conductivity coefficient such that temperature dependency does not occur in the temperature detecting circuit. 3. The resistance value of the first resistor or the second resistor is adjusted by processing an arbitrary resistor among a plurality of resistors provided in advance by a laser trimming method. The temperature detection circuit according to claim 1 or 2, wherein: 4. The temperature detection circuit according to claim 1, wherein the size of the diode is adjusted by a laser trimming technique. 5. The temperature detection circuit according to claim 1, wherein said first current source is constituted by a depletion-type transistor connected in saturation. 6. A circuit in which a temperature-dependent diode provided between a first voltage power supply and a second voltage power supply and a first current source are connected in series; A circuit in which a first resistor and a second current source provided in series with a second voltage power supply are connected in series; and a voltage at a connection point between the diode and the first current source is supplied to a first input; A comparator that compares the magnitude of both inputs with a voltage at a connection point between the first resistor and the second current source as a second input, and controls on / off of an output transistor based on a comparison result signal; An overheat protection circuit, wherein the second current source has a MOS transistor circuit configuration having no temperature dependency. 7. The overheat protection circuit according to claim 6, wherein said comparator has substantially hysteresis. 8. The second current source has a drain connected to the first resistor, a first MOS transistor having a source connected to the first power supply, and a drain connected to the first power supply. And the gate and the source are connected to the first MOS.
A depletion type MOS transistor connected to the gate of the transistor, a drain connected to the gate of the first MOS transistor, a gate and a source of the depletion type MOS transistor, and a gate connected to a source of the first MOS transistor And a second MOS transistor having a source connected to the second power supply, and the depressin type MOS transistor and the second MOS transistor are connected to the second MOS transistor.
8. The overheat protection circuit according to claim 6, wherein said current source has a conductivity coefficient such that temperature dependency does not occur. 9. The resistance value of the first resistor or the second resistor is adjusted by processing an arbitrary resistor among a plurality of parallel-connected resistors provided in advance by a laser trimming technique. The overheat protection circuit according to any one of claims 6 to 8, wherein: 10. The overheat protection circuit according to claim 6, wherein the size of the diode is adjusted by a laser trimming technique. 11. The overheat protection circuit according to claim 6, wherein said first current source is constituted by a depletion-type transistor connected in saturation. 12. The heating protection circuit according to claim 6, wherein the diode is arranged near an output transistor that supplies a power supply voltage to the IC. 13. The output transistor according to claim 1, wherein the output transistor is arranged to surround the diode.
2. The overheat protection circuit according to 2. 14. An IC circuit incorporating the temperature detection circuit according to any one of claims 1 to 5 or the heating protection circuit according to any one of claims 6 to 13. 15. A portable electronic device incorporating the temperature detection circuit according to any one of claims 1 to 5 or the heating protection circuit according to any one of claims 6 to 13. 16. A voltage regulator incorporating the temperature detection circuit according to claim 1 or the heat protection circuit according to claim 6. 17. A DC-DC converter incorporating a temperature detection circuit according to any one of claims 1 to 5 or a heating protection circuit according to any one of claims 6 to 13.
DC converter. 18. A battery pack incorporating the temperature detection circuit according to any one of claims 1 to 5 or the heating protection circuit according to any one of claims 6 to 13. 19. An on-vehicle electrical component incorporating the temperature detection circuit according to any one of claims 1 to 5 or the heating protection circuit according to any one of claims 6 to 13.