GB2051432A

GB2051432A - Power supply circuits

Info

Publication number: GB2051432A
Application number: GB8017149A
Authority: GB
Original assignee: Nilssen O K
Current assignee: Nilssen O K
Priority date: 1979-05-24
Filing date: 1980-05-23
Publication date: 1981-01-14
Also published as: JPS5625385A; FR2457596A1; DE3019876A1

Abstract

A push-pull inverter circuit, particularly suitable for use as a battery charger, employs a feedback means operable to regulate transistor switching frequency and oscillation in accordance with the magnitude of the output. This feedback means, preferably providing electrical isolation from the inverter circuit output, may comprise a variable impedance means utilized with a saturable core transformer for providing drive current to the transistors. Voltage developed across the impedance means as a result of transistor drive current determines the transformer saturation time and hence the transistor switching frequency. A transformer means having a significant magnetic flux leakage path between its primary and secondary windings is employed in a power supply to limit the transformer means current. <IMAGE>

Description

SPECIFICATION Power supply circuits This invention relates in general to electrical power supply circuits and, more particularly, to circuits particularly suitable for use as battery chargers.

As is well-known in the art, the main function of a push-pull inverter is to convert a direct or unidirectional input voltage into an alternating output voltage. Examples of inverter circuits believed to be representative of the state of the art appear in the following references including U.S. Patent Nos. 2,809,303, 2,997,664,3,119,058, 3,172,060, 3,219,906, 3,248,640, 3,265,953, 3,297,959,3,324,411, 3,348,120, 3,350,661, 3,351,840, 3,461,405, 3,467,852, 3,579,026, 3,624,481, 3,663,994, 3,691,450, 3,831,078, 3,913,036, 3,927,363, 4,016,475, 4,016,477, and 4,047,089.

Although the prior art is replete with inverter circuits, to the best of applicant's knowledge, inverter circuits have not in the past been utilized in power supply circuits such as battery chargers to provide efficient output regulation by way of feedback control of the inverter oscillation, particularly with safetyrequired electrical isolation between the power supply output and the feedback signals to the inverter.

Accordingly, an aim of the present invention is to provide electrical energy conversion and power supply devices preferably adaptable for use as a battery charger having means for controlling or regulating outputs in an efficient and cost-effective manner.

The present invention therefore provides an inverter power supply circuit connected to a DC input and being operable to supply an output to a load and having transistor switching means, said circuit including drive circuit means operable to supply current to the transistor switching means for effecting inverter circuit oscillation, and regulating means operable to regulate the drive circuit means to vary the inverter oscillation frequency in accordance with the magnitude of the power supply output.

Further features and advantages of the present invention will become apparent from the following description, which, when taken in conjunction with accompanying drawings, discloses preferred embodiments of the invention.

Figure 1 is a schematic diagram of a preferred embodiment of an inverter circuit according to the present invention; Figure 2 illustrates characteristic waveform diagrams of the collector voltage, base voltage and collector current for both of the switching transistors illustrated in Fig. 1 during certain operating conditions; Figure 3 is a schematic circuit diagram of another preferred embodiment of the present invention adapted for use as a battery charger; Figure 4 illustrates a preferred type of main transformer employed in the circuit of Fig. 3; Figure 5 is a schematic diagram illustrating the electrical equivalent of the transformer illustrated in Fig. 4; and Figure 6 illustrates an alternate form of a variable impedance means useful in conjunction with a battery charger circuit similar to that shown in Fig. 3.

An improved high efficiency inverter circuit according to one preferred embodiment of the present invention is illustrated schematically in Fig. 1 and is designated generally be reference numeral 1 0. The circuit 10 comprises a main transformer 11, a pair of relatively highpower switching transistors 1 2 and 13, a saturable current transformer 14 and a nonsaturable current transformer 1 5. The main transformer 11 has a primary winding 11 p and a secondary winding ills. The secondary winding I 1s is connected to deliver a high frequency alternating output voltage to a pair of terminals 1 7 and 1 8. The transistor 1 2 has a base 1 2b, a collector 1 2c and an emitter 1 2e; similarly, the transistor 1 3 has a base 1 3b, collector 1 3c and emitter 1 3e.

The inverter circuit 10 also comprises a resistor 19, a capacitor 20 and diodes 22, 23, 24, and 25. The transformer 14 has windings 26, 27, and 28 wound on a toroidal magnetic core 29. Similarly, the transformer 15 has windings 30, 31, and 32 wound on a toroidal magnetic core 33. The primary winding 11 p has a center tap 34 connected to a terminal 35 which is in turn connected to a positive source of DC voltage or B +. Similarly, the transformer 1 5 winding 32 has a center tap 36 connected to a negative terminal 37 or B - .The center tap 34 on the transformer 11 is connected by a lead 38 to one end of the resistor 19, and the other end of resistor 1 9 is connected by means of a lead 39 to the base 1 2b of transistor 1 2. The lead 39 is also connected to one end of a cathode 22c of diode 22, to an anode 24a of the diode 24 and to one end of winding 28 of transformer 1 5 through an impedance means designated generally by reference character Z for a purpose to be described later. The other end of winding 28 is connected by a lead 40 to the base 1 3b, to a cathode 23c of diode 23 and to an anode 25a of the diode 25. The two ends of winding 32 of transformer 1 5 are connected to cathodes 24c and 25c.The emitter 1 2e of transistor 1 2 is connected by means of a lead 42 to an anode 22a of diode 22 and to the B - terminal 37. Similarly, the emitter 1 3e of transistor 1 3 is connected by means of a lead 43 to an anode 23a of diode 23 and to the B - terminal 37.

One end of the primary winding 11 p of transformer 11 is connected by means of a lead 44 to one end of the winding 26 of transformer 14. Similarly, the other end of the winding 11 p of transformer 11 is connected by means of a lead 45 to one end of the winding 27 on the transformer 14. The other ends of the windings 26 and 27 are con nected by means of leads 46 and 47 to the windings 30 and 31, respectively, of the transformer 1 5. The other end of winding 30 is connected by means of the lead 48 to the collector 1 2c of transistor 12, and the other end of winding 31 is connected by means of a lead 49 to the collector 1 3c of transistor 13. The capacitor 20 is connected between the leads 48 and 49.

Disregarding the function of the impedance means Z for the moment, the basic operation of the inverter circuit Fig. 1 may be described in conjunction with the waveform diagram of Fig. 2. A positive starting bias signal is provided from B + through resistor 1 9 and lead 39 to the base 12b, and through impedance means Z, winding 28 and lead 40 to the base 13b, respectively. This bias current is effective to trigger one or the other of the transistors 1 2 or 1 3 into conduction. Assume that transistor 1 2 is the conducting transistor at the time t,; the base voltage Eb has been driven to a positive state, and the collector current lc rises.The collector voltage Ec is substantially zero or only slightly positive while the transistor 1 2 is conducting. Current from B + flows through the main transformer winding 11 p and lead 44 through winding 26 of transformer 14, through winding 30 of transformer 15, and through lead 48, collector 12c, emitter 1 2e and lead 42 to B -. The current flowing through winding 26 produces a changing flux in the core 29 of transformer 14. So long as this core 29 is not saturated, the winding 28 provides a positive feedback to the base 12b.The current path from one end of the winding 28 is through the impedance means Z, lead 39, base 12b, emitter 12e, and lead 42 to B -. A return current path to winding 28 is provided from B through lead 42, diode 23 and lead 40 to the other end of winding 28.

At some time t2 core 29 becomes saturated and positive feedback to the base 1 2b ceases; the winding 28 in effect becomes a short circuit. Until the core 29 saturates, the transformer 14 provides the dominant feedback to the base drive circuit. However, when the transformer core 29 does saturate, the transformer 1 5 then provides the dominant feedback control. The transformer 1 5 is designed so that the core 33 is not saturated by the same conditions that cause the transformer 14 to saturate. The winding 32 of transformer 1 5 provides a subtractive feedback through the diode 24 to the base 12b.The implication of this performance is that as soon as the transformer 14 saturates, the base drive current reverses, thereby rapidly evacuating the stored charge carriers from the base-emitter junction of transistor 1 2. As soon as the baseemitter junction of transistor 1 2 is evacuated of carriers, the current taken by winding 32 of transformer 1 5 starts flowing through the diodes 22 and 24 and center tap 36 to B It should be noted that the subtractive feedback which causes the rapid evacuation of the charge carriers from the base-emitter junction of transistor 1 2 prevents any positive voltage transients on the base 1 3b of the opposite transistor 1 3. The saturated core 29 of transformer 1 4 in effect provides a circuit through the winding 28 and the impedance means Z, between the bases 1 2b and 1 3b so that both bases remain negative.Due to the unidirectionally subtractive current from transformer 15, both bases will stay at a negative voltage for as long as current flows through winding 32 of transformer 15, which is for as long as collector current lc continues to flow. The implication of this is that it does not matter that the transistors 1 2 and 1 3 may have significant turn-off delays. The nonconducting transistor simply cannot be turned on until current has stopped flowing through the opposite transistor.

The capacitor 20 connected between the collectors 1 2c and 1 3c serves to restrain the rate of rise until time t4 of the collector voltage Ec after the conducting transistor 21 is turned off. As a result, the transistor 1 2 is turned off completely before its collector voltage rises to any significant level. This greatly minimizes power dissipation in the transistor during the turn-off transition. It should be noted that the capacitor 20 also bridges the windings 30 and 31 of the transformer 1 5 and provides a path for continuity of current flow through the current feedback transformers 14 and 15.

The main transformer 11 contains some amount of leakage inductance. This inductance in effect provides inertia to the current flow so that the collector voltage Ec on the transistor 1 2 that is being turned off could rise to a very high level. However, the ultimate voltage that can be reached by the collector of the off transistor cannot be more than twice the magnitude of B +. From the nature of the circuit, it may be noted that when the collector voltage Ec of one transistor has risen to a magnitude of twice B +, the voltage on the collector of the other transistor has fallen to zero. Also, the collector voltage cannot rise beyond twice the magnitude of B + because the other transistor 13, in com bina & on with its base to emitter diode, acts as a clamp.

In order to eliminate the turn-on losses entirely, the off-transistor should not be turned on until after its collector voltage has reached zero. This is precisely what is accomplished by the base drive circuits in the opera tion of the transformer 1 4 and 1 5.

The operation of transistor 1 3 can be described by reference to the lower or second set of waveform diagrams of Fig. 2, which are substantially identical to the upper set shown for transistor 12--only displaced in time.

Between time t3 and t4, the collector voltage Ec of transistor 1 3 decreases to slightly below zero. The rate of decline corresponds to the rate of rise of the collector voltage of the now off transistor 1 2. This rate of change is restrained by the charging of the capacitor 20, and continues until clamping takes place. At time t4, a negative current ic begins to flow between the base 1 3b and collector 1 3c and continues until time t5. The path for this current flow is from B - through diode 23, the base-collector junction of transistor 13, windings 31 and 27, and winding 1 1p of transformer 11 to B +. This results in a return of energy to the power supply during this time span.The transistor 1 3 and diode 23 in this mode function as a clamp to limit the magnitude of the voltage of the collector 1 2c of transistor 12.

At time t5, the base 1 3b is driven positive and transistor 1 3 begins to conduct in a positive direction. The path for this current flow lc is from B + through the right-hand half of winding 11 p, through windings 27 and 31, collector 1 3c and emitter 1 3e to B -. A positive feedback is provided from winding 28 of transformer 1 4 to the base 13b. The current lc continues to flow until time t7, which occurs shortly after the transformer 1 4 saturates at time t6, and positive feedback to the base 1 3b ceases.At this time, the transformer 1 5 takes over and supplies a subtractive feedback to the base 1 3b rapidly evacuating its charge carriers. The transistor 1 3 is turned off and its collector voltage begins to rise until time t8, completing the cycle of operation.

The alternate conduction of each transistor 1 2 and 1 3 produces current flow through winding 11 p of transformer 11 in opposite directions. This alternating current in winding 11 p is transformed into a high-frequency AC voltage at the secondary winding 11s which is supplied to the output terminals 1 7 and 1 8.

Regarding the operation of the impedance means Z in Fig. 1, voltage developed across the impedance means as a result of the positive feedback current through winding 28 will determine the saturation time of the transformer 1 4 and hence the switching frequency of transistors 1 2 and 1 3. In a saturable core transformer such as transformer 14, it is wellknown that the product of the applied voltage and the switching time to saturation is constant, but temperature dependent. Therefore, any increase in the value of the impedance means Z will result in an increased voltage in the feedback loop, causing the time for the core 27 of transformer 14 to reach saturation to decrease correspondingly.The shorter the feedback transformer saturation time, the higher the switching frequency of transistors 12, 1 3. The impedance means Z can comprise a resistor, a capacitor or an inductor; it can also consist of a non-linear device or a filament of an incandescent lamp. Of course, a resistive impedance will absorb energy and reduce the amount of feedback current available for the bases of transistors 1 2 and 1 3. If the impedance means Z is variable, the switching frequency of transistors 12, 1 3 will be modulated in accordance with the value of the impedance means. As the switching frequency of transistors 12, 1 3 increases, the waveform of the collector voltage Ec of each of these transistors tends to become more sinusoidal than its Fig. 2 counterpart.Further, the impedance means Z can regulate the drive to transistors 12, 1 3 for intermittent inverter circuit oscillation.

According to the present invention, the use of the impedance means Z to control the switching frequency and oscillation of transistors in a regulated-output push-pull inverter circuit is particularly suitable for use in the battery charger illustrated schematically in Fig. 3. In that figure, circuit elements identical to those depicted in Fig. 1 are designated by primed reference numerals. Because common elements function in essentially the same manner as their Fig. 1 counterparts, their operation will not be repeated in detail. Typical component types and values for a nominal 12 volt battery charger are indicated in Fig. 3.

It will be appreciated that the DC input to the circuit 10' may be conveniently supplied from an AC line through a conventional rectifying device (not shown) connected between the AC input and the DC input terminal 35' and 37'.

In the Fig. 3 embodiment of the present invention, the main transformer 11' has a frequency dependent output impedance and its secondary ills' comprises windings 50, 51. One end of winding 50 is connected by means of leads 52a and 52b to a center-tap lead 53 which is, in turn, connected to winding 51 by means of leads 54a and 54b. The center tap line 53 is connected to the output 18' through a circuit breaker 55. The other ends of windings 50, 51 are connected through diodes 56, 57, respectively, having anodes connected to a line 58 which is in turn connected to the negative output terminal 17'. Terminals 17' and 18' are connected to a battery (not shown) under charge. The connections to leads 52a, 52b, 54a and 54b will be explained presently.

The impedance means Z' comprises an isolating feedback transformer 59 having a torroidal core and a primary winding 61 having one end connected to line 39' through a line 62 and the other end connected to one end of winding 28' through a line 63. As will become apparent, the isolating transformer 59 serves to vary the value of the impedance of Z' in accordance with the magnitude of the battery charger output while providing electrical isolation between the output and the drive or feedback signals to the switching transistors 12' and 13'. This isolation is extremely important from a safety standpoint.

The isolating feedback transformer 59 has a secondary winding 64 having ends thereof connected to anodes of full-wave rectifying diodes 65, 66, which have cathodes connected to a collector lead 67 of a current limiting or regulating transistor 68, which has a base terminal 68b, an emitter 68e and a collector 68c. The emitter is connected to a ground lead 69 which is in turn connected through a switch 70 to a center tap 71 of the secondary winding 64 of the isolating feedback transformer 59. A resistor 72 is connected between the collector and base terminals of transistor 68 for providing a predetermined minimum bias to that transistor base terminal 68b.

A voltage limiting or regulating transistor 73 comprises a collector 73c which is connected directly to the base 68b of transistor 68. Transistor 73 also comprises an emitter terminal 73e which is connected to the ground lead 69 and a base terminal 73b which is connected to the ground lead 69 and a base terminal 73b which is connected to the ground lead 69 through a resistor 74.

A biasing means comprising a Zener diode 76 and a serially connected resistor 77 is connected to the positive output terminal 18' through a lead 78 to provide bias to the control element or base 68b of transistor 68.

Another biasing means including a Zener diode 79 provides bias to the control element 73b of transistor 73. The anode of the Zener diode 79 is connected to the ground lead 69 through a filtering capacitor 82. The cathode of Zener diode 79 is connected to the positive output lead 78 through a resistor 83 and a serially connected subtractive voltage means 84, the latter providing a subtractive reference voltage proportional to the battery charger output current for a purpose to be described.

The subtractive voltage means 84 comprises a toroidal core transfomer 86 having a winding 87 connected to leads 52a and 52b and another winding 88 connected to leads 54a and 54b. The ends of another winding 89 are connected through diodes 91, 92 to a lead 93, and the center tap of winding 89 is connected to a lead 94. A parallel resistor 96 and a filtering capacitor 97 are connected across leads 93 and 94. A subtractive voltage provided across leads 93 and 94 is proportional to the battery charging current.

In the operation of the battery charger circuit of Fig. 3, Zener diode 76 and resistor 77 provide bias to the base 68b of transistor 68.

The Zener diode 79, the resistor 83 and the subtractive voltage means 84 provide bias to the base 73b of the voltage limiting transistor 73. The current limiting transistor 68 is normally conductive for certain battery charging voltages; this transistor, operating in the active region, rapidly becomes non-conductive if the output voltage across terminals 17' and 18' decreases below the threshold voltage established by the Zener diode 76. Should the battery charging load increase, the output voltage at terminals 17' and 18' will decrease, causing a corresponding decrease in bias to the base terminal 68b of the current limiting transistor 68. This will cause the impedance reflected to the primary 61 of the isolating feedback transformer 59 to increase.

As the impedance means Z' increases in value, the time for the core 27' of the saturable core feedback transformer 14 will decrease, as previously described, in turn causing the switching frequency of transistors 12' and 13' to increase. Because the main transformer 11' has a frequency dependent output impedance, the current limiting impedance of that transformer will increase with the increase in switching frequency of transistors 12' and 13'. If the output terminals 17' and 18' are shorted, the isolating transformer 59 presents an impedance high enough such that the positive feedback current is reduced below a level enabling conduction of the transistors 12' and 13'; the resistor 72 provides a predetermined minimum bias to the current limiting transistor 68.The current limiting transistor 68 rapdily functions to regulate or limit the battery charging current to prevent damage to the circuit 10'.

Should the output voltage tend to increase, the bias to the base 73b of the voltage limiting transistor 73 will increase. The value of the bias to the base 73b of transistor 73 is determined by the resistor 83, the Zener diode 79 and by the subtractive voltage means 84. In charging batteries, higher voltage is preferred when the charging current is high.

The subtractive voltage means 84 reduces the reference voltage established by the Zener diode 79 by an amount in direct proportion to the charging current such that the output of the battery charger 10' matches the typical characteristics of the battery to be charged.

For example, in the case of a nominal 1 2 volt automobile lead-acid battery, when the charging current is high, the Fig. 3 circuit will regulate to approximately 13.5 or 14.0 volts; when the current is low, the circuit will regulate to approximately 1 2.5 volts. As transistor 73, operating in its active region, becomes conductive, it diverts bias from the current limiting transistor 68. Of course, as the bias to the transistor 68 decreases, the value of Z' will increase, as previously described, causing the switching frequency of transistors 12' and 13' to increase and the output impedance of transformer 11' to increase, thereby limiting the output voltage. If the transistor 73 be comes sufficiently conductive, the transistor 68 will become non-conductive.The transistor 73 is operable to control the output of the circuit 10' by regulating the inverter circuit for intermittent oscillation. When the output voltage at terminal 18' becomes sufficiently high, the charge on capacitor 82 will cause the inverter oscillation to terminate until the charge dissipates after sufficient reduction in the output voltage. This cycle will be repeated for "bang-bang" control of the power supply output.

Because battery chargers are typically used to supply charging current to batteries of different voltages, nominally values of 6 and 1 2 volts, each of Zener diodes 76 and 79 may be replaced by plural Zener diodes (not shown) each for a corresponding battery charger output voltage, and a selectively operable switching device (also not shown) operable to connect the Zener diodes appropriate for the voltage of the battery to be charged.

The switch 70 is operable, when opened, to cause the isolating transformer 59 to present to the feedback circuit a value of the impedance Z' sufficiently high such that the feedback current to the transistors 12' and 13' will be reduced to the point where the oscillation of circuit 10' terminates.

Fig. 4 illustrates a preferred main transformer 11' wherein the transformer comprises a significant magnetic flux leakage path, depicted by reference numeral 98, between the primary winding 11 p' and the secondary winding 1 Is'. The primary winding is not coupled to the secondary winding in a completely rigid fashion as is normally done with battery charger transformers. If the secondary 11s' becomes heavily loaded, which occurs, for example, in the cause of a short circuit, the magnetic flux path 98 would prevent excessive primary short circuit current. It is particularly important to provide windings of stranded wire, twisted together, to minimize power losses.

Fig. 5 illustrates an electrical equivalent to the transformer of Fig. 4. It is seen that a series impedance designated by reference numeral 99 provides a substantial effective series inductance to limit the output current through the secondary 11s' to a predetermined value. The series impedance 99 permits control of the circuit output as the switching frequency of transistors 12' and 13' is varied, as previously described.

Fig. 6 illustrates another alternate impedance means designated by the reference character Z" for use with the circuit of Fig. 3.

Leads 62 and 63 are connected to a primary winding 101 of a transformer 102, which has a secondary winding 103 having terminals connected to a shorting switch 1 04. When switch 104 is closed, full circuit output is obtained. When switch 104 is opened, the value of the impedance Z" will be increased to a value sufficiently high such that the feedback to the transistors 12' and 1 3' will be reduced to the point that the circuit will stop oscillating. Switch 104 may be controlled by a thermostat (not shown) which can be conveniently utilized to turn off the battery charger 10' if excessive temperatures, indicative of excessive charging current, are reached.

Claims

1. An inverter power supply circuit connected to a DC input and being operable to supply an output to a load and having transistor switching means, said circuit including drive circuit means operable to supply current to the transistor switching means for effecting inverter circuit oscillation, and regulating means operable to regulate the drive circuit means to vary the inverter oscillation frequency in accordance with the magnitude of the power supply output.

2. A supply circuit according to Claim 1 wherein the regulating means is operable to regulate the drive circuit means for intermittent inverter circuit oscillation.

3. A supply circuit according to Claim 1 or 2 including a main transformer comprising a primary winding in circuit with the transistor switching means and a secondary winding in circuit with the output and having a significant magnetic flux leakage path between the primary and secondary windings whereby to prevent excessive primary winding current.

4. A supply circuit according to any of claims 1-3, including frequency dependent impedance means in series with the output.

5. A supply circuit according to any of the preceding claims, wherein said transistor switching means comprises two alternately conducting switching transistors, each with a control element, said drive circuit means being operable to supply current to each control element for causing alternate transistor conduction, and said regulating means comprising feedback means in circuit with the output and being operable to vary the transistor switching frequency in accordance with the magnitude of the output.

6. A supply circuit according to Claims 4 and 5 wherein the feedback means is operable to vary the transistor switching frequency in accordance with the magnitude of the output voltage developed by said impedance means.

7. A supply circuit according to Claim 5 or 6 wherein the input and output thereof are electrically isolated and wherein the feedback means provides to the drive circuit means feedback signals which are electrically isolated from the power supply output.

8. A supply circuit according to any of Claims 5-7, wherein said feedback means is operable to regulate the drive circuit means for intermittent inverter circuit oscillation.

9. A supply circuit according to Claims 4 and 8 wherein the feedback means is operable to regulate the drive circuit means to vary the inverter circuit oscillation in accordance with the magnitude of the output voltage.

1 0. An inverter-type battery charger operable to convert power obtained from a DC voltage source to a battery charging output and having two alternately conducting switching transistors, each with a control element, said battery charger including drive circuit means comprising a saturable core feedback transformer operable to supply intermittent feedback current to each control element for causing alternate transistor conduction, and impedance means in circuit with the saturable core feedback transformer and the transistor control elements, whereby voltage developed across the impedance means as a result of feedback current determines the feedback transformer saturation time and hence the transistor switching frequency for regulation of the output.

11. The device of Claim 10 wherein the impedance means is variable in accordance with the magnitude of the output, whereby the transistor switching frequency is modulated in accordance with the value of the impedance means.

1 2. The device according to Claim 10 or 11 wherein the impedance means is operable to control the switching of the transistors for intermittent inverter circuit oscillation.

13. The device of Claim 10, 11 or 12 wherein the impedance means comprises an isolating feedback circuit means connected to the inverter circuit output.

1 4. The device of any of Claims 10 to 1 3 including a frequency dependent impedance means in series with the output.

1 5. An inverter circuit operable to convert power from a DC input to an AC output and having two alternately conducting switching transistors, each having a base-emitter junction containing stored charge carriers during transistor conduction, said circuit including first and second drive circuit means connected between the base-emitter junctions of the transistors, the first drive circuit means comprising a saturable core feedback transformer operable to supply intermittent positive feedback current for causing alternate transistor conduction, the second drive circuit means comprising a non-saturable transformer operable, upon saturation of the saturable core feedback transformer, to supply intermittent subtractive feedback current to rapidly turn off a conducting transistor by forcing evacuation of the charge carriers stored in its base-emitter junction, and impedance means in series with the saturable core feedback transformer and the transistor base-emitter junctions, whereby voltage developed across the impedance means as a result of positive feedback current determines the feed back transformer saturation time and hence the transistor switching frequency.

1 6. The inverter circuit of Claim 1 5 wherein the impedance means comprises a non-linear device.

1 7. The inverter circuit of Claim 1 5 or 1 6 wherein the value of the impedance means is temperature dependent.

1 8. An inverter power supply circuit connected to a DC input and being operable to provide a DC output, including two alternately conducting switching transistors, each having a control element, a main transformer having a frequency dependent output impedance and having primary and secondary windings in circuit with the transistors and the output, respectively, output rectifying means in circuit with the secondary of the main transformer and the output, first and second drive means connected to the transistor control elements, the first drive circuit means comprising a saturable core feedback transformer operable to supply intermittent positive feedback current for causing alternate transistor conduction, the second drive circuit means being operable to supply intermittent subtractive feedback current for rapidly turning off a conducting transistor, and impedance means in circuit with the saturable core feedback transformer and the transistor control elements for presenting a variable impedance of value determined in accordance with the magnitude of the output, whereby voltage developed across the impedance means as a result of positive feedback current modulates the transistor switching frequency in order to regulate the power supply output.

19. The supply circuit of Claim 18 wherein the impedance means comprises a transformer having a secondary winding in circuit with a shorting switch for selectively varying the value or the impedance means.

20. The supply circuit of Claim 1 9 wherein the shorting switch is thermostatically controlled.

21. The supply circuit of Claim 18 wherein the impedance means includes an isolating feedback transformer having primary and secondary windings, the primary winding being in circuit with the transistor control elements, full-wave rectifying means connected to the secondary of the isolating transformer, regulating transistor means connected to the output of the rectifying means, and means for supplying bias to the regulating transistor means to vary the conducting impedance thereof in accordance with the magnitude of the output voltage, whereby the regulating transistor means is operable to reflect to the primary winding of the isolating feedback transformer a variable impedance of value determined in accordance with the magnitude of the output voltage.

22. The supply circuit of Claim 21 including selectively operable switching means for opening the secondary winding of the isolating feedback transformer in order to increase the value of the impedance means to a level sufficiently high so as to terminate feedback current to the transistor control elements.

23. The supply circuit of Claim 21 or 22 wherein the isolating feedback transformer has a primary impedance sufficiently high such that, in the event of a short circuit at the power supply output, the positive feedback current to the switching transistors is reduced below the level necessary to cause conduction thereof.

24. The supply circuit of Claim 21, 22 or 23 wherein the regulating transistor means comprises current and voltage limiting transistor means in circuit with each other, the current limiting transistor means being operable to reflect to the primary winding of the isolating feedback transformer a variable impedance determined in accordance with the current limiting transistor means bias, the voltage limiting transistor means being operable to divert bias from the current limiting transistor means when the power supply output voltage exceeds a predetermined magnitude.

25. The supply circuit of Claim 24 wherein the means for supplying bias to the regulating transistor means includes means for establishing a threshold voltage, and means for providing a subtractive voltage inversely related to the magnitude of the power supply output current, whereby the voltage limiting transistor means is operable to divert bias from the current limiting transistor means when the power supply output voltage exceeds the voltage resultant of the threshold voltage reduced by the subtractive voltage.

26. The supply circuit of Claim 25 wherein the subtractive voltage means comprises a transformer having a primary winding in circuit with the main transformer and a secondary winding connected to rectifying and filtering means.

27. A battery charger operable to convert an AC input to a battery charging output, including transformer means having primary and secondary windings and a significant magnetic flux leakage path therebetween for limiting the transformer means current, and output rectifier means in circuit with the secondary winding of the transformer means.

28. An inverter power supply circuit connected to an input voltage and being operable to supply an output voltage, said circuit including inverter means in circuit with the input, frequency dependent output impedance means in circuit with the inverter means, and regulating means for regulating the inverter oscillation frequency in accordance with the magnitude of the power supply output.

29. The supply circuit of Claim 28 wherein the frequency dependent impedance means comprises a transformer having primary and secondary windings and a significant magnetic flux leakage path therebetween.

30. The supply circuit of Claim 28 or 29 wherein the regulating means is operable to regulate the inverter oscillation frequency and the output current in accordance with the magnitude of the output voltage.

31. The supply circuit of Claim 28, 29 or 30 wherein the input voltage is an AC voltage and the output voltage is a DC voltage, said supply circuit including input rectifying means connected to the input, and output rectifying means in circuit with the output impedance means.

32. The supply circuit of any of Claims 28 to 31 wherein said inverter means comprises switching transistor means, said supply circuit including drive circuit means operable to supply current to the switching transistor means for effecting oscillation of the inverter means, and said regulating means operable to regulate the drive circuit means to vary the inverter oscillation frequency in accordance with the magnitude of the output.

33. The supply circuit of Claim 32, including means for establishing a threshold voltage and additional means for providing a subtractive voltage inversely related to the magnitude of the power supply current, and wherein the regulating means is operable to regulate the drive circuit means to increase the inverter oscillation frequency in accordance with the excess in power supply output voltage beyond the resultant of the threshold voltage reduced by the subtractive voltage.

34. An inverter battery charger connected to an AC input and being operable to provide a DC battery charging output which is electrically isolated from the input, including input rectifying means in circuit with the input, inverter means in circuit with the input rectifying means and comprising two alternately conducting switching transistors, each having a control element, drive circuit means operable to supply current to each control element for causing alternate transistor conduction, feedback means in circuit with the output and being operable to provide to the drive circuit means feedback signals which are electrically isolated from the battery charger output in order to regulate the drive circuit means to vary the transistor switching frequency in accordance with the magnitude of the output, and output rectifying means in circuit with the inverter means.

35. The battery charger of Claim 34 including frequency dependent impedance means in series with the output.

36. The battery charger of Claim 34 or 35 wherein the feedback means comprises means for establishing a plurality of threshold voltages and manually operable switching means for selecting a desired one of the threshold voltages, the feedback means being operable to regulate the drive circuit means to increase the inverter oscillation frequency in accordance with the excess in battery charging output voltage beyond the selected threshold voltage.

37. The battery charger of Claim 34, 35 or 36 including a main transformer comprises ing a primary winding in circuit with the transistors and a secondary winding in circuit with the output rectifying means and having a significant magnetic flux leakage path between the primary and secondary windings whereby to prevent excessive primary winding current.

38. An inverter power supply circuit constructed substantially as herein described with reference to the embodiments shown in the accompanying drawings.