FR2877803A1 - METHOD AND DEVICE FOR RECEIVING A DEGRADE RADIONAVIGATION SIGNAL - Google Patents

Abstract

The invention relates to a method for receiving a radionavigation signal, which comprises, over a first determined time, a step of correlation calculations of the received signal sampled with a code of period T generated locally, the signal and the code being shifted by n relative to each other from one correlation to the next correlation. Prior to the correlation calculation step, it comprises a step of storing the signal in a memory for a second determined time and the signal is written in memory according to a write frequency fe and read according to a read frequency fl such that fl> fe with a view to the step of correlation calculations. The sampled signal comprising a real component I and an imaginary component Q, it comprises the steps consisting in adding the results of the correlation calculations of the same component and of the same offset n, and obtained over Tc code periods, Tc being an integer greater than 1, so as to obtain for each component, a coherent integration respectively designated ITC (n) and QTC (n), calculate the sum STC (n) = ITC (n) 2 + QTC (n) 2, and add the sums STC (n) or the same shift n, obtained over a time Tnc, Tnc = K x Tc, K being an integer greater than 1, so as to obtain for each shift n , a non-coherent integration C (n).

Description

METHOD AND DEVICE FOR RECEIVING A SIGNAL

RADIONAVIGATION DEGRADE

  The invention relates to the reception of a radionavigation signal from a satellite positioning system such as the GPS system (acronym for the English expression Global Positioning System).

  To function well, current receivers usually require direct satellite reception. Positioning o deteriorates rapidly especially in terms of accuracy and acquisition time, when the reception is disturbed as is the case inside a building or more generally in degraded environment.

  It briefly recalls the operation of the GPS system. It consists of a constellation of 28 satellites and a terrestrial network of ground-based reference stations. Each satellite gravitates 20 000 km from the earth with a period of revolution of 12 hours. They emit two signals, one at 1575.452 MHz for civil applications and the other at 1227.6 MHz for reserved access applications. In the following, we only consider the civil frequency. The signal emitted by a satellite consists of a 1575.452 MHz frequency carrier, modulated by a known spreading code and possibly by unknown data also known as databits. The satellites all transmit on the same frequency and the emitted signals are differentiated by their code.

  These codes have a period T, for example 1 ms and typically consist of 1023 chips.

  The positioning of the receiver is obtained by measuring the distance between a satellite and the receiver from the propagation time of the signal between this satellite and the receiver. In the receiver, a replica of the transmitted code is generated locally; the offset between the received signal and the local signal (i.e. the replica) corresponds to the desired propagation time. This offset is measured by phasing the received signal and the local signal; the phasing criterion corresponds to the maximization of the correlation function of the two signals, that is, to the search for a peak of correlation.

  Correlation calculations are performed from the real and imaginary components of the received signal.

  When the reception is degraded, the computation of a peak of correlation on a single period of code T is not possible: this computation is carried out by integrating over several periods of code, the correlations of the same component (real or imaginary) and the same offset between the signal and the code.

  This correlation calculation is generally carried out from half-code chip to half-code chip over an integration interval that can be varied. For an integration interval of 1 ms, a correlation computation time of about 2s is obtained. (2 x 1023 x 1 ms =, 2s). This calculation is multiplied by a factor P function of the drift of the local clock (or driver) of the receiver and the number of assumptions on the frequency of the signal to be considered to compensate for the Doppler effect. For a clock uncertainty of about 10 kHz, P = 21; we then obtain a computation time of approximately 2s x 21, ie 42s per code, that is to say by satellite.

  Taking an interval of 10ms (ie 10 code periods), we gain about 10 dB, but the calculation then lasts about 20s multiplied by a factor P -201, or about 4020 s per satellite. We finally won only 10 dB for a positioning time that has increased considerably.

  One solution for reducing this computation time is to perform 2 x 1023 multiplications in parallel instead of calculating them in series as before. We then obtain a computation time on an integration interval of 10 ms of about 2s (201 x 10 ms = 2s).

  In the following we place ourselves within the framework of this solution.

  Integrations are made from the real and imaginary components of the sampled signal. It is possible to perform a so-called coherent integration over a coherent time Tc possibly greater than T. A significant coherent time makes it possible to have a narrow bandwidth and to bring out the signal of the noise. On the other hand, it is not possible to greatly increase the coherent time. This one is very quickly limited by: the number of hypotheses of frequencies to be considered to compensate the Doppler effect, the potential acceleration of the mobile, the characteristics and the price of the pilot oscillator in the receiver.

  Because of these limitations, the consistent integration is less than 100ms and more typically set at 20ms, that is, the period of the databits. The reception sensitivity for a coherent time of 20 ms is located around -144 dBm according to the probabilities of absence and false detections of the radionavigation signal.

  To push back these limits, a so-called non-coherent integration performed on a non-coherent time Tnc is also generally performed, starting from the coherent integrations for the two components I and Q computed over a single period of code T. The non-coherent integration can be realized by means of a dedicated integrated circuit or a microprocessor.

  The non-coherent integration by means of a dedicated integrated circuit makes it possible to add only coherent integrations computed over a single period of code T. By using a microprocessor, the non-coherent integration of coherent integrations calculated on a duration greater than the T code period can only be performed by successive readings of coherent integrations and addition; the signal is then stored on 1 to 20 ms depending on the number of readings required, which requires a lot of resources of the microprocessor.

  An important object of the invention is therefore to be able to increase the integration time and consequently the sensitivity of the receiver, especially when the reception is degraded.

  To achieve this object, the invention proposes a method for receiving a radionavigation signal, which comprises, at a first determined time, a correlation calculation step of the sampled received signal with a locally generated period code T, the signal and the code being shifted by n relative to each other by a correlation to the next correlation.

  It is mainly characterized in that prior to the correlation calculation step, it comprises a step of storing the signal in a memory during a determined second time and in that the signal is written in memory according to a write frequency fe and read at a reading frequency f1 such that, for the correlation calculation step, the sampled signal having a real component I and an imaginary component Q, it comprises the steps of adding the results of the computation calculations. correlation of the same component and the same offset n, and obtained on Tc code periods, Tc being an integer greater than 1, so as to obtain for each component, a coherent integration respectively designated ITc (n) and QTc (n), compute the sum STc (n) = ITC (n) 2 + QTc (n) 2 and add the sums STc (n) or, JS, (n) with the same offset n, obtained over a time Tnc, Tnc = K x Tc, K being a superior integer 15 r to 1, so as to obtain for each shift n, a non-coherent integration C (n).

  Thus, one mixes a coherent integration on a Tc periods of code T and a noncoherent integration on a time Tnc, with Tnc = K x Tc.

  The coherent integration is calculated by integrating over several periods of code, the correlations of the same component (real or imaginary) and the same offset n between the signal and the code. The noncoherent integration is calculated by performing the sum over a time Tnc multiple of Tc (Tnc = K x Tc), amplitudes or powers on the two components of coherent integrations having the same offset n.

  According to one characteristic of the invention, the correlation calculation step is performed for several codes and the memorized signal is read for each code. Similarly, the correlation calculation step is performed for several frequency hypotheses considered to compensate for the Doppler effect and the memorized signal is read in memory for each of these hypotheses.

  The subject of the invention is also a receiver of a radionavigation signal, characterized in that it comprises a microprocessor capable of implementing the method as described.

  Finally, the invention also relates to a receiver of a radionavigation signal provided with a correlator capable of correlating the received signal which comprises a real component and an imaginary component, with a code of length N and of period generated locally, characterized in that it comprises connected to the correlator, a first memory and a first adder able to memorize 2N results from the correlator for the two components and to increment them respectively as and when the results provided by the correlator, for Tc periods of code, Tc being an integer greater than 1, to produce 2N coherent integrations respectively designated for each ITC and QTC component, / means for obtaining N amplitudes or sums powers STC (n) = ITC (n) 2 + QTC (n) 2, a second memory and a second adder able to memorize the N amplitudes or powers of the somrnes STc (n), and to increment them respectively during a duration Tnc, Tnc = K × Tc, K being an integer greater than 1, as the amplitudes or powers of the sums STc (n) are obtained, to realize N non-coherent integrations.

  Such a receiver makes it possible to mix non-coherent and coherent integrations, the latter being calculated over a period greater than the period of the code T and to achieve this mixing rapidly. With Tc = 10 ms, we gain more than 30% in signal processing time. The configuration flexibility between the coherent time and the number of coherent integrations makes it possible to optimize the detection power of the weak signals.

  Advantageously, it comprises connected to the correlator a third memory capable of storing the received signal at a write frequency fe and to restore it to the correlator at a read frequency f1 such that f1> fe.

  According to one characteristic of the invention, the correlator comprises a non-rotating shift register for the code and at least one non-rotating shift register for the signal.

  Other features and advantages of the invention will appear on reading the detailed description which follows, given by way of nonlimiting example and with reference to the appended drawings in which: FIG. 1 schematically represents an example of known correlator 5 with a rotating shift register for the code, and a non-rotating shift register for the signal, Fig. 2 schematically shows an example of a correlator with a non-rotating shift register for the code, and a slip shift register for the signal, FIG. 3 diagrammatically represents an example of improvement of the correlator of FIG. 2, FIG. 4 schematically represents an example of a receiver according to the invention able to implement a mix of coherent and non-coherent integrations with post-processing.

  According to one embodiment of the invention, a coherent integration is mixed on Tc periods of T code and a non-coherent integration over a time Tnc, with Tnc = K x Tc.

  The coherent integration is calculated by integrating over several periods of code, the correlations of the same component (real or imaginary) and the same offset n between the signal and the code.

  We will first detail these correlations on an example taking N = 2046 = 2 x 1023 because the correlation calculation is done from half-code chip to half-code chip. The code chips can be divided by 2 as in this example or by 3, 4, ..., even 3.3, etc. According to a first embodiment, from one correlation to another, the modification of the offset n is obtained by shifting the code, the signal remaining fixed. This technique proceeds in two phases using for each component of the signal, a structure such as that of Figure 1.

  The received signal whose frequency is compensated to take into account the Doppler effect, is loaded into a non-rotating shift register 1 and the code is loaded into a rotating shift register 2. The length N of the shift register represents the number code chips considered over the period of the code, ie 1 ms for example for the GPS. Then the signal remains fixed in its shift register 1 and the code performs N rotations on itself in the rotating shift register 2. A new correlation is calculated for each rotation, by means of N multipliers 3 and an adder 4 at N entries.

  The first phase loads the signal over a period of code. The second phase calculates a correlation with each rotation of the code on itself.

  The two real and imaginary components of the signal are processed by means of a shift register and can be for example expressed as follows: SRr (i, tk) are the samples of the real component of the signal at the instant ix 1 ms / 2046 + tk, SRi (i, tk) are the samples of the imaginary component of the signal at the instant ix 1 ms / 2046 + tk, i the index of the shift register varying from 0 to 2045, tk = to + kxl ms. tk represents the (k + 1) th period of code. Code (i, n) is the index code (i + n) modulo 2046, i the index of the shift register varying from 0 to 2045, n the delay index between the signal and the code, varying from 0 to 2045. At the k + 1th code period, the result of the correlation on the real component, provided by the adder for an offset n, is: E [Code (i, n) * SRr (i, tk) 1 = Code (0, n) * SRr (0, tk) + Code (1, n) * S Rr (1, tk) + ... r = o

  + Code (2044, n) * SRr (2044, tk) + Code (2045, n) * SRr (2045, tk) Similarly, at the k + 1 th code period, the result of the correlation on the imaginary component , supplied by the adder for an offset n, is: [Code (i, n) * SRi (i, tk)] = Code (0, n) * SRi (0, tk) + Code (1, n) * SRi (1, tk) + ... r = o

  + Code (2044, n) * SRi (2044, tk) + Code (2045, n) * SRi (2045, tk) According to another embodiment, from one correlation to another, the code is now fixed and only the signal is shifted: it slides in its shift register at each clock cycle. The slice of signal is always equal to a period of code but this slice is slippery.

  This embodiment is implemented using for each component of the signal, a structure such as that of FIG. 2.

  The code is entered in a non-rotating shift register 2 'and the component of the compensated signal of its Doppler is continuously fed into a sliding shift register 1. The length N of the shift register represents the number of chips of the code considered on the period of the code, ie l ms for example for the GPS.

  One of the advantages of this correlator lies in the parallelization of the two phases of the conventional solutions (loading of the signal in the shift register and calculation of the correlations). The signal processing time, therefore search satellites is divided by 2.

  According to another embodiment, the two structures for the real and imaginary components are combined as shown in the correlator example of FIG.

  This correlator 20 is provided with a non-rotating shift register 2 'for the code, common to the two components, of length N. The other elements were already present in the example of FIG. 3, namely a shift register 1 of length N for each real and imaginary component of the signal (respectively designated signal I and signal Q in the figure), 2N multipliers 3 (N for each component) and 2 adders 4 to N entered each (1 adder for each component).

  The 2 real and imaginary components of the signal can for example be expressed in the following way: SRr (i, tk, n) are the samples of the real component of the signal at the instant ix 1 ms / 2046 + tk, n, SRi (i, tk, n) are the samples of the imaginary component of the signal at the instant ix 1 ms / 2046 + tk, f, i the index of the shift register varies from 0 to 2045, tk, n = to + kxl ms + nxl ms / 2046. tk, n represents the (k + 1) th code period, n ranging from 0 to 2045 and represents the delay index between the signal and the code.

  At the k + 1th code period, the result of the correlation on the real component, supplied by the adder for an offset n, is: E [Code (i) * SRr (i, tk, n)] = Code (0) * SRr (0, tk n) + Code (1) * SRr (1, tk, n) + ... i = o

  + Code (2044) * SRr (2044, tk, n) + Code (2045) * SRr (2045, tk, n) Similarly, at the k + 1th code period, the result of the correlation on the imaginary component , supplied by the adder for an offset n, is: E [Code (i) * SRi (i, tk, n)] = Code (0) * SRi (O, tk, n) + Code (1) * SRi (1, tk, n) + ... i = o

  + Code (2044) * SRi (2044, tk, n) + Code (2045) * SRi (2045, tk, n) The process according to the invention proceeds in 3 steps. The first step calculates a coherent integration over a time Tc with its 2 real and imaginary components. The second step calculates an amplitude or a power from the coherent integrations on the 2 components. The third step calculates over a time Tnc the sum of the amplitudes or powers having the same offset. The 3 steps are repeated N times. In our example N = 2046. This mixing is implemented for example by means shown in FIG. 4.

  The coherent integration (1st step) is carried out during a time Tc, which is a multiple of the period of the code, in this case a multiple of 1 ms; it is performed separately on both the real and imaginary components.

  When the correlations were calculated from a rotating code and a fixed signal, the following results are obtained. For an offset n between the signal and the code, it results from the coherent integration, I (n, Tc) (or ITc (n)) for the real component and Q (n, Tc) (or QTC (n)) for the imaginary component: ## EQU1 ## = EE (Code (i, n) * SRi (i, tk)) with SRr (i, tk) and SRi (i, tk) at the instant ix 1 ms / 2046 + tk, tk = to + k x1 ms .

  Code (i, n) is the index code (i + n) modulo 2046 as previously indicated.

  When the correlations were calculated from a fixed code and a sliding signal, the following results are obtained. For an offset n between the signal and the code, it results from the coherent integration, I (n, Tc) (or ITc (n)) for the real component and Q (n, Tc) (or QTC (n)) for the imaginary component: TC-I 2045 I, 2.05 (n, Tc) = EE (Code (i) * SRr (i, tk ,,,)) k = 0 i = 0 TC-I 2045 Q = 02045 (n, Tc) = E [E (Code (i) * SRi (i, tk, n)) k = 0 i = 0 with SRr (i, tk, n) and SRi (i, tk, n) at l instant 1 ms / 2046 + tk, n, tk, n = to + kx 1 ms + nx 1 ms / 2046.

  This coherent integration can be obtained by means of a microprocessor capable of implementing the preceding calculations or by means of an adder 6 associated with a memory 7 with 2N boxes able to be read and then written such as a RAM memory. In known manner, the memory 7 is read and written as the results provided by the correlator 20.

  If a phase inversion due to a data bit change occurs during the coherent time, it may be difficult or impossible to detect the correlation peak. The coherent integration requires taking into account the phase of the data bits by means of a known device 5: the result of each correlation is multiplied by 1 according to the value of the databit on the code period during which the correlations are made.

  An amplitude or a power is then calculated from the 2 components of the coherent integrations (2nd step): ## EQU1 ## where S (n, Tc) '+ Q (n, Tc)' or S (n, Tc) = [I (n, Tc) '+ Q (n, Tc) 2 S (n, Tc) is also denoted by STc (n).

  This second step is obtained in known manner by a microprocessor or an integrated circuit 8.

  In the prior art, the non-coherent integration consists of calculating over the period of the code, the amplitude or the power of the correlations on the two components, and then summing them. According to the invention, the non-coherent integration (step 3) is calculated by performing the sum over a time Tnc multiple of Tc (Tnc = K × Tc), coherent amplitudes or powers of integrations having the same offset n. We obtain: C os (n) = jI (n, kTc) 2 + Q (n, kTc) 2 or C_os (n) = 1: [I (n, kTc) 2 + Q (n, kTc) 2 k = 0 k = This non-coherent integration can be obtained by means of a microprocessor able to implement the preceding calculations or by means of an adder 9 associated with a memory 10 with N boxes able to be read and written such that a RAM memory. In known manner, the memory 10 is read and then written as the results provided by the memory 7.

  These three steps are repeated for each shift n, that is to say N times. In our example N = 2046.

  According to one embodiment of the invention, designated real-time mode, the signal is sampled and processed for a satellite and a frequency assumption (considered to compensate for the Doppler effect) at a time. There is no memorization of the signal: the samples are directly used for the processing of the signal which lasts Tnc.

  According to a preferred mode designated post-processing mode, the signal is sampled and stored in memory 11 preferably during the noncoherent time Tnc which can reach 16 s. Then the samples are read into memory for each satellite and each Doppler frequency hypothesis; that is, the I and Q signals from the memory 11 are converted into baseband and compensated by a Doppler frequency by the element 12 before being inserted into the shift register of the correlator 20. Samples are read at a frequency much greater than their memory write frequency. This post-processing mode makes it possible to reduce the overall search time of the satellites as it appears on the following comparison.

  NBSAT: the number of BPDOPPLER satellites: the frequency band to be considered to compensate for the Doppler effect for satellite search Tnc: the noncoherent integration time Tc: the coherent integration time K: the ratio of read and write frequencies in post-processing T tr: Real-time global processing time Tpt: overall processing time post-processing Ona: Ttr = NBSAT x BPDOPPLER x Tc x Tnc Tpt = NBSAT x BPDOPPLER x Tc x Tnc / K + Tnc, Tpt = Ttr / K + Tnc Tnc is negligible compared to Ttr / K; typically K = 25 or more.

  In the example described, two-state signals (1) were considered; of course, the invention applies in the same way to signals with more than two states.

  So far it has been considered that only one code is generated locally. When it is desired to consider other codes, these can be considered in series by reprogramming the local code and the associated Doppler frequencies, as one changes code.

  They can be considered in parallel: it is then sufficient, according to a method known to those skilled in the art, to provide the receiver with parallel structures, that is to say to multiply the structure comprising the correlator (as well as the memory for post-processing if necessary, and / or integrators if necessary), by the number of local codes to consider.

  The received signal may be a satellite and / or pseudolite radionavigation signal.

  This is for example a positioning signal type GPS (acronym for the English expression Global Positioning System), or GALILEO or other positioning system.

2877803 14

Claims (12)

  A method of receiving a radionavigation signal, which comprises at a first determined time a correlation calculation step of the sampled received signal with a locally generated period code T, the signal and the code being shifted by n one with respect to the other of a correlation with the following correlation, characterized in that, prior to the correlation calculation step, it comprises a step of storing the signal in a memory during a second determined time and in this that the signal is written in memory according to a write frequency fe and read according to a reading frequency f1 such that f1 for the correlation calculation step, the sampled signal comprising a real component I and an imaginary component Q, it comprises the steps of adding the results of the correlation calculations of the same component and the same offset n, and obtained on Tc code periods, Tc being a integer greater than 1, so as to obtain for each component a coherent integration respectively designated ITC (n) and QTC (n), calculate the sum STC (n) = ITc (n) 2 + QTC (n) 2, and add sums STC (n) or \ DST. (n) of the same offset n, obtained over a time Tnc, Tnc = K × Tc, K being an integer greater than 1, so as to obtain for each shift n, a non-coherent integration C (n).   2. Method according to the preceding claim, characterized in that the correlation calculation step is performed for several codes and in that the memorized signal is read for each code.   3. Method according to one of the preceding claims, characterized in that the correlation calculation step is performed for several frequency hypotheses considered to compensate for the Doppler effect and in that the memorized signal is read in memory for each of these assumptions.   4. Method according to one of the preceding claims, characterized in that the second determined time is equal to Tnc.   5. Method according to one of the preceding claims, characterized in that on the first determined time, the signal is shifted from one correlation to the next, the code being fixed.   6. Method according to one of the preceding claims, characterized in that the first determined time is equal to the period of the code T. 7. Method according to one of the preceding claims, characterized in that the signal received is a satellite navigation signal and / or pseudolites.   8. Receiver of a radionavigation signal, characterized in that it comprises a microprocessor adapted to implement the method according to one of claims 1 to 7.   9. Receiver of a radionavigation signal provided with a correlator (20) capable of correlating the received signal which comprises a real component and an imaginary component, with a code of length N and of period generated locally, characterized in that it comprises connected to the correlator (20), a first memory (7) and a first adder (6) able to store 2N results from the correlator for the two components and to increment them respectively as and when the results provided by the correlator (20), during Tc code periods, Tc being an integer greater than 1, to produce 2N coherent integrations respectively designated for each ITC and QTC component, means (8) for obtaining N amplitudes or sums powers STc (n) = lrc (n) 2 + QTc (n) 2, a second memory (10) and a second adder (9) able to store the N amplitudes or powers of the sums STc (n), and to increment them respectively during a hard Let Tnc, Tnc = K x Tc, K being an integer greater than 1, as the amplitudes or powers of the sums STC (n) are obtained, to realize N non-coherent integrations.   10. Receiver according to the preceding claim, characterized in that it comprises connected to the correlator (20) a third memory (II) adapted to store the received signal at a write frequency fe and to restore it to the correlator at a frequency of reading fl such that fl> fe.   11. Receiver according to one of claims 9 to 10, characterized in that the signal comprising databits having a phase, it comprises between the correlator (20) and the first adder (6), a device (5) adapted to take account the phase of the databits.   12. Receiver according to one of claims 10 to 11, characterized in that the correlator comprises a non-rotating shift register (2 ') for the code and at least one non-rotating shift register (1) for the signal.