EP0715239A1

EP0715239A1 - High precision current mirror for low voltage supply

Info

Publication number: EP0715239A1
Application number: EP94830555A
Authority: EP
Inventors: Francesco Brianti; Valerio Pisati; Roberto Alini; David Moloney
Original assignee: STMicroelectronics SRL; SGS Thomson Microelectronics SRL
Current assignee: STMicroelectronics SRL
Priority date: 1994-11-30
Filing date: 1994-11-30
Publication date: 1996-06-05
Anticipated expiration: 2014-11-30
Also published as: DE69427479T2; EP0715239B1; DE69427479D1

Abstract

A high-precision current generating circuit (1), particularly intended for a low-impedance circuit user, comprises a current mirror formed by a first (M1) and a second (M2) transistor of the MOS type, and a voltage regulator (4) having two terminals respectively connected to the drain terminals (D1,D2) of the two transistors (M1,M2) to maintain equal values of drain-source voltage (Vds1 and Vds2) on said transistors.

Description

This invention relates to a high-precision current generating circuit.
The invention particularly concerns a high-precision current generating circuit for a low-impedance circuit user outlet, being of the type which comprises a current mirror having an input branch which is applied a reference current and an output branch which supplies a proportional current to the reference current.
As is well known, a current mirror configuration is a conventional circuit arrangement which is normally employed to generate a constant output current proportional to a given reference current.
Current mirrors have many applications in the field of integrated circuits, and are used in particular for biasing differential circuits in oscillator circuits, sample-and-hold circuits, and digital-to-analog converters.
In certain of these applications, such as high-resolution digital-to-analog converter circuits, it is highly important that the current from the generator be precise, constant over time, and unaffected by possible variations in load draw or variations in supply voltage.

Background Art

Current mirror generator circuits of conventional design are described in P. R. Gray and R. G. Meyer "Analysis and Design of Analog Integrated Circuits", Wiley, New York, 1984, pages 233-246 and 703-718.
Such circuits can be implemented, in general, either using transistors of the bipolar type or transistors of the MOS type.
In its simplest form, the current mirror described in the above-mentioned book comprises two transistors and a reference current generator. In its simplest form, the reference generator comprises a resistor connected in series with one of the two transistors.
This prior approach is quite straightforward circuit-wise, and is widely used with circuits which do not require high precision.
The relatively low precision of such a circuit can be explained in relation to its implementation with transistors of the MOS type. The output current from the generator is supplied by one of the two MOS transistors comprising the mirror, and as is well recognized, the internal resistance of a MOS transistor varies with the difference in potential between its drain and source terminals. Since the drain terminal of the transistor, which outputs the current, is usually connected to a load whose impedance may not be always constant, the voltage Vds between the drain and the source of the transistor may also vary during the normal operation of the circuit. This variation in voltage results in a variation in the internal resistance of the transistor, and consequently in the output current from the circuit.
Other circuit designs have been developed in order to provide current generators with enhanced precision.
One of these designs, known from the pertinent literature as a "cascode" current mirror configuration and described in the aforementioned book, enables the output resistance of the current generator to be increased so as to reduce the error due to any variations in the supply voltage. This prior approach, while representing an advance from the standpoint of precision, still has some shortcomings. Its particular cascode structure, for instance, reduces the useful dynamic range for the signal, making the circuit unsuited to low supply voltage applications, e.g. in the 2.5 to 3 volts range.
A further prior approach, also described in the aforementioned book, is the Wilson configuration of the current generating circuit. This is an improvement over the former arrangement as far as precision is concerned, but is again beset with the same drawbacks as the former approach in that it retains the cascode configuration. Also, this approach disallows the biasing of plural output branches on the basis of a single reference current and cannot provide plural output currents.
The underlying technical problem of this invention is to provide a current generating circuit which has such high-precision functional features, and constructional features of reduced silicon area occupation in the integrated circuit, as to overcome the aforementioned drawbacks besetting the prior art.
This technical problem is solved by a high-precision current generating circuit of the type which comprises a transistor current mirror as defined in Claim 1.
The features of this invention will be apparent from the following detailed description of embodiments thereof, with reference to the accompanying illustrative drawings, in which:

Figure 1 is a diagram showing schematically a current generating circuit according to this invention;
Figure 2 shows a first implementation of the current generating circuit of this invention;
Figure 3 is a detailed diagram showing a preferred embodiment of the current generating circuit in Figure 1;
Figure 4 shows a second implementation of the current generating circuit according to the invention;
Figure 5 shows a third implementation of the current generating circuit according to the invention; and
Figure 6 illustrates an exemplary application of the current generating circuit according to the invention.

Detailed Description

With reference in particular to the example of Figure 1, generally and schematically shown at 1 is a high-precision current generating circuit which embodies this invention.
The circuit 1 is a current mirror configuration, and comprises two transistors M1 and M2 of the N-channel MOS type. The first transistor M1 forms substantially the input branch 2 of the current mirror and has its source terminal S1 connected to a fixed potential reference, specifically the electric ground GND of the circuit. The drain terminal D1 of the transistor M1 is connected to a reference current Iref generator.
The output branch 3 of the current mirror comprises the second transistor M2. The source terminal S2 of the transistor M2 is connected to the circuit electric ground GND, its drain terminal D2 being the output terminal of the circuit.
The control terminals G1 and G2 of both transistors M1 and M2 are connected together.
Advantageously, a voltage regulator 4 is connected to the drain terminals D1 and D2 of the transistors M1 and M2 to maintain at said terminals a drain-source voltage value, Vds1 and Vds2, which is the same for both transistors.
The voltage regulator 4 comprises an operational amplifier OA having two inputs, one of the inverting (-) type and another of the non-inverting (+) type, and an output U.
The non-inverting (+) input of the amplifier OA is connected to the drain terminal D1 of the first transistor M1. The inverting (-) input is connected to the drain terminal D2 of the second transistor M2, and the output U is connected to the common node A between the gate terminals G1 and G2 of both transistors M1, M2.
A compensation capacitor C1 is connected between the output U and the non-inverting (+) input of the operational amplifier.
As regards the operation of the circuit in Figure 1, the reference current Iref being flowed through the first transistor M1 is mirrored into the current Iout from the second transistor M2. In fact, the two transistors have their source terminals S1 and S2, and gate terminals G1 and G2, connected together.
The provision of the voltage regulator 4, consisting of the operational amplifier OA, implies that the drain-source voltages Vds1 and Vds2 should be the same value for both transistors M1 and M2. Accordingly, since the transistors M1 and M2 also have the same gate-source voltage, they will be operating in exactly the same condition.
The bias voltage of the two transistors M1 and M2 is supplied from the output U of the operational amplifier OA.
The two feedback branches existing between the output U of the operational amplifier OA and the two (+) and (-) inputs will now be discussed. These two branches are formed essentially by the two transistors M1 and M2.
In particular, the output terminal U of the amplifier OA is connected to the gate terminal G1 of the transistor M1, and the drain terminal D1 of this transistor is connected to the non-inverting (+) input of the amplifier OA.
With the signal present on the drain terminal of a transistor always being shifted 180 degrees from the signal present on the gate terminal thereof, the feedback loop on this branch is bound to be a negative one. The capacitor C1, also feedback connected between the output U and the non-inverting (+) input of the amplifier OA, applies a typical compensation, referred to as "pole splitting", whereby the poles are split by Miller's Effect.
As for the feedback branch connected to the inverting (-) input of the amplifier OA, it is at once apparent that the feedback at this input is a positive one. In fact, the signal present on the amplifier output U is inverted by the transistor M2 before it is applied to the inverting (-) input.
It is therefore necessary that, to reduce the loop gain of such positive feedback to a value of less than one, the load connected to the output of this current generator, on the drain terminal D2 of the transistor M2, has a very low input impedance value.
Shown in Figure 2 is a wiring diagram for a first implementation of the current generating circuit in Figure 1. In particular, the same circuit arrangement as in the previous embodiment has now been implemented using P-channel MOS transistors.
The circuit 1 comprises two P-channel transistors M1 and M2, and a voltage regulator 4. The first transistor M3 forms the input branch 2 of the current mirror and has its source terminal S3 connected to a fixed potential reference, specifically the positive pole Vcc of the supply voltage generator, while its drain terminal D3 is connected to a reference current Iref generator.
The output branch 3 of the current mirror consists of a second P-channel transistor M4. The source terminal S4 of this transistor is connected to the positive pole Vcc of the supply voltage generator, while the transistor drain terminal D4 is coincident with the output terminal of the circuit.
The control terminals G3 and G4 of both transistors M3 and M4 are connected together.
Advantageously, a voltage regulator 4 is connected to the drain terminals D3 and D4 of the transistors M3 and M4, thereby maintaining on said terminals a drain-source voltage value, Vds3 and Vds4, which is the same for both transistors.
The voltage regulator 4 comprises an operational amplifier OA having two inputs, one of the inverting (-) type and another of the non-inverting (+) type, and an output U.
The non-inverting (+) input of the amplifier OA is connected to the drain terminal D3 of the first transistor M3. The inverting (-) input is connected to the drain terminal D4 of the second transistor M4, while the output U is connected to the common node A between the gate terminals G3 and G4 of both transistors M3 and M4.
A compensation capacitor C1 is connected between the output U and the non-inverting (+) input of the operational amplifier.
The circuit of Figure 2 operates in the same manner as the previously described circuit of Figure 1.
The voltage regulator 4, consisting of the operational amplifier OA, implies that the drain-source voltages, Vds3 and Vds4, at the two transistors M3 and M4 should be the same value.
Accordingly, with the transistors M3 and M4 at the same gate-source voltage, they will operate in exactly the same condition.
Similar to the circuit of Figure 1, the feedback existing at the inverting (-) input of the operational amplifier OA is a positive one. It is therefore necessary that, to reduce the loop gain of said positive feedback to a value of less than one, the load connected to the output of this current generator has a very low input impedance value.
The circuit of Figure 1 is illustrated in greater detail by Figure 3; in particular, an optional embodiment of the operational amplifier OA using transistors of the MOS type is shown.
This amplifier OA is formed by four MOS transistors denoted by M5, M6, M7 and M8, in a known circuit arrangement. The two P-channel transistors M5 and M6 form the differential input stage, while the drain terminal D8 of the N-channel transistor M8 is the output terminal U of the amplifier.
Shown in Figure 4 is a wiring diagram for a second implementation of the current generating circuit illustrated by Figure 1. In particular, a circuit arrangement is shown which uses bipolar transistors of the NPN type.
The circuit 1 comprises two transistors T1 and T2 and a voltage regulator 4. The first transistor T1 forms the input branch 2 of the current mirror and has its emitter terminal E1 connected to a fixed potential reference, specifically the electric ground GND of the circuit, while its collector terminal C1 is connected to a reference current Iref generator.
The output branch 3 of the current mirror consists of a second transistor T2. The emitter terminal E2 of this transistor is connected to the electric ground GND of the circuit, the transistor collector terminal C2 being the output terminal of the circuit.
The base terminals B1 and B2 of both transistors T1 and T2 are connected together.
Advantageously, a voltage regulator 4 is connected to the collector terminals C1 and C2 of the transistors T1 and T2 to maintain at said terminals a collector-emitter voltage value, Vce1 and Vce2, which is the same for both transistors.
The voltage regulator 4 comprises an operational amplifier OA having two inputs, one of the inverting (-) type and another of the non-inverting (+) type, and an output U.
The non-inverting (+) input of the amplifier OA is connected to the collector terminal C1 of the first transistor T1. The inverting (-) input is connected to the collector terminal C2 of the second transistor T2, while the output U is connected to the common node A between the base terminals B1 and B2 of both transistors T1 and T2.
A compensation capacitor C1 is connected between the output U of the operational amplifier and the non-inverting (+) input thereof.
The circuit of Figure 4 operates in a similar manner to the circuit of Figure 1.
The voltage regulator 4, consisting of the operational amplifier OA, implies that the collector-emitter voltages Vce1 and Vce2 at the two transistors T1 and T2 should be the same value. Accordingly, since the transistors T1 and T2 also have equal base-emitter voltages, they will be operating in similar conditions.
The feedback existing at the inverting (-) input of the operational amplifier OA is in this circuit positive, as in the circuit of Figure 1. It is necessary, therefore, that in order to reduce the loop gain of said positive feedback to a value of less than unity, the load connected to the output of this current generator has a very low input impedance value.
Figure 5 shows a wiring diagram for a third implementation of the current generating circuit illustrated by Figure 1. In particular, the same circuit arrangement has been embodied here using bipolar transistors of the PNP type.
The circuit comprises two transistors T3 and T4 and a voltage regulator 4. The first transistor T3 forms the input branch of the current mirror and has its emitter terminal E3 connected to a fixed potential reference, specifically the positive pole Vcc of the supply voltage generator, while its collector terminal C3 is connected to a reference current Iref generator.
The output branch of the current mirror consists of a second transistor T4. The emitter terminal E4 of this transistor is connected to the positive pole Vcc of the supply voltage generator, while the transistor collector terminal C4 is the output terminal of the circuit.
The base terminals B3 and B4 of both transistors T3 and T4 are connected together.
Advantageously, a voltage regulator 4 is connected to the collector terminals C3 and C4 of the transistors T3 and T4 to maintain, at said terminals, an equal collector-emitter voltage value, Vce3 and Vce4, for both transistors.
The non-inverting (+) input of the amplifier OA is connected to the collector terminal C3 of the first transistor T3. The inverting (-) input is connected to the collector terminal C4 of the second transistor T4, while the output U is connected to the common node A between the base terminals B3 and B4 of both transistors T3 and T4.
A compensation capacitor C1 is connected between the output U and the non-inverting (+) input of the operational amplifier.
The circuit of Figure 5 operates similar to the circuit of Figure 1.
The voltage regulator 4, comprising the operational amplifier OA, implies that the collector-emitter voltages, Vce3 and Vce4, at the two transistors T3 and T4 should be the same. Accordingly, since the transistors T3 and T4 also have the same base-emitter voltage, their conditions of operation will also be the same.
In a similar way to the circuit of Figure 1, this circuit also has a positive feedback present at the inverting (-) input of the operational amplifier OA, whereby the load connected to the output of this current generator is to have a very low input impedance value, if the loop gain of said positive feedback is to be a value of less than unity.
Figure 6 shows a wiring diagram for a digital-to-analog converter which employs a high-precision current generating circuit in accordance with this invention.
In particular, the output branch 30 of the current mirror circuit 10 comprises, additionally to the first transistor M12, two more transistors M13 and M14. The drain D13, D14 and source S13, S14 terminals of these transistors are respectively connected to the drain D12 and source S12 terminals of the first transistor M12, while their gate terminals G13 and G14 are connected, each through a switch sw1 and sw2, to the gate terminal G12 of the first transistor M12.
The two switches sw1 and sw2 are controlled by the digital input signal, and the currents flowed through the individual transistors M12, M13 and M14 are summed into the output current Iout from the D/A converter. In this particular embodiment, the digital input signal is a two-bit signal, but it would be quite easy, as can be evinced from the modular construction of the output branch 30, to multiply the number of the transistors provided to achieve enhanced resolution for the D/A converter.
Thus, the circuit of this invention provides a high-precision current generator which is quite simple circuit-wise. In addition, this circuit can be used in different technologies, bipolar and MOS transistors, and can operate on a low supply voltage.
A further advantage is that the output branch of this current generator can be easily duplicated to output several currents from a single reference current.

Claims

A high-precision current generating circuit (1), particularly intended for a low-impedance circuit user, being of the type which comprises a transistor current mirror having an input branch (2) and an output branch (3), wherein the input branch (2) comprises a first transistor (M1) and the output branch (3) comprises at least a second transistor (M2), both transistors having first and second terminals and a control terminal, the first terminals (S1,S2) of both transistors (M1,M2) being connected to a fixed potential reference (GND), the control terminals (G1,G2) of both transistors (M1,M2) being connected together, and the second terminal (D1) of the first transistor (M1) connected to a current (Iref) generator, characterized in that it comprises a voltage regulator (4) having at least a first (+) and a second (-) terminal respectively connected to the second terminal (D1) of the first transistor (M1) and the second terminal (D2) of the second transistor (M2) to maintain equal voltage values at said terminals.
A circuit according to Claim 1, characterized in that said voltage regulator (4) comprises an operational amplifier (OA) having a first input (+) connected to the second terminal (D1) of the first transistor (M1) and a second input (-) connected to the second terminal (D2) of the second transistor (M2), and an output (U) connected to the control terminals (G1,G2) of the first (M1) and the second (M2) transistors.
A circuit according to Claim 2, characterized in that the first input (+) of the operational amplifier (OA) is of the non-inverting type and the second input (-) is of the inverting type.
A circuit according to Claim 3, characterized in that a compensation capacitor (C1) is connected between the non-inverting (+) input and the output (U) of the operational amplifier (OA).
A circuit according to Claim 1, characterized in that the output branch (3) of said current mirror comprises at least a third transistor (M13) having first and second terminals and a control terminal, the first (S13) and second (D13) terminals being connected to the corresponding terminals (S2,D2) of the second transistor (M2) and the control terminal (G13) being connected to the control terminal (G2) of said second transistor (M2) through a first switch means (SW1).
A circuit according to any of the preceding claims, characterized in that said transistors are of the N-channel MOS type.
A circuit according to any of Claims 1 to 5, characterized in that said transistors are of the P-channel MOS type.
A circuit according to any of Claims 1 to 5, characterized in that said transistors are of the NPN bipolar type.
A circuit according to any of Claims 1 to 5, characterized in that said transistors are of the PNP bipolar type.