EP0689191A2 - Mobile transceiver - Google Patents

Abstract

The terminal has an estimator continuously forming estimated values (SNR(i)) of the signal-to-noise ratio of the speech signal. Power values, Px(i), of sample values of the speech signal (x(i)) are derived and smoothed. The minimus of a group of L successive smoothed power values is derived. The groups follow on successively without any gap. Each group contains at least as many smoothed power values as needed so that all power values belonging to any phoneme of the speech signal can be detected from any group. An actual value of the signal to noise ratio is then formed from the smoothed power values and the derived minimus. <IMAGE>

Description

The invention relates to a mobile radio terminal with a speech processing device.

In the field of speech processing, noise signals are often contained in speech signals to be processed, which leads to a reduction in speech quality and thus, in particular, to poor speech intelligibility. This problem occurs, for example, in the case of mobile radio terminals which are used in motor vehicles and have a hands-free device. Speech signals that are received by microphones of the hands-free system arranged in the motor vehicle contain, on the one hand, speech signal components that are generated by the respective user (voice source) of the mobile radio terminal within the motor vehicle, and, on the other hand, noise signal components that result from other ambient noises and while driving, essentially from the engine and Driving noises exist.

"IEEE Transactions on Acoustics, Speech, and Signal Processing, VOL. ASSP-29, No. 3, June 1981, pp. 582-587" describes an arrangement for the adaptive estimation of time delays of two strongly correlated signals in digital systems. One of the two signals is delayed by a controllable delay element. The delay values of the delay element are adaptively adapted to the correlated signals. The delay values are determined with the help of an algorithm, which is now referred to by experts as the LMS algorithm (Least Mean Square). This algorithm is based on minimizing the power or the square of error values which result from the formation of the difference between the delayed and the non-delayed signal. The core of the LMS algorithm is the recursive calculation of the Delay values using estimates for the gradient of the performance of the error values.

In the prior art cited above, the difference between two samples of two mutually time-shifted signals is formed to form the error values, one of the signals being delayed. The corresponding delay value is rounded to an integer multiple of a sampling interval of the signals. Convergence problems occur in such a way that strong oscillations of the rounded delay values occur when very small error values are reached. The delay values oscillate between two rounded delay values at a sampling interval.

The invention has for its object to improve the speech quality of the speech signals to be processed and to reduce convergence problems.

• zur Bildung von Gradientenschätzwerten durch Multiplikation von Fehlerwerten für zwei Sprachsignale mit den Ausgangswerten eines Digitalfilters, das eine Phasenverschiebung von 90 Grad bewirkt und zur Filterung eines der zwei Sprachsignale dient,
• zur rekursiven Ermittlung von Verzögerungsschätzwerten aus den Gradientenschätzwerten, wobei aus den Verzögerungsschätzwerten durch Rundung die Verzögerungswerte gebildet werden, die zur Einstellung der Verzögerungsmittel dienen und
• zur Bildung jeweils wenigstens eines Fehlerwertes für einen bestimmten Abtastzeitpunkt aus der Differenz zwischen einem Sprachsignalschätzwert, der zur Abschätzung des weiteren Sprachsignals zu einem gegenüber dem bestimmten Abtastzeitpunkt um den Verzögerungsschätzwert verschobenen Zeitpunkt dient und durch Interpolation von Abtastwerten des weiteren Sprachsignals gebildet wird, und dem Abtastwert eines anderen der zu verarbeitenden Sprachsignale zu dem bestimmten Abtastzeitpunkt

vorgesehen sind und daß eine Addiervorrichtung zum Addieren der gegeneinander zeitversetzten Sprachsignale vorgesehen ist.The object is achieved in that the speech processing device is provided for processing a first and at least one further speech signal consisting of noise and speech signal components and present as samples, that delay means are provided for delaying the sampled further speech signal, that control means

• to form gradient estimates by multiplying error values for two speech signals by the output values of a digital filter which causes a phase shift of 90 degrees and is used to filter one of the two speech signals,
• for the recursive determination of delay estimated values from the gradient estimated values, the delay estimated values which are used for setting the delay means being formed from the delay estimated values by rounding
• to form at least one error value for a specific sampling time from the difference between a speech signal estimate, which is used to estimate the further speech signal at a point in time shifted by the delay estimated value from the determined sampling time and is formed by interpolation of samples of the further speech signal, and the sampling value of another of the speech signals to be processed at the determined sampling time

are provided and that an adder for adding the mutually time-shifted speech signals is provided.

The gradient estimates serve to estimate the respective gradient of the performance of the error values or, in other words, the squared error values. The control means determine the delay estimates so that the performance of the error values is reduced. The convergence of the delay values determined from the delay estimated values is considerably improved since the delay estimated values have a higher resolution than the delay values due to the rounding. In this way, oscillations of the delay values are essentially avoided. The resolution of the delay values is chosen to be lower than the resolution of the delay estimated values in order to keep the technical effort involved in delaying the speech signals as low as possible. The signal / noise power ratio and the speech quality of a sum signal present at the output of the adding device are improved compared to the signal / noise power ratio and the speech quality of the individual speech signals.

In one embodiment of the invention, the digital filter is a digital Hilbert transformer.

A digital Hilbert transformer, which causes a phase shift of 90 degrees for all frequencies, has the transfer function of a low pass in terms of amount, so that the rounded delay values converge well, particularly for the low frequencies that are essential for a speech signal. The Hilbert transformer can also be replaced by a differentiator, for example, which also causes a phase shift of 90 degrees. However, a differentiator has a linearly increasing transfer function in terms of amount, so that in particular the low frequencies of a speech signal are suppressed, so that there is no convergence as good as that of a Hilbert transformer.

In another embodiment, means for smoothing the gradient estimates are provided.

This results in an improved estimate of the delay estimated values.

In a further embodiment, the speech processing device is provided for processing three speech signals.

Compared to a speech processing device for processing only two speech signals, the signal / noise power ratio and the speech quality of the sum signal present at the output of the adding device can be improved in this way.

The invention can also be designed in such a way that the use of a linear combination of error values is provided for determining a delay estimate for the further speech signal.

In this way, the stability of the speech processing device is increased.

For another embodiment of the invention, delay means are provided for delaying the first speech signal with a fixed delay time.

Without the delay means causing a fixed delay, there are only time offsets between the first and the further speech signal (s) adjustable, with which a leading of the first speech signal is effected. Depending on the position of a speech source generating the speech signal components in relation to microphones of the speech processing device that are used to convert the acoustic speech signals generated by the speech source into electrical speech signals, it must also be possible to set a lag of the first speech signal, which can be implemented in a simple manner with the aid of this configuration.

To further develop the invention, the speech processing device is integrated in a hands-free device.

In the case of hands-free devices in particular, there is the problem that received speech signals have disturbing noise signal components which impair the signal / noise power ratio and the speech quality of the speech signals. This problem occurs especially with mobile radio terminals if they are used in a very noisy environment, such as in an automobile. The use of the described invention therefore results in improved communication between the call participants, especially when used in hands-free devices.

Exemplary embodiments are explained in more detail below with reference to the drawings.

Fig. 1

eine Sprachverarbeitungsvorrichtung für zwei Sprachsignale,

Fig. 2

eine Steuervorrichtung zur Einstellung eines Zeitversatzes zwischen den beiden Sprachsignalen nach Fig. 1,

Fig. 3

eine Sprachverarbeitungsvorrichtung für drei Sprachsignale,

Fig. 4 und 5

Blockschaltbilder mit Steuervorrichtungen zur Einstellung von Zeitversätzen zwischen den drei Sprachsignalen nach Fig. 3,

Fig. 6 und 7

ein Blockschaltbild und ein Flußdiagramm zur Bestimmung des Signal-/ Rauschleistungsverhältnisses eines Sprachsignals,

Fig. 8

eine Einteilung von geglätteten Leistungswerten eines Sprachsignals in Gruppen und Untergruppen und

Fig. 9

ein Mobilfunkendgerät mit einer Sprachverarbeitungsvorrichtung nach Fig. 1 bis 8.

Show it:

Fig. 1

a speech processing device for two speech signals,

Fig. 2

1 a control device for setting a time offset between the two voice signals according to FIG. 1,

Fig. 3

a speech processing device for three speech signals,

4 and 5

3 block diagrams with control devices for setting time offsets between the three speech signals,

6 and 7

2 shows a block diagram and a flowchart for determining the signal / noise power ratio of a speech signal,

Fig. 8

a division of smoothed power values of a speech signal into groups and subgroups and

Fig. 9

a mobile radio terminal with a voice processing device according to FIGS. 1 to 8.

The speech processing device shown in FIG. 1 contains two microphones M1 and M2. These are used to convert acoustic to electrical voice signals, which are made up of speech and noise signal components. The speech signal components come from a single speech source (speaker), which as a rule has different distances from the two microphones M1 and M2. The speech signal components are thus highly correlated. The noise signal components of the two speech signals received by the microphones M1 and M2 are ambient noises generated by the individual speech source, which can be assumed to be uncorrelated or only slightly correlated with suitable microphone spacings in the range from 10 to 60 cm if the microphones reverberated in a so-called Environment such as in the car or in an office. If the speech source and speech processing device are located in a motor vehicle, for example, the noise signal components are caused in particular by engine and driving noises.

The microphone signals generated by the microphones M1 and M2 are digitized by analog-digital converters 1 and 2. The resulting digitized and thus present as samples x1 (i) and x2 (i) microphone signals are evaluated by a control device 3, which is used to control and set a delay element 4. The sampled microphone signals x1 (i) and x2 (i) are referred to below as microphone or speech signals. The delay element 4 delays the microphone signal x1 with delay values T1 that can be set by the control device 3. An adding device 5 adds the microphone signal x1 (i) delayed by the delay element 4 and the microphone signal x2 (i) delayed by a delay element 16 with a constant time delay T _max . The delay element 16 is provided in order to be able to set both a leading and a lagging of the microphone signal x1 (i) relative to the microphone signal x2 (i). A sum signal X (i) present at the output of the adding device 5 is a sampled speech signal, the signal / noise power ratio of which is increased compared to the signal / noise power ratios of the speech signals x1 (i) and x2 (i). Through a suitable setting of the delay time T1 of the delay element 4, the addition by the adder 5 increases the power of the voice signal components of the two voice signals x1 (i) and x2 (i) by approximately a factor of 4 and increases the power of the noise signal components only approximately caused by a factor of 2. This results in an improvement in the power-related signal / noise power ratio of approximately 3 dB.

Die Sprachsignalschätzwerte x1_int(i) sind Werte, die sich aus einer Interpolation von Abtastwerten des Sprachsignals x1(i) ergeben. Die Bestimmung der Sprachsignalschätzwerte x1_int(i) wird später erläutert. i ist eine Variable, die ganzzahlige Werte annehmen kann und mit der einerseits Abtastzeitpunkte der Sprachsignale x1(i) und x2(i) und andererseits auch Programmzyklen der programmierbaren und Steuermittel aufweisenden Steuervorrichtung 3 indiziert werden, wobei in einem Programmzyklus jeweils ein neuer Abtastwert per Sprachsignal verarbeitet wird.In Fig. 2 the operation of the control device 3 is explained in more detail using a block diagram. From the speech signal x2 (i) and speech signal estimated values x1 _int (i), error values e 1 (i) result from difference formation $${\text{e₁₂ (i) = x1}}_{\text{int}}\text{(i) - x2 (i)}$$

The speech signal estimates x1 _int (i) are values that result from an interpolation of samples of the speech signal x1 (i). The determination of the speech signal estimates x1 _int (i) will be explained later. i is a variable which can take integer values and with which, on the one hand, sampling times of the speech signals x1 (i) and x2 (i) and, on the other hand, also program cycles of the programmable control device 3 having control means 3, are indicated, with one new sample value per speech signal in each program cycle is processed.

Das die Werte x2_H(i) von x2(i) liefernde Digitalfilter 6 ist ein FIR-Filter der Ordnung K, das Koeffizienten h(0), h(1), ..., h(K) aufweist. Im vorliegenden Ausführungsbeispiel ist K gleich sechzehn, so daß das Digitalfilter 6 siebzehn Koeffizienten aufweist. Das Digitalfilter 6 besitzt dem Betrage nach die Übertragungsfunktion eines Tiefpasses. Es erzeugt weiterhin eine Phasenverschiebung von 90 Grad. Die feste Phasenverschiebung von 90 Grad ist die entscheidende Eigenschaft des Digitalfilters 6, der Verlauf des Betrages der Übertragungsfunktion ist für das Funktionieren der Sprachverarbeitungsvorrichtung nicht entscheidend. So kann das Digitalfilter 6 auch mit Hilfe eines Differenzierers realisiert werden, was allerdings zu einer Unterdrückung von niederfrequenten Anteilen von x2(i) und damit zu einer verringerten Leistungfähigkeit der Sprachverarbeitungsvorrichtung führen würde.A digital filter 6 carries out a Hilbert transformation of the sample values x2 (i):

The digital filter 6 supplying the values x2 _H (i) of x2 (i) is an FIR filter of the order K, which has coefficients h (0), h (1), ..., h (K). In the present exemplary embodiment, K is sixteen, so that the digital filter 6 has seventeen coefficients. The amount of the digital filter 6 has the transfer function of a low pass. It continues to produce a 90 degree phase shift. The fixed phase shift of 90 degrees is the decisive property of the digital filter 6, the course of the amount of the transfer function is not decisive for the functioning of the speech processing device. The digital filter 6 can thus also be implemented with the aid of a differentiator, which would, however, lead to a suppression of low-frequency components of x2 (i) and thus to a reduced performance of the speech processing device.

gebildet wird. N gibt die Anzahl der in die Berechnung eingehenden Abtastwerte von x1 an. N ist beispielsweise gleich 65. Die Multiplikation mit 1/P_x2(i) dient zur Vermeidung von Instabilitäten in der Steuervorrichtung 3 beim Steuern des Verzögerungsgliedes 4. Damit ergibt sich durch

ein auf die Kurzzeitleistung P_x2(i) normierter geschätzter Gradient grad(i) der Quadrate bzw. der Leistung der Fehlerwerte e₁₂(i) im Programmzyklus i.The output values x2 _H (i) are multiplied by the error values e₁₂ (i) and the reciprocal 1 / P _x2 (i) of a short-term power P _x2 (i), the short-term power P _x2 (i) after $${\text{P}}_{\text{x2}}{\text{(i) = P}}_{\text{x2}}\text{(i-1) + [x2 (i)] 2 - [x2 (iN)] 2}$$

is formed. N indicates the number of samples of x1 used in the calculation. N is, for example, equal to 65. The multiplication by 1 / P _x2 (i) serves to avoid instabilities in the control device 3 when the delay element 4 is controlled

an estimated on the short-term power P _x2 (i) graded gradient degree (i) of the squares or the power of the error values e₁₂ (i) in the program cycle i.

A function block 7 continuously forms estimated values SNR (i) of the associated signal / noise power ratio from the samples of the speech signal x2 (i), which are evaluated by a function block 8. An evaluation of the speech signal x1 (i) instead of the speech signal x2 (i) is also possible without the functionality of the speech processing device being restricted. The functioning of the function block 7 will be explained in more detail later with reference to FIGS. 6 to 8. Function block 8 carries out a threshold decision regarding the estimated values SNR (i). Only if the estimated values SNR (i) lie above a predefinable threshold is an intermediate memory 9 overwritten with the newly determined gradient estimated value grad (i). This case is symbolized by the closed position of a switch 11 which is controlled by the function block 8. The memory content (degree (i)) of the intermediate memory 9 is further processed by a functional unit 10. In the event that an estimated value SNR (i) lies below the predefinable threshold value, the buffer 9 is not overwritten with the newly determined gradient estimated value grad (i) and it retains its old memory content, which is symbolized by the open position of the switch 11 . The predefinable threshold, on which the opening and closing of the switch 11 by the function block 8 depends, is preferably between 0 and 10 dB.

The intermediate memory 9 supplies the gradient estimated values grad (i) stored in it to the functional unit 10, to which sample values of the speech signal x1 (i) are also supplied and which is used both for supplying the speech signal estimated values x1 _int (i) and for setting the delay element 4.

zu geglätteten ("smoothed") Gradientenschätzwerten sgrad(i) weiterverarbeitet. α ist eine Konstante, die im Ausführungsbeispiel den Wert 0,95 besitzt. Die Werte sgrad(i) werden von einem Funktionsblock 13 zur Adaption von Verzögerungsschätzwerten T1'(i) nach $$\text{T1'(i+1) = T1'(i) - µ * sgrad(i)}$$

verwendet. Die Bestimmung von Verzögerungsschätzwerten T1'(i) erfolgt damit rekursiv. µ ist ein konstanter Faktor bzw. Konvergenzparameter und liegt im Bereich

R_x2x2 bezeichnet eine Autokorrelationsfunktion des Sprachsignals x2(i) an der Stelle Null. Ein besonders vorteilhafter Wertebereich von µ ist im vorliegenden Ausführungsbeispiel 1,5 < µ < 3.The gradient estimated values grad (i) are followed by a function block 12 $$\text{degree (i) = α * degree (i-1) + (1-α) * degree (i)}$$

processed to smoothed gradient estimates sgrad (i). α is a constant that has the value 0.95 in the exemplary embodiment. The values sgrad (i) are followed by a function block 13 for adapting delay estimated values T1 '(i) $$\text{T1 '(i + 1) = T1' (i) - µ * sgrad (i)}$$

used. Delay estimated values T1 '(i) are thus determined recursively. µ is a constant factor or convergence parameter and is in the range

R _x2x2 denotes an autocorrelation function of the speech signal x2 (i) at the zero position. A particularly advantageous value range of μ in the present exemplary embodiment is 1.5 <μ <3.

The delay estimated values T1 '(i) can also be non-integer values, ie non-integer multiples of a sampling interval. A function block 14 rounds the delay estimated values T1 '(i) to integer delay values T1 (i) with which the delay device 4 is set. The rounding operation by function block 14 is necessary since values of the speech signal x1 (i) to be delayed by the delay element 4 are only available at the corresponding sampling times.

durch Interpolation dreier benachbarter Abtastwerte $\text{x1(i+T1(i)-1)}$

, $\text{x1(i+T1(i))}$

und $\text{x1(i+T1(i)+1)}$

des Sprachsignals x1 bildet. Der Funktionsblock 15 ist somit in der Lage, durch den Sprachsignalschätzwert x1_int(i) im Programmzyklus i einen Wert des Sprachsignals x1 zum Zeitpunkt i+T1(i), d.h. zu einem Zeitpunkt zwischen zwei Abtastzeitpunkten, zu bilden bzw. zu interpolieren. Die beschriebene Interpolation durch Funktionsblock 15 kann dadurch ersetzt werden, daß Funktionsblock 15 eine Tiefpaßfilterung der Abtastwerte x1(i) zur Interpolation von Werten zwischen den Abtastzeitpunkten durchführt.The functional unit 10 furthermore has a functional block 15 which tracks the speech signal estimated values x1 _int (i) $${\text{x1}}_{\text{int}}\text{(i) = x1 (i + T1 (i)) + 0.5 * [T1 '(i) - T1 (i)] * [x1 (i + T1 (i) +1)) - x1 (i + T1 (i) -1)]}$$

by interpolating three neighboring samples $\text{x1 (i + T1 (i) -1)}$

, $\text{x1 (i + T1 (i))}$

and $\text{x1 (i + T1 (i) +1)}$

of the speech signal x1. Function block 15 is thus able to use the speech signal estimate x1 _int (i) in program cycle i to form or interpolate a value of speech signal x1 at time i + T1 (i), ie at a time between two sampling times. The described interpolation by function block 15 can be replaced by function block 15 performing low-pass filtering of the sample values x1 (i) for the interpolation of values between the sample times.

Would be used to determine the error values e₁₂ (i) instead of the speech signal estimates x1 _int (i) the delayed samples of the speech signal x1 (i) present at the output of the delay element 4, as can be seen from "IEEE Transactions on Acoustics, Speech, and Signal Processing, VOL . ASSP-29, No. 3, June 1981, p. 582-587 "is known, the delay values T1 (i), with which the delay element 4 is set, would no longer converge when error values e 1 (i) = 0 were reached . There would be strong oscillations of the rounded delay values T1 (i). These would fluctuate between two delay values with the interval of a sampling interval. The corresponding true time delay between the speech signal components, which is determined by the different distances from the speaker to the microphones M1 and M2, would lie between these two delay values. In the present exemplary embodiment, such oscillations are avoided by using speech signal estimates x1 _int (i) when the error values are formed, by which the values of the speech signal x1 (i) are also used for delays by non-integer numbers Multiples of a sampling interval are available, ie also at times other than the sampling times i of the speech signal x1 (i).

The function block 12 used to smooth the gradient estimated values grad (i) brings about an improved determination of the delay estimated values T1 '(i).

The control device 3 adapts the delay estimates T1 '(i) or the delay values T1 (i) so that the square or the power of the error values e 1 (i) is reduced from one program cycle to the next. The convergence of T1 '(i) or T1 (i) is thus ensured.

FIG. 3 shows a speech processing device which works in principle like the speech processing device from FIG. 1 and now has three microphones M1, M2 and M3 for the delivery of microphone or speech signals. The microphone signals are fed to analog-to-digital converters 20, 21 and 22, which deliver digitized and thus sampled speech signals x1 (i), x2 (i) and x3 (i), which consist of speech and noise signal components. The speech signals x1 (i) and x3 (i) are supplied to adjustable delay elements 23 and 24. Analogously to FIG. 1, the speech signal x2 (i) is fed to a delay element 27 with a fixed delay time T _max. The output values of the delay elements 23, 24 and 27 are added to the sum signal X (i) by an adding device 25. A control device 26 evaluates the sample values of the speech signals x1 (i), x2 (i) and x3 (i) and derives rounded integer delay values T1 (i) and T3 () from these sample values analogously to the mode of operation of the control device 3 from FIGS. i) ab, which correspond to the integer multiples of a sampling interval of the sampled speech signals x1 (i), x2 (i) and x3 (i) and with which the delay elements 23 and 24 are set, so that an expansion from two to three microphone to be processed or voice signals is enabled.

FIG. 4 shows a first embodiment of the control device 26 from FIG. 3. Two functional units 10 are provided, the structure of which is identical to the structure of the functional unit 10 from FIG. 2 and which are used to set the delay elements 23 and 24 with the rounded time delay values T1 (i) and T3 (i).

und aus einer Differenz ${\text{x3}}_{\text{int}}\text{(i) - x2(i)}$

werden Fehlerwerte e₁₂(i) und e₃₂(i) gebildet.The upper functional unit 10 provides speech signal estimates x1 _int (i). The lower functional unit 10 supplies speech signal estimates x3 _int (i). From a difference ${\text{x1}}_{\text{int}}\text{(i) - x2 (i)}$

and from a difference ${\text{x3}}_{\text{int}}\text{(i) - x2 (i)}$

error values e₁₂ (i) and e₃₂ (i) are formed.

Here too, a digital filter 6 is provided, which has already been described in more detail in the explanations relating to FIG. 2, and which serves to receive the sample values x2 (i) and to supply values x2 _H (i) which are obtained by a Hilbert transformation of the Samples x2 (i) are generated. The values x2 _H (i) are multiplied on the one hand by the error values e₁₂ (i) and on the other hand by the error values e₃₂ (i). The first product x2 _H (i) * e₁₂ (i) is the upper, the second product x2 _H (i) * e₃₂ (i) is fed to the lower functional unit 10. The arrangement of the function blocks 7 and 8, the buffer 9 and the switch 11 is carried out analogously to FIG. 2 and is not shown in FIG. 4 for reasons of clarity.

FIG. 5 shows a version of the control device 26 that is expanded compared to FIG. 4. In contrast to FIG. 4, three digital filters 6 are now arranged instead of just one digital filter 6. These form the values x1 _H (i), x2 _H (i) and x3 _H (i) from the speech signal samples x1 (i), x2 (i) and x3 (i) by Hilbert transformation.

bebildet, die in ein erstes Produkt ${\text{0,3*e₁₃(i)*x3}}_{\text{H}}\text{(i)}$

eingehen. Ein zweites Produkt ergibt sich aus ${\text{0,7*e₁₂(i)*x2}}_{\text{h}}\text{(i)}$

. Die beiden Produkte entsprechen gewichteten Gradientschätzwerten der Quadrate der Fehlerwerte e₁₃(i) und e₁₂(i). Die Summe aus erstem und zweitem Produkt und damit eine Linearkombination der gewichteten Gradientschätzwerten wird der oberen Funktionseinheit 10 zugeführt.In the upper half of the block diagram shown in Fig. 5, error values e₁₃ (i) from the difference ${\text{x1}}_{\text{int}}\text{(i) -x2 (i)}$

illustrated in a first product ${\text{0.3 * e₁₃ (i) * x3}}_{\text{H}}\text{(i)}$

come in. A second product results from ${\text{0.7 * e₁₂ (i) * x2}}_{\text{H}}\text{(i)}$

. The two products correspond to weighted gradient estimates of the squares of the Error values e₁₃ (i) and e₁₂ (i). The sum of the first and second product and thus a linear combination of the weighted gradient estimated values is fed to the upper functional unit 10.

. Die Fehlerwerte e₃₂(i) werden durch die Differenz ${\text{x3}}_{\text{int}}\text{(i)-x2(i)}$

gebildet. Ein drittes Produkt ${\text{0,3*e₃₁(i)*x1}}_{\text{H}}\text{(i)}$

und ein viertes Produkt ${\text{0,7*e₃₂(i)*x2}}_{\text{H}}\text{(i)}$

werden aufaddiert und die sich ergebende Summe wird der unteren Funktionseinheit 10 zugeführt.Similarly, error values e₃₁ (i) and e₃₂ (i) are formed in the lower half of the block diagram shown in FIG. 5. The error values e₃₁ (i) result from the difference ${\text{x3}}_{\text{int}}\text{(i) -x1 (i)}$

. The error values e₃₂ (i) are the difference ${\text{x3}}_{\text{int}}\text{(i) -x2 (i)}$

educated. A third product ${\text{0.3 * e₃₁ (i) * x1}}_{\text{H}}\text{(i)}$

and a fourth product ${\text{0.7 * e₃₂ (i) * x2}}_{\text{H}}\text{(i)}$

are added up and the resulting sum is fed to the lower functional unit 10.

3, which contains a control device according to FIG. 4 or 5, a sum signal X (i) which is improved compared to the speech processing device with two microphones according to FIG. 1 can be generated. The signal / noise power ratio and thus the speech quality of the sum signal X (i) of the speech processing device according to FIG. 3 is further increased compared to the sum signal X (i) generated by the speech processing device according to FIG. 1. The control device according to FIG. 5 has an increased stability compared to the control device according to FIG. 4 when used in the speech processing device according to FIG. 3.

Both in FIG. 4 and in FIG. 5, for the sake of clarity, a representation of means (see function blocks 7 and 8, buffer store 9 and switch 11 in FIG. 2) has been dispensed with, which means that the speech processing is dependent on estimated values SNR ( i) for one of the microphone signals x1 (i), x2 (i) or x3 (i). Also for reasons of clarity, the normalization of products from error values and the output values of the digital filters 6 performing the Hilbert transformation to the power of an associated microphone signal (see 1 / P _x2 (i) in FIG. 2) is not shown. The extension of the control devices 26 according to FIGS. 4 and 5 by these two technical features result from their implementation in the control device 3 according to FIG. 2.

In order to increase the speech quality of the sum signals X (i) at the output of the adding devices 5 and 25 in FIGS. 1 and 3, the invention can be designed in such a way that the delay estimated values T1 '(i) and T3' (i) (e.g. For example, floating point numbers) to form the delay values T1 (i) and T3 (i) are not rounded to values which correspond to an integer multiple of a sampling interval (here: integers), but to values which correspond to a multiple of a fraction of a sampling interval . In particular, rounding the delay estimated values to multiples of a value which corresponds to a quarter or half of a sampling interval is advantageous. In this way, the resolution of the delay values is increased, which can thus be adjusted more precisely, so that the speech quality of the sum signals X (i) is further increased, since time differences from the speech source producing the speech signal components to the microphones M1, M2 and M3 are more precisely compensated can. When a speech signal is delayed by a multiple of a fraction of a sampling interval, interpolation or low pass filtering of speech signal samples is provided to produce speech signal values that lie between two speech signal samples. The interpolation or low-pass filtering can in particular be integrated into the delay means 4, 23 and 24.

With the help of FIGS. 6 and 7, the scheme is explained, on the basis of which the function block 7 from a sampled speech signal x (i), which consists of noise and speech signal components, the associated estimated values SNR (i) of the signal / noise power ratio, that is Ratio of the power of the speech signal components to the power of the noise signal components, determined. In FIG. 2, the sample values x2 (i) correspond to the sample values x (i). The function block 7 is shown in FIG. 6 on the basis of a block diagram. A functional block 30 is used to form power values P _x (i) of the samples x (i) by squaring the samples. Function block 30 also smoothes these power values P _x (i). The resulting smoothed power values P _{x, s} (i) are supplied to both function block 31 and function block 32. Function block 31 continuously determines estimated values P _n (i) for estimating the power of the noise signal component of the sampled values x (i), ie the power of the noise signal components of the sampled values x (i) is determined. From the smoothed power values P _{x, s} (i) and the estimated values P _n (i), the function block 32 continuously determines estimated values SNR (i) of the signal / noise power ratio of the sampled values x (i).

ein Kurzzeitleistungswert P_x(i) und dann durch $${\text{P}}_{\text{x,s}}{\text{(i) = α * P}}_{\text{x,s}}{\text{(i-1) + (1-α)*P}}_{\text{x}}\text{(i)}$$

ein neuer geglätteter Leistungswert gebildet wird. Mit Formel (1) wird ein Kurzzeitleistungswert P_x(i) einer Gruppe von N aufeinanderfolgenden Abtastwerten x(i) ermittelt. N ist hier beispielsweise gleich 128. Der Wert α aus Gleichung (2) liegt zwischen 0,95 und 0,98. Die Ermittlung von geglätteten Leistungswerten P_x,s(i) kann auch nur durch Gleichung (2) durchgeführt werden, wobei dann allerdings der Wert α ungefähr auf den Wert 0,99 zu erhöhen und P_x(i) durch x²(i) zu ersetzen ist.FIG. 7 shows a flow chart which explains the function of the function block 7 in more detail. The flow chart shows how estimated values SNR (i) of the corresponding signal / noise power ratio are formed from the sampled values x (i) of the speech signal x by a computer program. In an initialization block 33, at the beginning of the program described by FIG. 7, a counter variable Z is set to 0 and a variable P _{Mmin is set} to a value P _max . P _max is chosen so large that the smoothed power values P _{x, s} (i) are always smaller than P _max . P _max can, for example, be set to the maximum representable numerical value of a computer used to implement the program. A new sample value x (i) is read in in block 34. In block 35, a counter variable Z is increased by the value 1, after which a new smoothed power value P _{x, s} (i) is formed in block 36. It results from the fact that initially by $${\text{P}}_{\text{x}}{\text{(i) = P}}_{\text{x}}\text{(i-1) + x² (i) - x² (iN)}$$

a short-term power value P _x (i) and then through $${\text{P}}_{\text{x, p}}{\text{(i) = α * P}}_{\text{x, p}}{\text{(i-1) + (1-α) * P}}_{\text{x}}\text{(i)}$$

a new smoothed power value is formed. A short-term power value P _x (i) of a group of N successive sample values x (i) is determined using formula (1). N here is 128, for example. The value α from equation (2) is between 0.95 and 0.98. The determination of smoothed power values P _{x, s} (i) can also only be carried out using equation (2), in which case however the value α should be increased approximately to the value 0.99 and P _x (i) by x² (i) is replace.

A branch 37 then queries whether the smoothed power _value P _{x, s} (i) that has just been determined is less than P _Mmin . If this question is answered in the affirmative, ie P _{x, s} (i) is less than P _Mmin , block 38 _sets P _Mmin to the value of P _{x, s} (i). If the question of branch 37 is answered in the negative, block 38 is skipped. This means that the minimum of M smoothed power _values P _{x, s} is in P _Mmin after M program _cycles . Then the branch 39 is used to query whether the counter variable Z has a value greater than or equal to a value M. In this way it is determined whether M smoothed power values have already been processed.

bestimmt. Diese Operation stellt sicher, daß der vorläufige Schätzwert P_n(i) nicht größer als der aktuelle geglättete Leistungswert P_x,s(i) sein kann. Danach wird mit Block 41 nach der Formel $${\text{SNR(i) = [P}}_{\text{x,s}}{\text{(i) - min{c*P}}_{\text{n}}{\text{(i), P}}_{\text{x,s}}{\text{(i)}] / [c*P}}_{\text{n}}\text{(i)]}$$

ein aktueller Schätzwert SNR(i) des Signal-/Rauschleistungsverhältnisses des Sprachsignals x(i) ermittelt. Im Normalfall dient das Produkt c*P_n(i) zur Abschätzung der aktuellen Leistung des Rauschsignalanteils, und die Differenz ${\text{P}}_{\text{x,s}}{\text{(i)-c*P}}_{\text{n}}\text{(i)}$

dient zur Abschätzung der aktuellen Leistung des Sprachsignalanteils des Sprachsignals x(i). Die aktuelle Leistung des Sprachsignals wird durch den geglätteten Leistungswert P_x,s(i) geschätzt. Die Gewichtung mit einem Skalierungsfaktor c verhindert, daß durch P_n(i) die Rauschsignalleistung mit einem zu kleinen Wert abgeschätzt wird. Der Skalierungsfaktor c liegt typisch im Bereich von 1,3 bis 2. Durch die Minimumbildung in Block 41 bzw. Gleichung (4) wird sichergestellt, daß das nicht logarithmierte Signal-/ Rauschleistungsverhältnis SNR(i) auch dann positiv ist, wenn im Ausnahmefall c*P_n(i) größer als P_x,s(i) ist. Dann wird die Leistung des Rauschsignalanteils des Sprachsignals gleich der durch P_x,s(i) geschätzten Leistung des Sprachsignals gesetzt. Die durch P_x,s(i)-P_x,s(i) geschätzte Leistung des Sprachsignalanteils des Sprachsignals ist dann wie auch das nicht logarithmische Signal-/ Rauschleistungsverhältnis gleich Null. Das Programm wird nach der Berechnung des Schätzwertes SNR(i) mit dem Einlesen eines neuen Sprachsignalabtastwertes x(i) durch Block 34 fortgesetzt.If the question of branch 39 is answered in the negative, ie M smoothed power values have not yet been processed, the program is continued with block 40. There, a preliminary estimate P _n (i) of the noise signal power of the speech signal x is obtained $${\text{P}}_{\text{n}}{\text{(i) = min {P}}_{\text{x, p}}{\text{(i), P}}_{\text{n}}\text{(i)}}$$

certainly. This operation ensures that the preliminary estimate P _n (i) cannot be greater than the current smoothed power value P _{x, s} (i). Then with block 41 according to the formula $${\text{SNR (i) = [P}}_{\text{x, p}}{\text{(i) - min {c * P}}_{\text{n}}{\text{(i), P}}_{\text{x, p}}{\text{(i)}] / [c * P}}_{\text{n}}\text{(i)]}$$

a current estimate SNR (i) of the signal / noise power ratio of the speech signal x (i) is determined. In the normal case, the product c * P _n (i) is used to estimate the current power of the noise signal component, and the difference ${\text{P}}_{\text{x, p}}{\text{(i) -c * P}}_{\text{n}}\text{(i)}$

is used to estimate the current power of the speech signal component of the speech signal x (i). The current power of the speech signal is estimated by the smoothed power value P _{x, s} (i). The weighting with a scaling factor c prevents P _n (i) from estimating the noise signal power with a value that is too small. The scaling factor c is typically in the range from 1.3 to 2. The minimum formation in block 41 or equation (4) ensures that the non-logarithmic signal / noise power ratio SNR (i) is also positive if in exceptional cases c * P _n (i) is greater than P _{x, s} (i). Then the power of the noise signal component of the voice signal is set equal to the power of the voice signal estimated by P _{x, s} (i). The power of the speech signal component of the speech signal estimated by P _{x, s} (i) -P _{x, s} (i) is then equal to zero, as is the non-logarithmic signal / noise power ratio. After the calculation of the estimated value SNR (i), the program continues with the reading in of a new speech signal sample value x (i) by block 34.

die Komponenten eines Vektors minvec der Dimension W aktualisiert. Danach wird durch Verzweigung 43 abgefragt, ob die Komponenten minvec₁ bis minvec_w mit ansteigendem Vektorindex ansteigen, d.h. ob gilt: $${\text{minvec}}_{\text{j+1}}{\text{> minvec}}_{\text{j}}\text{für 1 ≦ j ≦ W-1}$$

Wird die Abfrage von Verzweigung 43 verneint, d.h. die zuletzt ermittelten in den Komponenten des Vektors minvec stehenden zuletzt ermittelten W Minima steigen nicht monoton an, wird durch Block 44 nach $${\text{P}}_{\text{n}}{\text{(i) = min{minvec}}_{\text{w}}{\text{, minvec}}_{\text{w-1}}\text{, ... , minvec₁}}$$

der vorläufige Schätzwert P_n(i) der Rauschsignalleistung aus den Minima der Komponenten des Vektors minvec, d.h aus dem Minimum der letzten $\text{L=W*M}$

aufeinanderfolgenden geglätteten Leistungswerte P_x,s(i), bestimmt. Bei einer Bejahung der durch Verzweigung 43 gestellten Frage, d.h. bei einem monotonen Ansteigen der zuletzt ermittelten in den Komponenten des Vektors minvec stehenden W Minima wird in Block 45 P_n(i) gleich P_Mmin gesetzt, so daß eine Anpassung der Abschätzung des Rauschsignalanteils beschleunigt erfolgt, da P_n(i) an dem Minimum des letzten (M < L) Werte bestimmt wird. Danach wird in Block 46 die Zählervariable Z wieder auf 0 gesetzt und P_Mmin erhält erneut den Wert P_max.If the query of branch 39 is answered in the affirmative, ie M smoothed sample values P _{x, s} (i) have been processed, in block 42 by

updated the components of a vector minvec of dimension W. Then it is queried by branch 43 whether the components minvec 1 to minvec _{w increase} with increasing vector index, ie whether: $${\text{minvec}}_{\text{j + 1}}{\text{> minvec}}_{\text{j}}\text{for 1 ≦ j ≦ W-1}$$

If the query of branch 43 is negated, ie the last W minima determined in the components of the vector minvec do not rise monotonously, block 44 follows $${\text{P}}_{\text{n}}{\text{(i) = min {minvec}}_{\text{w}}{\text{, minvec}}_{\text{w-1}}\text{, ..., minvec₁}}$$

the preliminary estimate P _n (i) of the noise signal power from the minima of the components of the vector minvec, ie from the minimum of the last $\text{L = W * M}$

successive smoothed power values P _{x, s} (i). If the question posed by branch 43 is answered in the affirmative, ie if the W minima found last in the components of the vector minvec increases monotonously, P _n (i) is set equal to P _Mmin in block 45, so that an adaptation of the estimation of the noise signal component is accelerated takes place since P _n (i) is determined at the minimum of the last (M <L) values. Then in block 46 the counter variable Z is reset to 0 and P _Mmin again receives the value P _max .

aufeinanderfolgenden geglätteten Leistungswerten P_x,s(i) W aufeinanderfolgende Untergruppen zusammengefaßt. Die Gruppen mit jeweils L Werten folgen lückenlos aufeinander und überlappen sich jeweils mit L-M gelätteten Leistungen P_x,s(i).Through the described program, M successive smoothed P _{x, s} (i) samples x (i) of the speech signal x are combined into a subgroup. Within such a subgroup, the minimum of the smoothed power values P _{x, s} (i) is determined by the operations carried out with branch 37 and block 38. The W minima determined last are stored in the components of the vector minvec. If the last W minima are not monotonically increasing (see branch 43), then a preliminary estimate P _n (i) of the power of the noise signal component is determined from the minimum of the minima of the last W subgroups, ie from the minimum of a group, according to block 44. They are each used to form a group $\text{L = W * M}$

successive smoothed power values P _{x, s} (i) W summarized successive sub-groups. The Groups with L values follow each other without gaps and overlap with LM smoothed powers P _{x, s} (i).

In the event that the minima of W successive subgroups increase monotonously (see branch 43), the minimum of the last subgroup with M smoothed power values P _x is determined by block 45 to estimate the current estimated value P _n (i) of the power of the noise signal component _{. s} (i) used. This shortens the time period with which monotonically increasing smoothed power values P _{x, s} (i) also cause a change in the estimated values SNR (i).

geglätteten Leistungswerten P_x,s(i) zusammengefaßt. Nach jeweils M geglätteten Leistungen P_x,s(i) wird aus dem Minimum der letzten W Untergruppenminima bzw. der letzten L geglätteten Leistungswerte P_x,s(i) der Wert P_n(i) bestimmt, der zur Abschätzung der Rauschsignalleistung dient. In Fig. 8 sind acht Gruppen mit jeweils L Abtastwerten x(i) dargestellt, die jeweils W = 4 Untergruppen mit M geglätteten Leistungswerten P_x,s(i) enthalten. Die acht Gruppen überlappen sich teilweise. So enthalten zwei aufeinanderfolgende Gruppen jeweils L-M gleiche geglättete Leistungswerte P_x,s(i). Auf diese Weise wird ein guter Kompromiß zwischen dem erforderlichen Rechenaufwand und der jeweiligen Verzögerungszeit erreicht, mit der eine Aktualisierung eines Schätzwertes P_n(i) der Rauschsignalleistung zur Aktualisierung eines Schätzwertes SNR(i) des Signal/ Rauschleistungsverhältnisses erfolgt. Eine Realisierung mit aneinandergrenzenden, d.h. sich nicht überlappenden Gruppen ist auch denkbar. Allerdings ist dann bei verringertem Rechenaufwand die Zeitspanne zwischen zwei Schätzwerten SNR(i) vergrößert, so daß die Reaktionszeit auf sich ändernde SNR des Sprachsignals x(i) vergrößert ist.8 illustrates how the smoothed power values P _{x, s are combined} in groups and subgroups. In each case, M smoothed power values P _{x, s} (i), which are present at sampling times i, are combined into a subgroup. The subgroups are contiguous. The minimum of the smoothed power values P _{x, s} (i) is determined for each subgroup. W subgroup minima are stored in the vector minvec. As a rule, ie with non-monotonically increasing W subgroups minima, W subgroups become a group with $\text{L = W * M}$

smoothed power values P _{x, s} (i) summarized. After M smoothed powers P _{x, s} (i), the value P _n (i) is determined from the minimum of the last W subgroup minima or the last L smoothed power values P _{x, s} (i), which is used to estimate the noise signal power. 8 shows eight groups each with L samples x (i), each of which contains W = 4 subgroups with M smoothed power values P _{x, s} (i). The eight groups partially overlap. Two successive groups each contain the same smoothed power values P _{x, s} (i). In this way, a good compromise is achieved between the required computational effort and the respective delay time with which an update of an estimated value P _n (i) of the noise signal power takes place in order to update an estimated value SNR (i) of the signal / noise power ratio. A realization with adjacent, ie non-overlapping groups is also conceivable. However, the time span between two estimated values SNR (i) is then increased with a reduced computing effort, so that the reaction time to changing SNR of the speech signal x (i) is increased.

The described speech processing device thus has an estimation device which is suitable for the continuous formation of estimated values SNR (i) of the signal / noise power ratio of noisy speech signals x (i). In particular, no speech pauses are required to estimate the noise signal power. The estimation device described uses the special time profile of smoothed power values of the speech signal x (i), which is characterized by peaks and intermediate areas with smaller smoothed power values P _{x, s} (i), their temporal expansion from the respective speech source, ie the respective speaker , depends. The areas between the peaks are used to estimate the power of the noise signal component. The groups with L smoothed power values P _{x, s} (i) must follow one another without gaps, ie they must either adjoin or overlap. Furthermore, it must be ensured that at least one value of an area lying between two peaks with smaller smoothed power values P _{x, s} (i) can be recorded by each group, ie each group must contain so many smoothed power values P _{x, s} (i) that at least all values belonging to any peak can be recorded. Since the most extended peaks can be estimated by the most extended phonemes of a speech signal, ie the vowels, the number L describing the group size can be derived from this. For a sampling rate of the speech signal of 8 kHz, a useful value of L is in the range between 3000 and 8000. An advantageous value for W is 4. With such a dimensioning, there is a good compromise between the computational effort and the speed of reaction of the function block 7.

FIG. 9 shows a use of the voice processing device from FIG. 3 in a mobile radio terminal 50. The language processing means 20 to 26 are combined in a function block 51, which forms the sum signal values X (i) from the microphone or speech signals generated by the microphones M1, M2 and M3. The microphones M1, M2 and M3 advantageously have a distance of 10 to 60 cm, so that in a so-called "reverberated" environment (eg car, office) the interference signal components of the speech signals supplied by the microphones M1, M2 and M3 are largely uncorrelated. This also applies when only two microphones are used as in FIG. 1. A function block 52 which processes the sum signal values X (i) combines all the other means of the mobile radio terminal 50 for receiving, processing and transmitting signals which are used for communication with a base station (not shown) serve, the transmission and reception of signals via an antenna 54 coupled to the function block 52. Furthermore, a loudspeaker 53 coupled to the function block 52 is provided. A user (speaker, listener) communicates acoustically with the mobile radio terminal 50 via the microphones M1 to M3 and the loudspeaker 53, which are part of a hands-free device integrated in the mobile radio terminal 50. The use of such a mobile radio terminal 50 is particularly advantageous in motor vehicles, since there the hands-free communication via the mobile radio terminal is particularly disturbed by engine or driving noise (noise).

Claims (8)

Mobile radio terminal with a speech processing device for processing a first (x2 (i)) and at least one further (x1 (i), x3 (i)) speech signals consisting of noise and speech signal components and present as samples with delay means (4, 23, 24) Delay of the sampled further speech signal (x1 (i), x3 (i)), with control means (3, 26) - to form gradient estimates (grad (i), sgrad (i)) by multiplying error values (e₁₂ (i), e₃₂ (i), e₁₃ (i), e₃₁ (i)) for two speech signals (e.g. x1 (i) and x2 (i)) with the output values of a digital filter (6), which causes a phase shift of 90 degrees and is used to filter one of the two speech signals (for example x2 (i)), - For the recursive determination of delay estimated values (T1 '(i), T3' (i)) from the gradient estimated values (grad (i), sgrad (i)), with the delay estimated values (T1 '(i), T3' (i) ) the delay values (T2 (i), T3 (i)) are formed by rounding, which serve to set the delay means (4, 23, 24) and - To form at least one error value (e₁₂ (i), e₃₂ (i), e₁₃ (i), e₃₁ (i)) for a certain sampling time (i) from the difference between a speech signal estimate (x1 _int (i), x3 _int (i)), which is used to estimate the further speech signal (x1 (i), x3 (i)) at a point in time shifted by the delay estimated value (T1 '(i), T3' (i)) compared to the determined sampling time (i) and is formed by interpolating samples of the further speech signal (x1 (i), x3 (i)) and the sample of another of the speech signals (x1 (i), x2 (i), x3 (i)) to be processed to the determined one Sampling time (i) and with an adding device (5, 25) for adding the mutually time-shifted speech signals (x1 (i), x2 (i), x3 (i)). Mobile radio terminal according to claim 1,
characterized by
that the digital filter (6) is a digital Hilbert transformer. Mobile radio terminal according to claim 2,
characterized by
that means (12) for smoothing the gradient estimates (grad (i)) are provided. Mobile radio terminal according to one of claims 1 to 3,
characterized by
that the speech processing device is provided for processing three speech signals (x1 (i), x2 (i), x3 (i)). Mobile radio terminal according to one of claims 1 to 4,
characterized by
that to determine a delay estimate (T1 '(i), T3' (i)) for the further speech signal (x1 (i), x3 (i)) the use of a linear combination of error values (e₁₂ (i) with e₁₃ (i), e₃₁ (i) with e₃₂ (i)) is provided. Mobile radio terminal according to one of claims 1 to 5,
characterized by
that delay means (16, 27) are provided for delaying the first speech signal (x2 (i)) with a fixed delay time (T _max ). Mobile radio terminal according to one of claims 1 to 6,
characterized by
that the speech processing device is integrated into a hands-free device (M1, M2, M3, 51, 52, 53). Speech processing device for processing a first (x2 (i)) and at least one further (x1 (i), x3 (i)) consisting of noise and speech signal components and present as samples speech signals with delay means (4, 23, 24) for delaying the sampled further speech signal (x1 (i), x3 (i)) and with control means (3, 26) - to form gradient estimates (grad (i), sgrad (i)) by multiplying error values (e₁₂ (i), e₃₂ (i), e₁₃ (i), e₃₁ (i)) for two speech signals (e.g. x1 (i) and x2 (i)) with the output values of a digital filter (6), which causes a phase shift of 90 degrees and is used to filter one of the two speech signals (for example x2 (i)), - For the recursive determination of delay estimated values (T1 '(i), T3' (i)) from the gradient estimated values (grad (i), sgrad (i)), with the delay estimated values (T1 '(i), T3' (i) ) by rounding to integer multiples of a sampling interval of the speech signal samples (x1 (i), x2 (i), x3 (i)) the delay values (T2 (i), T3 (i)) are formed, which are used to set the delay means (4, 23, 24) serve and - To form at least one error value (e₁₂ (i), e₃₂ (i), e₁₃ (i), e₃₁ (i)) for a certain sampling time (i) from the difference between a speech signal estimate (x1 _int (i), x3 _int (i)), which is used to estimate the further speech signal (x1 (i), x3 (i)) compared to the one determined Sampling time (i) serves the time shifted by the delay estimated value (T1 '(i), T3' (i)) and is formed by interpolation of samples of the further speech signal (x1 (i), x3 (i)) and the sample value of another the speech signals to be processed (x1 (i), x2 (i), x3 (i)) at the determined sampling time (i) and with an adding device (5, 25) for adding the mutually time-shifted speech signals (x1 (i), x2 (i), x3 (i)).

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