CN116054854A - Low-complexity phased array self-interference digital domain suppression method - Google Patents
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Abstract
The invention discloses a low-complexity phased array self-interference digital domain suppression method, which comprises the following steps: s1, a baseband transmitting signal s (n) is converted into digital and analog and then is transmitted by a beam shaper a t Weighted by M t Transmitting by a local transmitting array antenna formed by isotropic array elements; s2, transmitting signals pass through a cross coupling channel H SI Coupled to the local receive array antenna to form self-interference; s3, contain M r The signals received by the local receiving array antennas of the receiving array elements are transmitted by the receiving beam shaper a r And (3) weighting and combining, obtaining a baseband receiving signal y (n) after analog-to-digital conversion, and carrying out self-interference reconstruction and self-interference digital domain suppression. The invention can realize the switching of the wave beam by multiplying a fixed weight matrix after estimating the self-interference reconstruction coefficient at the beginning direction of the wave beamAnd then directly updating the self-interference reconstruction coefficient to realize stable self-interference cancellation performance with lower complexity.
Description
Technical Field
The invention relates to phased array self-interference suppression, in particular to a low-complexity phased array self-interference digital domain suppression method.
Background
Phased array systems are gradually moving toward high integration. On the same platform, a plurality of phased arrays with different functions such as reconnaissance, interference, communication and the like are generally integrated to meet the multi-aspect functional requirements of the platform. How to solve the problem of electromagnetic interference between different functional phased arrays is one of the key challenges faced by current multi-phased array integrated platforms. The traditional half-duplex technology divides channel resources into time domain, frequency domain, code domain, space domain or the combination thereof, and avoids the problem of electromagnetic interference through different channel access modes. However, in the environment of lack of channel resources today, the lower channel resource utilization of the conventional half duplex technology obviously cannot meet the throughput rate of the multi-phased array integrated platform. Therefore, the simultaneous co-frequency transceiving technology becomes one of the effective ways to solve the electromagnetic compatibility requirement of the integrated platform with high frequency spectrum efficiency
Self-interference cancellation is the key for realizing the simultaneous same-frequency transceiving of the phased array. In general, the self-interference power of electromagnetic coupling between the transmit-receive phased arrays is very high, if cancellation is not performed, the expected signal can be submerged, so that the expected signal cannot be correctly demodulated, and the receive phased array radio frequency front end can be blocked, so that the receive phased array radio frequency front end cannot work normally. The existing self-interference cancellation technology can be divided into propagation domain cancellation, analog domain cancellation and digital domain cancellation according to different processing domains. The main purpose of the self-interference cancellation in the propagation and analog domains is to reduce the power of the received self-interference and prevent the saturation of the receiving element. However, residual self-interference still needs to be suppressed to near the receiver noise floor via digital domain cancellation.
The studies of the self-interference digital domain suppression of phased arrays by the literature currently exist can be divided into two categories according to the different types of phased arrays. In the first type, in the digital phased array, each transmitting antenna is provided with an observation channel, so that a reference signal sampled by the front end of the radio frequency is output to realize digital self-interference cancellation. However, this self-interference cancellation structure based on a digital phased array requires one observation channel for each transmitting element, which brings about huge resource consumption for the system in terms of size, weight, complexity and power consumption, especially for large phased arrays. In the second category, in analog or hybrid phased arrays, the reference signal is baseband coupled from the transmitter to achieve digital self-interference cancellation. The reference signal acquisition mode has certain advantages in terms of complexity and engineering implementation difficulty. Unfortunately, in such a scenario, the self-interference cancellation technique based on channel estimation adopted at present is still very complex in the scenario of the abundant number of multipaths, such as phased array electromagnetic coupling. How to further reduce the complexity of the self-interference cancellation algorithm without losing the self-interference cancellation performance is yet to be studied.
On the other hand, the above-described studies on phased array self-interference digital domain suppression generally assume that the beam pointing is fixed. However, phased arrays all have scanning behavior, in which case the widely used adaptive self-interference cancellation technique faces two main problems. Firstly, the self-interference propagation channel changes too fast due to the change of the beam direction, so that the self-interference propagation channel cannot be tracked in time, and the instantaneous performance of self-interference cancellation can be greatly reduced. Second, when the beam direction changes, the self-interference cancellation performance needs to converge again, resulting in an undesirable self-interference cancellation effect during the convergence time. When the convergence time is longer than the beam pointing duration, digital self-interference cancellation will be ineffective and even introduce additional interference.
Disclosure of Invention
The invention aims to overcome the defects of the prior art, and provides a low-complexity phased array self-interference digital domain suppression method which can realize direct updating of the self-interference reconstruction coefficient after beam switching by multiplying a fixed weight matrix after estimating the self-interference reconstruction coefficient at the initial direction of the beam and realize stable self-interference cancellation performance with lower complexity.
The aim of the invention is realized by the following technical scheme: a low-complexity phased array self-interference digital domain suppression method comprises the following steps:
s1, in the case of beam scanning, makingDifferent beam directives (wave bits) are distinguished by the mark p, and the baseband transmitting signal s (n) is converted into digital-to-analog conversion and then transmitted by a transmitting beam shaper a t Weighted by M t Transmitting by a local transmitting array antenna formed by isotropic array elements;
s2, transmitting signals pass through a cross coupling channel H SI Coupled to the local receive array antenna to form self-interference;
s3, contain M r The signals received by the local receiving array antennas of the receiving array elements are transmitted by the receiving beam shaper a r And (3) weighting and combining, obtaining a baseband receiving signal y (n) after analog-to-digital conversion, and carrying out self-interference reconstruction and self-interference digital domain suppression.
The beneficial effects of the invention are as follows: according to the self-interference reconstruction method, after the self-interference reconstruction coefficient is estimated at the initial direction of the wave beam, the self-interference reconstruction coefficient can be directly updated after the wave beam is switched by multiplying a fixed weight matrix, stable self-interference cancellation performance is realized with lower complexity, no iteration process exists in self-interference cancellation, and the instant cancellation performance reduction caused by the wave beam switching in the traditional self-adaptive self-interference cancellation is avoided; the self-interference reconstruction coefficient is calculated by only one matrix multiplication update, so that the calculation complexity is low; the reference signal coupling self-transmitter baseband of the self-interference cancellation method is convenient to obtain and low in implementation complexity; the self-interference coupling channel information is not needed, and the realization is simple.
Drawings
FIG. 1 is a flow chart of the method of the present invention;
FIG. 2 is a schematic block diagram of a self-interference digital domain suppression system of the present invention;
fig. 3 is a flow chart of self-interference digital domain suppression.
Detailed Description
The technical solution of the present invention will be described in further detail with reference to the accompanying drawings, but the scope of the present invention is not limited to the following description.
As shown in fig. 1, a low complexity phased array self-interference digital domain suppression method includes the following steps:
s1, in the case of beam scanning, the different beams are distinguished using the mark pThe baseband transmitting signal s (n) is digital-to-analog converted and then transmitted by the beam shaper a t Weighted by M t Transmitting by a local transmitting array antenna formed by isotropic array elements;
fig. 2 is a schematic block diagram of a self-interference digital domain suppression system according to the present invention in an embodiment of the present application. In the case of beam scanning, we use the marker p to distinguish between different beam orientations. In the transmitting chain, the baseband transmitting signal s (n) is converted into digital-to-analog conversion and then transmitted by a transmitting beam shaper a t Weighted and made up of M t And emitting by the root isotropic array element. Without loss of generality, it is assumed that the transceiver arrays are all uniform rectangular arrays. When the wave beam points to the p-th wave bit, the baseband equivalent model of the transmitting array element transmitting signals of the m-th row and the n-th column of the local transmitting array antenna can be expressed as
wherein ,
is the transmit beamforming weight vector +.>
Is a component of the group. Consider non-adaptive beamforming, +.>
Can be written as +.>
wherein ,
represents the Kronecker product, M t x and Mt y Representing the number of transmitting array elements along the x-axis and the y-axis, respectively, assuming a spacing of +.>
Wavelength lambda->
Indicating the normalized inter-element distance, furthermore, < >>
and />
Is defined as
wherein ,
and />
Is the azimuth and pitch angle of the transmit array at the p-th wave position. Thus (S)>
Can be expressed as
S2, transmitting signals pass through a cross coupling channel H SI Coupled to the local receive array antenna to form self-interference;
the transmitted signal passes through the cross-coupling channel H SI Coupled to the local receive array, forming self-interference. Physical spatial placement of pseudo-local transmit and receive array antennas spaced apart along the y-axis
Spacing>
The included angle is theta; the distance between the (m, n) th transmitting array element and the (u, v) th receiving array element is
wherein 
Is a wavelength normalization value;
considering that the transmit and receive arrays are typically on the same platform, self-interference propagation is near-field propagation. Thus, we use spherical wave models to characterize line-of-sight propagation between transmitting and receiving elements. Then, the mutual coupling channel gain of the transmitting array element of the mth row and the nth column of the local transmitting array antenna and the receiving array element of the mth row and the nth column of the local receiving array antenna can be written as
Wherein, kappa is to ensure 10lg 10 E{||H SI ||}＝-P h,dB And the normalized coefficient, P, of the call h,dB Representing spatial isolation in dB. Thus, the coupling self-interference from the transmitting array element of the mth row and the nth column of the local transmitting array antenna to the receiving array element of the mth row and the nth column of the local receiving array antenna when the wave beam points to the p-th wave bit can be written as
wherein ,τ(m,n)→(u,v) Representing the normalized propagation delay from the (m, n) th transmitting element to the (u, v) th receiving element. In general, the propagation delay τ (m,n)→(u,v) By delays of integer multiples of the sampling period
Delay of sum fractional times of sampling period +.>
Composition is prepared. S (n- τ) according to an ideal fractional delay filter model (m,n)→(u,v) ) Can be expressed as
wherein ,
is an ideal fractional delay filter with a fractional delay of +.>
Coefficient at the time, defined as
It can be seen that (9) characterizes the relationship between a signal with fractional delay and a signal with integer delay. With pairs of rectangular windows of length 2M+1
After windowing, (9) can be rewritten as
wherein ,DI max Is the maximum integer propagation delay between two transceiver element pairs, M characterizes the early arrival time of the reference signal relative to the received self-interference. Thus, the path signal vector s (n) and the tap signal vector s tap (n) has the following mapping relation
s(n)＝P T s tap (n) (12)
Wherein the mapping matrix
Is defined as
P＝[p -M p -M+1 … p -M+L-1 ] T (13)
wherein
Tap sample vector s tap Is based on integer sampling period, defined as
s tap (n)＝[s(n+M) s(n+M-1) … s(n+M-L+1)] T (15)
S3, contain M r The signals received by the local receiving array antennas of the receiving array elements are transmitted by the receiving beam shaper a r And (3) weighting and combining, obtaining a baseband receiving signal y (n) after analog-to-digital conversion, and carrying out self-interference reconstruction and self-interference digital domain suppression.
At the receiving array plane, M r The received signal of the root receiving array element is received by the beam shaper a r Weighted combination, analog-to-digital conversion to obtain baseband received signal y (n) which is obtained by self-interference y SI (n) and far-end desired signal y DS (n) composition. Therefore, the baseband received self-interference when the beam is directed to the p-th wave bit can be written as
wherein ,
representing the path coefficient vector, s (n) representing the path signal vector, respectively defined as
Also consider the receive beamforming to be non-adaptive beamforming, then
And->
In the same form. Combining (12) and (16), the tap model received from the disturbance can be written
For ease of analysis, we will N at the p-th wave position s The self-interference sample vector received during a sampling period is expressed as
Based on the above system model->
Can be expressed as
wherein ,
furthermore, the->
Defined as->
Tap sample matrix S tap Is defined as
S tap ＝[s tap (n) s tap (n-1) … s tap (n-N s )] T (22)
Similarly, the self-interference model at the p-1 st wave position can be written
Wherein Λ is a diagonal matrix representing the path coefficient w at time p-1 p-1 Path coefficient w to time p p Is used to change the amount of change in (a). Normally self-interference coupling path gain H SI Invariable, Λ is mainly caused by the change of the wave bit direction, which we call the shaping coefficient increment matrix. Considering non-adaptive beamforming, the diagonal elements of Λ may be written as
wherein
The optimal self-interference reconstruction coefficients can be written under the criterion of minimizing the residual interference power
Under the condition of P full rank, the method can be further written
It can be seen that if it has been estimated that
Then can be from->
Update direct get +.>
In other words. The reconstruction coefficients at the p-th beam position may be calculated from the reconstruction coefficients at the p-1 st beam position without the need for information of the self-interference coupled channel. Further, considering non-adaptive beamforming, pΛ (P H P) -1 P H May be pre-computed for invocation. Therefore, the implementation of (27) requires L 2 This requires fewer complex multiplication operations than are required by conventional RLS algorithms. Thus, this approach is referred to as a low complexity phased array self-interference digital domain suppression approach.
Specifically, on the basis of estimating the optimal self-interference reconstruction coefficient at the initial wave position, the self-interference reconstruction coefficient at the next wave position is updated through simple matrix multiplication, so that stable phased array self-interference digital domain suppression is realized with lower complexity under the condition of wave beam scanning. Firstly, obtaining initial wave position self-interference reconstruction coefficients at the beginning direction of a wave beam through a traditional coefficient estimation algorithm
Then, an update weight matrix pΛ (P H P) -1 P H The method comprises the steps of carrying out a first treatment on the surface of the And finally, obtaining an optimal self-interference reconstruction coefficient under the current wave position according to a formula (27), and performing self-interference digital domain suppression. The flow chart of the method is shown in fig. 3, and the specific flow steps are as follows:
a1: in an offline module, calculating a mapping matrix P according to the actual physical space placement position of the receiving and transmitting phased array;
a2: in the off-line module, the inverse of the mapping matrix autocorrelation is calculated (P H P) -1 ；
A3. If the current wave position is directed to be initialThe wave bit, namely p=1, is estimated by using the traditional LS coefficient estimation algorithm to obtain the initial wave bit self-interference reconstruction coefficient
I.e. < ->
Then directly jumping to A7;
a4: if the current wave bit direction is not the initial wave bit, namely p is not equal to 1, in the weight matrix calculation module, according to the wave beam forming coefficient information provided by the wave controller
And->
Calculating a beamforming coefficient increment matrix lambda; />
Beamforming coefficient delta matrix
Here, multiplication with/expression matrix elements and division of matrix elements, diagonal elements of Λ can be written in non-adaptive beamforming
A5: in the weight matrix calculation module, the updated weight matrix pΛ (P H P) -1 P H ；
A6: in the coefficient updating module, the formula is adopted
Calculating the self-interference reconstruction coefficient of the current wave position +.>
A7. By using the obtained
Reconstructing self-interference with the transmitted baseband information, i.e. +.>
And the self-interference obtained by subtracting the reconstruction from the baseband received signal y (n) is utilized to complete self-interference cancellation, so that self-interference digital domain suppression is realized.
While the foregoing description illustrates and describes a preferred embodiment of the present invention, it is to be understood that the invention is not limited to the form disclosed herein, but is not to be construed as limited to other embodiments, but is capable of use in various other combinations, modifications and environments and is capable of changes or modifications within the spirit of the invention described herein, either as a result of the foregoing teachings or as a result of the knowledge or skill of the relevant art. And that modifications and variations which do not depart from the spirit and scope of the invention are intended to be within the scope of the appended claims.
Claims (6)
1. A low-complexity phased array self-interference digital domain suppression method is characterized by comprising the following steps of: the method comprises the following steps:
s1, under the condition of beam scanning, different beam orientations are distinguished by using a mark p, and a baseband transmission signal s (n) is converted into a digital-to-analog conversion and then is transmitted by a transmission beam shaper a t Weighted by M t Transmitting by a local transmitting array antenna formed by isotropic array elements;
s2, transmitting signals pass through a cross coupling channel H SI Coupled to the local receive array antenna to form self-interference;
s3, contain M r The signals received by the local receiving array antennas of the receiving array elements are transmitted by the receiving beam shaper a r And (3) weighting and combining, obtaining a baseband receiving signal y (n) after analog-to-digital conversion, and carrying out self-interference reconstruction and self-interference digital domain suppression.
2. The low complexity phased array self-interference digital domain suppression method of claim 1, wherein: the local transmitting array antenna and the local receiving array antenna are uniform rectangular array antennas.
3. A low complexity phased array self-interference digital domain suppression method according to claim 2, characterized by: in the step S1, when the beam is directed to the p-th wave position, the baseband equivalent model of the transmitting signals of the transmitting array elements of the m-th row and the n-th column of the local transmitting array antenna is expressed as
wherein ,
is the transmit beamforming weight vector +.>
In consideration of non-adaptive beamforming, +.>
Expressed as:
wherein ,
represents the Kronecker product, +.>
and />
Representing the number of transmitting array elements along the x-axis and the y-axis, respectively, assuming a spacing of +.>
Wavelength lambda->
Representing normalized inter-element distances; in addition, in the case of the optical fiber,
and />
Is defined as
wherein ,
and />
Is the azimuth and pitch angle of the transmitting array at the p-th wave position, thus +.>
Expressed as:
4. a low complexity phased array self-interference digital domain suppression method according to claim 2, characterized by: the step S2 includes:
s201, arranging physical space of local transmitting and receiving array antennas at intervals along y-axis
Spacing>
The included angle is theta; the distance between the (m, n) th transmitting array element and the (u, v) th receiving array element is
wherein 
Is a wavelength normalization value;
the local transmitting and receiving array antennas are arranged on the same platform, self-interference propagation is near-field propagation, spherical wave model is used for representing line-of-sight propagation between transmitting and receiving elements, and the gain of a mutual coupling channel between the transmitting array element of the mth row and the nth column of the local transmitting array antenna and the receiving array element of the mth row and the nth column of the local receiving array antenna is as follows:
wherein, kappa is to ensure 10lg 10 E{||H SI ||}＝-P h,dB And the normalized coefficient, P, of the call h,dB Represents the spatial isolation in dB;
s202, when the wave beam points to the p-th wave bit, calculating the coupling self-interference from the transmitting array element of the m-th row and n-th column of the local transmitting array antenna to the receiving array element of the u-th row and v-th column of the local receiving array antenna as
wherein ,τ(m,n)→(u,v) Representing the normalized propagation delay from the (m, n) th transmitting element to the (u, v) th receiving element, propagation delay τ (m,n)→(u,v) By delays of integer multiples of the sampling period
Delay of sum fractional times of sampling period +.>
Composition, s (n- τ) according to an ideal fractional delay filter model (m,n)→(u,v) ) Represented as
wherein ,
is an ideal fractional delay filter with a fractional delay of +.>
Coefficient at the time, defined as
With pairs of rectangular windows of length 2M+1
After windowing, obtain
wherein ,
is the maximum integer propagation delay between two transceiver element pairs, M characterizes the early arrival time of the reference signal relative to the received self-interference, and thus the path signal vector s (n) and the tap signal vector s tap (n) has the following mapping relation
s(n)＝P T s tap (n)
Wherein the mapping matrix
Is defined as
P＝[p -M p -M+1 … p -M+L-1 ] T
wherein
Tap sample vector s tap Is based on integer sampling period, defined as
s tap (n)＝[s(n+M) s(n+M-1) … s(n+M-L+1)] T 。
5. A low complexity phased array self-interference digital domain suppression method according to claim 2, characterized by: the step S3 includes:
s301 comprises M r The signals received by the local receiving array antennas of the receiving array elements are transmitted by the receiving beam shaper a r Weighting and combining, and obtaining a baseband receiving signal y (n) after analog-to-digital conversion;
s302, setting the baseband receiving signal to be formed by self-interference y SI (n) and far-end desired signal y DS (n) baseband received self-interference when beam is directed to the p-th wave position as
wherein ,
representing a path coefficient vector, s (n) representing a path signal vector, defined as:
s303, considering the received beam forming to be non-adaptive beam forming, then
And->
In the same form, the tap model received from the disturbance is expressed as:
will N at the p-th wave position s The self-interference sample vector received during a sampling period is expressed as
wherein ,
furthermore, the->
The definition is as follows:
tap sample matrix S tap Is defined as
S tap ＝[s tap (n) s tap (n-1) … s tap (n-N s )] T
Writing the self-interference model at the p-1 wave position
Wherein Λ is a diagonal matrix representing the path coefficient w at time p-1 p-1 Path coefficient w to time p p Is set from the change of the interference coupling path gain H SI Invariably, Λ is caused by a change in the wave position direction, called the shaping coefficient delta matrix, considering non-adaptive beamforming,
diagonal meta-writing of Λ
wherein
S304, under the criterion of minimizing residual interference power, determining the optimal self-interference reconstruction coefficient as
Under the condition of P full rank, obtain
If it has estimated
Then from->
Update direct get +.>
I.e. the reconstruction coefficients at the p-th beam position are calculated from the reconstruction coefficients at the p-1 st beam position and no information of the self-interference coupling channel is needed;
considering non-adaptive beamforming, pΛ (P H P) -1 P H And (5) performing pre-calculation for calling, and completing self-interference reconstruction and self-interference digital domain suppression.
6. The low complexity phased array self-interference digital domain suppression method of claim 5, wherein: the self-interference reconstruction and self-interference digital domain suppression process comprises the following steps:
A1. calculating a mapping matrix P according to the actual physical space placement position of the receiving and transmitting phased array;
A2. the inverse of the mapping matrix autocorrelation is calculated (P H P) -1 ；
A3. If the current wave bit direction is the initial wave bit, namely p=1, the initial wave bit self-interference reconstruction coefficient is obtained by utilizing the traditional LS coefficient estimation algorithm
I.e. < ->
Then directly jumping to A7;
A4. if the current wave bit direction is not the initial wave bit, i.e. p +.1, according to the information of the wave beam forming coefficient
And->
Calculating a beamforming coefficient increment matrix lambda;
beamforming coefficient delta matrix
Here, multiplication with/expression matrix elements and division of matrix elements, diagonal elements of Λ can be written in non-adaptive beamforming
A5. An update weight matrix pΛ (P H P) -1 P H ；
A6. According to the formula
Calculating the self-interference reconstruction coefficient of the current wave position +.>
A7. By using the obtained
Reconstructing self-interference with the transmitted baseband information, i.e. +.>
And the self-interference obtained by subtracting the reconstruction from the baseband received signal y (n) is utilized to complete self-interference cancellation, so that self-interference digital domain suppression is realized. />
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