CN114938252A - Bidirectional interference suppression method applied to multi-user filtering multi-sound underwater acoustic communication - Google Patents

Bidirectional interference suppression method applied to multi-user filtering multi-sound underwater acoustic communication Download PDF

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CN114938252A
CN114938252A CN202210551936.8A CN202210551936A CN114938252A CN 114938252 A CN114938252 A CN 114938252A CN 202210551936 A CN202210551936 A CN 202210551936A CN 114938252 A CN114938252 A CN 114938252A
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CN114938252B (en
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李海森
孙琳
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Harbin Engineering University
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B13/00—Transmission systems characterised by the medium used for transmission, not provided for in groups H04B3/00 - H04B11/00
    • H04B13/02—Transmission systems in which the medium consists of the earth or a large mass of water thereon, e.g. earth telegraphy
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
    • H04L25/03082—Theoretical aspects of adaptive time domain methods
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2689—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation
    • H04L27/2691—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation involving interference determination or cancellation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03592—Adaptation methods
    • H04L2025/03598—Algorithms
    • H04L2025/03611—Iterative algorithms
    • H04L2025/03649—Algorithms using recursive least square [RLS]
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
    • Y02D30/00—Reducing energy consumption in communication networks
    • Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks

Abstract

The invention discloses a bidirectional interference suppression method applied to multi-user filtering multi-sound underwater acoustic communication, which comprises the following steps: the method adopts a bidirectional parallel interference suppression structure to process co-channel interference and intersymbol interference in multi-user filtering multi-tone underwater acoustic communication, a forward structure is used for processing a sub-band signal after filtering multi-tone demodulation, and a reverse structure is used for processing a time reversal form of the sub-band signal. The forward and reverse structures of the method adopt time reversal technology for preprocessing, adopt continuous interference cancellation and decision feedback equalization for post processing, and combine the results of the two-way processing to obtain the final decision value of the user sequence. The invention has the beneficial effects that: extra diversity gain is obtained through bidirectional processing, error propagation can be avoided, lower error rate and output mean square error than those of a traditional one-way interference suppression method can be obtained, and reliability is higher.

Description

Bidirectional interference suppression method applied to multi-user filtering multi-sound underwater acoustic communication
Technical Field
The invention relates to the technical field of multi-user underwater acoustic communication, in particular to a method which can avoid error propagation and can improve the performance of co-channel interference and intersymbol interference suppression in multi-user filtering multi-sound underwater acoustic communication through extra diversity gain obtained by bidirectional processing.
Background
The multi-carrier modulation is used in multi-user underwater acoustic communication, so that the code element range influenced by intersymbol interference (ISI) caused by underwater acoustic channel multipath expansion and co-channel interference (CCI) caused by channel correlation can be reduced, the communication performance is improved, and the complexity of interference suppression is reduced. Orthogonal Frequency Division Multiplexing (OFDM) is a multicarrier technology that is most widely studied and applied in the field of underwater acoustic communications. However, OFDM sub-bands overlap each other and are sensitive to frequency offset, and complex frequency offset estimation and compensation techniques need to be employed. Compared with OFDM, multi-user filtered multi-tone (FMT) can be implemented not only by dividing a symbol range of sub-bands to reduce CCI and ISI effects and by fast fourier transform, but also is not sensitive to frequency offset because passbands of FMT sub-bands do not overlap each other and have strong spectral suppression, and thus has been recently applied to multi-user underwater acoustic communications. In multi-user FMT underwater acoustic communication, although the code element range influenced by CCI and ISI is reduced by dividing sub-frequency bands, because the bandwidth of the sub-frequency bands is still larger than the relevant bandwidth of a channel, shortened CCI and ISI still exist in a signal demodulated by a receiving end FMT, and the suppression is needed. Document [1] (Li H S, Sun L, Du W D, et al, multiple-input multiple-output communicating filtered multiple modulation [ J ] Applied Acoustics,2017,119(4):29-38.) proposes an interference suppression method combining continuous interference cancellation and adaptive equalization, which has a good effect, but the used equalizer is a linear equalizer, so its interference suppression performance is limited. Theoretically, if the decision value that the equalizer has outputted is accurate, replacing the linear equalization in the method of document [1] with a Decision Feedback Equalizer (DFE) will achieve better performance. However, DFEs are prone to error propagation problems when the error rate of the decided values is high.
Disclosure of Invention
Aiming at the defects of the prior art, the invention provides a bidirectional interference suppression method applied to multi-user filtering multi-sound underwater acoustic communication.
In order to realize the purpose, the technical scheme adopted by the invention is as follows:
a bidirectional interference suppression method applied to multi-user filtering multi-sound underwater acoustic communication comprises the following steps:
and at the receiving end of the multi-user FMT underwater acoustic communication, processing the sub-band signals after FMT demodulation by using a bidirectional interference suppression method. The forward structure is used to process the FMT demodulated sub-band signal and the reverse structure is used to process the time-reversed version of the sub-band signal. The forward and reverse structures of the method adopt Time Reversal (TR) technology for preprocessing, continuous interference cancellation and DFE are adopted for post-processing, and the final decision value of the user sequence is obtained by combining the results of the bidirectional processing.
Further, the bidirectional interference suppression method includes the following steps:
s1: in the forward interference suppression, firstly, preprocessing a sub-band signal after FMT demodulation by using a TR (transmitter-receiver) technology to realize pre-compression of CCI (coherent interference) and ISI (intersymbol interference), and then, performing post-processing on the CCI and ISI residual after TR by using an interference cancellation and DFE (differential frequency coefficient);
s2: in the reverse interference suppression, firstly, storing the sub-band signals after FMT demodulation according to the received signals by using a buffer, firstly, outputting the signals entering the buffer at last, and finally, performing time reversal processing on the signals entering the buffer at first, then, preprocessing the reversed FMT sub-band signals by using a TR (transmitter and receiver) technology, and performing post-processing by using interference cancellation and DFE (differential feedback) coefficients;
s3: storing the estimated value of the sub-band signal output by the DFE in the inversion processing according to the mode that the buffer is used for storing the signal output by the DFE in the inversion structure, the signal which firstly enters the buffer is output at last, and the signal which finally enters the buffer is output at first, performing time inversion processing, and performing equal gain combination on the processed signal and the sub-band signal output by the DFE in the forward processing;
s4: and judging the signals output by the equal gain combination, wherein the judgment value is used for estimating the interference in the next interference cancellation of the sub-band signals of other users.
Further, the interference cancellation and adaptive equalization in forward interference suppression and reverse interference suppression are performed sequentially and iteratively, and are stopped when the system performance is stable.
Further, S1 specifically includes:
the TR processed signal in the forward interference suppression is:
Figure BDA0003650388780000031
in the above formula, α is the user number, m is the sub-band number, q (α,γ),(m,m) (lT, nT) represents the m-th sub-band sequence d of the gamma user at the transmitting end γ,m (lT) and the mth TR output signal y in the α -th receiver forward structure α,m Combined channel response between (nT), d α,m (lT) denotes the symbol sequence transmitted by the mth user on the mth subband, d α,m (nT) represents d when l ═ n α,m The value of (lT), T denotes the symbol interval and N denotes the number of users.
The estimated values of CCI in the forward structure are:
Figure BDA0003650388780000032
in the above formula
Figure BDA0003650388780000033
Shows the subband sequence d after the bi-directional interference suppression process α,m (nT).
The signal after interference cancellation is:
Figure BDA0003650388780000034
q (α,α),(m,m) (lT, nT) represents q when γ ═ α (α,γ),(m,m) Value of (lT, nT), q (α,α),(m,m) (nT, nT) represents q when l ═ n (α,α),(m,m) The value of (lT, nT).
Residual ISI is suppressed using a DFE whose output estimates are:
Figure BDA0003650388780000041
where a (nT) and b (nT) represent the coefficients of the DFE feedforward and feedback filters, respectively, N 1 And N 2 Number of non-causal and causal taps for feedforward filters, N f Is the number of taps of the feedback filter,
Figure BDA0003650388780000042
representing the decision value, z, of the forward DFE α,m (nT) represents the signal of the interference cancellation output in the forward configuration.
Further, S2 specifically includes:
the signal processed by TR in the reverse interference suppression is expressed as:
Figure BDA0003650388780000043
in the above formula
Figure BDA0003650388780000044
M-th sub-band sequence d representing the y-th user at the transmitting end γ,m (lT) output signal of mth TR in the α -th receiver inverse structure
Figure BDA0003650388780000045
The combined channel response between the two channels,
Figure BDA0003650388780000046
indicating that the mth user transmits the sequence d on the mth sub-band γ,m The time reversed version of (lT) is,
Figure BDA0003650388780000047
indicating that the alpha-th user transmits the sequence d on the m-th sub-band α,m The time reversed version of (lT) is,
Figure BDA0003650388780000048
when l is equal to n
Figure BDA0003650388780000049
The value of (c).
The signal after the reverse interference cancellation is:
Figure BDA00036503887800000410
in the above formula
Figure BDA00036503887800000411
When gamma is alpha
Figure BDA00036503887800000412
The value of (a) is,
Figure BDA00036503887800000413
when l is equal to n
Figure BDA00036503887800000414
The value of (a) is,
Figure BDA00036503887800000415
decision value representing a sequence of subbands after a bi-directional interference suppression process
Figure BDA00036503887800000416
Time reversed version of (c).
The DFE outputs estimates as:
Figure BDA0003650388780000051
in the above formula
Figure BDA0003650388780000052
And
Figure BDA0003650388780000053
representing the coefficients of the DFE feedforward and feedback filters in the inversion process respectively,
Figure BDA0003650388780000054
represents the decision values of the DFE in an inverted configuration,
Figure BDA0003650388780000055
representing the output signal of the interference cancellation in the inverted configuration.
Further, the gain combination in S4 specifically includes:
Figure BDA0003650388780000056
in the above formula, rho represents a merging coefficient, and the value range of rho is 0 to 1,
Figure BDA0003650388780000057
representing the estimated value output after the forward structure processing,
Figure BDA0003650388780000058
representing DFE output estimates in an inverse configuration
Figure BDA0003650388780000059
Time reversed version of (3).
Compared with the prior art, the invention has the advantages that:
extra diversity gain is obtained through bidirectional processing, the problem that the DFE is easy to generate error propagation when the communication performance is poor can be avoided, the error rate and the output mean square error which are lower than those of the traditional one-way interference suppression method can be obtained, and the reliability is higher.
Drawings
FIG. 1 is a functional block diagram of multi-user FMT underwater acoustic communications with two-way interference suppression according to an embodiment of the present invention;
FIG. 2 is a schematic block diagram of FMT modulation and demodulation in accordance with an embodiment of the present invention;
FIG. 3 is a block diagram of a bi-directional interference suppression scheme according to an embodiment of the present invention;
FIG. 4 is a combined channel response graph after TR processing according to an embodiment of the present invention;
FIG. 5 is a signal-to-interference ratio diagram after TR processing in accordance with an embodiment of the present invention;
FIG. 6 is a graph of performance after TR processing in accordance with an embodiment of the present invention;
FIG. 7 is a comparison graph of the performance of the sub-bands of the bi-directional interference suppression method and the two uni-directional interference suppression methods according to the embodiment of the present invention;
fig. 8 is a comparison graph of the overall performance of the bi-directional interference suppression method and the two uni-directional interference suppression methods according to the embodiment of the present invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention will be further described in detail below with reference to the accompanying drawings by way of examples.
Architecture for a communication system
For the convenience of analysis, the principle of the proposed method is analyzed below by taking a communication system with 2-array elements for transmitting 2-channel user signals and 2-array elements for receiving as an example, and in practical application, the proposed method is applicable to the case of multi-array element transmission and multi-array element reception. Fig. 1 shows a schematic block diagram of the system, and the schematic block diagram of FMT modulation and demodulation in fig. 1 is shown in fig. 2.
As can be seen from fig. 1 and 2, the mth subband signal output by the pth FMT demodulation can be represented as
Figure BDA0003650388780000061
In the formula (1), d α,i (nT) represents the signal transmitted by the alpha array element in the ith sub-band, T represents the symbol interval, g t (nT c ) And g r (nT c ) Representing the discrete-time responses of the transmit and receive filters respectively,
Figure BDA0003650388780000062
discrete-time form, w, representing the response of the underwater acoustic channel between the alpha-transmitting and beta-receiving array elements β (kT c ) Representing the noise received by the beta-th receiving element, T c T/K denotes the interpolated symbol interval, and K denotes the multiple of the interpolation.
The FMT demodulated signal at the right end of equation (1) includes the signal transmitted on the mth sub-band, the intercarrier interference generated by other sub-band signals on the mth sub-band signal, and the noise 3 part. The sub-carriers of FMT are not overlapped with each other, have strong frequency spectrum suppression and are insensitive to frequency offset, and under the condition that the two sides of the transceiver are fixed, the influence of inter-carrier interference on the communication performance is very slight and can be ignored. When the power of the transmission signal is high, the influence of noise on the communication performance is small. The method focuses mainly on the suppression of CCI and ISI, and the communication structure in question is based on the fact that both the transmitting and receiving parties are fixed, thus ignoring the effects of noise and inter-carrier interference, equation (1) can be expressed as
Figure BDA0003650388780000071
As can be seen from equation (2), the discrete time response of the ith subchannel between the alpha-th transmitting array element and the beta-th receiving array element is
Figure BDA0003650388780000072
As can be seen from equation (3), since the channel response differs for each sub-band, the receiving end needs to process the signal output by FMT demodulation separately.
Forward interference suppression
Fig. 3 shows a schematic block diagram of bi-directional interference suppression, taking the mth subband sequence of the α -th user as an example. As can be seen from fig. 3, the method includes 2 parallel interference suppression structures, wherein the upper half of fig. 3 is a forward structure for demodulating the output sub-signal of FMT
Figure BDA0003650388780000073
Processing is carried out, the lower half is an inverse structure for demodulating a time-reversed version of the output sub-signal for FMT
Figure BDA0003650388780000074
And (6) processing.
As can be seen from FIG. 3, the signal processed by TR in the forward architecture can be represented as
Figure BDA0003650388780000075
In the formula (4), q (α,γ),(m,m) (lT, nT) represents the combined channel response after TR processing, and is expressed as
Figure BDA0003650388780000076
The formula (5) represents
Figure BDA0003650388780000077
For underwater acoustic channels
Figure BDA0003650388780000078
Time reversed version of (c).
Formula (4) can be further represented as
Figure BDA0003650388780000079
The right end of equation (6) has term 1 as the desired signal, and terms 2 and 3 represent the residual CCI and ISI after TR processing, respectively, and post-processing is required.
The proposed method employs interference cancellation to suppress the residual CCI. For interference cancellation, the CCI needs to be estimated first. Regardless of the error of the channel estimation, the estimated value of CCI in the forward structure is
Figure BDA0003650388780000081
In formula (7)
Figure BDA0003650388780000082
And representing the decision value of the sub-band sequence after the bidirectional interference suppression processing.
The signal after interference cancellation is
Figure BDA0003650388780000083
The proposed method suppresses residual ISI using a DFE which outputs an estimate of
Figure BDA0003650388780000084
In equation (9), a (nT) and b (nT) represent the coefficients of the DFE feedforward and feedback filters, respectively, N 1 And N 2 Number of non-causal and causal taps for feedforward filters, N f Is the number of taps of the feedback filter,
Figure BDA0003650388780000085
representing the decision values of the forward DFE.
Reverse interference suppression
The signal processed by TR in the reverse interference suppression can be expressed as
Figure BDA0003650388780000086
In the formula (10)
Figure BDA0003650388780000087
Representing a time-reversed version of the sequence transmitted by the gamma user on the ith subband,
Figure BDA0003650388780000088
represents the combined channel response after TR processing in the reverse structure, and has the expression
Figure BDA0003650388780000089
Due to the matched filtering characteristics of the TR technique, and the matching of the transmit and receive filters of the FMT, when α ═ γ
Figure BDA0003650388780000091
Accordingly, formula (10) can be further represented as
Figure BDA0003650388780000092
In the inverse configuration, the proposed method still uses interference cancellation and DFE to handle CCI and ISI, respectively, as represented by the 2 nd and 3 rd terms at the right end of equation (12). The signal after the reverse interference cancellation is
Figure BDA0003650388780000093
The DFE output estimate in the inverse configuration is
Figure BDA0003650388780000094
In formula (14)
Figure BDA0003650388780000095
And
Figure BDA0003650388780000096
representing the coefficients of the DFE feedforward and feedback filters in the inversion process respectively,
Figure BDA0003650388780000097
representing the decision values of the DFE in an inverted configuration.
Merging decisions
The output signal of DFE in reverse structure is reversed in time and combined with the output signal of DFE in forward structure to obtain
Figure BDA0003650388780000098
In the formula (15), ρ represents a combination coefficient, and its value range is 0 to 1. Since the combined channel response after TR processing has symmetry, the proposed method sets the value of ρ to 0.5.
In the proposed method, interference cancellation in forward and reverse architectures and DFE processing sequences, iterations, are performed until the overall system performance is stable.
Concrete underwater acoustic communication computing example
The underwater acoustic communication method provided by the invention has been verified in the experiment of a channel pool of Harbin engineering university, and a specific calculation example is given below to illustrate the effectiveness of the invention.
The channel water pool is 45m long, 5m deep and 6m wide, sound-absorbing wedges are fully distributed around the water pool, and the bottom of the water pool is a sand bottom. The distance between the 2 transmitting array elements and the water surface is 1.5m and 2m, the distance between the 2 receiving array elements and the water surface is 0.7m and 0.9m, and the communication distance between the two transmitting and receiving parties is 15 m.
In the experiment, signals transmitted by 2 array elements consist of Hamming windowed chirp signals of 8-16kHz, guard time intervals and information signals. The communication band of the information signal is 8-16kHz, divided into 4 sub-bands based on the FMT principle. The roll-off factor for the transmit and receive filters is set to 0.5 for each band, the number of transmitted symbols is 1600, BPSK mapping is used, with the first 250 and last 250 symbols used for training the DFE in bi-directional interference suppression. In the experiment, the filter coefficient used for channel estimation of each sub-band was 61, the number of causal and non-causal taps of the DFE feedforward filter was 23, the number of taps of the feedback filter was 8, and the adaptive algorithms used for channel estimation and DFE were Recursive Least Squares (RLS) algorithms, with a forgetting factor set to 0.999.
First, the performance after TR treatment was analyzed, as represented by TR treatment in the forward structure. To quantify the impact of CCI and ISI after TR treatment, signal-to-interference ratios are defined
Figure BDA0003650388780000101
In formula (16)
Figure BDA0003650388780000102
And SIR α,m Respectively representing TR-processed signals with respect to ISI, CCI, andpower ratio of total interference, σ α,m And σ γ,m Respectively, the average power of the mth sub-band transmission sequence for the mth and the gamma users.
Figure BDA0003650388780000103
And SIR α,m Higher interference indicates better interference rejection and less residual interference.
Fig. 4 shows a continuous-time version of the combined channel response after TR processing. In FIG. 4, q (1,1),(m,m) (t) and q (2 ,2),(m,m) (t) is normalized with respect to its maximum value, q (1,2),(m,m) (t) relative to q (1,1),(m,m) The maximum value of (t) is normalized, q (2,1),(m,m) (t) relative to q (2,2),(m,m) The maximum value of (t) is normalized. Fig. 5 shows the signal-to-interference ratio calculated based on equation (16).
By performing a joint analysis on fig. 4 and 5, the following three points can be obtained. First, q in FIG. 4 (1,2),(m,m) (t) is significantly greater than q (2,1),(m,m) (t), this indicates that the CCI of User2 to User1 is significantly greater than the CCI of User1 to User2, which is similar to that in FIG. 5
Figure BDA0003650388780000111
The results are consistent. Second, fig. 4 shows q and q in addition to the 4 th sub-band (1,1),(m,m) (t) comparison of q (2 ,2),(m,m) The side lobe compression of (t) is more pronounced, as in FIG. 5
Figure BDA0003650388780000112
Are consistent. Third, in fig. 5, except for the 1 st sub-band of user 1
Figure BDA0003650388780000113
And
Figure BDA0003650388780000114
that is, the effect of residual ISI after TR processing is greater than residual CCI.
FIG. 6 shows the performance indexes of bit error rate and output mean square errorThe performance of each sub-band after TR processing is shown. It can be seen from fig. 6 that both the bit error rate and the output mean square error for user 1 are higher than for user 2 for sub-bands 1-3, since there is more residual CCI and ISI in the user 1 signal, which is comparable to the SIR in fig. 5 1,m ＜SIR 2,m And m is 1,2,3 is identical; for the 4 th sub-band, the bit error rate and output mean square error of user 1 are lower than those of user 2, which is similar to SIR in FIG. 5 1,m ＞SIR 2,m And m is also equal to 4.
Fig. 7 and 8 show the performance of the sub-bands after the proposed method of bi-directional interference cancellation and DFE, respectively, and the total performance after serial-to-parallel conversion. For comparison, the performance of 2 single direction interference suppression methods is given in fig. 7 and 8. The 2 comparison methods are respectively as follows: first, the LE-based one-way interference suppression method adopted in document [1 ]; second, DFE-based one-way interference suppression methods.
As can be seen from FIGS. 7 and 8, the following document [1]]Compared with the LE-based unidirectional interference suppression method, the error rate and the output mean square error of the method are obviously reduced, and the performance is obviously improved; compared with the DFE-based one-way interference suppression method, the error rate is reduced to 10 after the interference suppression treatment by the two methods from the aspect of the error rate -3 Even the error rate of some sub-bands is reduced to 0, so the error rate of the method is lower, but the difference is not obvious, the output mean square error of the method is obviously lower and the performance is obviously improved by analyzing the output mean square error. In addition, as can be seen from the constellation diagram of fig. 8, compared with 2 comparison methods, the estimation values of the symbols processed by the method are more intensively distributed around the expected values-1 and 1, and the performance is better.
It will be appreciated by those of ordinary skill in the art that the examples described herein are intended to assist the reader in understanding the manner in which the invention is practiced, and it is to be understood that the scope of the invention is not limited to such specifically recited statements and examples. Those skilled in the art can make various other specific changes and combinations based on the teachings of the present invention without departing from the spirit of the invention, and these changes and combinations are within the scope of the invention.

Claims (6)

1. A bidirectional interference suppression method applied to multi-user filtering multi-tone underwater acoustic communication is characterized by comprising the following steps:
processing a sub-band signal demodulated by a multi-user filtered multi-tone (FMT) underwater Acoustic Communication (ACS) by using a bidirectional interference suppression method at a receiving end of the FMT ACS; the forward structure is used for processing the sub-band signals after FMT demodulation, and the reverse structure is used for processing the time reversal form of the sub-band signals; the forward and reverse structures of the proposed method both adopt Time Reversal (TR) technology for preprocessing, adopt continuous interference cancellation and Decision Feedback Equalizer (DFE) for post-processing, and combine the results of the two-way processing to obtain the final decision value of the user sequence.
2. The method of claim 1, wherein the method comprises: the bidirectional interference suppression method comprises the following steps:
s1: in the forward interference suppression, firstly, a sub-band signal after FMT demodulation is preprocessed by using a TR (transmitter-receiver) technology, so that pre-compression of co-channel interference (CCI) and intersymbol interference (ISI) is realized, and then, the CCI and ISI remained after TR are subjected to post-processing by using interference cancellation and DFE (differential frequency transmitter);
s2: in the reverse interference suppression, firstly, storing the sub-band signals after FMT demodulation according to the received signals by using a buffer, outputting the signals entering the buffer at first and outputting the signals entering the buffer at first, performing time reversal processing, then preprocessing the reversed FMT sub-band signals by using a TR (transmitter-receiver) technology, and performing post-processing by using an interference cancellation and DFE (digital feedback coefficient);
s3: storing the estimation value of the sub-band signal output by the DFE in the reverse processing according to the mode that the buffer is used for storing the signal output by the DFE in the reverse structure, the signal which firstly enters the buffer is output at last, and the signal which finally enters the buffer is output at first, carrying out time reversal processing, and carrying out equal gain combination on the processed signal and the sub-band signal output by the DFE in the forward processing;
s4: and judging the signals output by the equal gain combination, wherein the judgment value is used for estimating the interference in the next interference cancellation of the sub-band signals of other users.
3. The method of claim 2, wherein the method comprises: interference cancellation and adaptive equalization in forward interference suppression and reverse interference suppression are performed sequentially and iteratively, and are stopped when system performance is stable.
4. The method of claim 2, wherein the method comprises: s1 specifically includes:
the signal processed by TR in forward interference suppression is:
Figure FDA0003650388770000021
in the above formula, α is the user number, m is the sub-band number, q (α,γ),(m,m) (lT, nT) represents the m-th sub-band sequence d of the gamma user at the transmitting end γ,m (lT) and the mth TR output signal y in the forward structure of the alpha-th user receiver α,m Combined channel response between (nT), d α,m (lT) denotes the symbol sequence transmitted by the mth user on the mth subband, d α,m (nT) represents d when l ═ n α,m (lT), T represents a symbol interval, and N represents a number of users;
the estimated values of CCI in the forward structure are:
Figure FDA0003650388770000022
in the above formula
Figure FDA0003650388770000023
Shows the subband sequence d after the bi-directional interference suppression process α,m (nT) a decision value;
the signal after interference cancellation is:
Figure FDA0003650388770000024
q (α,α),(m,m) (lT, nT) represents q when γ ═ α (α,γ),(m,m) Value of (lT, nT), q (α,α),(m,m) (nT, nT) represents q when l ═ n (α,α),(m,m) (lT, nT);
using the DFE to suppress residual ISI, the DFE outputs an estimate of:
Figure FDA0003650388770000031
where a (nT) and b (nT) represent the coefficients of the DFE feedforward and feedback filters, respectively, N 1 And N 2 Number of non-causal and causal taps for feedforward filters, N f Is the number of taps of the feedback filter,
Figure FDA0003650388770000032
representing the decision value, z, of the forward DFE α,m (nT) represents a signal of an interference cancellation output in a forward configuration.
5. The method of claim 2, wherein the method comprises: s2 specifically includes:
the signal processed by TR in the reverse interference suppression is expressed as:
Figure FDA0003650388770000033
in the above formula
Figure FDA0003650388770000034
M-th sub-band sequence d representing the y-th user at the transmitting end γ,m (lT) output signal of mth TR in the α -th receiver inverse structure
Figure FDA0003650388770000035
The combined channel response between the two channels,
Figure FDA0003650388770000036
indicating that the mth user transmits the sequence d on the mth sub-band γ,m The time reversed version of (lT) is,
Figure FDA0003650388770000037
indicating that the alpha-th user transmits the sequence d on the m-th sub-band α,m The time reversed version of (lT) is,
Figure FDA0003650388770000038
when l is equal to n
Figure FDA0003650388770000039
A value of (d);
the signal after the reverse interference cancellation is:
Figure FDA00036503887700000310
in the above formula
Figure FDA00036503887700000311
When gamma is alpha
Figure FDA00036503887700000312
The value of (a) is,
Figure FDA00036503887700000313
when l is equal to n
Figure FDA00036503887700000314
The value of (a) is,
Figure FDA00036503887700000315
decision value representing a sequence of subbands after a bi-directional interference suppression process
Figure FDA00036503887700000316
Time reversed version of (a);
the DFE outputs estimates as:
Figure FDA0003650388770000041
in the above formula
Figure FDA0003650388770000042
And
Figure FDA0003650388770000043
representing the coefficients of the DFE feedforward and feedback filters in the inversion process respectively,
Figure FDA0003650388770000044
represents the decision values of the DFE in an inverted configuration,
Figure FDA0003650388770000045
representing the output signal of the interference cancellation in the inverted configuration.
6. The method of claim 2, wherein the method comprises: the gain combination in S4 specifically includes:
Figure FDA0003650388770000046
in the above formula, rho represents a merging coefficient, and the value range of rho is 0 to 1,
Figure FDA0003650388770000047
representing the estimated value of the output after the forward structure processing,
Figure FDA0003650388770000048
representing DFE output estimates in an inverse configuration
Figure FDA0003650388770000049
Time reversed version of (3).
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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20130208768A1 (en) * 2010-06-07 2013-08-15 University Of Delaware Underwater acoustic multiple-input/multiple-output (mimo) communication systems and methods
CN103684703A (en) * 2013-12-20 2014-03-26 中国船舶重工集团公司第七一五研究所 Space-time-frequency code and passive time reversal reception based underwater acoustic MIMO (Multiple Input Multiple Output) communication method
CN105187350A (en) * 2015-08-06 2015-12-23 哈尔滨工程大学 Time reversal acoustic communication method based on filtered multitone modulation
US20160086484A1 (en) * 1999-08-09 2016-03-24 Kamilo Feher Base Station Devices and Automobile Wireless Communication Systems
US10361802B1 (en) * 1999-02-01 2019-07-23 Blanding Hovenweep, Llc Adaptive pattern recognition based control system and method

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US10361802B1 (en) * 1999-02-01 2019-07-23 Blanding Hovenweep, Llc Adaptive pattern recognition based control system and method
US20160086484A1 (en) * 1999-08-09 2016-03-24 Kamilo Feher Base Station Devices and Automobile Wireless Communication Systems
US20130208768A1 (en) * 2010-06-07 2013-08-15 University Of Delaware Underwater acoustic multiple-input/multiple-output (mimo) communication systems and methods
CN103684703A (en) * 2013-12-20 2014-03-26 中国船舶重工集团公司第七一五研究所 Space-time-frequency code and passive time reversal reception based underwater acoustic MIMO (Multiple Input Multiple Output) communication method
CN105187350A (en) * 2015-08-06 2015-12-23 哈尔滨工程大学 Time reversal acoustic communication method based on filtered multitone modulation

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