CN113517874B - Fast response automatic gain control circuit for transimpedance amplifier - Google Patents
Fast response automatic gain control circuit for transimpedance amplifier Download PDFInfo
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- CN113517874B CN113517874B CN202110786524.8A CN202110786524A CN113517874B CN 113517874 B CN113517874 B CN 113517874B CN 202110786524 A CN202110786524 A CN 202110786524A CN 113517874 B CN113517874 B CN 113517874B
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03G—CONTROL OF AMPLIFICATION
- H03G3/00—Gain control in amplifiers or frequency changers without distortion of the input signal
- H03G3/20—Automatic control
- H03G3/30—Automatic control in amplifiers having semiconductor devices
- H03G3/3036—Automatic control in amplifiers having semiconductor devices in high-frequency amplifiers or in frequency-changers
- H03G3/3042—Automatic control in amplifiers having semiconductor devices in high-frequency amplifiers or in frequency-changers in modulators, frequency-changers, transmitters or power amplifiers
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/45—Differential amplifiers
- H03F3/45071—Differential amplifiers with semiconductor devices only
- H03F3/45076—Differential amplifiers with semiconductor devices only characterised by the way of implementation of the active amplifying circuit in the differential amplifier
- H03F3/45179—Differential amplifiers with semiconductor devices only characterised by the way of implementation of the active amplifying circuit in the differential amplifier using MOSFET transistors as the active amplifying circuit
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
- Y02D30/00—Reducing energy consumption in communication networks
- Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks
Abstract
The invention discloses a quick response automatic gain control circuit for a transimpedance amplifier, which comprises the transimpedance amplifier, a transimpedance amplifier copying circuit and a control voltage generating circuit, wherein the output ends of the transimpedance amplifier and the transimpedance amplifier copying circuit are respectively connected to two input ends of the control voltage generating circuit, and the output end of the control voltage generating circuit is connected to the gain control end of the transimpedance amplifier. The fast response automatic gain control circuit for the transimpedance amplifier can quickly respond and establish a steady state when the input optical power changes, can keep the stable gain and small enough output jitter of the transimpedance amplifier when a service signal passes, and can be used for a burst mode; and compared with the prior art, the method can provide a larger transimpedance gain adjustment range, thereby widening the dynamic range of the transimpedance amplifier.
Description
Technical Field
The invention relates to the technical field of transimpedance amplifiers, in particular to a fast response automatic gain control circuit for a transimpedance amplifier.
Background
In an optical fiber communication system, a Trans-impedance amplifier (TIA) is located at the forefront end of a receiving link, and functions to convert and amplify a weak photocurrent signal generated by a photodiode into a voltage signal, and output the voltage signal to a subsequent circuit for processing. The performance of the TIA greatly affects the performance of the entire receive chain.
The input dynamic range is an important indicator of TIA, which is defined as the ratio of saturated input optical power to the received optical sensitivity, i.e. the ratio of the maximum and minimum input optical power within a certain allowable error rate range. The sensitivity is mainly determined by the noise performance, the smaller the equivalent input noise of the TIA is, the better the sensitivity is, and the larger transimpedance gain is adopted under the general condition to be beneficial to reducing the equivalent input noise; the saturated input optical power is mainly determined by factors such as the distortion degree of the output signal, and the larger the trans-impedance is, the more easily saturation occurs in the case of a large signal.
To solve this contradiction, an automatic gain control (AGC, automatic GainControl) circuit is usually added to the TIA, i.e. the transimpedance gain is adjusted according to the magnitude of the input optical power, and a larger transimpedance is maintained when the input optical power is smaller, and the transimpedance is reduced when the input optical power is larger, so as to improve the distortion of the output voltage signal, thereby widening the dynamic range of the TIA.
AGC typically includes two part functions, one for amplitude detection and one for gain adjustment. Since TIA is a single-ended input single-ended output amplifier, the direct current component and alternating current component in the input current are amplified in the same proportion, so monitoring the change of the direct current component of the output voltage can reflect the change of the input current amplitude. Conventional AGC generally uses a low-pass filter to obtain average values Vout, avg of TIA output voltage in a time domain, as shown in fig. 1, and uses a TIA replica circuit with an input current of 0 to generate a reference voltage, and an error amplifier to amplify an error between the average value of TIA output voltage and the reference voltage, so as to implement an amplitude detection function. The gain adjusting function is realized by connecting an NMOS as an active resistor and a passive resistor in parallel, and controlling the grid electrode of the NMOS by adopting the output voltage of an error amplifier. When the input optical power is increased, the average value of the TIA output voltage is reduced, the error amplifier output voltage is increased, and the active resistance provided by the NMOS is reduced, so that the transimpedance gain is reduced.
In the above AGC method, the cut-off frequency ω of the RC low-pass filter LPF =1/(RC) determines the bandwidth and response time of the AGC loop. On the one hand, if omega LPF The low-frequency alternating current component in the data signal is not attenuated by the low-pass filter and is superposed on Vout and avg, so that the transimpedance cannot be kept stable, and the output signal is dithered; on the other hand, a large RC determines that the AGC loop requires a long convergence time to establish steady state. In the continuous communication mode, the low frequency cut-off frequency of the AGC loop is typically set at several tens kHz, meaningMeaning that when the input optical power changes, it takes tens of us for the AGC loop to re-stabilize.
In an optical line terminal (OLT, opticalLineTerminal) of a passive optical network (PON, passiveOpticalNetwork) system, a signal processed by a receiving end is in Burst-Mode, and optical power of the signal received in Burst is different from one optical network unit (ONU, opticalNetworkUnit) to another. Depending on the communication standard, burst mode receiving systems generally need to establish stable operating conditions within tens to hundreds of ns, meaning that the TIA requires a fast AGC response, and thus conventional AGC loops cannot be adapted for burst mode.
In order to achieve a fast AGC response, the gate of the NMOS is controlled by a bias circuit generating a dc level Vgate, as shown in fig. 2. When the input optical power is 0 or smaller, the voltage difference between Vgate and the output voltage Vout does not reach the threshold voltage Vth of NMOS, the NMOS is in a cut-off area, and the transconductance of TIA is determined by a passive resistor; when the input optical power increases, the current flowing through the passive resistor increases, so that the output voltage Vout decreases, and the gate-source voltage Vgs of the NMOS increases; when Vgs exceeds Vth, the NMOS is conducted, the NMOS is in a linear region and is equivalent to an active resistor, and the transconductance of the TIA is parallel connection of a passive resistor and the active resistor, so that the total transconductance is reduced; the larger the input optical power, the larger the Vgs of the NMOS and the lower the transconductance gain of the TIA. The on-resistance of the NMOS can be changed along with the gate-source voltage of the NMOS, so that the method can realize quick response.
However, in practical applications, the output amplitude of TIA generally needs to be within 0.2V to ensure that no significant signal distortion occurs. At lower optical powers, vgate must be biased at the critical point where the NMOS is turned off to on to switch the NMOS between the off and linear regions. On the one hand, due to factors such as temperature variation and process deviation, the accurate setting of the bias voltage has great difficulty. On the other hand, the overdrive voltage of NMOS does not exceed 0.2V over the entire dynamic range, it is difficult to adjust to a sufficiently small transimpedance gain at large input power, so that it is difficult to achieve a large TIA dynamic range.
Disclosure of Invention
The invention aims to provide a fast response automatic gain control circuit for a transimpedance amplifier, which can quickly respond to the establishment of a steady state when the input optical power changes, can keep stable gain and small enough output jitter of the transimpedance amplifier when a service signal passes, can be used for a burst mode, and can provide a larger transimpedance gain adjustment range, thereby widening the dynamic range of the transimpedance amplifier.
In order to solve the above-described problems, the present invention provides a fast response automatic gain control circuit for a transimpedance amplifier, comprising:
the output ends of the transimpedance amplifier and the transimpedance amplifier replica circuit are respectively connected to two input ends of the control voltage generation circuit, and the output end of the control voltage generation circuit is connected to the gain control end of the transimpedance amplifier;
the control voltage generating circuit comprises a bandwidth-adjustable low-pass filter, a transconductance amplifier, a current source Q1, a current source Q2, a transistor M2 and a resistor R0, wherein the input end of the bandwidth-adjustable low-pass filter is used as one input end of the control voltage generating circuit and is connected with the output end of the transimpedance amplifier, the output end of the bandwidth-adjustable low-pass filter is connected to the negative input end of the transconductance amplifier, the positive input end of the transconductance amplifier is a direct current reference voltage Vref, one end of the resistor R0 is used as the other input end of the control voltage generating circuit and is connected with the output end of the transimpedance amplifier, the other end of the resistor R0 is connected with the output end of the transconductance amplifier, the source electrode of the transistor M2 is connected to the output end of the transconductance amplifier and the positive end of the current source Q1, the grid electrode and the drain electrode are connected to the negative end of the current source Q2, and the grid electrode and the drain electrode voltage of the transistor M2 are used as the output end of the control voltage generating circuit.
As a further improvement of the present invention, the transimpedance amplifier includes an inverting amplifier-a, a feedback resistor Rf0 and a transistor M1, the input end of the inverting amplifier-a is the input end of the transimpedance amplifier, the output end of the inverting amplifier-a is the output end of the transimpedance amplifier, the input end and the output end of the inverting amplifier-a are respectively connected with two ends of the feedback resistor Rf0, and the input end and the output end of the inverting amplifier-a are also respectively connected with the drain electrode and the source electrode of the transistor M1.
As a further improvement of the present invention, both the transistor M1 and the transistor M2 are NMOS.
As a further development of the invention, the direct reference voltage Vref at the positive input of the transconductance amplifier is generated by a level shift of the output voltage of the transimpedance amplifier replica circuit or by an additional bias circuit.
As a further improvement of the present invention, the emitter follower for realizing the level shift is the same as the emitter follower in the bandwidth-adjustable low-pass filter, and the level shift is realized by two resistor voltage division.
As a further development of the invention, the current source Q1 and the current source Q2 have equal currents I0.
As a further development of the invention, the direct reference voltage Vref at the positive input of the transconductance amplifier is lower than the output voltage of the replica circuit of the transimpedance amplifier.
As a further improvement of the present invention, the output voltage Vout of the transimpedance amplifier replica circuit, 0 is equal to the output voltage at which the transimpedance amplifier input current is 0.
As a further improvement of the invention, the bandwidth-adjustable low-pass filter has a high-low cut-off frequency; the low-pass cut-off frequency of the bandwidth adjustable low-pass filter is adjusted to be low after the gain is stable.
As a further improvement of the invention, the transconductance amplifier is a linear differential voltage-single ended current amplifier.
The invention has the beneficial effects that:
the fast response automatic gain control circuit for the transimpedance amplifier can quickly respond and establish a steady state when the input optical power changes, can keep the stable gain and small enough output jitter of the transimpedance amplifier when a service signal passes, and can be used for a burst mode; and compared with the prior art, the method can provide a larger transimpedance gain adjustment range, thereby widening the dynamic range of the transimpedance amplifier.
The foregoing description is only an overview of the present invention, and is intended to be implemented in accordance with the teachings of the present invention, as well as the preferred embodiments thereof, together with the following detailed description of the invention, given by way of illustration only, together with the accompanying drawings.
Drawings
FIG. 1 is a prior art automatic gain control circuit for detecting a DC component of an output voltage of a transimpedance amplifier;
FIG. 2 is a prior art fast response automatic gain control circuit;
FIG. 3 is a block diagram of a fast response AGC circuit for a transimpedance amplifier in accordance with a preferred embodiment of the present invention;
FIG. 4 is a second block diagram of a fast response AGC circuit for a transimpedance amplifier in accordance with a preferred embodiment of the present invention;
fig. 5 is a block diagram of a bandwidth-tunable low-pass filter, level-shifting and transimpedance amplifier in a preferred embodiment of the present invention.
Detailed Description
The present invention will be further described with reference to the accompanying drawings and specific examples, which are not intended to be limiting, so that those skilled in the art will better understand the invention and practice it.
As shown in fig. 3, a fast response automatic gain control circuit for a transimpedance amplifier according to a preferred embodiment of the present invention includes a transimpedance amplifier, a transimpedance amplifier replica circuit, and a control voltage generation circuit, wherein the output terminals of the transimpedance amplifier and the transimpedance amplifier replica circuit are respectively connected to two input terminals of the control voltage generation circuit, and the output terminal of the control voltage generation circuit is connected to the gain control terminal of the transimpedance amplifier.
The control voltage generating circuit comprises a bandwidth-adjustable low-pass filter, a transconductance amplifier, a current source Q1, a current source Q2, a transistor M2 and a resistor R0, wherein the input end of the bandwidth-adjustable low-pass filter is used as one input end of the control voltage generating circuit and is connected with the output end of the transimpedance amplifier, the output end of the bandwidth-adjustable low-pass filter is connected to the negative input end of the transconductance amplifier, the positive input end of the transconductance amplifier is a direct current reference voltage Vref, one end of the resistor R0 is used as the other input end of the control voltage generating circuit and is connected with the output end of the transimpedance amplifier, the other end of the resistor R0 is connected with the output end of the transconductance amplifier, the source electrode of the transistor M2 is connected to the output end of the transconductance amplifier and the positive end of the current source Q1, the grid electrode and the drain electrode are connected to the negative end of the current source Q2, and the grid electrode and the drain electrode voltage of the transistor M2 are used as the output end of the control voltage generating circuit.
In some embodiments, the transimpedance amplifier includes an inverting amplifier-a, a feedback resistor Rf0 and a transistor M1, where an input terminal of the inverting amplifier-a is an input terminal of the transimpedance amplifier, an output terminal of the inverting amplifier-a is an output terminal of the transimpedance amplifier, the input terminal and the output terminal of the inverting amplifier-a are respectively connected to two ends of the feedback resistor Rf0, and the input terminal and the output terminal of the inverting amplifier-a are also respectively connected to a drain electrode and a source electrode of the transistor M1.
In some embodiments, transistor M1 and transistor M2 are both NMOS; current source Q1 and current source Q2 have equal current I0; the direct current reference voltage Vref at the positive input of the transconductance amplifier is lower than the output voltage of the replica circuit of the transimpedance amplifier.
As shown in fig. 4, in one embodiment, the dc reference voltage Vref at the positive input of the transimpedance amplifier is generated by level shifting the output voltage of the transimpedance amplifier replica circuit; in other embodiments, the direct current reference voltage Vref at the positive input of the transconductance amplifier may be generated by an additional biasing circuit.
The transimpedance amplifier replica circuit is a replica of the transimpedance amplifier, comprises an inverting amplifier and a feedback resistor, and has the same topology and proportional circuit parameters as the inverting amplifier-A and the feedback resistor Rf0 in the transimpedance amplifier, so that the output voltage Vout of the transimpedance amplifier replica circuit is equal to the output voltage when the input current of the transimpedance amplifier is 0.
In the present invention, the output signal Vout, avg of the bandwidth-adjustable low-pass filter is the average of the transimpedance amplifier output signal Vout in the time domain. Optionally, the transconductance amplifier is a linear differential voltage-single-ended current amplifier, and has a transconductance gain Gm, and generates an output current iagc=gm (Vref-Vout, avg), which flows through a resistor R0 to generate a potential difference between the source of the transistor M2 and the output terminal of the transimpedance amplifier replica circuit, where the output voltage of the automatic gain control circuit is vagc=vout, 0+r0×iagc+vgs2, where Vgs2 is the gate-source voltage of the transistor M2, and is determined by the current I0 of the current sources Q1, Q2 and the size and process parameters of the transistor M2.
When the input optical power is 0 or less, vout, avg > Vref, iagc is a negative value, the automatic gain control circuit generates a lower control voltage Vagc to make the gate-source voltage of the transistor M1 be Vgs 1=vagc-vout=vout, and 0+r0×iagc+vgs2-Vout be lower than the on threshold voltage Vth of the transistor M1, thereby causing the transistor M1 to be turned off and making the gain of the transimpedance amplifier be equal to the resistance value Rf0 of the feedback resistor.
When the input optical power increases, vout, avg decreases, iagc becomes positive and increases proportionally, vagc increases, transistor M1 goes from the cut-off region into the linear region, corresponding to the active resistance, whose source-drain resistance is rmos=1/[ μ ] n C ox (W/L)(Vagc-Vout-Vth)]. Thus, as the input optical power increases, the gain Rf0// Rmos of the transimpedance amplifier decreases, thereby achieving automatic gain regulation. Compared with the prior art in fig. 2, since the gate control voltage of the active resistor NMOS is varied, a larger Rmos adjustment range can be provided, and thus a large dynamic range can be realized by using the automatic gain adjustment circuit of the present invention.
The key point of realizing the quick response is that the bandwidth-adjustable low-pass filter has a high-low cut-off frequency; the AGC loop has a relatively high low-pass cut-off frequency when the optical power burst is input, so that the time constant for establishing the AGC loop is relatively small, and stable gain can be established within tens of ns; after the gain is stable, the low-pass cut-off frequency of the bandwidth-adjustable low-pass filter is adjusted to be in a low-grade state, so that the bandwidth of the AGC loop is reduced, and when the low-frequency component of the input signal passes through, the transimpedance gain can be kept stable, thereby ensuring that the jitter of the output signal of the transimpedance amplifier does not exceed the index requirement.
As shown in fig. 5, in one embodiment, the emitter follower used to implement the level shift is identical to the emitter follower in the bandwidth-adjustable low-pass filter, and the level shift is implemented by the voltage division of the resistors Re1 and Re2, then vref=α (Vout, 0-Vbe) where α=re 2/(Re 1+re 2) when the input optical power is 0 or small<1, and Vout, avg=vout-Vbe, thus Vref<Vout, avg. Resistor R1 in low pass filter<<R2, when the input optical power bursts, the switch is closed, the parallel resistance of R1 and R2 is approximately equal to R1, and the low-pass cut-off frequency is omega LPF1 =1/(R1C 0), giving the AGC loop a fast response; after the gain stabilizes, the switch is turned off and the low-pass cut-off frequency becomes ω LPF1 =1/(R2C 0), making the AGC loop have a smaller loop bandwidth. The values of R1 and R2 are respectively determined by the response time index and the low-frequency cutoff frequency index of the transimpedance amplifier. Alternatively, the transconductance amplifier is a differential voltage-single ended current amplifier, and may be implemented by a differential amplifier with emitter degeneration resistance as shown in the figure, with high linearity in the variation range of Vout, avg. In other embodiments of the present invention, the bandwidth-adjustable low-pass filter, the level shift and the transimpedance amplifier may have other structures, which are not described herein.
In the fast response automatic gain control circuit for the transimpedance amplifier, the active resistor NMOS is used as a part of the transimpedance gain, the grid control voltage of the active resistor NMOS and the time domain average value of the output voltage of the transimpedance amplifier are in linear relation, and a larger gain adjustment range can be provided, so that the transimpedance amplifier can realize a large dynamic range by using the fast response automatic gain control circuit.
The automatic gain control circuit adopts a low-pass filter with high and low cut-off frequencies, so that the AGC loop can quickly respond when the input optical power bursts, and meanwhile, the AGC loop is ensured to be switched to a lower low-frequency cut-off frequency when the service signal passes, so that the transimpedance amplifier can keep stable gain and small enough output jitter.
The above embodiments are merely preferred embodiments for fully explaining the present invention, and the scope of the present invention is not limited thereto. Equivalent substitutions and modifications will occur to those skilled in the art based on the present invention, and are intended to be within the scope of the present invention. The protection scope of the invention is subject to the claims.
Claims (10)
1. A fast response automatic gain control circuit for a transimpedance amplifier comprising:
the output ends of the transimpedance amplifier and the transimpedance amplifier replica circuit are respectively connected to two input ends of the control voltage generation circuit, and the output end of the control voltage generation circuit is connected to the gain control end of the transimpedance amplifier;
the control voltage generating circuit comprises a bandwidth-adjustable low-pass filter, a transconductance amplifier, a current source Q1, a current source Q2, a transistor M2 and a resistor R0, wherein the input end of the bandwidth-adjustable low-pass filter is used as one input end of the control voltage generating circuit and is connected with the output end of the transimpedance amplifier, the output end of the bandwidth-adjustable low-pass filter is connected to the negative input end of the transconductance amplifier, the positive input end of the transconductance amplifier is a direct current reference voltage Vref, one end of the resistor R0 is used as the other input end of the control voltage generating circuit and is connected with the output end of the transimpedance amplifier, the other end of the resistor R0 is connected with the output end of the transconductance amplifier, the source electrode of the transistor M2 is connected to the output end of the transconductance amplifier and the positive end of the current source Q1, the grid electrode and the drain electrode are connected to the negative end of the current source Q2, and the grid electrode and the drain electrode voltage of the transistor M2 are used as the output end of the control voltage generating circuit.
2. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein the transimpedance amplifier comprises an inverting amplifier-a, a feedback resistor Rf0 and a transistor M1, the input terminal of the inverting amplifier-a is the input terminal of the transimpedance amplifier, the output terminal of the inverting amplifier-a is the output terminal of the transimpedance amplifier, the input terminal and the output terminal of the inverting amplifier-a are respectively connected to both ends of the feedback resistor Rf0, and the input terminal and the output terminal of the inverting amplifier-a are also respectively connected to the drain and the source of the transistor M1.
3. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 2, wherein transistor M1 and transistor M2 are both NMOS.
4. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein the direct current reference voltage Vref at the positive input of the transimpedance amplifier is generated by level shifting the output voltage of the replica circuit of the transimpedance amplifier or by an additional bias circuit.
5. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 4, wherein the emitter follower for achieving level shifting is identical to the emitter follower in the bandwidth-adjustable low-pass filter, and level shifting is achieved by two resistor voltage division.
6. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein current source Q1 and current source Q2 have equal current I0.
7. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein the direct current reference voltage Vref at the positive input of the transimpedance amplifier is lower than the output voltage of the replica circuit of the transimpedance amplifier.
8. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein the output voltage Vout of the transimpedance amplifier replica circuit, 0 is equal to the output voltage at which the transimpedance amplifier input current is 0.
9. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein the bandwidth adjustable low pass filter has a high and low cut-off frequency; the low-pass cut-off frequency of the bandwidth adjustable low-pass filter is adjusted to be low after the gain is stable.
10. The fast response automatic gain control circuit for a transimpedance amplifier according to claim 1, wherein the transconductance amplifier is a linear differential voltage-single ended current amplifier.
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| CN117294264B (en) * | 2023-11-22 | 2024-03-12 | 成都明夷电子科技有限公司 | Low-noise transimpedance amplifier for optical receiver |
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| CN113517874A (en) | 2021-10-19 |
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