CN109617844B - Carrier synchronization method and system - Google Patents

Carrier synchronization method and system Download PDF

Info

Publication number
CN109617844B
CN109617844B CN201910020334.8A CN201910020334A CN109617844B CN 109617844 B CN109617844 B CN 109617844B CN 201910020334 A CN201910020334 A CN 201910020334A CN 109617844 B CN109617844 B CN 109617844B
Authority
CN
China
Prior art keywords
pilot signal
carrier
doppler
signal
phase
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Active
Application number
CN201910020334.8A
Other languages
Chinese (zh)
Other versions
CN109617844A (en
Inventor
司江勃
阮奇
李赞
关磊
颜灵恩
齐佩汉
程梓豪
王丹阳
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Xidian University
Original Assignee
Xidian University
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Xidian University filed Critical Xidian University
Priority to CN201910020334.8A priority Critical patent/CN109617844B/en
Publication of CN109617844A publication Critical patent/CN109617844A/en
Application granted granted Critical
Publication of CN109617844B publication Critical patent/CN109617844B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/004—Arrangements for detecting or preventing errors in the information received by using forward error control
    • H04L1/0056—Systems characterized by the type of code used
    • H04L1/0059—Convolutional codes
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/004—Arrangements for detecting or preventing errors in the information received by using forward error control
    • H04L1/0056—Systems characterized by the type of code used
    • H04L1/0059—Convolutional codes
    • H04L1/006—Trellis-coded modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/20—Modulator circuits; Transmitter circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/003—Arrangements for allocating sub-channels of the transmission path
    • H04L5/0048—Allocation of pilot signals, i.e. of signals known to the receiver
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H04L2027/0024—Carrier regulation at the receiver end
    • H04L2027/0026—Correction of carrier offset

Landscapes

  • Engineering & Computer Science (AREA)
  • Signal Processing (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

Abstract

The embodiment of the invention discloses a carrier synchronization method and a carrier synchronization system, wherein the method comprises the steps of carrying out convolutional code coding with a code rate of 1/2 on input single-bit information by a sending end to obtain a coded output sequence with a length of L, modulating the coded output sequence to obtain modulated data with a length of L + N, estimating Doppler frequency offset and Doppler change rate of carrier waves in pilot signals according to instantaneous phases of the pilot signals in the received signals by a receiving end, compensating the received signals according to Doppler frequency offset estimation values of the carrier waves and Doppler change rate estimation values of the carrier waves, carrying out Viterbi decoding on the compensated signals, and carrying out the following processing until L th moment in the decoding process, namely obtaining survivor paths of each state at the next moment corresponding to the current moment to obtain an instantaneous phase value at the next moment corresponding to the current moment, and selecting a survivor path with the smallest path metric value for L th moments to carry out backtracking to obtain a sequence which is closest to the coded output sequence.

Description

Carrier synchronization method and system
Technical Field
The embodiment of the invention relates to the technical field of wireless communication, in particular to a method and a system for carrier synchronization.
Background
In high dynamic environments such as mobile communication or military communication, doppler frequency offset and doppler frequency change rate are generated due to the relatively high-speed movement of two communication parties, so that the error performance of a receiving end is rapidly deteriorated, and the communication quality is seriously reduced. Based on this, how to accurately estimate and eliminate the doppler frequency offset and the doppler change rate in a high dynamic environment, and further obtain ideal carrier synchronization is an urgent problem to be solved.
It should be noted that the conventional carrier synchronization method may include: Data-Aided (DA) algorithms, Non-Data-Aided (NDA) algorithms, and Code-Aided (CA) algorithms. The DA algorithm has wide estimation range, the estimation precision is related to the known sequence length, extra bandwidth is occupied, and the system efficiency is low in short burst communication; the NDA algorithm comprises two ways of phase-locked loop and blind estimation, the frequency band utilization rate is high, but the signal-to-noise ratio is high, and the synchronization range is narrow; the CA algorithm utilizes soft decision to realize parameter estimation, has low requirement on signal-to-noise ratio, high operation complexity and limited estimation range.
In addition, some researchers also provide an open-loop capture auxiliary phase-locked loop method, namely, firstly, the open-loop capture reduces the range of parameters to be estimated, and then, the phase-locked loop is used for realizing signal tracking, however, under the condition of low signal-to-noise ratio, when the power of an input signal is lower than the loop threshold, the phase-locked loop is difficult to converge and cannot work correctly, and some researchers also utilize convolutional codes in the carrier synchronization stage and carry out phase-locked loop tracking and Viterbi (Viterbi) decoding at a receiving end, the method effectively utilizes coding information, but because two times of hard demodulation are involved and the decoding characteristic of the convolutional codes cannot be fully utilized, the bit error rate of a receiver is higher.
Disclosure of Invention
In order to solve the above technical problems, embodiments of the present invention are expected to provide a method and a system for carrier synchronization, which can improve phase tracking accuracy and reduce system error rate.
The technical scheme of the invention is realized as follows:
in a first aspect, an embodiment of the present invention provides a method for carrier synchronization, where the method includes:
the transmitting end carries out convolutional code coding with code rate of 1/2 on the input single-bit information to obtain a coded output sequence with length of L;
the transmitting end modulates the coded output sequence according to a set modulation strategy to obtain modulated data with the length of L + N;
a receiving end estimates the Doppler frequency offset and the Doppler change rate of a carrier wave in a pilot signal according to the instantaneous phase of the pilot signal in a received signal;
the receiving end compensates the received signal according to the Doppler frequency offset estimation value of the carrier in the pilot signal and the Doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal;
the receiving end carries out Viterbi decoding on the compensated signal, and in the decoding process, the following processing is carried out until L th time aiming at each current time:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment;
for the L th time, the receiving end selects the survivor path with the minimum path metric value to backtrack, and the sequence closest to the coded output sequence is obtained.
In a second aspect, an embodiment of the present invention provides a system for carrier synchronization, where the system includes a sending end and a receiving end;
the sending end comprises a first communication interface, a first memory and a first processor; wherein the content of the first and second substances,
the first communication interface is used for receiving and sending signals in the process of receiving and sending information with other external network elements;
the first memory for storing a computer program operable on the first processor;
the first processor, when executing the computer program, is configured to perform the following steps:
carrying out convolutional code coding with the code rate of 1/2 on the input single-bit information to obtain a coded output sequence with the length of L, and modulating the coded output sequence according to a set modulation strategy to obtain modulation data with the length of L + N;
the receiving end includes: a second communication interface, a second memory, and a second processor;
the second communication interface is used for receiving and sending signals in the process of receiving and sending information with other external network elements;
the second memory for storing a computer program operable on a second processor;
the second processor, when executing the computer program, is configured to perform the following steps:
estimating Doppler frequency offset and Doppler change rate of a carrier in a pilot signal according to the instantaneous phase of the pilot signal in a received signal; and the number of the first and second groups,
compensating the received signal according to the Doppler frequency offset estimation value of the carrier in the pilot signal and the Doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal; and the number of the first and second groups,
performing Viterbi decoding on the compensated signal, and performing the following processing for each current time until L th time in the decoding process:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment; and the number of the first and second groups,
and for the L th time, selecting a survivor path with the minimum path metric value for backtracking to obtain a sequence closest to the coded output sequence.
The embodiment of the invention provides a method and a system for carrier synchronization; the effects of improving the phase tracking precision and reducing the system error rate can be achieved by combining the pilot signal and the Viterbi decoding to carry out carrier synchronization.
Drawings
Fig. 1 is a schematic diagram of a baseband model of a communication system according to an embodiment of the present invention;
fig. 2 is a flowchart illustrating a method for carrier synchronization according to an embodiment of the present invention;
fig. 3 is a schematic diagram illustrating a specific signal flow of Viterbi decoding according to an embodiment of the present invention;
fig. 4 is a schematic design diagram of a third-order phase-locked loop according to an embodiment of the present invention;
fig. 5 is a schematic diagram of a system for carrier synchronization according to an embodiment of the present invention;
fig. 6 is a schematic diagram of another system component for carrier synchronization according to an embodiment of the present invention;
FIG. 7 is a comparison diagram of simulation performance provided by an embodiment of the present invention;
FIG. 8 is a comparison diagram of simulation performance provided by an embodiment of the present invention;
fig. 9 is a comparison diagram of simulation performance according to another embodiment of the present invention.
Detailed Description
The technical solution in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.
Referring to fig. 1, which shows a schematic baseband model of a communication system according to an embodiment of the present invention, it can be seen that a transmitting end Tx generates a transmission sequence or a transmission signal x and transmits the transmission sequence or the transmission signal x to a communication channel H. In the embodiment of the present invention, the communication channel H has a high dynamic characteristic, that is, both communication parties are in a relatively high-speed moving state, such as satellite communication, high-speed mobile platform communication, and the like, after the transmission sequence x passes through the communication channel H, the reception signal y at the receiving end Rx may be represented as y ═ Hx + n, where n represents noise caused by the communication channel.
Example one
Based on the baseband model illustration shown in fig. 1, a method for carrier synchronization provided in an embodiment of the present invention may be applied to a transmitting end Tx and a receiving end Rx in the baseband model shown in fig. 1, and referring to fig. 2, the method may include:
s201, the sending end carries out convolutional code coding with code rate of 1/2 on the input single-bit information to obtain a coded output sequence with length of L;
for S201, in a possible implementation manner, the sending end performs convolutional coding with a code rate of 1/2 on the input single-bit information to obtain a coded output sequence, which may include:
carrying out convolutional code coding with the code rate of 1/2 on the input single-bit information to obtain a two-bit coding sequence, wherein the length of the two-bit coding sequence is L, and the output expressions of the two-bit coding sequence are g respectively1(x)＝1+x+x2And g2(x)＝1+x2Where x denotes a delay of a single bit of information of the input, x2A second delay representing the input single bit information.
S202: the sending end modulates the coded output sequence according to a set modulation strategy to obtain modulation data;
for S202, in a possible implementation manner, the modulation data with a length of L + N, which is obtained by the transmitting end modulating the coded output sequence according to the set modulation policy, may include:
adding an all-zero sequence with the length of N before the coded output sequence to obtain a signal with the length of L + N and added with a pilot sequence;
QPSK modulation is carried out on the signal added with the pilot frequency sequence to obtain a four-phase modulation signal; where the "00" sequence maps to 1/4 π, the "01" sequence maps to 3/4 π, the "10" sequence maps to 5/4 π, and the "11" sequence maps to 7/4 π.
S203: the receiving end estimates the Doppler frequency offset and the Doppler change rate of the carrier wave in the pilot signal according to the instantaneous phase of the pilot signal in the received signal;
for S203, in a possible implementation manner, the estimating, by the receiving end, a doppler shift of a carrier in the pilot signal and a doppler change rate of the carrier according to an instantaneous phase of the pilot signal in the received signal includes:
the receiving end extracts a pilot signal of a received signal to obtain an instantaneous phase of the pilot signal;
and aiming at the instantaneous phase of the pilot signal, the receiving end estimates the Doppler frequency offset and the Doppler change rate of the carrier in the pilot signal according to the minimum mean square error criterion to obtain the Doppler frequency offset estimation value and the Doppler change rate estimation value of the carrier in the pilot signal.
It should be noted that the received signal is a signal obtained after the modulated data received by the receiving end passes through a communication channel, in this embodiment of the present invention, the received signal may be represented as a signal obtained by a communication system having ideal symbol timing synchronization and negligible intersymbol interference, and the receiving end may output a sample value through matched filtering, based on which the received signal is a signal obtained by passing the modulated data through the communication channel
Figure BDA0001940552120000051
For this implementation, preferably, the extracting, by the receiving end, a pilot signal of a received signal to obtain an instantaneous phase of the pilot signal may include:
for the received signal
Figure BDA0001940552120000052
The first N data are subjected to-1/4 pi phase rotation to obtain pilot signals after phase rotation; wherein, ckA modulated signal representing energy normalization; t represents the period of the symbol; n iskRepresenting zero-mean complex white Gaussian noise introduced by a communication channel, in-phase components and orthogonal components of which are independent of each other, and the variance is N in mean0/2, △ fT representing the unknown normalized carrier Doppler frequency offset, △ aT2Representing an unknown normalized carrier doppler rate of change offset; understandably, the phasesThe pilot signal after bit rotation contains Doppler frequency offset and Doppler change rate;
carrying out differential operation on the pilot signal after the phase rotation according to a formula 1 to obtain the instantaneous phase of the pilot signal;
rk+1e-π/4*conj(rke-π/4) (1)
wherein, conj (r)ke-π/4) Is represented by rke-π/4Complex conjugation of (a).
For this implementation, preferably, the estimating, according to a minimum mean square error criterion, a doppler shift of a carrier in the pilot signal and a doppler change rate of the carrier with respect to the instantaneous phase of the pilot signal to obtain a doppler shift estimated value of the carrier in the pilot signal and a doppler change rate estimated value of the carrier in the pilot signal may include:
for the instantaneous phase of the pilot signal, obtaining a doppler frequency offset estimation value of a carrier in the pilot signal by using formula 2 and obtaining a doppler change rate estimation value of a carrier in the pilot signal by using formula 3:
Figure BDA0001940552120000061
Figure BDA0001940552120000062
wherein the content of the first and second substances,
Figure BDA0001940552120000063
for instantaneous phase, calculate coefficients αN，βN，γNRespectively, as follows:
Figure BDA0001940552120000064
it should be noted that the Minimum Mean square Error criterion (MMSE) belongs to a pilot signal open loop acquisition method. High dynamic environment when doppler frequency shift and doppler rate of change shift are presentLower, the baseband signal phase of the receiving end
Figure BDA0001940552120000065
Can be expressed as:
Figure BDA0001940552120000066
wherein the content of the first and second substances,
Figure BDA0001940552120000067
representing the phase value of the carrier at time t, △ f representing the Doppler frequency shift of the carrier, △ a representing the Doppler rate shift of the carrier, and for the remainder R (t, △ t), when △ t goes to 0, the term is infinitely small on the third order of △ t1,t2,...,tN(t1<t2<...<tN) Sampled value of time phase
Figure BDA0001940552120000071
And (3) performing differential calculation of the phase, and after phase ambiguity is eliminated, obtaining a phase error calculation formula as follows:
Figure BDA0001940552120000072
Figure BDA0001940552120000073
based on MMSE criterion, in order to obtain the minimum error e (N), let e (N) pair
Figure BDA0001940552120000074
And
Figure BDA0001940552120000075
calculating the partial derivatives and making them be 0, and obtaining the following products by sorting:
Figure BDA0001940552120000076
Figure BDA0001940552120000077
wherein the content of the first and second substances,
Figure BDA0001940552120000078
s204: the receiving end compensates the received signal according to the Doppler frequency offset estimation value of the carrier in the pilot signal and the Doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal;
for S204, in a possible implementation manner, the compensating the received signal according to the doppler frequency offset estimation value of the carrier in the pilot signal and the doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal may include:
by compensation formula
Figure BDA0001940552120000079
And compensating the received signal to obtain a compensated signal.
S205, the receiving end carries out Viterbi decoding on the compensated signal, and in the decoding process, the following processing steps are carried out until L:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment;
for S205, in a possible implementation manner, the acquiring a survivor path of each state at a next time corresponding to the current time may include:
setting sequence {k-3,ck-2,ck-1Denoted as the transmitted information sequence up to the current time kT, Sk∈ {0, 1., Q-1} represents a state node set at the current time, and comprises Q nodes;
setting up
Figure BDA0001940552120000081
Expressed as an information sequence entering the mth node from a certain survivor path, the branch metric from the mth node at the current moment to a certain node at the next moment is obtained as
Figure BDA0001940552120000082
Branch metric value according to each state
Figure BDA0001940552120000083
Obtaining the path metric values of all the states from the beginning of decoding to the next time
Figure BDA0001940552120000084
And selecting a minimum value from the path metric values of each state, and deleting other paths to obtain survival paths of all the states from the next moment.
It should be noted that Viterbi decoding is a maximum likelihood decoding algorithm of convolutional codes, and in the embodiment of the present invention, a specific signal flow process of Viterbi decoding may be as shown in fig. 3, in the absence of a synchronization error, a signal passes through an Additive White Gaussian Noise (AWGN) channel to a receiving end, where a complex baseband signal received by the receiving end is denoted as r (t) s (t | α) + n (t);
wherein α is represented as a sequence of transmitted symbols and N (t) is a double sideband power spectral density of N0A complex white gaussian noise signal.
In the case of the transmit waveform s (t), the prior probability of the received signal is:
Figure BDA0001940552120000085
by maximizing a priori probability information, i.e.The transmitted symbol sequence estimate can be obtained by the idea of Maximum likelihood (M L, Maximum L ikelihood) as
Figure BDA0001940552120000086
Simplifying the transmit symbol sequence estimate can result in:
Figure BDA0001940552120000087
when estimating the transmitted sequence, the complexity increases exponentially with the length L of the sequence, and the computational complexity can be reduced in an iterative manner to simplify the computation
Figure BDA0001940552120000091
Then in the nth symbol period, the path metric is calculated as:
Figure BDA0001940552120000092
wherein the content of the first and second substances,
Figure BDA0001940552120000093
called the branch metric at the current time, which is calculated by the equation:
Figure BDA0001940552120000094
by continuously calculating the state path metric of each state reached at each moment, selecting the path record with the maximum path metric of each state for storage, namely storing the survivor path, and tracing back after recording all the survivor paths at L T moment, namely the ending moment, the Viterbi decoding of the transmitted symbol sequence can be completed, so that the code sequence obtained by demodulation is a sequence with the maximum prior probability.
For S205, in a possible implementation manner, the acquiring an instantaneous phase value of a next time corresponding to the current time may include:
will be described asThe phase shift amount of the previous time is determined as
Figure BDA0001940552120000095
After the phase offset passes through a third-order phase-locked loop, an instantaneous phase value at the next moment is obtained
Figure BDA0001940552120000096
Wherein, the instantaneous phase value calculation formula at the next moment is expressed as:
Figure BDA0001940552120000097
gamma is the third order phase locked loop gain.
It should be noted that, because the conventional second-order phase-locked loop has a steady-state phase difference in a high dynamic environment, the third-order phase-locked loop according to the embodiment of the present invention eliminates the steady-state phase difference in the high dynamic environment, and the third-order phase-locked loop according to the embodiment of the present invention may be used for tracking a residual carrier phase offset.
In the embodiment of the present invention, the system function of the three-order phase-locked loop structure is:
Figure BDA0001940552120000098
wherein, ω isnFor the loop natural angular frequency, a and b are the loop filter design parameters in the phase-locked loop.
In a specific implementation, for the design of a third-order phase-locked loop, that is, for the design of a loop filter, referring to fig. 4, a bilinear transformation method is used to implement the transformation from an analog domain to a digital domain, and parameters of the loop filter can be obtained as follows:
Figure BDA0001940552120000101
Figure BDA0001940552120000102
Figure BDA0001940552120000103
wherein, KdIs the phase discriminator gain, K0Is the gain of a Numerically Controlled Oscillator (NCO).
S206, for the L th moment, the receiving end selects the survivor path with the minimum path metric value to backtrack, and the sequence closest to the coding output sequence is obtained.
It is to be understood that, until S206, the Viterbi decoding process is completed, so that the demodulated code sequence is a sequence with the maximum prior probability, so that the receiving end can complete carrier synchronization according to the sequence closest to the coded output sequence. Since the carrier synchronization method shown in fig. 2 combines the pilot signal and Viterbi decoding, the effects of improving the phase tracking accuracy and reducing the system bit error rate can be achieved.
Example two
Based on the same inventive concept of the foregoing embodiment, referring to fig. 5, it shows a composition of a system 5 for carrier synchronization provided in the embodiment of the present invention, where the system 5 includes a transmitting end 51 and a receiving end 52; wherein, the transmitting end 51 comprises a coding part 511 and a modulation part 512; wherein the content of the first and second substances,
the encoding part 511 is configured to perform convolutional coding with a code rate of 1/2 on the input single-bit information to obtain a coded output sequence with a length of L;
the modulation part 512 is configured to modulate the coded output sequence according to a set modulation strategy to obtain modulated data with a length of L + N;
the receiving end 52 includes: an estimation section 521, a compensation section 522, and a decoding section 523; wherein, the estimating part 521 is configured to estimate the doppler frequency offset of the carrier wave in the pilot signal and the doppler change rate of the carrier wave according to the instantaneous phase of the pilot signal in the received signal;
the compensation part 522 is configured to compensate the received signal according to the doppler frequency offset estimation value of the carrier in the pilot signal and the doppler change rate estimation value of the carrier in the pilot signal, so as to obtain a compensated signal;
the decoding part 523 is configured to perform Viterbi decoding on the compensated signal, and perform the following processing for each current time until L th time in the decoding process:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment;
for the L th time, the receiving end selects the survivor path with the minimum path metric value to backtrack, and the sequence closest to the coded output sequence is obtained.
For the above scheme, in a possible implementation manner, the encoding part 511 is configured to perform convolutional coding with a code rate of 1/2 on the input single-bit information to obtain a two-bit encoding sequence, where the length of the two-bit encoding sequence is L, and output expressions of the two-bit encoding sequence are g respectively1(x)＝1+x+x2And g2(x)＝1+x2Where x denotes a delay of a single bit of information of the input, x2A second delay representing the input single bit information.
For the above scheme, in a possible implementation manner, the modulating part 512 is configured to add an all-zero sequence with a length of N before encoding the output sequence, so as to obtain a signal with a length of L + N to which a pilot sequence is added;
QPSK modulation is carried out on the signal added with the pilot frequency sequence to obtain a four-phase modulation signal; where the "00" sequence maps to 1/4 π, the "01" sequence maps to 3/4 π, the "10" sequence maps to 5/4 π, and the "11" sequence maps to 7/4 π.
With regard to the above scheme, in a possible implementation manner, the estimation part 521 is configured to:
extracting a pilot signal of a received signal to obtain an instantaneous phase of the pilot signal;
and aiming at the instantaneous phase of the pilot signal, estimating the Doppler frequency offset and the Doppler change rate of the carrier in the pilot signal according to a minimum mean square error criterion to obtain the Doppler frequency offset estimation value and the Doppler change rate estimation value of the carrier in the pilot signal.
In the above implementation, preferably, the estimating part 521 is configured to:
for the received signal
Figure BDA0001940552120000111
The first N data are subjected to-1/4 pi phase rotation to obtain pilot signals after phase rotation; wherein, ckA modulated signal representing energy normalization; t represents the period of the symbol; n iskRepresenting zero-mean complex white Gaussian noise introduced by a communication channel, in-phase components and orthogonal components of which are independent of each other, and the variance is N in mean0/2, △ fT representing the unknown normalized carrier Doppler frequency offset, △ aT2Representing an unknown normalized carrier doppler rate of change offset; understandably, the phase-rotated pilot signal contains doppler frequency offset and doppler change rate;
carrying out differential operation on the pilot signal after the phase rotation according to a formula 1 to obtain the instantaneous phase of the pilot signal;
rk+1e-π/4*conj(rke-π/4) (1)
wherein, conj (r)ke-π/4) Is represented by rke-π/4Complex conjugation of (a).
In the above implementation, preferably, the estimating part 521 is configured to:
for the instantaneous phase of the pilot signal, obtaining a doppler frequency offset estimation value of a carrier in the pilot signal by using formula 2 and obtaining a doppler change rate estimation value of a carrier in the pilot signal by using formula 3:
Figure BDA0001940552120000121
Figure BDA0001940552120000122
wherein the content of the first and second substances,
Figure BDA0001940552120000123
for instantaneous phase, calculate coefficients αN，βN，γNRespectively, as follows:
Figure BDA0001940552120000124
with regard to the above solution, in a possible implementation manner, the compensation part 522 is configured to:
by compensation formula
Figure BDA0001940552120000125
And compensating the received signal to obtain a compensated signal.
For the above solution, in a possible implementation manner, the decoding section 523 is configured to:
setting sequence {k-3,ck-2,ck-1Denoted as the transmitted information sequence up to the current time kT, Sk∈ {0, 1., Q-1} represents a state node set at the current time, and comprises Q nodes;
setting up
Figure BDA0001940552120000126
Expressed as an information sequence entering the mth node from a certain survivor path, the branch metric from the mth node at the current moment to a certain node at the next moment is obtained as
Figure BDA0001940552120000127
Branch metric value according to each state
Figure BDA0001940552120000128
Obtaining the path metric values of all the states from the beginning of decoding to the next time
Figure BDA0001940552120000131
And selecting a minimum value from the path metric values of each state, and deleting other paths to obtain survival paths of all the states from the next moment.
For the above solution, in a possible implementation manner, the decoding section 523 is configured to:
determining the phase offset of the current time as
Figure BDA0001940552120000132
After the phase offset passes through a third-order phase-locked loop, an instantaneous phase value at the next moment is obtained
Figure BDA0001940552120000133
Wherein, the instantaneous phase value calculation formula at the next moment is expressed as:
Figure BDA0001940552120000134
gamma is the third order phase locked loop gain.
It is understood that in this embodiment, "part" may be part of a circuit, part of a processor, part of a program or software, etc., and may also be a unit, and may also be a module or a non-modular.
In addition, each component in the embodiment may be integrated in one processing unit, or each unit may exist alone physically, or two or more units are integrated in one unit. The integrated unit can be realized in a form of hardware or a form of a software functional module.
Based on the understanding that the technical solution of the present embodiment essentially or a part contributing to the prior art, or all or part of the technical solution may be embodied in the form of a software product stored in a storage medium, and include several instructions for causing a computer device (which may be a personal computer, a server, or a network device, etc.) or a processor (processor) to execute all or part of the steps of the method of the present embodiment. And the aforementioned storage medium includes: a U-disk, a removable hard disk, a Read Only Memory (ROM), a Random Access Memory (RAM), a magnetic disk or an optical disk, and other various media capable of storing program codes.
Based on the above structure of the system 5, referring to fig. 6, it shows another composition of the system 5 for carrier synchronization provided in the embodiment of the present invention, where the system 5 includes a sending end 51 and a receiving end 52;
the transmitting end 51 comprises a first communication interface 601, a first memory 602 and a first processor 603; the various components are coupled together by a first bus system 604. It is understood that the first bus system 804 is used to enable connection communications between these components. The first bus system 804 includes a power bus, a control bus, and a status signal bus in addition to a data bus. For clarity of illustration, however, the various buses in the transmit side 51 are labeled as a first bus system 804 in FIG. 5; wherein the content of the first and second substances,
the first communication interface 601 is configured to receive and transmit signals in a process of receiving and transmitting information with other external network elements;
the first memory 602 for storing a computer program operable on the first processor 603;
the first processor 603, configured to execute the following steps when running the computer program:
carrying out convolutional code coding with the code rate of 1/2 on the input single-bit information to obtain a coded output sequence with the length of L, and modulating the coded output sequence according to a set modulation strategy to obtain modulation data with the length of L + N;
and the receiving end 52 includes: a second communication interface 605, a second memory 606, and a second processor 607; the various components are coupled together by a second bus system 608. It will be appreciated that the second bus system 608 is used to enable communications for connections between these components. The second bus system 608 includes a power bus, a control bus, and a status signal bus in addition to a data bus. For clarity of illustration, however, the various buses in the receiver 52 are labeled in FIG. 6 as the second bus system 608;
the second communication interface 605 is configured to receive and transmit signals in a process of receiving and transmitting information with other external network elements;
the second memory 606 for storing a computer program capable of running on the second processor 607;
the second processor 607, configured to execute the following steps when running the computer program:
estimating Doppler frequency offset and Doppler change rate of a carrier in a pilot signal according to the instantaneous phase of the pilot signal in a received signal; and the number of the first and second groups,
compensating the received signal according to the Doppler frequency offset estimation value of the carrier in the pilot signal and the Doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal; and the number of the first and second groups,
performing Viterbi decoding on the compensated signal, and performing the following processing for each current time until L th time in the decoding process:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment; and the number of the first and second groups,
and for the L th time, selecting a survivor path with the minimum path metric value for backtracking to obtain a sequence closest to the coded output sequence.
It is to be understood that the first Memory 602 and the second Memory 606 in embodiments of the present invention may be either volatile Memory or non-volatile Memory, or may include both volatile and non-volatile Memory, wherein non-volatile Memory may be Read-Only Memory (ROM), Programmable Read-Only Memory (PROM), Erasable Programmable Read-Only Memory (erase PROM, EPROM), Electrically Erasable Programmable Read-Only Memory (EEPROM), or flash Memory volatile Memory may be Random Access Memory (RAM) which serves as external cache Memory, by way of exemplary but not limiting illustration, many forms of RAM are available such as Static random access Memory (Static SRAM), dynamic random access Memory (dynamic RAM, DRAM), Synchronous dynamic random access Memory (syncronous DRAM, SDRAM), Double Data Rate Synchronous dynamic random access Memory (Double Data RAM), SDRAM (Synchronous DRAM), or other types of RAM suitable for Direct access systems including but not limited to RAM 602 and DRAM, and the other types of RAM intended for Direct access systems include the first and second RAM described herein.
And the first processor 603 and the second processor 607 may be integrated circuit chips having signal processing capability. In implementation, the steps of the above method may be performed by integrated logic circuits of hardware or instructions in the form of software in the first processor 603 and the second processor 607. The first Processor 603 and the second Processor 607 may be general purpose processors, Digital Signal Processors (DSPs), Application Specific Integrated Circuits (ASICs), Field Programmable Gate Arrays (FPGAs) or other programmable logic devices, discrete Gate or transistor logic devices, discrete hardware components. The various methods, steps and logic blocks disclosed in the embodiments of the present invention may be implemented or performed. A general purpose processor may be a microprocessor or the processor may be any conventional processor or the like. The steps of the method disclosed in connection with the embodiments of the present invention may be directly implemented by a hardware decoding processor, or implemented by a combination of hardware and software modules in the decoding processor. The software module may be located in ram, flash memory, rom, prom, or eprom, registers, etc. storage media as is well known in the art. The storage medium is located in the first memory 602 and the second memory 606, and the first processor 603 and the second processor 607 read the information in the first memory 602 and the second memory 606, and complete the steps of the above method in combination with the hardware thereof.
For a hardware implementation, the Processing units may be implemented within one or more Application Specific Integrated Circuits (ASICs), Digital Signal Processors (DSPs), Digital Signal Processing Devices (DSPDs), Programmable logic devices (P L D), Field-Programmable Gate arrays (FPGAs), general purpose processors, controllers, microcontrollers, microprocessors, other electronic units configured to perform the functions described herein, or a combination thereof.
For a software implementation, the techniques described herein may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The software codes may be stored in a memory and executed by a processor. The memory may be implemented within the processor or external to the processor.
Specifically, when the first processor 603 and the second processor 607 are further configured to run the computer program, the steps performed by the transmitting end and the receiving end in the technical solution shown in fig. 2 are correspondingly performed, which is not described herein again.
EXAMPLE III
Based on the carrier synchronization method and/or the carrier synchronization system described in the foregoing embodiments, the present embodiment verifies the technical solutions described in the foregoing embodiments through specific simulation examples.
In the present simulation example, the simulation conditions and contents are as follows:
matlab was used to compare the simulation analysis of the previous examples with the existing carrier tracking algorithm. The simulation parameters are set as follows: the conventional communication signal is QPSK modulated, the code rate is 1/2 convolutional code coding, the coding length is 1000, the cycle number is 100000, and the range of the signal-to-noise ratio is 2dB to 8 dB. The specific simulation content is as follows:
a. simulation analysis shows that when the normalized doppler frequency offset is zero, the system bit error rate performance changes with the normalized doppler change rate, the parameter setting is that the normalized doppler frequency △ fT is 0, and the normalized doppler change rate is △ aT2＝10-4，△aT2＝2×10-4，△aT2＝3×10-4，△aT2＝4×10-4，△aT2＝5×10-4。
b. Simulation analysis is carried out on the curve that the error rate performance of the system changes along with the normalized Doppler frequency shift when the normalized Doppler change rate is zero in the embodiment, and the parameter setting is the normalized Doppler change rate △ aT2The normalized doppler frequency offset is △ fT-1 × 10-3，△fT＝2×10-3，△fT＝4×10-3，△fT＝6×10-3，△fT＝8×10-3。
c. The parameter setting is normalized doppler frequency shift △ fT-1 × 10-3Normalized Doppler Change Rate △ aT2＝10-4。
The specific simulation results and analysis are as follows:
a. referring to fig. 7, the abscissa represents the signal-to-noise ratio range, and the ordinate represents the system error rate, it can be seen that when the normalized doppler frequency offset is zero, the error rate performance of the carrier synchronization method proposed in the foregoing embodiment deteriorates with the increase of the normalized doppler change rate, that is, when the normalized doppler change rate is greater than 4 × 10-3When time comes, loop tracking is prone to loss of lock, so tracking can deteriorate.
b. Referring to fig. 8, the abscissa represents the signal-to-noise ratio range, and the ordinate represents the system error rate, it can be seen that when the normalized doppler change rate is zero, the error rate performance of the carrier synchronization method proposed in the foregoing embodiment deteriorates with the increase of the normalized doppler frequency offset, that is, when the normalized doppler frequency offset is greater than the normalized doppler frequency offset, the doppler frequency offset exceeds the loop bandwidth of the phase-locked loop, so the tracking performance deteriorates.
c. Referring to fig. 9, the abscissa represents the snr range, the ordinate represents the system error rate, the symbol MMSEP LL in the figure represents the conventional MMSE algorithm assisted single-pll carrier tracking method, M L P LL represents the conventional maximum likelihood estimation assisted single-pll carrier tracking method, MMSE PSP represents the carrier synchronization method proposed in the previous embodiment, and qpside L represents the theoretical error rate curve of QPSK signal-5Compared with the traditional method, the signal-to-noise ratio of the method provided by the invention is improved by 4 dB. It can be seen that the carrier synchronization method proposed by the foregoing embodiment is significantly superior to the conventional method.
It should be noted that: the technical schemes described in the embodiments of the present invention can be combined arbitrarily without conflict.
The above description is only for the specific embodiments of the present invention, but the scope of the present invention is not limited thereto, and any person skilled in the art can easily conceive of the changes or substitutions within the technical scope of the present invention, and all the changes or substitutions should be covered within the scope of the present invention. Therefore, the protection scope of the present invention shall be subject to the protection scope of the appended claims.

Claims (9)

1. A method of carrier synchronization, the method comprising:
the transmitting end carries out convolutional code coding with code rate of 1/2 on the input single-bit information to obtain a coded output sequence with length of L;
the transmitting end modulates the coded output sequence according to a set modulation strategy to obtain modulated data with the length of L + N;
a receiving end estimates the Doppler frequency offset and the Doppler change rate of a carrier wave in a pilot signal according to the instantaneous phase of the pilot signal in a received signal;
the receiving end compensates the received signal according to the Doppler frequency offset estimation value of the carrier in the pilot signal and the Doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal;
the receiving end carries out Viterbi decoding on the compensated signal, and in the decoding process, the following processing is carried out until L th time aiming at each current time:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment;
for the L th time, the receiving end selects the survivor path with the minimum path metric value to backtrack, and obtains the sequence closest to the coding output sequence,
wherein, the receiving end estimates the Doppler frequency offset and Doppler change rate of the carrier in the pilot signal according to the instantaneous phase of the pilot signal in the received signal, and the method comprises the following steps:
the receiving end extracts the pilot signal of the received signal to obtain the instantaneous phase of the pilot signal,
the receiving end extracts a pilot signal of a received signal to obtain an instantaneous phase of the pilot signal, including:
the receiving end pair receives the signal
Figure FDA0002450603580000011
The first N data are subjected to-1/4 pi phase rotation to obtain pilot signals after phase rotation; wherein, ckA modulated signal representing energy normalization; t represents the period of the symbol; n iskRepresenting zero-mean complex white Gaussian noise introduced by a communication channel, in-phase components and orthogonal components of which are independent of each other, and the variance is N in mean02; Δ fT represents the unknown normalized carrier doppler frequency offset; Δ aT2Representing an unknown normalized carrier doppler rate of change offset;
the receiving end carries out differential operation on the pilot signal after the phase rotation according to a formula 1 to obtain the instantaneous phase of the pilot signal;
rk+1e-π/4*conj(rke-π/4) (1)
wherein, conj (r)ke-π/4) Is represented by rke-π/4Complex conjugation of (a).
2. The method of claim 1, wherein the receiving end estimates the doppler shift of the carrier and the doppler change rate of the carrier in the pilot signal according to the instantaneous phase of the pilot signal in the received signal, further comprising:
and aiming at the instantaneous phase of the pilot signal, the receiving end estimates the Doppler frequency offset and the Doppler change rate of the carrier in the pilot signal according to the minimum mean square error criterion to obtain the Doppler frequency offset estimation value and the Doppler change rate estimation value of the carrier in the pilot signal.
3. The method of claim 2, wherein the receiving end estimates the doppler shift of the carrier and the doppler change rate of the carrier in the pilot signal according to a minimum mean square error criterion for the instantaneous phase of the pilot signal, and obtains the doppler shift estimate of the carrier and the doppler change rate estimate of the carrier in the pilot signal, including:
for the instantaneous phase of the pilot signal, obtaining a doppler frequency offset estimation value of a carrier in the pilot signal by using formula 2 and obtaining a doppler change rate estimation value of a carrier in the pilot signal by using formula 3:
Figure FDA0002450603580000021
Figure FDA0002450603580000022
wherein the content of the first and second substances,
Figure FDA0002450603580000023
for instantaneous phase, calculate coefficients αN，βN，γNAre respectively shown as follows：
Figure FDA0002450603580000024
βN＝(2N+1)(8N+11)，γN＝(N+1)(N+2)。
4. The method of claim 1, wherein the receiving end compensates the received signal according to the doppler shift estimation value of the carrier in the pilot signal and the doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal, and the method comprises:
by compensation formula
Figure FDA0002450603580000025
And compensating the received signal to obtain a compensated signal.
5. The method according to claim 1, wherein the obtaining the survivor path of each state at the next time corresponding to the current time comprises:
setting sequence {k-3,ck-2,ck-1Denoted as the transmitted information sequence up to the current time kT, Sk∈ {0, 1., Q-1} represents a state node set at the current time, and comprises Q nodes;
setting up
Figure FDA0002450603580000031
Expressed as an information sequence entering the mth node from a certain survivor path, the branch metric from the mth node at the current moment to a certain node at the next moment is obtained as
Figure FDA0002450603580000032
Branch metric value according to each state
Figure FDA0002450603580000033
Obtaining the path degree of all the states from the beginning of decoding to the next timeMagnitude of
Figure FDA0002450603580000034
And selecting a minimum value from the path metric values of each state, and deleting other paths to obtain survival paths of all the states from the next moment.
6. The method of claim 1, wherein obtaining the instantaneous phase value for the next time corresponding to the current time comprises:
determining the phase offset of the current time as
Figure FDA0002450603580000035
After the phase offset passes through a third-order phase-locked loop, an instantaneous phase value at the next moment is obtained
Figure FDA0002450603580000036
Wherein, the instantaneous phase value calculation formula at the next moment is expressed as:
Figure FDA0002450603580000037
gamma is the third order phase locked loop gain.
7. The method of claim 1, wherein the transmitting end performs convolutional code coding with a code rate of 1/2 on the input single-bit information to obtain a coded output sequence, and the method comprises:
carrying out convolutional code coding with the code rate of 1/2 on the input single-bit information to obtain a two-bit coding sequence, wherein the length of the two-bit coding sequence is L, and the output expressions of the two-bit coding sequence are g respectively1(x)＝1+x+x2And g2(x)＝1+x2Where x denotes a delay of a single bit of information of the input, x2A second delay representing the input single bit information.
8. A system for carrier synchronization is characterized in that the system comprises a sending end and a receiving end;
the sending end comprises a first communication interface, a first memory and a first processor; wherein the content of the first and second substances,
the first communication interface is used for receiving and sending signals in the process of receiving and sending information with other external network elements;
the first memory for storing a computer program operable on the first processor;
the first processor, when executing the computer program, is configured to perform the following steps:
carrying out convolutional code coding with the code rate of 1/2 on the input single-bit information to obtain a coded output sequence with the length of L, and modulating the coded output sequence according to a set modulation strategy to obtain modulation data with the length of L + N;
the receiving end includes: a second communication interface, a second memory, and a second processor;
the second communication interface is used for receiving and sending signals in the process of receiving and sending information with other external network elements;
the second memory for storing a computer program operable on a second processor;
the second processor, when executing the computer program, is configured to perform the following steps:
estimating Doppler frequency offset and Doppler change rate of a carrier in a pilot signal according to the instantaneous phase of the pilot signal in a received signal; and the number of the first and second groups,
compensating the received signal according to the Doppler frequency offset estimation value of the carrier in the pilot signal and the Doppler change rate estimation value of the carrier in the pilot signal to obtain a compensated signal; and the number of the first and second groups,
performing Viterbi decoding on the compensated signal, and performing the following processing for each current time until L th time in the decoding process:
obtaining the survival path of each state at the next moment corresponding to the current moment,
acquiring an instantaneous phase value of the next moment corresponding to the current moment; and the number of the first and second groups,
for the L th time, selecting the survivor path with the minimum path metric value for backtracking to obtain the sequence closest to the coded output sequence,
estimating the Doppler frequency offset and Doppler change rate of a carrier wave in a pilot signal according to the instantaneous phase of the pilot signal in a received signal, wherein the method comprises the following steps:
extracting a pilot signal of a received signal, obtaining an instantaneous phase of the pilot signal,
the extracting a pilot signal of a received signal to obtain an instantaneous phase of the pilot signal includes:
for the received signal
Figure FDA0002450603580000051
The first N data are subjected to-1/4 pi phase rotation to obtain pilot signals after phase rotation; wherein, ckA modulated signal representing energy normalization; t represents the period of the symbol; n iskRepresenting zero-mean complex white Gaussian noise introduced by a communication channel, in-phase components and orthogonal components of which are independent of each other, and the variance is N in mean02; Δ fT represents the unknown normalized carrier doppler frequency offset; Δ aT2Representing an unknown normalized carrier doppler rate of change offset;
carrying out differential operation on the pilot signal after the phase rotation according to a formula 1 to obtain the instantaneous phase of the pilot signal;
rk+1e-π/4*conj(rke-π/4) (1)
wherein, conj (r)ke-π/4) Is represented by rke-π/4Complex conjugation of (a).
9. The system according to claim 8, wherein the second processor, when running the computer program, is further configured to perform the steps of any of claims 2 to 6;
the first processor, when executing the computer program, is further configured to perform the steps of claim 7.
CN201910020334.8A 2019-01-09 2019-01-09 Carrier synchronization method and system Active CN109617844B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN201910020334.8A CN109617844B (en) 2019-01-09 2019-01-09 Carrier synchronization method and system

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN201910020334.8A CN109617844B (en) 2019-01-09 2019-01-09 Carrier synchronization method and system

Publications (2)

Publication Number Publication Date
CN109617844A CN109617844A (en) 2019-04-12
CN109617844B true CN109617844B (en) 2020-08-07

Family

ID=66018382

Family Applications (1)

Application Number Title Priority Date Filing Date
CN201910020334.8A Active CN109617844B (en) 2019-01-09 2019-01-09 Carrier synchronization method and system

Country Status (1)

Country Link
CN (1) CN109617844B (en)

Families Citing this family (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110768919B (en) * 2019-09-11 2022-09-16 武汉船舶通信研究所（中国船舶重工集团公司第七二二研究所） Carrier synchronization method and device for satellite communication system
CN111510411B (en) * 2019-11-25 2023-05-30 南京中科晶上通信技术有限公司 Carrier phase synchronization processing method, device, terminal and storage medium
CN113055327B (en) * 2019-12-26 2022-11-29 鹤壁天海电子信息系统有限公司 Signal demodulation method and device and signal demodulator
CN111600628A (en) * 2020-05-12 2020-08-28 北京理工大学 Self-adaptive filtering narrow-band interference suppression method with Doppler first-order change rate
CN111624402B (en) * 2020-05-31 2022-09-02 西南电子技术研究所（中国电子科技集团公司第十研究所） Method for accurately estimating carrier frequency of weak PM signal
CN113721270A (en) * 2021-07-28 2021-11-30 江苏师范大学 Satellite signal carrier synchronization method and system
CN113852581A (en) * 2021-09-16 2021-12-28 电子科技大学 Phase noise estimation and elimination method for single carrier THP-FTN system
CN114189417B (en) * 2021-12-07 2023-10-17 北京零壹空间电子有限公司 Carrier frequency synchronization method, carrier frequency synchronization device, computer equipment and storage medium
CN116155668B (en) * 2023-04-20 2023-07-14 北京中天星控科技开发有限公司 Anti-frequency offset carrier recovery method, system and storage medium

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN1494794A (en) * 2001-01-19 2004-05-05 �����ɷ� Frequency searcher and frequency-locked data demodulator using programmable rotor
CN101984562A (en) * 2010-11-09 2011-03-09 大连工业大学 Narrow-band signal gain estimation method

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN1131614C (en) * 2002-02-26 2003-12-17 东南大学 Ruike and equalization cascade receiving method under the code division multiple address low band-spreading ratio and its equipment
JP4352035B2 (en) * 2005-06-21 2009-10-28 株式会社東芝 OFDM demodulator, method and program
CN101692661A (en) * 2009-10-19 2010-04-07 上海奇微通讯技术有限公司 High-efficient differential interference rejection circuit and high-efficient differential interference rejection method
CN105763500A (en) * 2014-12-20 2016-07-13 西安飞东电子科技有限责任公司 Frequency deviation, time delay and phase deviation combined synchronization method of continuous phase modulation signals

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN1494794A (en) * 2001-01-19 2004-05-05 �����ɷ� Frequency searcher and frequency-locked data demodulator using programmable rotor
CN101984562A (en) * 2010-11-09 2011-03-09 大连工业大学 Narrow-band signal gain estimation method

Also Published As

Publication number Publication date
CN109617844A (en) 2019-04-12

Similar Documents

Publication Publication Date Title
CN109617844B (en) Carrier synchronization method and system
JP3100447B2 (en) Adaptive equalizer and receiver
US8090060B2 (en) Demodulation technique for GFSK and DPSK
JP4098096B2 (en) Spread spectrum receiver
US7136445B2 (en) Phase tracker for linearly modulated signal
KR20120016294A (en) Improved sync detection and frequency recovery for satellite systems
US20170180171A1 (en) Frequency synchronization of convolutionally coded gfsk signals
CN102413089B (en) Shannon limit coding GMSK demodulation method used for satellite communication system
US8571139B2 (en) Method for tracking the phase of a signal modulated by continuous phase modulation and synchronization device implementing said method
CN108683620B (en) Phase noise resistant blind carrier recovery method suitable for high-order modulation mode
JP2008530951A (en) Demodulator and receiver for pre-encoded partial response signals
US7415078B2 (en) Demodulation for phase modulation
CN113132285A (en) Digital demodulation system and method
CN110519200B (en) Polarization code auxiliary carrier synchronization system and method under low signal-to-noise ratio environment
CN109714286B (en) Carrier frequency offset estimation method for Pi/8D8PSK demodulation
US6721366B1 (en) Phase tracking apparatus and method for continuous phase modulated signals
TWI407737B (en) Method and apparatus for phase reference tracking of digital phase modulated signals in the receiver
CN109818894B (en) GMSK signal detection method and detection device under multipath channel
CN111262594A (en) LDPC code assisted carrier synchronization system, medium, communication system
US8811546B2 (en) Adaptive reference symbol method and apparatus for a receiver
KR100945532B1 (en) Apparatus and method for estimating phase error using variable step size
JP4970283B2 (en) High memory efficiency sliding window addition
JP4268498B2 (en) Phase error correction circuit, receiving apparatus using the same, and phase error correction method
JPH09214574A (en) Phase detector for data synchronization device and its operating method
US6940927B2 (en) Simplified symbol timing tracking circuit for a CPM modulated signal

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant